Power conversion device

Abstract

A reactor (2), switching elements (3a-3d, 5a-5d), a first capacitor (6), and a second capacitor (4) are provided between an AC power source (1) and a DC load (7). The controller (8) controls the operation of the switching elements (3a-3d, 5a-5d). The controller (8) controls the voltage of the first capacitor (6) and the voltage of the second capacitor (4) to predetermined command values ​​and switches between a first control mode of controlling the switching elements (3a-3d, 5a-5d) at a constant switching frequency during an AC cycle of the AC power source (1) and a second control mode of controlling the switching elements (3a-3d, 5a-5d) at a frequency lower than the switching frequency of the first control mode, based on load information for the DC load (7).

Description

Technical Field

[0001] The present application relates to a power conversion device. Background Art

[0002] Conventional power conversion devices include an example in which a reactor, an inverter consisting of a plurality of switching elements and capacitors, and a converter consisting of a plurality of switching elements and smoothing capacitors are connected in series with an AC power supply. Furthermore, when power conversion is performed between the AC voltage of the AC power supply and the DC voltage of the smoothing capacitor, the switching elements of the inverter and the switching elements of the converter are controlled in such a manner that the charging and discharging operations of the DC capacitors are performed within one switching cycle of the inverter and the charging and discharging amounts are equalized (for example, see Patent Document 1 below).

[0003] In the above-mentioned conventional power conversion device, the DC capacitor controls the switching action of each switching element in such a way that the charge and discharge amounts thereof are equal within one switching cycle of the inverter, regardless of the long cycle of the AC power supply. Therefore, the charge and discharge amounts themselves are reduced, and the ripple voltage can be suppressed. Therefore, the capacitance required in the DC capacitor is reduced, and the device can be miniaturized.

[0004] Prior art literature

[0005] Patent Literature

[0006] Patent Document 1: Japanese Patent No. 6129450 Summary of the Invention

[0007] However, in the conventional technology shown in Patent Document 1, PWM (Pulse Width Modulation) is performed at a constant switching frequency during the AC cycle regardless of the load condition. This leads to an increase in switching losses in the power conversion device and high-frequency losses (iron loss and copper loss) in the reactor, resulting in a reduction in circuit efficiency in the light load range.

[0008] The present application discloses a technology for solving the above-mentioned problems, and its purpose is to suppress the switching loss and high-frequency loss (iron loss, copper loss) of the inductor generated in the power conversion device more than before by reducing the total number of switching times of multiple switching elements according to the load state, and provide a high-efficiency power conversion device in a wide range.

[0009] The power conversion device disclosed in the present application is provided with at least one inductor, a plurality of switching elements and a first capacitor between an AC power source and a DC load, and a second capacitor is provided between the inductor and the first capacitor, and the power conversion device is provided with a controller for controlling the switching action of the switching element. The power conversion device performs power conversion between the AC voltage of the AC power source and the voltage of the first capacitor, wherein the controller controls the voltage of the first capacitor and the voltage of the second capacitor to predetermined command values ​​while switching between a first control mode of controlling the switching element at a constant switching frequency during an AC cycle of the AC power source and a second control mode of controlling the switching element at a frequency lower than the switching frequency of the first control mode based on load information for the DC load.

[0010] According to the power conversion device disclosed in the present application, the total number of switching times of multiple switching elements can be reduced according to the state of the DC load, so the switching loss generated in the power conversion device and the high-frequency loss (iron loss, copper loss) of the inductor can be suppressed, and an efficient device can be provided in a wide range. BRIEF DESCRIPTION OF THE DRAWINGS

[0011] Figure 1 This is a circuit diagram of the power conversion device according to the first embodiment.

[0012] Figure 2 This is a block diagram of the controller according to the first embodiment.

[0013] Figure 3 A is an explanatory diagram showing the magnitude relationship among the AC voltage, the voltage of the second capacitor, and the voltage of the first capacitor according to the first embodiment.

[0014] Figure 3 B is an explanatory diagram showing the magnitude relationship among the AC voltage, the voltage of the second capacitor, and the voltage of the first capacitor according to the first embodiment.

[0015] Figure 4 This shows the main operation waveform during the continuous SW operation according to the first embodiment and the switching pattern in each operation range.

[0016] Figure 5 The main operation waveform during the simple SW boost operation according to the first embodiment and the switching pattern in each operation range are shown.

[0017] Figure 6 These are other main operation waveforms during the simple SW boost operation according to the first embodiment and the switching patterns in each operation range.

[0018] Figure 7The main operation waveform during the simple SW step-down operation according to the first embodiment and the switching pattern in each operation range are shown.

[0019] Figure 8 1 and 2 show other main operation waveforms during the simple SW step-down operation according to the first embodiment and the switching patterns in each operation range.

[0020] Figure 9 This is a block diagram of the control operation determiner according to the first embodiment.

[0021] Figure 10 This is a block diagram of the second capacitor voltage command value calculator according to the first embodiment.

[0022] Figure 11 This is a block diagram of the second capacitor voltage controller according to the first embodiment.

[0023] Figure 12 This is a block diagram of the continuous switching controller according to the first embodiment.

[0024] Figure 13 A is a switching pattern in the operating range R1 according to the first embodiment.

[0025] Figure 13 B is a switching pattern in the operating range R2 according to the first embodiment.

[0026] Figure 13 C is a switching pattern in the operating range R3 according to the first embodiment.

[0027] Figure 14 A is a diagram for explaining the operation of the duty totalizer according to the first embodiment.

[0028] Figure 14 B is a diagram for explaining the operation of the duty totalizer according to the first embodiment.

[0029] Figure 15 This is a block diagram of the simplified SW boost controller according to the first embodiment.

[0030] Figure 16 This is a block diagram of the SW carrier calculator A according to the first embodiment.

[0031] Figure 17 This is a block diagram of the simplified SW step-down controller according to the first embodiment.

[0032] Figure 18 This is a block diagram of the SW carrier calculator B according to the first embodiment.

[0033] Figure 19 This is a circuit diagram of a power conversion device according to the second embodiment.

[0034] Figure 20 This is a block diagram of a controller according to the second embodiment.

[0035] Figure 21 This is a block diagram of a control operation determiner according to the second embodiment.

[0036] Figure 22 This is a circuit diagram of a power conversion device according to the third embodiment.

[0037] Figure 23 This is a circuit diagram of a power conversion device according to a fourth embodiment.

[0038] Figure 24 This is a circuit diagram of a power conversion device according to the fifth embodiment.

[0039] Figure 25 This is a block diagram of a controller according to the fifth embodiment.

[0040] Figure 26 This shows the main operation waveform during the continuous SW operation according to the fifth embodiment and the switching pattern in each operation range.

[0041] Figure 27 The main operation waveforms and switching patterns in each operation range during the simple SW step-up operation according to the fifth embodiment are shown.

[0042] Figure 28 These are other main operation waveforms during the simple SW boost operation according to the fifth embodiment and the switching patterns in each operation range.

[0043] Figure 29 This is the switching pattern in each operating range according to the fifth embodiment.

[0044] Figure 30 This is a circuit diagram of a power conversion device according to a sixth embodiment.

[0045] Figure 31 This is a diagram showing an example of hardware of a controller in the embodiment.

[0046] (Explanation of Symbols)

[0047] 100, 200, 300, 400, 500, 600: power conversion device; 1: AC power supply; 2: reactor; 3: single-phase inverter; 4: second capacitor; 5: single-phase converter; 6: first capacitor; 7: load; 8, 32, 35, 37, 40, 44: controller; 3a, 3b, 3c, 3d, 5a, 5b, 5c, 5d, 39a, 39b, 39c, 39d: semiconductor switch element; G3a, G3b, G3c, G3d, G5a, G5b, G5c, G5d, G39a, G39b, G39c, G39d: Gate signal; 9, 33: Control action determiner; 10: Second capacitor voltage command value calculator; 11: Second capacitor voltage controller; 12, 41: Main controller; 13: Continuous switching controller; 14: Simple switching boost controller; 15: Simple switching buck controller; 16, 42: Gate signal Generator; 17: Area determiner; 18, 34: SW action determiner; 19: Action range R3 period calculator; 20: Vsubref calculator; 21: FFDuty calculator; 22: Duty totalizer; 23: Action range determiner; 24: On-time calculator; 25: SW carrier calculator A; 26: Action range R1 / SW carrier calculator A; 27: Action range R2 / SW carrier calculator A; 28: SW carrier calculator B; 29: Action range R1 / SW carrier calculator B; 30: Action range R2 / SW carrier calculator B; 31: Action range R3 / SW carrier calculator B; 3aa, 3dd, 5aa, 5cc, 5dd, 38a, 38b, 40a, 40b: Diodes; 36: Diode bridge; 38: Diode rectifier branch; 39: Multi-stage drive branch; 1000: Processor; 1010: Storage device. DETAILED DESCRIPTION

[0048] Implementation method 1.

[0049] Figure 1 This is a circuit diagram of the power conversion device according to the first embodiment.

[0050] The power conversion device 100 of this first embodiment includes a main circuit for converting the AC voltage and AC power of a single-phase AC power source 1 into a DC voltage and DC power and outputting them to a load 7, and a controller 8 for controlling the main circuit. The main circuit is configured by sequentially connecting a reactor 2, a single-phase inverter 3, and a single-phase converter 5 between the AC power source 1 and a first capacitor 6.

[0051] The single-phase inverter 3 includes a first branch having a pair of semiconductor switching elements 3a and 3b connected in series, a second branch having a pair of semiconductor switching elements 3c and 3d connected in series, and a second capacitor 4 connected between the first branch and the second branch in parallel with the first branch and the second branch.

[0052] The single-phase converter 5 includes a third leg having a pair of semiconductor switching elements 5a and 5b connected in series, and a fourth leg having a pair of semiconductor switching elements 5c and 5d connected in series. The third leg and the fourth leg are connected in parallel with each other.

[0053] Reactor 2 is connected between the P busbar of AC power supply 1 and the midpoint of the first leg of single-phase inverter 3. Furthermore, the midpoint of the third leg of single-phase converter 5 is connected to the midpoint of the second leg of single-phase inverter 3, and the midpoint of the fourth leg of single-phase converter 5 is connected to the N busbar of AC power supply 1. Furthermore, regarding first capacitor 6, the upper ends of the third and fourth legs of single-phase converter 5 and load 7 are connected to the P-side terminal, while the lower ends of the third and fourth legs of single-phase converter 5 and load 7 are connected to the N-side terminal.

[0054] In addition, the semiconductor switching elements used in the single-phase inverter 3 and the single-phase converter 5 are preferably IGBTs (Insulated Gate Bipolar Transistors) with diodes connected in anti-parallel, MOSFETs (Metal Oxide Semiconductor Field Effect Transistors) with diodes connected between the source and drain, or cascaded GaN-HEMTs (Gallium nitride-High Mobility Transistors). In addition, the feedback diode can be a diode built into the IGBT, MOSFET, or GaN-HEMT, or a diode can be provided externally. In addition to resistive loads and battery loads, the load 7 can also be a motor or compressor connected via an inverter (not shown). In addition, the first capacitor 6 and the second capacitor 4 can be composed of aluminum electrolytic capacitors or film capacitors.

[0055] The AC voltage Vac of the AC power source 1, the voltage Vdc of the first capacitor 6 (hereinafter also referred to as the output voltage) and the voltage Vsub of the second capacitor 4 are detected by voltage sensors (not shown) and input to the controller 8. Furthermore, the AC current iac of the AC power source 1 is detected by a current sensor (not shown) and input to the controller 8.

[0056] In the controller 8, based on the load information (i.e., the above-mentioned voltage information and current information) detected and input by the above-mentioned sensors, the gate signals G3a, G3b, G3c, G3d of the single-phase inverter 3 and the gate signals G5a, G5b, G5c, G5d of the single-phase converter 5 are generated to control the power conversion device 100.

[0057] Figure 2 This is a block diagram of the controller according to the first embodiment.

[0058] The controller 8 is composed of a control operation determination unit 9, a second capacitor voltage command value calculation unit 10 (in Figure 2 The voltage command value calculator 10 is referred to as the second capacitor voltage controller 11 (in Figure 2 It is composed of a voltage controller 11 , a main controller 12 and a gate signal generator 16 .

[0059] Here, the control action determiner 9 outputs a control selection signal SelectSignal (in the input power) for selecting a control action to be performed in the main controller 12 based on the load information, that is, the magnitude relationship between the AC voltage Vac and the output voltage Vdc, and the information of the input power calculated based on the AC voltage Vac and the AC current iac. Figure 2 SS in Chinese).

[0060] The second capacitor voltage command value calculator 10 calculates and outputs a voltage command value Vsub* of the second capacitor 4 according to the operating state based on the AC voltage Vac and the voltage command value (hereinafter also referred to as output voltage command value) Vdc* of the first capacitor 6 .

[0061] The second capacitor voltage controller 11 calculates and outputs a control signal DutyVsub for controlling the voltage Vsub of the second capacitor 4 to be equal to the voltage command value Vsub*. In the following, command values ​​serving as control targets are denoted by *.

[0062] The main controller 12 is composed of three components: a continuous switching controller 13, a simple switching boost controller 14, and a simple switching buck controller 15. In this application, switches are sometimes referred to simply as "SW." In particular, the continuous switching controller 13 is referred to simply as "continuous SW controller 13," the simple switching boost controller 14 is referred to simply as "simple SW boost controller 14," and the simple switching buck controller 15 is referred to simply as "simple SW buck controller 15."

[0063] The main controller 12 determines which of the three controllers 13, 14, and 15 to use for control based on the control selection signal SS input from the control operation determiner 9. Furthermore, based on the voltage and current information Vac, Vdc, Vsub, and iac detected by the sensors and the control signal DutyVsub calculated by the second capacitor voltage controller 11, the main controller 12 generates a duty cycle signal Dutytotal for power factor control and voltage control, and a carrier signal Carrier for comparison with the duty cycle signal Dutytotal.

[0064] In addition, the gate signal generator 16 controls and calculates the gate signal through PWM (Pulse Width Modulation) based on the duty cycle signal Dutytotal and the carrier signal Carrier, and outputs gate signals 3a~3d and 5a~5d that turn on / off the gates of the semiconductor switching elements 3a~3d of the single-phase inverter 3 and the semiconductor switching elements 5a~5d of the single-phase converter 5.

[0065] Next, the operation of the power conversion device 100 according to the first embodiment will be described.

[0066] The power conversion device 100 switches the AC current iac input from the AC power supply 1 while coordinating the single-phase inverter 3 and the single-phase converter 5, controls the AC current iac flowing through the inductor 2 to a high power factor while boosting or lowering the voltage, and uses the first capacitor 6 to smooth the power and supply DC power to the load 7.

[0067] In particular, in the first embodiment, the continuous switching controller 13 , the simple switching step-up controller 14 , and the simple switching step-down controller 15 constituting the main controller 12 perform three control operations: continuous switching control, simple switching step-up control, and simple switching step-down control.

[0068] The continuous SW control by the continuous SW controller 13 corresponds to the first control mode in the claims, and the simple SW step-up control by the simple SW step-up controller 14 and the simple SW step-down control by the simple SW step-down controller 15 correspond to the second control mode in the claims.

[0069] First, use Figure 3 Describes the overview of each control operation.

[0070] exist Figure 3 A and B show the magnitude relationship among the AC voltage Vac, the voltage Vsub of the second capacitor 4, and the voltage (output voltage) Vdc of the first capacitor 6 in a system half cycle. Figure 3A shows the waveform when the voltage is increased. Figure 3 B shows the waveform during voltage reduction. In the power conversion device 100 according to Embodiment 1, there are two operating ranges during voltage reduction, designated R1 and R2, and three operating ranges during voltage reduction, designated R1, R2, and R3. Operation is performed while switching the switching pattern within each range. Furthermore, the same operation is performed within the same designated range during voltage reduction and voltage increase.

[0071] Figure 4 This shows the main operating waveform during continuous switching operation and the switching pattern in each operating range.

[0072] In the continuous SW control, gate driving is performed at a switching frequency of several kHz or more, and the AC current iac has a sinusoidal waveform close to a power factor of "1". Figure 4 The gate signal waveforms for one switching cycle in the operating ranges R1 and R2 are shown. In the single-phase inverter 3, the first and second legs switch with an offset of half a drive cycle. In the single-phase converter 5, the fourth leg does not switch, while the third leg switches. The gate signals of the upper and lower switching elements in each leg are inversely spaced.

[0073] By switching in this manner, reactor 2 is magnetized and demagnetized twice during each switching cycle, and second capacitor 4 is charged and discharged once. This allows second capacitor 4 to be charged and discharged with the time constant of the switching cycle, allowing it to be constructed with a small capacitor. Furthermore, synchronous rectification can be performed by turning on the gate signal in the fourth leg of single-phase converter 5 at the timing that current flows through the parallel diode.

[0074] In the figure, VG3a to VG3d represent the voltages of gate signals G3a to G3d applied to the gates of the switching elements 3a to 3d of the single-phase inverter 3, and VG5a to VG5d represent the voltages of gate signals G5a to G5d applied to the gates of the switching elements 5a to 5d of the single-phase converter 5. Furthermore, Csub represents the capacitance of the second capacitor 4, Chg represents the charge state, and DisChg represents the discharge state (the same applies to the following figures).

[0075] Continuous switching control differs from the other two control modes, simple switching step-up control and simple switching step-down control, by employing PWM control at frequencies exceeding several kHz, enabling high-voltage operation. Therefore, continuous switching control is effective under conditions of heavy loads with high currents or when a high voltage step-up ratio is required.

[0076] also, Figure 4The gate signal pattern shown is an example. It is a switching pattern in which the reactor 2 is excited and demagnetized twice and the second capacitor 4 is charged and discharged once during each switching element is switched once. Figure 4 Switching mode shown.

[0077] Figure 5 as well as Figure 6 This shows the main operating waveform during simple SW step-up operation and the switching pattern in each operating range.

[0078] Simple SW boost control switches on and off only a few times during the AC half-cycle. This number of switching cycles is less than with continuous SW control. While this reduces the power factor, it significantly reduces the losses associated with high-frequency drive. Simple SW boost control switches between switching and non-switching within operating range R1, depending on input power conditions. Figure 5 It is the action of the switch in the action range R1. Figure 6 This is an operation that does not switch in operating range R1. When the power is low, switching in operating range R2 alone can fully ensure the power factor. However, when the power increases, switching in operating range R2 alone increases the current peak, causing a deterioration in the power factor and an increase in losses.

[0079] In the case of simplified SW boost control, the first and second legs of the single-phase inverter 3 and the third leg of the single-phase converter 5 are switched, while the fourth leg is not switched. The gate signals of the upper and lower switching elements of each leg are inverted. Furthermore, similar to the case of continuous SW control, the gate signal of the fourth leg of the single-phase converter 5 can be turned on at the timing that current flows through the parallel diode, thereby performing synchronous rectification.

[0080] In the simplified SW boost control, within the operating ranges R1 and R2, the voltage Vdc of the first capacitor 6 tracks the on-time of the command value at a user-specified switching frequency, causing the gate signal to turn on. After the specified switching frequency, a path connecting the AC power supply 1 and the load 7 is formed so that no charge or discharge occurs in the second capacitor 4, resulting in free resonance operation between the reactor 2 and the first capacitor 6.

[0081] also, Figure 5 、 Figure 6 The gate signal pattern shown is an example, and is a pattern in which the reactor 2 is excited and demagnetized twice and the second capacitor 4 is charged and discharged once during each switching element is switched once, and is not limited to the above. Figure 5 、 Figure 6The switching frequency set by the user can be any number, but a trade-off exists where a higher switching frequency results in a power factor closer to 1 but increases circuit loss.

[0082] Figure 7 as well as Figure 8 This shows the main operating waveform during simple SW step-down operation and the switching pattern in each operating range.

[0083] In the simplified SW step-down control, similar to the simplified SW step-up control, switching is performed only a few times within an AC half-cycle. This low switching frequency significantly reduces losses associated with high-frequency driving. Furthermore, in the simplified SW step-down control, switching or non-switching can be selected within the operating range R1 based on the step-down ratio of the voltage Vdc of the first capacitor 6. Figure 7 It is the action of the switch in the action range R1. Figure 8 When the step-down ratio is low, switching in the operating range R2 alone can adequately control the power factor and voltage. However, when the step-down ratio increases, voltage control does not operate normally in the operating range R2 alone.

[0084] In the case of simplified SW step-down control, the first and second legs of the single-phase inverter 3 and the third leg of the single-phase converter 5 are switched, while the fourth leg is not switched. The gate signals of the upper and lower switching elements of each leg are inverted. Furthermore, similar to continuous SW control, the gate signal of the fourth leg of the single-phase converter 5 can be turned on at the timing that current flows through the parallel diode, thereby performing synchronous rectification.

[0085] Unlike continuous switching control and simple switching boost control, simplified step-down control includes an operating range R3. In operating range R3, under the conditions of Vac > Vdc > Vsub, it is necessary to drive the system in a switching pattern where reactor 2 is excited and demagnetized once, while second capacitor 4 is charged and discharged once, during each switching operation of each switching element. Therefore, the number of switching cycles required is doubled compared to operating ranges R1 and R2. Therefore, in simplified step-down control, the total number of charge and discharge cycles in operating ranges R1 and R2 must equal the number of charge and discharge cycles in operating ranges R3.

[0086] exist Figure 7 In the case of switching in the operating range R1 shown, if switching is performed once in the operating range R1 and once in the operating range R2, the minimum number of switching times in the AC half cycle becomes 8 times, and the number of times that the user can set becomes a multiple of 8. Figure 8In the illustrated operating range R1, when operating without switching, the minimum number of switching cycles is 4, and the number of switching cycles that can be set by the user is a multiple of 4. When the number of switching cycles is increased to 8 as described above, the switching frequency becomes 400 Hz, which reduces high-frequency loss compared to the case of continuous SW control operation.

[0087] also, Figure 7 as well as Figure 8 The gate signal pattern shown is an example. In operating ranges R1 and R2, the reactor 2 is excited and reset twice, and the second capacitor 4 is charged and discharged once, each time each switching element switches once. The user-set switching frequency can be any multiple of 8 for switching in operating range R1, or any multiple of 4 for non-switching in operating range R2. A higher switching frequency results in a power factor closer to unity, but circuit loss increases, resulting in a trade-off.

[0088] Next, the configuration and operation of the controller 8 according to the first embodiment will be described.

[0089] Figure 9 It is a block diagram of the control operation determiner 9.

[0090] The control action determiner 9 is input with the AC voltage Vac detected by the sensor not shown, the voltage (output voltage) Vdc of the first capacitor 6, and the AC current iac. The area determiner 17 of the control action determiner 9 determines whether the output voltage Vdc is stepped up or stepped down relative to the AC voltage Vac. In addition, the SW action determiner 18 calculates the action power converted by the power conversion device 100, compares it with the power value determined in advance by the user, and determines whether to perform continuous SW action control or simple SW action control. Moreover, the control selection signal SelectSignal obtained by summing up the various signals is output to the main controller 12. In addition, Figure 9 In the table, RMS represents the effective value, and SQRT2 represents the square root of 2.

[0091] Figure 10 2 is a block diagram of the second capacitor voltage command value calculator 10 .

[0092] The second capacitor voltage command value calculator 10 calculates the voltage command value Vsub* of the second capacitor 4 based on the operating state of the AC voltage Vac and the output voltage command value Vdc*. The relationship between the AC voltage Vac and the output voltage command value Vdc* is as follows: Figure 3 As shown, it is roughly divided into Figure 3 A or Figure 3 B these two modes.

[0093] That is, in Figure 3 In A, it is set so that Vsub*=Vdc* / 2 is always achieved. Figure 3 In B, the voltage command value Vsub* of the second capacitor 4 is set based on the magnitude relationship between the AC voltage Vac and the output voltage command value Vdc* so that the period of the operating range R3 and the period of the operating range R2 are equal.

[0094] In the power converter of the first embodiment, the condition that the output voltage command value Vdc* becomes the same as the effective value of the AC voltage Vac becomes the voltage drop limit. Therefore, the lower limit value of the voltage command value Vsub* of the second capacitor 4 is half of the effective value of the AC voltage Vac.

[0095] By making the voltage command value Vsub* of the second capacitor 4 variable during the step-down operation, the power factor improvement effect can be enhanced compared to the conditional expression of Vsub*=Vdc* / 2 when switching is performed only in the operation range R2 and the operation range R3.

[0096] exist Figure 10 The operation range R3 period calculator 19 performs the following equation (1). In addition, the Vsubref calculator 20 performs the following equation (2). Here, ω = 2π·fac, fac is the frequency of the AC power supply. In addition, Figure 10 In the figure, ÷ represents a divider and MUX represents a multiplexer.

[0097] T3=1 / ω·arcsin(Vdc* / √2Vac) (1)

[0098] Vsubref=Vdc*-√2Vac·sin(ω·T3) (2)

[0099] Figure 11 2 is a block diagram of the second capacitor voltage controller 11 .

[0100] The second capacitor voltage controller 11 calculates the deviation between the voltage Vsub of the second capacitor 4 detected by the sensor outside the figure and the voltage command value Vsub* of the second capacitor 4 calculated by the second capacitor voltage command value calculator 10, and controls the second capacitor 4 by tracking the voltage command value Vsub* of the second capacitor 4 through proportional integral control (PI control). Figure 11 In the equation, PI stands for proportional-integral controller and ÷ stands for divider.

[0101] The main controller 12 is composed of three components: a continuous switching controller 13, a simple switching step-up controller 14, and a simple switching step-down controller 15. Based on the control selection signal SelectSignal output from the control action determiner 9, the main controller 12 switches between the three control actions. It also generates a duty cycle signal Dutytotal corresponding to each control action and a carrier signal Carrier to be compared with the duty cycle signal Dutytotal, and outputs the generated signal to the gate signal generator 16.

[0102] Figure 12 is a block diagram of the continuous SW controller 13 .

[0103] The continuous switching controller 13 operates when a signal corresponding to the step-up continuous switching operation is input as the control selection signal SelectSignal.

[0104] The continuous switching controller 13 calculates the deviation between the detected voltage Vdc of the first capacitor 6 and the voltage command value Vdc* for the first capacitor 6, performs proportional-integral control (PI control) using a proportional-integral controller (PI controller), and outputs an AC current command value Iac*. The AC current command value Iac* is multiplied by a phase-locked loop (PLL) waveform obtained from the AC voltage Vac via a PLL (Phased Locked Loop) circuit to determine the instantaneous value iac* of the AC current command value. The deviation from the detected AC current iac is proportionally controlled (P control) using a proportional controller (P controller). The power factor control duty ratio DutyPFC is calculated by adding the theoretical duty ratio DutyPFCFF calculated by the FFDuty calculator 21. The duty totalizer 22 then adds or subtracts the control signal DutyVsub calculated and output by the second capacitor voltage controller 11, and outputs the duty ratio signal Dutytotal to the gate signal generator 16.

[0105] The FFDuty calculator 21 selects a calculation equation based on the operating range conditions determined by the operating range determiner 23. Furthermore, in the continuous switching controller 13, the carrier signal Carrier is a fixed value, thus forming a sawtooth or triangular wave with a user-set frequency. Furthermore, the FFDuty calculator 21 calculates the theoretical duty cycle DutyPFCFF for a switching mode in which the AC current iac increases and decreases twice within one switching cycle, and the second capacitor 4 charges and discharges once each.

[0106] By applying feedforward (FF) control to the control duty cycle, the responsiveness of high power factor control can be improved. In the power conversion device 100 of this embodiment 1, the control duty cycle is switched according to the operating range, so the introduction of feedforward control can suppress abrupt changes during switching.

[0107] The following equations (3) to (6) show an example of the theoretical duty ratio DutyPFCFF in the power conversion device 100 of the first embodiment. Figure 13 A, B, and C show switching patterns in the respective operating ranges R1 to R3 of the first embodiment.

[0108] exist Figure 13 In the action range R1 of A, there are five modes: A1, B1, D1, E1, and F1. Figure 13 In the action range R2 of B, there are five modes: A2, B2, C2, D2, and F2. Figure 13 The operating range R3 of C includes four modes: A3, B3, C3, and F3.

[0109] Formula (3) becomes the reference Figure 4 The action consisting of the modes A1 and E1 in the action range R1, equation (4) becomes the reference Figure 4 The action consisting of the modes A1 and F1 in the action range R1, equation (5) becomes the reference Figure 4 The action composed of the modes C2 and D2 in the action range R2, equation (6) becomes the reference Figure 4 The operation consists of the modes B2 and D2 in the operation range R2.

[0110] Furthermore, any switching pattern may be used as long as the AC current iac increases and decreases twice and the second capacitor 4 is charged and discharged once each within one switching cycle.

[0111] DutyPFCFF1=(Vsub-Vac) / Vsub (3)

[0112] DutyPFCFF2=(Vdc-Vac-Vsub) / (Vdc-Vsub) (4)

[0113] DutyPFCFF3=(Vdc-Vac) / Vsub (5)

[0114] DutyPFCFF4=(Vdc-Vac) / (Vdc-Vsub) (6)

[0115] Figure 14 A and Figure 14 B is used to illustrate the composition Figure 12FIG. 1 is a diagram showing the operation of the duty totalizer 22 of the continuous switching controller 13 .

[0116] The duty totalizer 22 adds and subtracts the power factor control duty ratio DutyPFC obtained by the above calculation and the control signal DutyVsub calculated by the second capacitor voltage controller 11, and outputs the duty ratio signal Dutytotal to the next-stage gate signal generator 16.

[0117] The operating range determiner 23 determines the operation of R1 to R3 based on the magnitude relationship among the detected AC voltage Vac, the voltage Vdc of the first capacitor 6 , and the voltage Vsub of the second capacitor 4 , and outputs a signal to the FFDuty calculator 21 .

[0118] exist Figure 14 A shows an example of the operation within the operation range R1. Figure 14 B shows an example of operation within operating range R2. By adding and subtracting the power factor control duty ratio DutyPFC and the control signal DutyVsub at the same timing as shown in the figure, the charge and discharge amount of the second capacitor 4 can be varied without changing the amount of change in DutyPFC within one switching cycle, thereby controlling the voltage of the second capacitor 4 to be constant.

[0119] In continuous switching control, control is performed using switching operations at frequencies of several kHz or higher, thereby increasing the voltage step-up ratio and boosting the voltage Vdc of the first capacitor 6. When the load 7 comprises an inverter, a motor, and a compressor, a higher voltage Vdc of the first capacitor 6 can increase the inverter modulation rate, reducing losses in the inverter or motor / compressor, and achieving higher system efficiency.

[0120] Figure 15 1 is a block diagram of the simplified SW boost controller 14 .

[0121] The simple SW boost controller 14 operates when a signal corresponding to the boost simple switching operation is input as the control selection signal SelectSignal.

[0122] The simplified SW boost controller 14 calculates the deviation between the voltage Vdc of the first capacitor 6 detected by the sensor and the voltage command value Vdc* of the first capacitor 6, performs proportional integral control (PI control) using a proportional integrator (PI device), and then normalizes the result, thereby calculating the duty ratio required to control the output voltage Vdc.

[0123] Specifically, on-time calculator 24 first calculates the on-time Ton within the AC half-cycle based on the duty cycle. By multiplying gain K by the number of SW cycles within operating range R1, it derives the proportional amount TSW1 within operating range R1 within on-time Ton. Gain K ranges from 0 to 1 and is arbitrarily set by the user in consideration of the power factor. Alternatively, a pre-set table can be used to adjust the gain according to load conditions.

[0124] The SW carrier calculator A25 generates a carrier signal Carrier1 in the operating range R1 and a carrier signal Carrier2 in the operating range R2, and outputs them to the gate signal generator 16. The carrier signals Carrier1 and Carrier2 may be sawtooth waves or triangular waves.

[0125] Here, the switching number N1 of the operating range R1 input to the SW carrier calculator A25 is determined to be "0" or "1" by the multiplexer (MUX) according to the input power. When the input power is above a predetermined value, the power factor deteriorates and the circuit loss decreases when no switching is performed in the operating range R1. When the switching number N1 is "0", Figure 6 The action shown, in the case of "1", becomes Figure 5 The number of switching times N2 and the delay time Tdl2 within the operating range R2 can be set arbitrarily by the user in consideration of the power factor. Alternatively, a table can be set in advance so that the values ​​can be changed according to the load conditions.

[0126] In the simple SW boost control operation, the duty cycle of the operating range R1 can be calculated based on the ratio TSW1 of the operating range and its carrier signal Carrier1. The duty totalizer 22 adds or subtracts the power factor control duty cycle DutyPFC, which is a fixed value of 0.5, and its control signal DutyVsub to the duty cycle of the operating range R2, and outputs the result as the duty cycle signal Dutytotal to the gate signal generator 16. The operation of the duty totalizer 22 is the same as that of the continuous SW control. In addition, Figure 15 In the figure, MUX stands for multiplexer.

[0127] Figure 16 This is a block diagram of the SW carrier calculator A25 constituting the simplified SW boost controller 14 .

[0128] The SW carrier calculator A25 includes an operating range R1 / SW carrier calculator A26 and an operating range R2 / SW carrier calculator A27. The SW carrier calculator A25 generates carrier signals Carrier1 and Carrier2 for operating ranges R1 and R2, respectively, based on the switching counts N1 and N2 for each operating range R1 and R2, and the switching delay time Tdl2 for operating range R2.

[0129] Specifically, the operating range R1 / SW carrier calculator A26 calculates the switching period T1 for operating range R1, then converts the period to a sawtooth or triangular wave, outputting it as carrier signal Carrier 1. Switching period T1 is calculated by performing the following equation (7). Here, N1 is the number of switching operations within operating range R1 and is either "0" or "1."

[0130] T1=N1 / ω·arcsin(Vsub* / √2Vac) (7)

[0131] Furthermore, the operating range R2 / SW carrier calculator A27 calculates the switching period TSW2 within the operating range R2 using the following equation (8) and converts it into the SW carrier. The SW carrier, after incorporating the user-set delay time Tdl2 into the operating range R2, is then output as carrier signal Carrier2. Here, N2 represents the number of switching operations within the operating range R2 and is an integer greater than or equal to 1.

[0132] TSW2=2·(Ton-TSW1) / N2(8)

[0133] Compared to continuous switching operation, simple SW boost control uses far fewer switching cycles, resulting in a lower step-up ratio. Therefore, it is the control of choice when low boost conditions are set as the load conditions for the power converter. Furthermore, simple SW boost control only requires a few switching cycles per AC half-cycle, significantly reducing losses in semiconductor devices and reactors.

[0134] Figure 17 This is a block diagram of the simple SW step-down controller 15.

[0135] The simple SW step-down controller 15 operates when a signal corresponding to the step-down simple switching operation is input as the control selection signal SelectSignal.

[0136] The simplified switching step-down controller 15 calculates the deviation between the detected voltage Vdc of the first capacitor 6 and the voltage command value Vdc* for the first capacitor 6. This deviation is then normalized using a proportional-integrator (PI controller) through proportional-integral control (PI control). This calculation then calculates the duty cycle required to control the output voltage Vdc. Similar to the continuous switching controller 13, the power factor control duty cycle DutyPFCFF is then added to the theoretical duty cycle DutyPFCFF obtained from the FFDuty calculator 21 to determine the power factor control duty cycle DutyPFC. Furthermore, the duty totalizer 22 adds and subtracts the power factor control duty cycle DutyPFC from the control signal DutyVsub, outputting the resulting duty cycle signal Dutytotal to the gate signal generator 16.

[0137] The operation of the duty totalizer 22 is the same as that of the continuous switching controller 13, so its description is omitted.

[0138] The SW carrier operator B28 uses the voltage Vdc of the first capacitor 6, the AC voltage Vac, the switch judgment signal Sj1 of the action range R1 described later, and the switching number N3 of the action range R3 to calculate the carrier signal in each action range, and uses the output signal from the action range determiner 23 to output the carrier signal Carrier in each action range of the multiplexer (MUX) to the gate signal generator 16.

[0139] Here, the switch determination signal Sj1 of the operating range R1 is a signal determined by a multiplexer (MUX) to be "0" or "1" based on the voltage command value Vdc* of the first capacitor 6, and switches between a switching mode and a non-switching mode in the operating range R1. When the voltage command value Vdc* is below a predetermined value, if switching is not performed in the operating range R1, deterioration of the voltage constant control occurs. When the switch determination signal Sj1 is "0", it becomes Figure 8 The action shown, in the case of "1", becomes Figure 7 The user can arbitrarily set the switching frequency N3 within the operating range R3 in consideration of the power factor. Alternatively, a table can be set in advance to adjust the switching frequency N3 according to load conditions.

[0140] Figure 18 This is a block diagram of the SW carrier calculator B28 constituting the simplified SW step-down controller 15 .

[0141] SW carrier calculator B28 is comprised of an operating range R1 / SW carrier calculator B29, an operating range R2 / SW carrier calculator B30, and an operating range R3 / SW carrier calculator B31. It generates carrier signals Carrier 1, Carrier 2, and Carrier 3, respectively, based on the switching determination signal Sj1 for operating range R1 and the number of switching times N3 for operating range R3.

[0142] That is, the operating range R1 / SW carrier calculator B29 calculates the period T1 and switching period TSW1 of the operating range R1, then changes the period to a sawtooth wave or a triangular wave and outputs it as the carrier signal Carrier1. The period T1 is calculated using the following equation (9), and the switching period TSW1 is calculated using the following equation (10). Here, Sj1 is the switching determination signal for the operating range R1, which is either "0" or "1." N3 is the number of switching times in the operating range R3. When switching is performed in the operating range R1, it is an integer that is a multiple of "4", and when switching is not performed in the operating range R1, it is an integer that is a multiple of "2".

[0143] T1=1 / ω·arcsin(Vsub* / √2Vac) (9)

[0144] TSW1=Sj1·N3 / 4·T1 (10)

[0145] Furthermore, the operating range R2 / SW carrier calculator B30 calculates the period T2 and switching period TSW2 of the operating range R2, then converts the period into a sawtooth wave or a triangular wave, outputting it as carrier signal Carrier2. The period T2 is calculated using the following equation (11), and the switching period TSW2 is calculated using equations (12) or (13). Equation (12) is used when the switch determination signal for operating range R1 is "1," and equation (13) is used when the switch determination signal for operating range R1 is "0."

[0146] T2=1 / ω·arcsin(Vdc / √2Vac)-T1 (11)

[0147] TSW2=N3 / 4·T2 (12)

[0148] TSW2=N3 / 2·T2 (13)

[0149] The operating range R3 / SW carrier calculator B31 calculates the switching period TSW3 of the operating range R3, then changes the period to a sawtooth wave or a triangle wave and outputs it as the carrier signal Carrier 3. The switching period TSW3 is calculated by performing the following equation (14).

[0150] TSW3={1 / (2·fac)-2·(T1+T2)} / N3 (14)

[0151] In the simple SW step-down control, similarly to the simple SW step-up control, the number of switching operations in an AC half cycle is reduced to only a few times, so the losses in semiconductor devices and reactors can be significantly reduced.

[0152] The gate signal generator 16 calculates gate pulses based on the duty cycle signal Dutytotal and the carrier signal Carrier through PWM (Pulse Width Modulation) control, and outputs gate signals corresponding to each operating range to the semiconductor switching elements 3a~3d of the single-phase inverter 3 and the semiconductor switching elements 5a~5d of the single-phase converter 5.

[0153] As described above, in this embodiment 1, by selecting, according to the load state, continuous SW control for controlling the power factor and voltage at a constant switching frequency in the AC half cycle, simple SW step-up control for controlling the power factor and voltage by switching only a few times in the AC half cycle, and simple SW step-down control, the capacitance of the second capacitor 4 is a small capacitor as in the conventional technology, and at the same time, the circuit loss can be reduced over a wider range than in the conventional technology.

[0154] Implementation method 2.

[0155] Figure 19 1 is a circuit diagram showing a power conversion device 200 according to Embodiment 2. Figure 20 This is a block diagram of the controller 32 in the power conversion device 200 , and components corresponding to or equivalent to those in the first embodiment are denoted by the same reference numerals.

[0156] The main circuit configuration of the power conversion device 200 of this second embodiment is basically the same as that of the power conversion device 100 of the first embodiment. Furthermore, the components of the controller 32 are the same as those of the controller 8 of the first embodiment, but the control operation determination unit 33 constituting the controller 32 is different from that of the first embodiment. Therefore, detailed description of the components other than the control operation determination unit 33 will be omitted here.

[0157] Figure 21 It is a block diagram of the control operation determiner 33.

[0158] The control operation determiner 33 outputs a control selection signal SelectSignal (in the output state) for selecting the control to be performed in the main controller 12 based on the magnitude relationship between the AC voltage Vac and the output voltage Vdc and the magnitude of the output voltage Vdc. Figure 21 SS in Chinese).

[0159] Specifically, the control operation determiner 33 receives inputs of the AC voltage Vac and the output voltage Vdc. The region determiner 17 determines whether the output voltage Vdc is stepped up or down relative to the AC voltage Vac. Furthermore, the switching operation determiner 34 compares the output voltage Vdc with a value predetermined by the user to determine whether continuous switching operation, simple switching step-down operation, or simple switching step-up operation is in effect. The control selection signal SelectSignal, obtained by summing these signals, is output to the main controller 12.

[0160] The operation of the components other than the control operation determiner 33 is the same as that of the first embodiment, and thus description thereof is omitted. The main operation of the power conversion device 300 is the same as that of the power conversion device 100 according to the first embodiment, and thus description thereof is omitted.

[0161] As described above, the power conversion device 200 according to the second embodiment determines the control action by the control action determiner 33 based on the output voltage Vdc, and thus can exhibit an effective effect for power conversion devices that perform an operation to make the output voltage Vdc variable under the same power conditions.

[0162] Implementation method 3.

[0163] Figure 22 1 is a circuit diagram showing a power conversion device 300 according to the third embodiment, and components corresponding to or equivalent to those in the first embodiment are denoted by the same reference numerals.

[0164] The third embodiment is characterized by a different structure of the single-phase converter 5 from that of the first embodiment. Specifically, the single-phase converter 5 of the third embodiment includes a third branch formed by connecting a diode 5aa and a semiconductor switching element 5b in series, and a fourth branch formed by connecting a diode 5cc and a semiconductor switching element 5d in series. These third and fourth branches are connected in parallel with each other.

[0165] Furthermore, the midpoint of the third leg of the single-phase converter 5 is connected to the midpoint of the second leg of the single-phase inverter 3, and the midpoint of the fourth leg of the single-phase converter 5 is connected to the N busbar of the AC power supply 1. Furthermore, with respect to the first capacitor 6, the upper ends of the third and fourth legs of the single-phase converter 5 are connected to the P-side terminal of the load 7, while the lower ends of the third and fourth legs of the single-phase converter 5 are connected to the N-side terminal of the load 7.

[0166] The configuration of the controller 35 may adopt the configuration of the controller 8 of the power conversion device 100 according to the first embodiment, or the configuration of the controller 32 of the power conversion device 200 according to the second embodiment.

[0167] Since the remaining components are the same as those in Embodiment 1, detailed descriptions thereof will be omitted. Furthermore, the main operations of the power conversion device 300 are the same as those of the power conversion device 100 according to Embodiment 1. Furthermore, the operations of the controller 35 are the same as those of the controller 8 in Embodiment 1 or the controller 32 in Embodiment 2, and thus descriptions thereof will be omitted.

[0168] As described above, the power conversion device 300 involved in embodiment 3 cannot perform synchronous rectification operation because diodes 5aa and 5cc are used in the upper branch of the single-phase converter 5, but the recovery characteristics during switching are improved, so it is possible to achieve reduced switching losses and stable circuit driving.

[0169] Implementation method 4.

[0170] Figure 23 This is a circuit diagram of a power conversion device 400 according to the fourth embodiment, and components corresponding to or equivalent to those in the first embodiment are denoted by the same reference numerals.

[0171] The power conversion device 400 of this fourth embodiment includes a main circuit and a controller 37 for converting the AC voltage and AC power of a single-phase AC power source 1 into a DC voltage and DC power and outputting them to a load 7. The main circuit is composed of a single-phase inverter 3, a single-phase converter 5, a diode bridge 36, and a reactor 2, which are connected in series between the AC power source 1 and a first capacitor 6.

[0172] The single-phase inverter 3 includes a first branch having a diode 3aa and a semiconductor switching element 3b connected in series, a second branch having a semiconductor switching element 3c and a diode 3dd connected in series, and a second capacitor 4 located between the first and second branches and connected in parallel with the first and second branches.

[0173] The single-phase converter 5 includes a third leg having a diode 5aa and a semiconductor switching element 5b connected in series, and a fourth leg having a diode 5cc and a diode 5dd connected in series. The third leg and the fourth leg are connected in parallel with each other.

[0174] The diode bridge 36 is connected between the AC power supply 1 and the reactor 2 . The reactor 2 is connected between the upper output terminal of the diode bridge 36 and the midpoint of the first leg of the single-phase inverter 3 .

[0175] Furthermore, the midpoint of the third leg of the single-phase converter 5 is connected to the midpoint of the second leg of the single-phase inverter 3, and the midpoint of the fourth leg of the single-phase converter 5 is connected to the lower output terminal of the diode bridge 36. Furthermore, with respect to the first capacitor 6, the upper ends of the third and fourth legs of the single-phase converter 5 are connected to the load 7 at its P-side terminal, and the lower ends of the third and fourth legs of the single-phase converter 5 are connected to the load 7 at its N-side terminal.

[0176] Furthermore, the semiconductor switching elements used in the single-phase inverter 3 and the single-phase converter 5 are preferably IGBTs (Insulated Gate Bipolar Transistors) with diodes connected in antiparallel, MOSFETs (Metal Oxide Semiconductor Field Effect Transistors) with diodes connected between the source and drain, or cascaded GaN-HEMTs (Gallium Nitride-High Mobility Transistors). Furthermore, the feedback diode may be a diode built into the IGBT, MOSFET, or GaN-HEMT, or a separate external diode may be provided.

[0177] The configuration of the controller 37 may adopt the configuration of the controller 8 of the power conversion device 100 according to the first embodiment, or the configuration of the controller 32 of the power conversion device 200 according to the second embodiment.

[0178] Since the other components are the same as those in Embodiment 1, detailed descriptions thereof will be omitted. Furthermore, the main operations of power conversion device 400 are the same as those of power conversion device 100 according to Embodiment 1. Furthermore, the operations of controller 37 are the same as those of controller 8 in Embodiment 1 or controller 32 in Embodiment 2, and thus descriptions thereof will be omitted.

[0179] As described above, the power conversion device 400 according to the fourth embodiment can reduce the number of semiconductor switching elements compared to the previous embodiments 1 to 3 by adding the diode bridge 36. Therefore, the circuit can be configured at a relatively low cost.

[0180] Implementation method 5.

[0181] Figure 24 This is a circuit diagram of a power conversion device 500 according to the fifth embodiment, and components corresponding to or equivalent to those in the first embodiment are denoted by the same reference numerals.

[0182] The power conversion device 500 of the fifth embodiment includes a main circuit and a controller 40 for converting AC voltage and AC power of a single-phase AC power supply 1 into DC voltage and DC power and outputting the converted power to a load 7 .

[0183] The main circuit includes a reactor 2 , a diode rectifier branch 38 , and a multi-stage drive branch 39 connected in this order between an AC power source 1 and a first capacitor 6 , and further includes a second capacitor 4 .

[0184] The diode rectifier branch 38 is composed of a pair of diodes 38a and 38b connected in series. Furthermore, the multi-stage drive branch 39 is composed of four semiconductor switching elements 39a to 39d connected in series. Furthermore, a second capacitor 4 is connected between the connection point between the pair of semiconductor switching elements 39a and 39b and the connection point between the pair of semiconductor switching elements 39c and 39d.

[0185] Reactor 2 is connected between the P busbar of AC power supply 1 and the midpoint of diode rectifier branch line 38. Furthermore, the upper end of diode rectifier branch line 38 and the upper end of multi-stage drive branch line 39 are connected to the P-side terminal of first capacitor 6, while the lower end of diode rectifier branch line 38 and the lower end of multi-stage drive branch line 39 are connected to the N-side terminal of first capacitor 6.

[0186] The N bus bar of the AC power supply 1 is connected to the connection point between the pair of semiconductor switching elements 39 a and 39 b in the upper arm and the pair of semiconductor switching elements 39 c and 39 d in the lower arm of the multi-stage drive branch line 39 .

[0187] The semiconductor switching elements 39a to 39d used in the multi-stage drive branch 39 are preferably IGBTs (Insulated Gate Bipolar Transistors) with diodes connected in antiparallel, MOSFETs (Metal Oxide Semiconductor Field Effect Transistors) with diodes connected between the source and drain, or cascaded GaN-HEMTs. The feedback diodes may be built into the IGBTs, MOSFETs, or GaN-HEMTs, or they may be externally provided.

[0188] The AC voltage Vac of the AC power source 1, the voltage Vdc of the first capacitor 6, and the voltage Vsub of the second capacitor 4 are detected by voltage sensors (not shown) and input to the controller 40. Furthermore, the AC current iac of the AC power source 1 is detected by a current sensor (not shown) and input to the controller 40.

[0189] The controller 40 generates gate signals G39 a , G39 b , G39 c , and G39 d for the multi-stage drive branch line 39 based on load information (voltage information and current information) detected and input by the sensors, and controls the power conversion device 500 .

[0190] Figure 25 This is a block diagram of a controller according to the fifth embodiment, and components corresponding to or equivalent to those in the first embodiment are denoted by the same reference numerals.

[0191] The controller 40 is composed of a control operation determiner 9 , a second capacitor voltage controller 11 , a main controller 41 , and a gate signal generator 16 .

[0192] Here, the main controller 41 is composed of two controllers: the continuous switching controller 13 and the simple switching boost controller 14. The main controller 41 determines which of the two controllers 13 and 14 to use for control based on the SelectSignal (abbreviated in the figure as SS) input from the control action determiner 9. Furthermore, based on the voltage and current information Vac, Vdc, and Vsub detected by the sensors and the control signal DutyVsub calculated by the second capacitor voltage controller 11, the main controller 41 generates a duty cycle signal Dutytotal for power factor control and voltage control, as well as a carrier signal Carrier for comparison with the duty cycle signal Dutytotal.

[0193] The gate signal generator 42 calculates the gate signal through PWM (Pulse Width Modulation) control based on the duty cycle signal Dutytotal and the carrier signal Carrier, and outputs the gate signal for turning on / off each gate of the semiconductor switching elements 39a to 39d of the multi-stage drive branch 39.

[0194] Next, the operation of the power conversion device 500 according to the fifth embodiment will be described.

[0195] The power conversion device 500 switches the AC current iac input from the AC power supply 1 using a multi-stage drive branch line 39 to control the AC current iac flowing through the inductor 2 to a high power factor while boosting the voltage, and uses the first capacitor 6 to smooth the power and supply DC power to the load 7.

[0196] The operation of the fifth embodiment is different from that of the first embodiment. In the main controller 41, two control operations are performed: continuous SW control and simple SW boost control. Therefore, the relationship diagram of each voltage is simply Figure 3 Case A. In the fifth embodiment, there is no voltage step-down operation, so the voltage command value Vsub* of the second capacitor 4 always satisfies the relationship of Vsub*=Vdc* / 2.

[0197] Figure 26 This shows the main operating waveform during continuous switching operation and the switching pattern in each operating range.

[0198] In continuous SW control, the gate drive is performed at a switching frequency of several kHz or more, and the AC current iac becomes a sinusoidal waveform close to the power factor "1". Figure 26 , gate signal waveforms in one switching cycle in the operating range R1 and the operating range R2 are shown respectively.

[0199] exist Figure 26 In the positive half-cycle, the semiconductor switching elements 39a and 39b of the upper arm of the multi-stage drive branch 39 switch, while the semiconductor switching elements 39c and 39d of the lower arm operate to conduct current through the parallel diode. Furthermore, in the negative half-cycle, the operations of the semiconductor switching elements 39a and 39b of the upper arm and the semiconductor switching elements 39c and 39d of the lower arm are reversed from those in the positive half-cycle. Furthermore, by turning on the semiconductor switching elements during the conduction of current through the parallel diode, synchronous rectification can be achieved.

[0200] By switching the semiconductor switching elements 39a-39d of the multi-stage drive branch 39 in this manner, reactor 2 is magnetized and demagnetized twice during each switching cycle, and second capacitor 4 is charged and discharged once. Thus, second capacitor 4 is charged and discharged with the time constant of the switching cycle, allowing it to be constructed with a small capacitor. The effects of continuous switching control have been previously described in the first embodiment and are therefore omitted.

[0201] also, Figure 26 The gate signal pattern shown is an example, and is a pattern in which the reactor 2 is excited and demagnetized twice and the second capacitor 4 is charged and discharged once during each semiconductor switching element 39a to 39d is switched once. Figure 26 pattern.

[0202] Figure 27 as well as Figure 28 This shows the main operating waveform during simple SW step-up operation and the switching pattern in each operating range.

[0203] In simple SW boost control, switching is performed only a few times within an AC half cycle. This is less frequent than in continuous SW control, so while the power factor decreases, the losses associated with high-frequency drive can be significantly reduced.

[0204] In the simple SW step-up control, switching operation and non-switching operation can be selected in the operation range R1 according to the conditions of input power and power factor. Figure 27 is a diagram showing the operation when the switch is in the operation range R1. Figure 28 This diagram shows the operation when no switching is performed in operating range R1. When power is low, switching in operating range R2 alone can adequately maintain the power factor. However, when power increases, operating range R2 alone increases the current peak, deteriorating the power factor and increasing losses.

[0205] Even in the case of simple SW boost control, during the positive half-cycle, the semiconductor switching elements 39a and 39b of the upper arm of the multi-stage drive branch 39 switch, while the semiconductor switching elements 39c and 39d of the lower arm operate to conduct current through the parallel diode. During the negative half-cycle, the operations of the semiconductor switching elements 39a and 39b of the upper arm and the semiconductor switching elements 39c and 39d of the lower arm are reversed from those in the positive half-cycle. Alternatively, by turning on the semiconductor switching elements during the conduction of current through the parallel diode, synchronous rectification can be achieved.

[0206] In the simplified SW boost control, in operating ranges R1 and R2, the gate signal is turned on for an on-time that allows the output voltage Vdc to track the command value at a user-specified number of switching cycles. After the specified number of switching cycles, a path connecting the AC power supply 1 and the load 7 is formed so that no charge or discharge occurs in the second capacitor 4, resulting in free resonance operation between the reactor 2 and the first capacitor 6.

[0207] also, Figure 27 、 Figure 28 The gate signal pattern shown is an example, and is a pattern in which the reactor 2 is excited and demagnetized twice and the second capacitor 4 is charged and discharged once during each element is switched once. Figure 27 、 Figure 28 The switching times set by the user can be any number, but a trade-off exists where a higher switching frequency results in a power factor closer to 1 but increases circuit loss.

[0208] Next, the operation of the controller 40 according to the fifth embodiment will be described. In the fifth embodiment, components denoted by corresponding reference numerals as those in the first embodiment operate in the same manner, and therefore their description will be omitted.

[0209] In the fifth embodiment, there is no voltage reduction condition compared to the first embodiment, so the voltage command value Vsub* of the second capacitor 4 always satisfies the relationship of Vsub*=Vdc* / 2.

[0210] In the controller 40 , the control action may be determined by using either the control action determiner 9 of the first embodiment or the control action determiner 33 of the second embodiment.

[0211] The main controller 41 is composed of two components: a continuous SW controller 13 and a simple SW boost controller 14. It switches between two actions based on a control selection signal SelectSignal (abbreviated as SS in the figure) output from a control action determiner 9, while generating a duty cycle signal Dutytotal corresponding to each action and a carrier signal Carrier compared with the duty cycle signal Dutytotal, and outputs the signals to the next-stage gate signal generator 16.

[0212] Figure 29 The switching patterns in the operating ranges R1 and R2 of the fifth embodiment are shown.

[0213] In the operating range R1, there are four modes: A1, D1, E1, and F1. In the operating range R2, there are four modes: A2, B2, C2, and D2. As in the first embodiment, the theoretical duty ratio DutyPFCFF is calculated by combining the switching patterns.

[0214] In the gate signal generator 42, gate pulses are calculated through PWM (Pulse Width Modulation) control based on the duty cycle signal Dutytotal and the carrier signal Carrier, and gate signals corresponding to each operating range are output to the semiconductor switching elements 39a~39d of the multi-stage drive branch line 39.

[0215] As described above, the power conversion device 500 according to the fifth embodiment uses the controller 40 to control the on / off state of the semiconductor switching elements of the multi-stage drive branch 39. As in the first embodiment, a low-capacitance capacitor similar to that used in the prior art can be used as the second capacitor 4, while also reducing circuit losses over a wider range than in the prior art. Furthermore, the reduction in the number of components compared to the first embodiment, which eliminates the inability to perform a step-down operation, allows for lower costs and reduced losses.

[0216] Implementation method 6.

[0217] Figure 30 This is a circuit diagram of a power conversion device 600 according to the sixth embodiment, and components corresponding to or equivalent to those in the fifth embodiment are denoted by the same reference numerals.

[0218] The structure of the main circuit of the power conversion device 600 of this embodiment 6 is slightly different from that of the embodiment 5. Specifically, in this embodiment 6, the main circuit includes a diode bridge 36, a reactor 2, and a multi-stage drive branch line 39 connected in this order between the AC power supply 1 and the first capacitor 6, and also includes a second capacitor 4.

[0219] The multi-stage drive branch 39 is constructed by connecting four elements in series: a pair of diodes 40a and 40b in the upper arm and a pair of semiconductor switching elements 39c and 39d in the lower arm. Furthermore, a second capacitor 4 is connected between the connection point between the pair of diodes 40a and 40b in the upper arm and the connection point between the pair of semiconductor switching elements 39c and 39d in the lower arm.

[0220] Reactor 2 is connected between the upper end of diode bridge 36 and the midpoint of multi-stage drive branch line 39. The upper end of multi-stage drive branch line 39 is connected to the P-side terminal of first capacitor 6, and the lower end of multi-stage drive branch line 39 is connected to the N-side terminal of first capacitor 6.

[0221] The remaining components of the main circuit are the same as those in Embodiment 5, and thus their description is omitted. Furthermore, the configuration of the controller 44 is the same as that in Embodiment 5. Furthermore, the operation of the controller 44, which generates gate signals G39c and G39d for the multi-stage drive branch 39 based on input load information (voltage and current information) to control the on / off state of the semiconductor switching elements 39c and 39d, is also the same as that in Embodiment 5, and thus their description is omitted.

[0222] As described above, the power conversion device 600 according to the sixth embodiment can reduce the number of semiconductor switching elements compared to the fifth embodiment by adding the diode bridge 36. Therefore, the circuit can be configured at a relatively low cost.

[0223] In addition, the controllers 8, 32, 35, 37, 40, and 44 in the above embodiment are examples of hardware. Figure 31 The system is shown as comprising a processor 1000 and a storage device 1010. Although not shown, the storage device 1010 includes a volatile storage device such as a random access memory and a non-volatile auxiliary storage device such as a flash memory.

[0224] Alternatively, an auxiliary storage device such as a hard disk may be provided in place of the flash memory. Processor 1000 executes a program input from storage device 1010. In this case, the program is input from the auxiliary storage device to processor 1000 via the volatile storage device. Furthermore, processor 1000 can output data such as computation results to the volatile storage device of storage device 1010, or store data in the auxiliary storage device via the volatile storage device.

[0225] This application describes various exemplary embodiments and examples, but various features, modes, and functions described in one or more embodiments are not limited to application in specific embodiments and can be applied to the embodiments alone or in various combinations.

[0226] Therefore, numerous modifications not shown are contemplated within the scope of the technology disclosed in this application, including, for example, modifying, adding, or omitting at least one component, or extracting at least one component and combining it with components from other embodiments.

Claims

1. A power conversion device comprising at least one reactor, a plurality of switching elements, and a first capacitor disposed between an AC power source and a DC load, and a second capacitor disposed between the reactor and the first capacitor, the power conversion device comprising a controller for controlling the switching operation of the switching element, the power conversion device performing power conversion between the AC voltage of the AC power source and the voltage of the first capacitor, and capable of switching a current path through the second capacitor and a current path not through the second capacitor to output a plurality of voltages, wherein: The controller controls the voltage of the first capacitor and the voltage of the second capacitor to predetermined command values ​​and switches between a first control mode for controlling the switching element at a constant switching frequency during an AC cycle of the AC power source and a second control mode for controlling the switching element at a frequency lower than the switching frequency of the first control mode, based on load information for the DC load. When performing a boost operation based on the second control method, the controller excites and demagnetizes the reactor twice and charges and discharges the second capacitor once each during one switching operation of the switching element.

2. The power conversion device according to claim 1, wherein: The controller has: a control action determiner for determining a control action to be performed in the controller according to the load information; a second capacitor voltage command value calculator for calculating a voltage command value for the second capacitor; a second capacitor voltage controller for controlling the voltage of the second capacitor to the voltage command value calculated by the second capacitor voltage command value calculator; a main controller that performs a control operation of the switching element based on the first control method and a control operation of the switching element based on the second control method; as well as The gate signal generator generates a gate signal for driving the switching element according to the duty ratio and the carrier signal generated by the controller.

3. The power conversion device according to claim 1, wherein: When the controller performs a voltage reduction operation based on the second control method, during the time when the AC voltage becomes lower than the voltage of the first capacitor and higher than the voltage of the second capacitor, the inductor is excited and reset twice during the period when the switching element is switched once, and the second capacitor is charged and discharged once respectively.

4. The power conversion device according to claim 2, wherein: When the main controller performs the control operation based on the second control method, the main controller controls the switching element so that a period for charging the second capacitor and a period for discharging the second capacitor coincide with each other.

5. The power conversion device according to claim 2, wherein: The second capacitor voltage command value calculator determines the voltage command value of the second capacitor based on the voltage command value of the first capacitor and the AC voltage input from the AC power supply.

6. A power conversion device comprising at least one reactor, a plurality of switching elements, and a first capacitor provided between an AC power source and a DC load, a second capacitor provided between the reactor and the first capacitor, the power conversion device including a controller for controlling the switching operation of the switching elements, the power conversion device performing power conversion between the AC voltage of the AC power source and the voltage of the first capacitor, wherein: The controller controls the voltage of the first capacitor and the voltage of the second capacitor to predetermined command values, and switches between a first control mode for controlling the switching element at a constant switching frequency during an AC cycle of the AC power source and a second control mode for controlling the switching element at a frequency lower than the switching frequency of the first control mode, based on load information for the DC load. The controller has: a control action determiner for determining a control action to be performed in the controller according to the load information; a second capacitor voltage command value calculator for calculating a voltage command value for the second capacitor; a second capacitor voltage controller for controlling the voltage of the second capacitor to the voltage command value calculated by the second capacitor voltage command value calculator; a main controller that performs a control operation of the switching element based on the first control method and a control operation of the switching element based on the second control method; as well as a gate signal generator for generating a gate signal for driving the switching element according to the duty cycle and the carrier signal generated by the controller; The main controller is composed of a continuous SW controller that performs a voltage step-up operation according to the first control method, a simple SW step-up controller that performs a voltage step-up operation according to the second control method, and a simple SW step-down controller that performs a voltage step-down operation according to the second control method. When performing a boost operation according to the second control method, the simple SW boost controller determines whether to perform a switching operation or a non-switching operation in a range where the AC voltage is lower than the voltage of the second capacitor based on the input power calculated based on the AC current and AC voltage input from the AC power supply.

7. The power conversion device according to claim 2 or 6, wherein: The control action determiner determines a control action to be performed in the main controller based on input power calculated based on the AC current and AC voltage input from the AC power supply.

8. The power conversion device according to claim 2 or 6, wherein: The control action determiner determines a control action to be performed in the main controller based on an output voltage to the DC load.

9. The power conversion device according to claim 6, wherein: When performing a boost operation according to the second control method, the simple SW boost controller changes the number of switching operations according to the power factor of the AC power source and the input power.

10. The power conversion device according to claim 6, wherein: When performing a boost operation according to the second control method, the simple SW boost controller changes the timing of starting switching according to the power factor of the AC power supply while the AC voltage of the AC power supply is within a range equal to or greater than the voltage of the second capacitor.

11. The power conversion device according to claim 6, wherein: When performing a voltage step-down operation according to the second control method, the simple SW step-down controller determines whether to perform switching or not to perform switching when the AC voltage of the AC power source is within a range below the voltage of the second capacitor based on the voltage of the first capacitor.

12. The power conversion device according to claim 6, wherein: When the simple SW step-down controller performs a step-down operation according to the second control method, the number of switching operations is changed according to the power factor of the AC power source and the input power.

13. A power conversion device comprising at least one reactor, a plurality of switching elements, and a first capacitor provided between an AC power source and a DC load, a second capacitor provided between the reactor and the first capacitor, the power conversion device including a controller for controlling the switching operation of the switching elements, the power conversion device performing power conversion between the AC voltage of the AC power source and the voltage of the first capacitor, wherein: The controller controls the voltage of the first capacitor and the voltage of the second capacitor to predetermined command values, and switches between a first control mode for controlling the switching element at a constant switching frequency during an AC cycle of the AC power source and a second control mode for controlling the switching element at a frequency lower than the switching frequency of the first control mode, based on load information for the DC load. The controller has: a control action determiner for determining a control action to be performed in the controller according to the load information; a second capacitor voltage command value calculator for calculating a voltage command value for the second capacitor; a second capacitor voltage controller for controlling the voltage of the second capacitor to the voltage command value calculated by the second capacitor voltage command value calculator; a main controller that performs a control operation of the switching element based on the first control method and a control operation of the switching element based on the second control method; as well as a gate signal generator for generating a gate signal for driving the switching element according to the duty cycle and the carrier signal generated by the controller; The second capacitor voltage command value calculator determines the voltage command value of the second capacitor based on the voltage command value of the first capacitor and the AC voltage input from the AC power supply. The second capacitor voltage command value operator determines the voltage command value of the second capacitor so that the time when the AC voltage becomes above the voltage of the first capacitor and the time when the AC voltage becomes below the voltage of the first capacitor and above the voltage of the second capacitor are consistent in the AC half cycle under the condition that the voltage command value of the first capacitor becomes below the peak value of the AC voltage.

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