Method for estimating rotor position of an ac motor based on inverter pulse width modulation wave excitation

Abstract

The application provides an AC motor rotor position estimation method based on inverter pulse width modulation wave excitation. A two-level voltage source inverter is modulated by a space vector voltage, and the voltage vector average values during the zero and non-zero vector action periods in the two adjacent PWM periods are calculated. Meanwhile, the current during the zero voltage vector action period at the start / half / end time of each period is measured, the average current slope in the two "half PWM periods" in each period is calculated, and the rotor position calculation formula is obtained to estimate the rotor position of the AC motor in the zero speed to high speed range. The application directly extracts the rotor position information from the stator inductance matrix by using the salient pole effect of the motor itself, realizes the rotor position estimation in the zero speed to high speed range, and is not affected by the change of the counter electromotive force amplitude. The application does not need to inject a high frequency signal, avoids additional loss, and performs current sampling at the start / end time and half cycle time of the PWM period, greatly simplifies the calculation of the current change rate.

Description

Technical Field

[0001] The present invention relates to the field of electrical engineering technology, and in particular to a method for estimating the rotor position of an AC motor in a wide speed range based on fundamental pulse width modulation waveform excitation of a voltage source inverter. Background Art

[0002] High-performance closed-loop control of AC motors typically relies on mechanical position sensors (such as photoelectric encoders or resolvers) to measure rotor speed or angle. However, mechanical position sensors are significantly affected by the operating environment, such as motor vibration and ambient temperature. Not only are they prone to pulse loss, resulting in large measurement errors, but they are also susceptible to damage, resulting in limited reliability. Furthermore, installing a mechanical position sensor requires the introduction of additional interfaces in the control circuit, making the control system more susceptible to interference. Mechanical position sensors are typically mounted on the motor shaft, increasing the motor's size and the shaft's moment of inertia, thus impacting the dynamic performance of the speed control system. Finally, high-precision position sensors are expensive, increasing the cost of the drive system. Due to these shortcomings, researchers both domestically and internationally have proposed using software to estimate the angle or speed of the AC motor rotor, thereby achieving sensorless operation of the AC motor.

[0003] Sensorless position control technologies for AC motors can be primarily categorized into two types: back-EMF-based rotor position estimation and saliency-effect-based rotor position estimation. Back-EMF-based rotor position estimation generally estimates the back-EMF directly or indirectly, and can also integrate it to obtain the flux. The flux contains information about the rotor's speed and angle, hence the name flux observation. The key to this approach lies in designing a suitable flux calculation model. Currently, methods such as model reference adaptive identification, full-order or reduced-order flux observers, sliding mode observers, and extended Kalman filters are commonly used. Salient-pole-effect-based AC motor rotor position estimation utilizes the principle that the motor's saliency effect is modulated by the rotor position. By injecting high-frequency signals or transient switching signals, the induced high-frequency current or voltage signals are processed to obtain rotor or flux angle signals. A prior art method for AC motor rotor position estimation has also been proposed, utilizing fundamental pulse-width modulation (PWM) excitation from a two-level inverter. This method can accurately estimate the rotor position over a wide speed range.

[0004] However, rotor position estimation techniques based on back-EMF are primarily suitable for medium- and high-speed motor operation. At low speeds, the back-EMF amplitude is small and difficult to extract, making accurate flux calculation difficult. Furthermore, at low speeds, the flux model is significantly affected by motor parameters, resulting in poor signal interference resistance. Furthermore, these methods cannot operate stably at zero speed.

[0005] Rotor position estimation techniques based on high-frequency signal injection differ from those based on back-EMF. High-frequency signal injection-based rotor position estimation techniques are not affected by the small back-EMF amplitude during low-speed operation and can be used for position estimation at zero and low speeds. However, these methods require the injection of additional high-frequency signals, resulting in additional losses and noise, while also limiting the control bandwidth. Therefore, they are generally suitable for use at low speeds.

[0006] Traditional rotor position estimation techniques based on fundamental-wave PWM excitation utilize only the excitation of non-zero voltage vectors within the PWM cycle. This method can estimate the rotor position of AC motors at zero and low speeds. Alternatively, by also considering the excitation of zero voltage vectors within the PWM cycle, this method can also estimate the rotor position of AC motors from zero speed to medium and high speeds. However, this method requires multiple current sampling during several voltage vector periods. This current sampling is asynchronous with the current loop sampling, placing high demands on the bandwidth and sampling accuracy of the analog-to-digital converter (ADC), increasing the difficulty and cost of implementation.

[0007] As can be seen, traditional AC motor sensorless control methods based on the motor fundamental wave model typically do not perform well at low and zero speeds and are only suitable for medium and high speeds. Meanwhile, AC motor sensorless control methods based on high-frequency injection are generally only suitable for low and zero speed operation. To address these issues, a sensorless control method for AC motors based on pulse-width modulation (PWM) transient excitation of a voltage source inverter has been proposed, which can estimate the rotor position of AC motors from zero to high speeds. However, this method relies on sampling the current rate of change during zero and non-zero voltage vectors within each PWM cycle. This current rate of change sampling is often asynchronous with the current sampling of the current loop control. This results in asynchrony between the implementation of the position estimation method and the current loop control process, placing high demands on the computing speed of the microprocessor or digital signal processor, the implementation of the sensorless method, and the conversion rate of the analog-to-digital converter (ADC) used for current sampling. Summary of the Invention

[0008] In view of the defects in the prior art, the object of the present invention is to provide a method for estimating the rotor position of an AC motor based on inverter pulse width modulation wave excitation.

[0009] According to one aspect of the present invention, a method for estimating the rotor position of an AC motor based on inverter pulse width modulation wave excitation is provided. The method directly utilizes the salient pole effect of the AC motor and the fundamental pulse width modulation (PWM) signal of a two-level voltage source inverter to extract the rotor position information in the stator inductance matrix from the voltage equation in the α-β coordinate system through mathematical transformation.

[0010] Space vector voltage modulation is adopted for the two-level voltage source inverter, and the average value of the voltage vector during the period of zero vector and non-zero vector applied in two adjacent PWM cycles is calculated respectively. At the same time, by measuring the current during the period of zero voltage vector at the beginning / half / end of each cycle in the two adjacent PWM cycles, the average current slope in the two "half PWM cycles" in each PWM cycle is calculated to obtain a rotor position calculation formula. Based on this formula, the rotor position of the AC motor in the range from zero speed to high speed is estimated.

[0011] Preferably, it includes:

[0012] Obtain the average voltage in the α-β coordinate system of two half cycles within any PWM cycle, and obtain the average current slope in the α-β coordinate system of the two half cycles of the PWM cycle;

[0013] Obtaining an average voltage in an α-β coordinate system of two half cycles in another PWM cycle adjacent to the PWM cycle, and obtaining an average current slope in an α-β coordinate system of two half cycles in the another PWM cycle;

[0014] Obtaining a rotor position estimation formula based on the half-cycle average voltage and half-cycle average current slopes of the two adjacent PWM cycles;

[0015] Based on the rotor position estimation formula, a phase-locked loop or an observer is used to estimate the rotor angle.

[0016] Preferably, obtaining the average voltage in the α-β coordinate system of two half cycles within any PWM cycle and obtaining the average current slope in the α-β coordinate system of the two half cycles of the PWM cycle include:

[0017] The voltage equation of the three-phase permanent magnet synchronous motor in the three-phase coordinate system is transformed by coordinate transformation, and each variable is transformed from the three-phase coordinate system to the α-β coordinate system to obtain the voltage equation of the three-phase permanent magnet synchronous motor in the α-β coordinate system:

[0018]

[0019] Among them, [u α u β ] T is the α-β component of the stator voltage, [i α i β ] T is the α-β component of the stator current, R is the stator phase resistance, ψ f is the rotor flux amplitude, θ is the rotor flux angle, ω=dθ / dt is the synchronous angular velocity, L1=(L d +Lq ) / 2,L2=(L d -L q ) / 2,L d and L q are the direct-axis and quadrature-axis inductances of the permanent magnet synchronous motor in the synchronous rotating coordinate system;

[0020] Assume that in a certain PWM cycle, the voltage vector generated by the inverter is V 01 -V I -V II -V 02 -V III -V IV -V 01 , where V 01 and V 02 is the zero voltage vector, V I ~V IV is a non-zero voltage vector, V I ~V IV and V II ~V III They can be the same or different. When the PWM frequency is much higher than the fundamental frequency of the motor, it is approximately assumed that the rotor angle of the motor remains basically unchanged within one PWM cycle. Then, from formula (1), the average voltage equation in the first half of the PWM cycle is obtained:

[0021]

[0022] in, is the voltage vector V 01 、V I 、V II and V 02 The α-β component of the average value in the first half of the PWM period, is the average value of the α-β component of the current in the first half of the PWM cycle, is the α-β component of the average value of the current change rate during the first half of the PWM period;

[0023] The process of obtaining the average voltage equation in the first half of the PWM cycle is the same as that in the first half of the PWM cycle. The average voltage equation in the second half of the PWM cycle is obtained as follows:

[0024]

[0025] in, is the voltage vector V 07 、V III 、V IV and V 01 The α-β component of the average value in the second half of the PWM cycle, is the average value of the α-β component of the current in the second half of the PWM cycle, is the α-β component of the average value of the current change rate during the second half of the PWM period.

[0026] Preferably, measuring Voltage vector V 01 、V I 、V II and V 02 The α-β component of the average value in the first half of the PWM cycle is calculated as follows:

[0027]

[0028] Among them, t I V I Actual action time, t II V II Actual action time, T s is the PWM period, V I_αβ and V II_αβ V I and V II The α-β component of

[0029] Preferably, the α-β component of the average value of the current change rate in the first half of the PWM cycle is measured, The calculation method used is:

[0030] First, according to the three-phase current collected at the beginning and end of this half PWM cycle, they are transformed into the α-β coordinate system to obtain the current components at the two moments respectively. That is, the current component at the beginning of half PWM cycle and That is, the current component at the end of half a PWM cycle; then use the two-point method to find:

[0031]

[0032] Preferably, obtaining the average voltage in the α-β coordinate system of two half cycles in another PWM cycle adjacent to the PWM cycle, and obtaining the average current slope in the α-β coordinate system of two half cycles in the another PWM cycle, includes:

[0033] Assuming that the average current in the first half and the second half of a PWM cycle are approximately equal, the following equation is obtained:

[0034]

[0035] make:

[0036]

[0037]

[0038] Then we have:

[0039]

[0040] Considering the voltage equation of another PWM cycle adjacent to or separated by one or more PWM cycles from the PWM cycle, we obtain:

[0041]

[0042] in:

[0043]

[0044]

[0045] in, and They are the α-β components of the average voltage values ​​of the first and second half cycles in the PWM cycle, and It is the α-β component of the average value of the current change rate during the first and second half cycles.

[0046] Preferably, the rotor position estimation formula is obtained based on the half-cycle average voltage and half-cycle average current slopes of the two adjacent PWM cycles;

[0047] From the formula (4) and formula (7), we can get:

[0048]

[0049] in,

[0050] By measuring or calculating, the quantities in equations (5a), (5b), (8a), and (8b) are obtained and substituted into equation (9), thus obtaining csin2θ and ccos2θ. The rotor angle θ is then obtained by performing an inverse tangent operation:

[0051] Preferably, within one cycle of the PWM wave used to estimate the rotor position, the action time of at least one non-zero voltage vector should not be shorter than a certain threshold value t min , its operation process includes:

[0052] The first step, during the first PWM cycle used to detect the rotor position:

[0053] If the first non-zero voltage vector V I Action time t I Less than t min :V IAt the beginning, the PWM wave of the corresponding phase shifts to the left (t min -t I ), so that the action time of the first non-zero voltage vector is extended to t min , the PWM waveforms of the other two phases remain unchanged;

[0054] If the first non-zero voltage vector V I Action time t I Greater than or equal to the threshold t min :V I The PWM wave of the corresponding phase at the start time is shifted to the left by k*t min (k>0), the PWM waveforms of the other two phases remain unchanged;

[0055] Step 2: In the PWM cycle adjacent to the PWM cycle selected in step 1:

[0056] If the second non-zero voltage vector V II Action time t II Less than a threshold t min :V II At the end of the phase, the PWM wave of the corresponding phase shifts to the right (t min -t II ), so that the action time of the second non-zero voltage vector is extended to t min , the PWM waveforms of the other two phases remain unchanged;

[0057] If the second non-zero voltage vector V II Action time t II Greater than or equal to the threshold t min :V II The PWM wave of the corresponding phase at the end moment is shifted to the right by k*t min , the PWM waveforms of the other two phases remain unchanged.

[0058] According to a second aspect of the present invention, a terminal is provided, comprising a memory, a processor, and a computer program stored in the memory and executable on the processor, wherein the processor may be configured to execute any one of the methods described above when executing the program.

[0059] According to a third aspect of the present invention, there is provided a computer-readable storage medium having a computer program stored thereon, which can be used to perform any one of the methods described above when the program is executed by a processor.

[0060] Compared with the prior art, the present invention has at least one of the following beneficial effects:

[0061] The AC motor rotor position estimation method based on inverter pulse width modulation wave excitation in the embodiment of the present invention does not require the design of a flux observer. It directly utilizes the salient pole effect of the motor itself (referring to the inconsistency between the d-axis inductance and the q-axis inductance of the permanent magnet synchronous motor) to extract the rotor position information from the stator inductance matrix. The rotor position can be estimated in the zero-speed to high-speed range, and the estimation is not affected by changes in the back-electromotive force amplitude.

[0062] The AC motor rotor position estimation method based on inverter pulse width modulation wave excitation in the embodiment of the present invention does not need to inject high-frequency signals into the motor, thus avoiding the extra loss caused by injecting high-frequency signals and reducing noise.

[0063] The AC motor rotor position estimation method based on inverter pulse width modulation wave excitation in the embodiment of the present invention avoids multiple current sampling within the voltage vector. It only needs to sample the current at the start / end time and half-cycle time of a PWM cycle, which greatly simplifies the calculation of the current change rate. The implementation of the motor position sensorless control method based on PWM excitation does not need to rely on high-performance hardware and processors, and the software method of position estimation is also more convenient to implement. BRIEF DESCRIPTION OF THE DRAWINGS

[0064] Other features, objects and advantages of the present invention will become more apparent upon reading the detailed description of non-limiting embodiments with reference to the following drawings:

[0065] Figure 1 Flowchart of a method for estimating the rotor position of an AC motor based on inverter pulse width modulation wave excitation in an embodiment of the present invention;

[0066] Figure 2 The PWM waveforms are the original PWM waveforms and the PWM waveforms after the first step of modification within one switching period Ts in the first sector in a preferred embodiment of the present invention;

[0067] Figure 3 The original PWM waveform and the PWM waveform modified in the second step within one switching period Ts in the first sector in a preferred embodiment of the present invention;

[0068] Figure 4 This is the waveform of the motor rotor position estimation at 0 rpm in the simulation test of the present invention;

[0069] Figure 5 This is the motor rotor position estimation at 400 rpm in the simulation test of the present invention;

[0070] Figure 6 This is the motor rotor position estimation at 1200 rpm in the simulation test of the present invention. DETAILED DESCRIPTION

[0071] The present invention will be described in detail below with reference to specific embodiments. The following embodiments will help those skilled in the art to further understand the present invention, but are not intended to limit the present invention in any form. It should be noted that, for those skilled in the art, several variations and improvements can be made without departing from the scope of the present invention. These all fall within the scope of protection of the present invention.

[0072] Based on the existing position sensorless control method for AC motors using pulse width modulation (PWM) transient excitation based on a voltage source inverter, the present invention eliminates the step of measuring the current slope during a single voltage vector in each PWM cycle. Instead, current sampling is performed only at the beginning and end of each PWM cycle and at half-cycle times to estimate the rotor position of the AC motor from zero speed to high speed. This allows the current sampling required for rotor position estimation to be synchronized with the current sampling required for the original current loop control of the AC motor, greatly simplifying the implementation process of such position estimation methods and having high application value.

[0073] Based on the above inventive concept, the present invention provides an embodiment, a rotor position estimation method, which directly utilizes the saliency effect of the AC motor itself and the fundamental pulse width modulation (PWM) signal of the two-level voltage source inverter to extract the rotor position information in the stator inductance matrix from the voltage equation in the α-β coordinate system.

[0074] Space vector voltage modulation is used for the two-level voltage source inverter. The average voltage vector during the action of the zero vector and non-zero vector applied in the two adjacent (adjacent or non-adjacent) PWM cycles is calculated respectively. At the same time, by measuring the current of the two zero voltage vectors in each cycle of the two adjacent PWM cycles at the start / end time of each PWM cycle and the half time of the PWM cycle, the average current slope in the two "half PWM cycles" in each PWM cycle is calculated. The rotor position calculation formula is obtained to estimate the rotor position of the AC motor in the range from zero speed to high speed.

[0075] This embodiment directly utilizes the fundamental pulse-width modulation signal (PWM signal) of the voltage source inverter itself to stimulate the salient pole effect of the AC motor, thereby directly extracting rotor position information from the stationary coordinate system. This novel method overcomes the difficulties of rotor position estimation methods based on back-EMF, such as instability at zero speed and low speeds, low accuracy, and susceptibility to motor parameters. It also avoids the injection of high-frequency signals, reducing additional losses and noise. Furthermore, this embodiment avoids multiple current measurements during the action of a single non-zero voltage vector. Instead, phase currents are sampled synchronously (with PWM) during the zero vector period of a PWM cycle, significantly reducing the requirements for ADC bandwidth and resolution.

[0076] Furthermore, in a preferred embodiment of the present invention, a specific process of an AC motor rotor position estimation method is provided, including:

[0077] Step 1: Obtain the average voltage in the α-β coordinate system of two half cycles within any PWM cycle; obtain the average current slope in the α-β coordinate system of two half cycles of the PWM cycle;

[0078] Step 2: Obtain the average voltage in the α-β coordinate system of two half cycles in another PWM cycle adjacent to the PWM cycle; obtain the average current slope in the α-β coordinate system of two half cycles in another PWM cycle;

[0079] Step 3: Based on the average voltage and average current slopes of two half-cycles of two adjacent PWM cycles obtained in Step 1 and Step 2, a rotor position estimation formula is obtained;

[0080] Step 4: Estimate the rotor angle based on the rotor position estimation formula using a phase-locked loop or observer.

[0081] In a preferred embodiment of the present invention, taking the rotor position estimation of a three-phase permanent magnet synchronous motor as an example, a preferred implementation process of step 1 is provided, which is specifically as follows:

[0082] By transforming the voltage equation of the three-phase permanent magnet synchronous motor in the three-phase coordinate system through coordinate transformation, the variables are transformed from the three-phase coordinate system to the α-β coordinate system, and the voltage equation of the three-phase permanent magnet synchronous motor can be obtained:

[0083]

[0084] Among them, [u α u β ] T is the α-β component of the stator voltage, [i α i β ] T is the α-β component of the stator current, R is the stator phase resistance, ψ f is the rotor flux amplitude, θ is the rotor (flux) angle, ω=dθ / dt is the synchronous angular velocity, L1=(L d +L q ) / 2,L2=(L d -L q ) / 2,L d and L q are the direct axis (d-axis) and quadrature axis (q-axis) inductances of the permanent magnet motor in the synchronous rotating coordinate system, respectively.

[0085] Assume that in a certain PWM cycle, the applied voltage vector is V01 -V I -V II -V 02 -V III -V IV -V 01 , where V 01 and V 02 is the zero voltage vector, V I ~V IV is a non-zero voltage vector, V I ~V II Can be with V III ~V IV Same or different. When the PWM frequency is much higher than the fundamental frequency of the motor, it can be approximately considered that the rotor angle of the motor remains basically unchanged within a PWM cycle. Then, from formula (1), the average voltage equation in the first half of the PWM cycle can be obtained:

[0086]

[0087] in, is the voltage vector V 01 、V 07 、V I and V II The α-β component of the average value in the first half of the PWM period, is the average value of the α-β component of the current in the first half of the PWM cycle, is the average value of the α-β components of the current change rate in the first half of the PWM cycle. They are calculated as follows:

[0088]

[0089] Among them, t I V I Actual action time, t II V II Actual action time, T s is the PWM period, VI _αβ and VII _αβ V I and V II The α-β component of

[0090] The α-β components of the current average value do not need to be calculated. The average value of the α-β components of the current change rate in the first half of the PWM cycle can be obtained by first transforming the three-phase current collected at the beginning and end of this half cycle into the α-β coordinate system to obtain the current components at the two moments. (current component at the beginning of a half cycle) and (current component at the end of the half cycle), and then use the two-point method to find:

[0091]

[0092] Similarly, the average voltage equation for the second half of the PWM cycle can be obtained as:

[0093]

[0094] in, is the voltage vector V 07 、V III 、V IV and V 01 The α-β component of the average value in the second half of the PWM cycle, is the average value of the α-β component of the current in the second half of the PWM cycle, is the α-β component of the average current change rate during the second half of the PWM cycle. Similarly, the average voltage and average current change rate of the second half of the cycle can be calculated using the same formula.

[0095] In a preferred embodiment of the present invention, a preferred implementation process of step 2 is provided, which is as follows:

[0096] From equations (2)-(3), and assuming that the average current values ​​in the first and second half of a PWM cycle are approximately equal, we can obtain the following equation:

[0097]

[0098] make:

[0099]

[0100]

[0101] but:

[0102]

[0103] Similarly, considering the voltage equation of another PWM cycle adjacent to the above PWM cycle (adjacent to or separated by one or more PWM cycles), we can obtain:

[0104]

[0105] in:

[0106]

[0107]

[0108] and and They are the α-β components of the average voltage values ​​of the first and second half cycles in the PWM cycle, and It is the average value of the α-β components of the current change rate in the first and second half cycles.

[0109] The above embodiment synchronously realizes current sampling and current sampling required for the original current loop control of the AC motor, greatly simplifies the implementation process of the position estimation method, and has high application value.

[0110] In a preferred embodiment of the present invention, a preferred implementation process of step 3 is provided, which is specifically as follows:

[0111] From equation (4) and time (7), we can get the following equation (9):

[0112]

[0113] in, It can be seen that as long as the quantities in equations (5a), (5b), (8a) and (8b) can be obtained by measurement or calculation, and substituted into equation (9), csin2θ and ccos2θ can be obtained, and the rotor angle θ can be obtained by inverse tangent operation, such as:

[0114] In a preferred embodiment of the present invention, in order to overcome measurement noise, equation (9) can be input into a phase-locked loop (PLL) or an observer to obtain a smoother estimated rotor angle.

[0115] In order to use (9) for position estimation, the denominator in formula (9) must not be zero. At the same time, in order to achieve a sufficiently large signal-to-noise ratio, in a preferred embodiment of the present invention, the action time of at least one non-zero voltage vector in one cycle of the PWM wave used to estimate the rotor position should not be shorter than a certain threshold value t min There are different ways to implement this condition. One of them is as follows, which can ensure that the average current slope of a certain amplitude is measured while satisfying the non-zero denominator in formula (9). This method is completed in two steps:

[0116] The first step, such as Figure 2 As shown, in the first PWM cycle used to detect the rotor position:

[0117] If the first non-zero voltage vector V I Action time t I Less than t min :V I At the beginning, the PWM wave of the corresponding phase shifts to the left (t min -t I ), so that the action time of the first non-zero voltage vector is extended to t min, the PWM waveforms of the other two phases remain unchanged;

[0118] If the first non-zero voltage vector V I Action time t I Greater than or equal to the threshold t min :V I The PWM wave of the corresponding phase at the start time is shifted to the left by k*t min (k>0), the PWM waveforms of the other two phases remain unchanged.

[0119] Figure 2 In FIG, PWMA, PWMB and PWMC are the original three-phase PWM waveforms, and PWMA', PWMB' and PWMC' are the three-phase PWM waveforms after the first step of change. Figure 2 Taking the first sector as an example, the other sectors can also use the strategy of the first step to modify the PWM waveform. Trig1 to Trig3 are sampling moments within a PWM cycle, which can meet the requirements of synchronous sampling.

[0120] The second step is Figure 3 As shown, in the PWM period adjacent to the PWM period selected in the first step:

[0121] If the second non-zero voltage vector V II Action time t II Less than a certain threshold t min :V II At the end of the phase, the PWM wave of the corresponding phase shifts to the right (t min -t II ), so that the action time of the second non-zero voltage vector is extended to t min , the PWM waveforms of the other two phases remain unchanged;

[0122] If the second non-zero voltage vector V II Action time t II Greater than or equal to the threshold t min :V II The PWM wave of the corresponding phase at the end moment is shifted to the right by k*t min , the PWM waveforms of the other two phases remain unchanged.

[0123] Figure 3 In FIG, PWMA, PWMB and PWMC are the original three-phase PWM waveforms, and PWMA', PWMB' and PWMC' are the three-phase PWM waveforms after the second step change. Figure 3 Taking the first sector as an example, other sectors can also use the strategy of the second step to modify the corresponding PWM waveform. Trig1 to Trig3 are sampling moments within a PWM cycle, which can meet the requirements of synchronous sampling.

[0124] Based on the same inventive concept, the present invention provides a terminal, comprising a memory, a processor, and a computer program stored in the memory and executable on the processor. When the processor executes the program, the terminal can be used to execute any one of the methods.

[0125] Based on the same inventive concept, the present invention provides a computer-readable storage medium having a computer program stored thereon. When the program is executed by a processor, it can be used to perform any one of the methods.

[0126] Based on the AC motor wide speed range rotor position estimation method based on voltage source inverter fundamental pulse width modulation waveform excitation in the above embodiment, a position sensorless control simulation model based on classic 7-segment SVPWM excitation was built in Matlab / Simulink. Synchronous current sampling was used to calculate and measure the voltage average vector and average current slope within a half cycle to verify the effectiveness of the position estimation method proposed in the above embodiment of the present invention in zero speed and high speed ranges. The simulated motor and control parameters are shown in Table 1:

[0127] Table 1 Motor and control parameters

[0128]

[0129] In the simulation, the position signal estimated by equation (9) is input into a phase-locked loop (PLL) to obtain new position signals sin2θ and cos2θ and the rotor angle θ. The present invention is not limited to the use of a PLL; other methods, such as an observer, can also be used to obtain new position signals sin2θ and cos2θ and the rotor angle θ.

[0130] (1) The motor runs at zero speed

[0131] The motor speed is set to 0 rpm (electrical frequency 0 Hz), and the rotor position is estimated using the method designed by the present invention. The waveform during the simulation is as follows: Figure 4 As shown, the waveforms from top to bottom are sin2θ and cos2θ (the dashed line represents sin2θ, and the solid line represents cos2θ), the estimated rotor position and the actual rotor position (in electrical radians), and the position estimation error (in electrical radians). It should be noted that the estimated rotor position is close to 2π, which is very close to the actual rotor position, that is, 0 radians.

[0132] (2) The motor runs at 400 rpm

[0133] The motor is set to run at 400 rpm (electrical frequency 20 Hz), and the rotor position is estimated using the method designed by the present invention. The waveform during the simulation is as follows: Figure 5As shown, the waveforms from top to bottom are sin2θ and cos2θ (the dotted line represents sin2θ, the solid line represents cos2θ), the estimated rotor position and the actual rotor position (electrical radians), and the position estimation error (electrical radians). Figure 5 It means that when the motor runs at 400 rpm, the maximum error does not exceed 0.05 radians.

[0134] (3) The motor runs at 1200 rpm

[0135] The motor is set to run at 1200 rpm (electrical frequency 60 Hz), and the rotor position is estimated using the method designed by the present invention. The waveform during the simulation is as follows: Figure 6 As shown, the waveforms from top to bottom are sin2θ and cos2θ (the dotted line represents sin2θ, the solid line represents cos2θ), the estimated rotor position and the actual rotor position (electrical radians), and the error between the actual position and the estimated position (electrical radians).

[0136] The above simulations prove that the method proposed in the embodiment of the present invention can effectively estimate the angle of the permanent magnet synchronous motor rotor at zero speed and medium and high speeds.

[0137] It should be noted that the present invention is not only applicable to rotor position estimation for three-phase permanent magnet synchronous motors, but can also be applied to position estimation for other types of three-phase AC motors, such as induction motors and synchronous reluctance motors. Furthermore, the present invention is not limited to three-phase two-level voltage source converters; it is also applicable to other voltage source converters that can drive these three-phase AC motors.

[0138] The above describes specific embodiments of the present invention. It should be understood that the present invention is not limited to the specific embodiments described above, and those skilled in the art may make various modifications or variations within the scope of the claims without affecting the essence of the present invention. The above preferred features may be used in any combination as long as they do not conflict with each other.

Claims

1. A method for estimating the rotor position of an AC motor based on inverter pulse width modulation wave excitation, characterized in that: For the voltage equation of the AC motor in the α-β coordinate system, the rotor position information in the stator inductance matrix is ​​extracted from the voltage equation through mathematical transformation by directly utilizing the salient pole effect of the AC motor itself and the fundamental pulse width modulation (PWM) signal of the two-level voltage source inverter; wherein: Obtain the average voltage in the α-β coordinate system of two half cycles within any PWM cycle, and obtain the average current slope in the α-β coordinate system of the two half cycles of the PWM cycle; Obtaining an average voltage in an α-β coordinate system of two half cycles in another PWM cycle adjacent to the PWM cycle, and obtaining an average current slope in an α-β coordinate system of two half cycles in the another PWM cycle; The rotor position estimation formula is obtained based on the half-cycle average voltage and half-cycle average current slopes of two adjacent PWM cycles; Based on the rotor position estimation formula, using a phase-locked loop or an observer, the rotor angle is estimated; The rotor position estimation formula is obtained based on the half-cycle average voltage and half-cycle average current slope of the two adjacent PWM cycles; in, L1=(L d +L q ) / 2,L2=(L d -L q ) / 2,L d and L q are the direct-axis and quadrature-axis inductances of the permanent magnet synchronous motor in the synchronous rotating coordinate system; By measuring or calculating, the following quantities are obtained: in, and They are the α-β components of the average voltage values ​​of the first and second half cycles in the PWM cycle respectively; and, is the α-β component of the average value of the current change rate during the first and second half cycles; in, and They are the α-β components of the average voltage values ​​of the first and second half cycles in the PWM cycle, and is the α-β component of the average value of the current change rate during the first and second half cycles; Substituting into equation (9), we can obtain csin2θ and ccos2θ, and obtain the rotor angle θ through the inverse tangent operation:

2. The method for estimating the rotor position of an AC motor based on inverter pulse width modulation wave excitation according to claim 1, characterized in that: The step of obtaining the average voltage in the α-β coordinate system of two half cycles within any PWM cycle and obtaining the average current slope in the α-β coordinate system of the two half cycles of the PWM cycle includes: The voltage equation of the three-phase permanent magnet synchronous motor in the three-phase coordinate system is transformed by coordinate transformation, and each variable is transformed from the three-phase coordinate system to the α-β coordinate system to obtain the voltage equation of the three-phase permanent magnet synchronous motor in the α-β coordinate system: Among them, [u α u β ] T is the α-β component of the stator voltage, [i α i β ] T is the α-β component of the stator current, R is the stator phase resistance, ψ f is the rotor flux amplitude, θ is the rotor angle, ω=dθ / dt is the synchronous angular velocity, L1=(L d +L q ) / 2,L2=(L d -L q ) / 2,L d and L q are the direct-axis and quadrature-axis inductances of the permanent magnet synchronous motor in the synchronous rotating coordinate system; Assume that in a certain PWM cycle, the voltage vector generated by the inverter is V 01 -V I -V II -V 02 -V III -V IV -V 01 , where V 01 and V 02 is the zero voltage vector, V I ~V IV is a non-zero voltage vector, V I ~V IV and V II ~V III They can be the same or different. When the PWM frequency is much higher than the fundamental frequency of the motor, it is approximately assumed that the rotor angle of the motor remains basically unchanged within one PWM cycle. Then, from formula (1), the average voltage equation in the first half of the PWM cycle is obtained: in, is the voltage vector V 01 、V I 、V II and V 02 The α-β component of the average value in the first half of the PWM period, is the average value of the α-β component of the current in the first half of the PWM cycle, is the α-β component of the average value of the current change rate during the first half of the PWM period; The process of obtaining the average voltage equation in the first half of the PWM cycle is the same as that in the first half of the PWM cycle. The average voltage equation in the second half of the PWM cycle is obtained as follows: in, is the voltage vector V 02 、V III 、V IV and V 01 The α-β component of the average value in the second half of the PWM cycle, is the average value of the α-β component of the current in the second half of the PWM cycle, is the α-β component of the average value of the current change rate during the second half of the PWM period.

3. The method for estimating the rotor position of an AC motor based on inverter pulse width modulation wave excitation according to claim 2, characterized in that: Measured voltage vector V 01 、V I 、V II and V 02 α-β components of the average value during the first half of the PWM period The calculation method used is: Among them, t I V I Actual action time, t II V II Actual action time, T s is the PWM period, V I_αβ and V II_αβ V I and V II The α-β component of 4. The method for estimating the rotor position of an AC motor based on inverter pulse width modulation wave excitation according to claim 2, characterized in that: Measure the α-β component of the average value of the current change rate in the first half of the PWM cycle, The calculation method used is: First, according to the three-phase current collected at the beginning and end of this half PWM cycle, they are transformed into the α-β coordinate system to obtain the current components at the two moments: the current component at the beginning of the half PWM cycle and the current component at the end of half a PWM cycle Then use the two-point method to find:

5. The method for estimating the rotor position of an AC motor based on inverter pulse width modulation wave excitation according to claim 2, characterized in that: The step of obtaining an average voltage in an α-β coordinate system of two half cycles in another PWM cycle adjacent to the PWM cycle, and obtaining an average current slope in an α-β coordinate system of two half cycles in the another PWM cycle, includes: Assuming that the average current in the first half and the second half of a PWM cycle are approximately equal, the following equation is obtained: make: Then we have: Considering the voltage equation of another PWM cycle adjacent to or separated by one or more PWM cycles from the PWM cycle, we obtain: in: in, and They are the α-β components of the average voltage values ​​of the first and second half cycles in the PWM cycle, and It is the α-β component of the average value of the current change rate during the first and second half cycles.

6. The method for estimating the rotor position of an AC motor based on inverter pulse width modulation wave excitation according to claim 1, characterized in that: In one cycle of the PWM wave used to estimate the rotor position, the action time of at least one non-zero voltage vector should not be shorter than the threshold value t min , its operation process includes: The first step, during the first PWM cycle used to detect the rotor position: If the first non-zero voltage vector V I Action time t I Less than t min :V I At the beginning, the PWM wave of the corresponding phase shifts to the left by t min -t I , so that the action time of the first non-zero voltage vector is extended to t min , the PWM waveforms of the other two phases remain unchanged; If the first non-zero voltage vector V I Action time t I Greater than or equal to the threshold t min :V I The PWM wave of the corresponding phase at the start time is shifted to the left by k*t min , k>0, the PWM waveforms of the other two phases remain unchanged; Step 2: In the PWM cycle adjacent to the PWM cycle selected in step 1: If the second non-zero voltage vector V II Action time t II Less than a certain threshold t min :V II At the end of the phase, the PWM wave of the corresponding phase shifts to the right by t min -t II , so that the action time of the second non-zero voltage vector is extended to t min , the PWM waveforms of the other two phases remain unchanged; If the second non-zero voltage vector V II Action time t II Greater than or equal to the threshold t min :V II The PWM wave of the corresponding phase at the end moment is shifted to the right by k*t min , the PWM waveforms of the other two phases remain unchanged.

7. A terminal comprising a memory, a processor, and a computer program stored in the memory and executable on the processor, wherein: When the processor executes the program, it can be used to perform the method according to any one of claims 1 to 6.

8. A computer-readable storage medium having a computer program stored thereon, characterized in that: When the program is executed by a processor, it can be used to perform the method according to any one of claims 1 to 6.

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