Full-order terminal sliding mode control method for brushless doubly-fed motor independent generation system

Abstract

The application discloses a full-order terminal sliding mode control method for a brushless doubly-fed motor independent power generation system, and belongs to the field of motor control. The application solves the problem that the traditional full-order terminal sliding mode control algorithm does not simultaneously consider the matching and non-matching uncertainty caused by motor parameter perturbation in the motor model, resulting in low system output voltage precision, poor system anti-interference ability and control effect. According to the brushless doubly-fed motor mathematical model, the application considers the matching and non-matching uncertainty caused by brushless doubly-fed motor parameter perturbation, determines the voltage loop state equation of the non-matching power winding outer ring voltage subsystem and the current loop state equation of the matching control winding inner ring current subsystem, and designs corresponding inner and outer loop full-order terminal sliding mode controllers for the subsystems to compensate the uncertainty, so that the tracking error of the power winding voltage amplitude converges to zero, and the tracking error of the control winding current amplitude converges to zero. The application is mainly suitable for independent power generation control when the brushless doubly-fed motor is connected to a balanced load.

Description

Technical Field

[0001] This invention belongs to the field of motor control. Background Technology

[0002] As a new type of AC induction motor, the brushless doubly-fed induction motor has broad application prospects in fields such as wind power generation and ship shaft-driven power generation. Compared with traditional doubly-fed induction motors, the brushless doubly-fed induction motor reduces the number of brushes and slip rings, making it more economical and reliable. Its typical structure consists of two sets of stator windings with different pole pairs and one set of rotor windings. The two sets of stator windings are the control winding and the power winding, respectively, and their magnetic fields are independent and uncoupled.

[0003] When brushless doubly-fed induction generators (DFIGs) operate in wind power generation and ship shaft-driven power generation, their rotor speed fluctuates significantly, and motor parameters are affected by electromagnetic saturation, temperature changes, and noise. Traditional PI control exhibits poor robustness, easily leading to large output voltage fluctuations. Novel control methods such as model predictive control have been applied to DFIG control in recent years to improve robustness and power generation accuracy. However, most of these methods require precise motor parameters, increasing the computational demands on the motor system and placing higher demands on controller hardware, thus raising the cost of the motor system. Furthermore, due to the presence of voltage and current loops in the DFIG control structure, perturbations of motor parameters such as resistance and inductance generate uncertainties in both the outer voltage loop and the inner current loop. This allows DFIGs to be considered as second-order nonlinear systems with both matched and unmatched uncertainties. However, existing research has not considered this aspect, resulting in steady-state errors in the output voltage of DFIGs and poor anti-interference capabilities.

[0004] Since a brushless doubly-fed induction generator (DFIG) is a typical nonlinear and multi-coupled system, existing technologies do not fully consider the system's matching and unmatching uncertainties. The system is sensitive to external disturbances and internal parameter perturbations, resulting in large output voltage ripples and poor power quality. Therefore, a robust and fast control method is urgently needed. Sliding mode control, as a typical nonlinear control method, has strong robustness to internal parameter perturbations and external disturbances, and is therefore widely used in motor control, aerospace, and robotics. However, traditional sliding mode controllers, due to the switching function in the control law, cause system chattering, which manifests as high system noise and poor output voltage accuracy in motor-generator systems. Full-order terminal sliding mode control can significantly improve the system's speed and robustness, effectively eliminating system chattering. However, traditional full-order terminal sliding mode control algorithms have large amplitudes of output voltage, current, and power, and reach steady state slowly, resulting in poor control performance. Furthermore, the matching and unmatching uncertainties caused by brushless doubly-fed induction generator system parameter perturbations are not considered in the motor model, leading to unsatisfactory control performance in terms of accuracy and dynamics of existing control methods. Scholars have already used sliding mode control to solve the unmatched uncertainty problem in other motor models. However, traditional full-order terminal sliding mode control algorithms do not simultaneously consider the matched and unmatched uncertainties caused by motor parameter perturbations in the motor model, resulting in low output voltage accuracy, poor system anti-interference capability, and poor control performance. These problems urgently need to be solved. Summary of the Invention

[0005] The purpose of this invention is to solve the problem that traditional full-order terminal sliding mode control algorithms do not simultaneously consider the matching and non-matching uncertainties caused by motor parameter perturbations in the motor model, resulting in low system output voltage accuracy, poor system anti-interference capability and control effect. This invention provides a full-order terminal sliding mode control method for a brushless doubly-fed motor independent power generation system.

[0006] A full-order terminal sliding mode control method for a brushless doubly-fed induction generator independent power generation system, including:

[0007] Based on the mathematical model of the brushless doubly fed motor, considering the matching and non-matching uncertainties caused by parameter perturbation of the brushless doubly fed motor, the voltage loop state equation of the outer loop voltage subsystem of the non-matching power winding and the current loop state equation of the inner loop current subsystem of the matching control winding are determined.

[0008] For the voltage loop state equation of the unmatched power winding outer loop voltage subsystem, an outer loop full-order terminal sliding mode controller is designed to compensate for the unmatched uncertainty in the independent power generation system of the brushless doubly fed motor, so that the tracking error of the power winding voltage amplitude of the brushless doubly fed motor converges to zero.

[0009] For the current loop state equation of the inner loop current subsystem of the matching control winding, an inner loop full-order terminal sliding mode controller is designed to compensate for the matching uncertainty in the independent generator system of the brushless doubly fed motor, so that the tracking error of the control winding current amplitude of the brushless doubly fed motor converges to zero.

[0010] Preferably, the specific process for designing the outer loop full-order terminal sliding mode controller is as follows:

[0011] Step A1: Based on the voltage loop state equation of the unmatched power winding outer loop voltage subsystem, design the outer loop voltage sliding surface and the outer loop composite reaching law;

[0012] Step A2: Substitute the voltage loop state equation of the unmatched power winding outer loop voltage subsystem into the designed outer loop voltage sliding surface, differentiate it, and then combine it with the outer loop composite reaching law to determine the overall control law of the outer loop full-order terminal sliding controller.

[0013] Preferably, in step A1, the outer loop voltage sliding surface is:

[0014]

[0015] In the formula, s u For the outer loop voltage sliding surface, e u The error between the output voltage amplitude of the power winding and the given value is given by c0, which is a constant, and q and p are both odd numbers, satisfying 0.

[0016]

[0017] In the formula, Let be the first-order derivative of the sliding surface of the outer loop voltage, defined as the outer loop composite reaching law, where c1, c2, a, η, k, and q′ are all constants, and sat(s) u ) for s u The saturation function.

[0018] Preferably, the overall control law of the outer loop full-order terminal sliding mode controller in step A2 is:

[0019] u = u eq +u n ;

[0020] in,

[0021]

[0022] u is the overall control law of the outer loop full-order terminal sliding mode controller. eq and u n ​Let c0 be the equivalent control law and the actual control law of the outer-loop full-order terminal sliding mode controller, respectively. c0 is a constant, and q and p are both odd numbers, satisfying 0. u The error between the output voltage amplitude of the power winding and the given value is given by τ, where t is the upper bound of the integral, τ is the sliding mode surface parameter, and c1, c2, a, η, k, and q′ are all constants. u It is the sliding surface of the outer loop voltage.

[0023] Preferably,

[0024] Where Δ is the boundary value of the boundary layer, and both k and Δ are constants.

[0025] Preferably, the specific process for designing the inner-loop full-order terminal sliding mode controller is as follows:

[0026] Step B1: Based on the current loop state equation of the inner loop current subsystem of the matched control winding, design the inner loop current sliding surface and the inner loop composite reaching law.

[0027] Step B2: Substitute the current loop state equation of the matching control winding inner loop current subsystem into the designed inner loop current sliding surface, differentiate it, and then combine it with the inner loop composite reaching law to determine the overall control law of the inner loop full-order terminal sliding controller.

[0028] Preferably, in step B1, the inner loop current sliding surface is:

[0029]

[0030] In the formula: s i For the inner loop current sliding surface, e i To control the error between the winding output current and the current setpoint, C2 = diag(c 21 ,c 22 C2 represents the positive definite diagonal matrix of the designed inner loop current sliding surface, c 21 and c 22 These are the first and second parameters in C2, respectively. Both q and p are odd numbers, satisfying 0

[0031]

[0032] In the formula, Let be the first-order derivative of the sliding mode surface of the inner loop current, defined as the inner loop composite reaching law, where c1, c2, a, η, k, and q′ are all constants, and sat(s) u ) for s u The saturation function.

[0033] ​​Preferably, the overall control law of the inner-loop full-order terminal sliding mode controller in step B2 is:

[0034] u′=u′ eq +u′ n

[0035] in,

[0036]

[0037] u' is the overall control law of the inner-loop full-order terminal sliding mode controller, u' eq and u' n These are the equivalent control law and the actual control law of the inner-loop full-order terminal sliding mode controller, respectively. F 00 T is the correlation matrix of the inner loop with respect to the dq-axis current of the control winding. 10 To control the autocorrelation matrix of the winding dq-axis current, where T 10 =R 20 / σ 20 L 20 R 20 To control the initial value of the winding resistance, σ 20 To control the combined inductance parameters of the rotor and rotor windings, L 20 To control the initial value of the winding inductance, i 2dq To control the dq-axis current of the winding, K u0 K is an intermediate variable. u0 =ω2L 1r0 L 2r0 [ω2L 1r0 L 2r0 I 2E +ω2(L 2 1r0 -L 10 L r0 )i 1d -R 10 L 1r0 i 1q ], ω2 is the electrical angular frequency on the control winding side, L 1r0 L represents the initial mutual inductance between the power winding and the rotor winding. 2r0 To control the initial value of the mutual inductance between the rotor winding and the winding, I 2E L represents the amplitude of the control winding current at the equilibrium point. 10 L is the initial value of the self-inductance of the power winding. r0 i is the initial value of the self-inductance of the rotor winding. 1d R is the d-axis current of the power winding. 10 Let i be the initial resistance value of the power winding. 1qLet ΔI be the q-axis current of the power winding. 2ref To control the error of the winding d-axis current reference value, t is the upper bound of integration, τ is the sliding mode surface parameter, and c0, c1, c2, a, η, k0, k', and q' are all constants, and sat(s i ) for s i The saturation function, s i It is the sliding surface of the inner loop current.

[0038] Preferably,

[0039] Where Δ is the boundary value of the boundary layer, and both k and Δ are constants.

[0040] Preferably, the voltage loop state equation is:

[0041]

[0042] Among them, e u i represents the error between the output voltage amplitude of the power winding and the given value. 2d To control the d-axis current of the winding, i 2d As the overall control law of the outer loop full-order terminal sliding mode controller, ΔK u For custom parameter K u The parameter perturbation, ΔK u K is a bounded value. u0 K is an intermediate variable. u0 =ω2L 1r0 L 2r0 [ω2L 1r0 L 2r0 I 2E +ω2(L 2 1r0 -L 10 L r0 )i 1d -R 10 L 1r0 i 1q ], ω2 is the electrical angular frequency on the control winding side, L 1r0 L represents the initial mutual inductance between the power winding and the rotor winding. 2r0 To control the initial value of the mutual inductance between the rotor winding and the winding, I 2E L represents the amplitude of the control winding current at the equilibrium point. 10 L is the initial value of the self-inductance of the power winding. r0 i is the initial value of the self-inductance of the rotor winding. 1d R is the d-axis current of the power winding. 10 Let i be the initial resistance value of the power winding. 1q e is the q-axis current of the power winding. idTo control the error between the winding d-axis current and the reference value, δ1 represents the gain uncertainty, and it satisfies the boundary condition: |δ1|=|ΔK u / K u0 | < 1;

[0043] The current loop state equation is:

[0044]

[0045] Among them, i 2dq To control the dq axis current of the winding, For i 2dq The differential, T 10 To control the autocorrelation matrix of the winding dq-axis current, δ2 is a constant, δ2=1-L 2 2r / (L2L r L2 is the self-inductance of the control winding, L r For the self-inductance of the rotor winding, d 2dq =[d 2d ,d 2q ] T d 2dq To control the d-axis component of the winding current 2d and q-axis component d 2q The cross disturbance vector between, u 2dq To control the dq axis voltage of the winding, u 2dq It also serves as the overall control law for the inner loop full-order terminal sliding mode controller.

[0046] Advantages of this invention:

[0047] For brushless doubly-fed induction generator (DFIG) independent power generation systems, environmental changes due to temperature and frequency variations lead to matched and unmatched uncertainties in the system. The full-order terminal sliding mode control method for DFIG independent power generation systems proposed in this invention compensates for these matched and unmatched uncertainties, offsets control errors caused by parameter perturbations, improves the accuracy of output voltage, current, and power, reduces ripple, and enhances the system's anti-interference capability and robustness under environmental changes. Furthermore, because sliding mode control itself accelerates error convergence, the output voltage and current of the DFIG system can quickly track the given values, accelerating the arrival at steady state and resulting in better control performance and significantly improved output power quality. Attached Figure Description

[0048] Figure 1 This is a flowchart illustrating the full-order terminal sliding mode control method for a brushless doubly-fed motor independent power generation system.

[0049] Figure 2This is a schematic diagram of a brushless doubly-fed motor independent power generation system.

[0050] Figure 2 Middle,U 1ref U1 is the reference value for the voltage amplitude of the power winding, and I is the output voltage amplitude of the power winding. 2E To control the feedforward compensation of the winding d-axis current, i 2dref To control the reference value of the d-axis current of the winding, i 2qref To control the reference value of the winding q-axis current, i 2d To control the actual value of the d-axis current of the winding, i 2q To control the actual value of the winding q-axis current, D 2d To control the feedforward compensation value of the winding d-axis current, D 2q To control the feedforward compensation value of the winding q-axis current, u 2a To control the voltage of phase a of winding, u 2b To control the voltage across the b-axis of the winding, u 2c To control the c-axis voltage of the winding, u DC The DC bus voltage is represented by CW, the control winding side is represented by PW, and the power winding side is represented by i. 2abc To control the three-phase currents abc of the windings, θ2 is the electrical angle of the motor, ω r ω1 is the rotor angular velocity, ω2 is the electromagnetic angular velocity on the control winding side, and u is the rotor angular velocity. 1abc For the three-phase abc voltage on the power winding side, i 1abc For the three-phase abc current on the power winding side, i 1d i is the d-axis current of the power winding. 1q Let θ be the q-axis current of the power winding. r ω is the rotor angular velocity, and ω1 is the electromagnetic angular velocity on the power winding side. Detailed Implementation

[0051] The technical solutions of the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of the present invention, and not all embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those skilled in the art without creative effort are within the scope of protection of the present invention.

[0052] It should be noted that, unless otherwise specified, the embodiments and features described in the present invention can be combined with each other.

[0053] This invention aims to address the problems of low output voltage accuracy and poor system anti-interference capability caused by the matching and non-matching uncertainties resulting from insufficient consideration of motor parameter perturbations in existing full-order terminal sliding mode control algorithms. A novel full-order terminal sliding mode control method for brushless doubly-fed motor independent power generation systems is proposed.

[0054] See Figure 1 and Figure 2 This embodiment describes a full-order terminal sliding mode control method for a brushless doubly-fed induction generator independent power generation system, comprising:

[0055] Based on the mathematical model of the brushless doubly fed motor, considering the matching and non-matching uncertainties caused by parameter perturbation of the brushless doubly fed motor, the voltage loop state equation of the outer loop voltage subsystem of the non-matching power winding and the current loop state equation of the inner loop current subsystem of the matching control winding are determined.

[0056] For the voltage loop state equation of the unmatched power winding outer loop voltage subsystem, an outer loop full-order terminal sliding mode controller is designed to compensate for the unmatched uncertainty in the independent power generation system of the brushless doubly fed motor, so that the tracking error of the power winding voltage amplitude of the brushless doubly fed motor converges to zero.

[0057] For the current loop state equation of the inner loop current subsystem of the matching control winding, an inner loop full-order terminal sliding mode controller is designed to compensate for the matching uncertainty in the independent generator system of the brushless doubly fed motor, so that the tracking error of the control winding current amplitude of the brushless doubly fed motor converges to zero.

[0058] Furthermore, the specific process for designing the outer loop full-order terminal sliding mode controller is as follows:

[0059] Step A1: Based on the voltage loop state equation of the unmatched power winding outer loop voltage subsystem, design the outer loop voltage sliding surface and the outer loop composite reaching law;

[0060] Furthermore, the voltage loop state equation is:

[0061]

[0062] Among them, e u i represents the error between the output voltage amplitude of the power winding and the given value. 2d To control the d-axis current of the winding, i 2d It also serves as the overall control law for the outer-loop full-order terminal sliding mode controller, ΔK u For custom parameter K u The parameter perturbation, ΔK u K is a bounded value. u0 K is an intermediate variable. u0 =ω2L 1r0 L 2r0 [ω2L 1r0 L 2r0 I 2E +ω2(L 2 1r0 -L 10 L r0 )i1d -R 10 L 1r0 i 1q ], e id To control the error between the winding d-axis current and the reference value, δ1 represents the gain uncertainty, and it satisfies the boundary condition: |δ1|=|ΔK u / K u0 |<1.

[0063] In step A1, the outer loop voltage sliding surface is:

[0064]

[0065] In the formula, s u For the outer loop voltage sliding surface, e u The error between the output voltage amplitude of the power winding and the given value is given by c0, which is a constant, and q and p are both odd numbers, satisfying 0.

[0066] The outer ring compounding convergence law is:

[0067]

[0068] In the formula, Let be the first-order derivative of the sliding surface of the outer loop voltage, defined as the outer loop composite reaching law, where c1, c2, a, η, k, and q′ are all constants, and sat(s) u ) for s u The saturation function. Specifically, c1 = 600, a = 2 / 3, η = 3 / 5, k = 30, c2 = 600.

[0069]

[0070] Here, Δ represents the boundary value of the boundary layer, and both k and Δ are constants. Switching control is used outside the boundary layer, while linearization control is used inside the boundary layer.

[0071] Step A2: Substitute the voltage loop state equation of the unmatched power winding outer loop voltage subsystem into the designed outer loop voltage sliding surface, differentiate it, and then combine it with the outer loop composite reaching law to determine the overall control law of the outer loop full-order terminal sliding controller.

[0072] The overall control law of the outer loop full-order terminal sliding mode controller in step A2 is:

[0073] u = u eq +u n ;

[0074] in,

[0075] ​

[0076] u is the overall control law of the outer loop full-order terminal sliding mode controller. eq and u n Let c0 be the equivalent control law and the actual control law of the outer-loop full-order terminal sliding mode controller, respectively. c0 is a constant, and q and p are both odd numbers, satisfying 0. u The error between the output voltage amplitude of the power winding and the given value is given by τ, where t is the upper bound of the integral, τ is the sliding mode surface parameter, and c1, c2, a, η, k, and q′ are all constants. u This is the outer loop voltage sliding surface. Specifically, c0 = 600, c1 = 600, a = 2 / 3, η = 3 / 5, k = 30, c2 = 600, q′ = 800.

[0077] Furthermore, to prove that the power winding output voltage stably tracks the voltage setpoint, the Lyapunov stability criterion is used to prove stability, where the Lyapunov function is V1 = 0.5S. u 2 Proof is required. Through a series of proofs, it can be concluded that:

[0078]

[0079] The above equation shows the outer loop voltage sliding surface s of the outer loop subsystem of the unmatched power winding output voltage amplitude. u Under the action of the overall control law u of the designed outer-loop full-order terminal sliding mode controller, the tracking error e will be output from the system. u =U 1ref -U1 will asymptotically converge to the symmetric point, i.e., S. u =0,e u =0. V1 is the Lyapunov function definition of the outer loop voltage, and η1 is the Lyapunov function stability coefficient of the outer loop.

[0080] The specific process for designing the inner-loop full-order terminal sliding mode controller is as follows:

[0081] Step B1: Based on the current loop state equation of the inner loop current subsystem of the matched control winding, design the inner loop current sliding surface and the inner loop composite reaching law.

[0082] In step B1, the inner loop current sliding surface is:

[0083]

[0084] In the formula: s i For the inner loop current sliding surface, e i To control the error between the winding output current and the current setpoint, C2 = diag(c 21 ,c​22 C2 represents the positive definite diagonal matrix of the designed inner loop current sliding surface, c 21 and c 22 These are the first and second parameters in C2, respectively. Both q and p are odd numbers, satisfying 0 21 =6,c 22 =13;

[0085] The inner ring compound reaching law is:

[0086]

[0087] In the formula, Let c1, c2, a, η, k, and q′ be the first derivative of the sliding surface of the inner loop current, defined as the inner loop composite reaching law. c1, c2, a, η, k, and q′ are all constants. Specifically, c1 = 600, a = 2 / 3, η = 3 / 5, k = 30, c2 = 600, q′ = 800, and sat(s u ) for s u The saturation function.

[0088] Step B2: Substitute the current loop state equation of the matching control winding inner loop current subsystem into the designed inner loop current sliding surface, differentiate it, and then combine it with the inner loop composite reaching law to determine the overall control law of the inner loop full-order terminal sliding controller.

[0089] The overall control law of the inner-loop full-order terminal sliding mode controller in step B2 is:

[0090] u′=u′ eq +u′ n

[0091] in,

[0092]

[0093] u' is the overall control law of the inner-loop full-order terminal sliding mode controller, u' eq and u' n These are the equivalent control law and the actual control law of the inner-loop full-order terminal sliding mode controller, respectively. F 00 T is the correlation matrix of the inner loop with respect to the dq-axis current of the control winding. 10 The autocorrelation matrix for controlling the dq-axis current of the winding, where T 10 =R 20 / σ 20 L 20 R 20 To control the initial value of the winding resistance, σ 20 ​To control the combined inductance parameters of the rotor and rotor windings, L 20 To control the initial value of the winding inductance, i 2dq To control the dq-axis current of the winding, K u0 K is an intermediate variable. u0 =ω2L 1r0 L 2r0 [ω2L 1r0 L 2r0 I 2E +ω2(L 2 1r0 -L 10 L r0 )i 1d -R 10 L 1r0 i 1q ], ω2 is the electrical angular frequency on the control winding side, which is also the rotational speed of the dq coordinate system, L 1r0 L represents the initial mutual inductance between the power winding and the rotor winding. 2r0 To control the initial value of the mutual inductance between the rotor winding and the winding, I 2E L represents the amplitude of the control winding current at the equilibrium point. 10 L is the initial value of the self-inductance of the power winding. r0 i is the initial value of the self-inductance of the rotor winding. 1d R is the d-axis current of the power winding. 10 Let i be the initial resistance value of the power winding. 1q Let ΔI be the q-axis current of the power winding. 2ref To control the error of the winding d-axis current reference value, t is the upper bound of integration, τ is the sliding mode surface parameter, and c0, c1, c2, a, η, k0, k', and q' are all constants, and sat(s) i ) for s i The saturation function, s i The sliding surface is the inner loop current. c0 = 600, k0 = 800.

[0094] Furthermore,

[0095] Here, Δ represents the boundary value of the boundary layer, and both k and Δ are constants. Switching control is used outside the boundary layer, while linearization control is used inside the boundary layer.

[0096] Furthermore, the state equation for the current loop is:

[0097]

[0098] Among them, i 2dq To control the dq axis current of the winding, For i 2dq The differential, T10 Let δ2 be the autocorrelation matrix controlling the dq-axis current of the winding, where δ2 is a constant, δ2=1-L 2 2r / (L2L r L2 is the self-inductance of the control winding, L r For the self-inductance of the rotor winding, d 2dq =[d 2d ,d 2q ] T d 2dq To control the d-axis component of the winding current 2d and q-axis component d 2q The cross disturbance vector between, u 2dq To control the dq axis voltage of the winding, u 2dq It also serves as the overall control law for the inner loop full-order terminal sliding mode controller.

[0099] To prove that the output current of the control winding stably tracks the current setpoint i 2dref The Lyapunov function of the inner loop is V² = 0.5S. i 2 Proof is required. Through a series of proofs, it can be concluded that:

[0100]

[0101] In the formula: η2 is a constant greater than 0, η2 is the Lyapunov function stability coefficient of the inner loop, and V2 is the definition of the Lyapunov function of the inner loop current;

[0102] The above equation shows that the state trajectory of the inner loop current subsystem of the matching control winding on the control winding side will, under the action of the overall control law u' of the designed inner loop full-order terminal sliding mode controller, start from any initial state s. i (0)≠0 to reach the ideal sliding surface s i (0) = 0, and then maintain full-order sliding dynamics on the sliding surface, s i (0) represents the inner loop current sliding surface s when time is 0. i The initial value, the total error e of the inner loop current tracking idq Its derivative will be along the sliding surface s i (0) = 0 converges to the symmetric point, when the tracking error e id If its derivative converges to 0 in a finite time, then it satisfies the Lyapunov stability criterion. The output current tracking error of the matched control winding inner loop current subsystem will asymptotically converge to zero. Where, e idq By e id and e iq Composition, e idq For d-axis current tracking error, e iqFor q-axis current tracking error, e id =i 2d -i 2dref e iq =i 2q -i 2qref .

[0103] Principle Analysis: This invention first fully considers environmental changes caused by temperature and frequency variations, leading to matching and non-matching uncertainties in the system. It collects the three-phase stator voltage on the power winding side and the three-phase stator current on the control winding side of the brushless doubly-fed motor; establishes a mathematical model of the brushless doubly-fed motor; calculates the errors in the three-phase stator voltage of the power winding and the three-phase stator current of the control winding; and fully considers internal parameter perturbations during actual operation, establishing an outer-loop subsystem for the power winding stator voltage and an inner-loop subsystem for the control winding stator current. For the designed voltage outer-loop subsystem, a full-order terminal sliding mode controller is designed, and a virtual control law is designed using backstepping to compensate for the non-matching uncertainties in the system. For the designed current inner-loop subsystem, a full-order terminal sliding mode controller is designed, enabling the output current to stably track the given value. This method improves the system's dynamic performance, control accuracy, and robustness to internal parameter perturbations and external disturbances. This invention is applicable to the independent generation control of a brushless doubly-fed motor connected to a balanced load.

[0104] Combination Figure 2 For brushless doubly-fed induction generator (DFIG) independent power generation systems, environmental changes due to temperature and frequency variations lead to matching disturbances such as d in the inner loop current state equation. 2dq That is, the d-axis component of the control winding current. 2d The cross disturbance vector between the q-axis and the d-axis components, as well as the unmatched uncertainty disturbance in the outer loop voltage state equation, specifically the d-axis current i of the control winding. 2d The full-order terminal sliding mode control method for a brushless doubly-fed induction generator independent power generation system proposed in this invention compensates for the matching and non-matching uncertainties in the system, thus offsetting the control error e caused by parameter perturbations. u and e i This improved the output voltage u. 1abc Current i 2abc The precision of its pulsation, i.e., U1-U 1ref i 2d -i 2dref and i 2q -i 2qref Smaller size improves the system's anti-interference capability and robustness under environmental changes. Simultaneously, due to the inherent ability of sliding mode control to accelerate error convergence, the output voltage U1 of the brushless doubly-fed motor system can quickly track the given value U. 1ref The output current i of the brushless doubly fed motor2d i 2q It can quickly track a given value i. 2dref and i 2qref Where U1 is the three-phase output voltage u of the motor. 1abc voltage amplitude, i 2d and i 2q The motor output current i 1abc After coordinate transformation, the d-axis and q-axis components of this invention accelerate the speed of reaching steady state, have better control effect, and significantly improve the quality of output power.

[0105] While the invention has been described herein with reference to specific embodiments, it should be understood that these embodiments are merely examples of the principles and applications of the invention. Therefore, it should be understood that many modifications can be made to the exemplary embodiments, and other arrangements can be designed without departing from the spirit and scope of the invention as defined by the appended claims. It should be understood that different dependent claims and features described herein can be combined in ways different from those described in the original claims. It is also understood that features described in conjunction with individual embodiments can be used in other described embodiments.

Claims

1. A full-order terminal sliding mode control method for a brushless doubly-fed induction generator independent power generation system, characterized in that, include: Based on the mathematical model of the brushless doubly fed motor, considering the matching and non-matching uncertainties caused by parameter perturbation of the brushless doubly fed motor, the voltage loop state equation of the outer loop voltage subsystem of the non-matching power winding and the current loop state equation of the inner loop current subsystem of the matching control winding are determined. For the voltage loop state equation of the unmatched power winding outer loop voltage subsystem, an outer loop full-order terminal sliding mode controller is designed to compensate for the unmatched uncertainty in the independent power generation system of the brushless doubly fed motor, so that the tracking error of the power winding voltage amplitude of the brushless doubly fed motor converges to zero. For the current loop state equation of the matching control winding inner loop current subsystem, an inner loop full-order terminal sliding mode controller is designed to compensate for the matching uncertainty in the independent generator system of brushless doubly fed motor, so that the tracking error of the control winding current amplitude of brushless doubly fed motor converges to zero. The specific process for designing the outer loop full-order terminal sliding mode controller is as follows: Step A1: Based on the voltage loop state equation of the unmatched power winding outer loop voltage subsystem, design the outer loop voltage sliding surface and the outer loop composite reaching law; The outer loop voltage sliding surface is: ； In the formula, is the outer - loop voltage sliding surface, is the error between the amplitude of the output voltage of the power winding and the given value, is a constant, both q and p are odd numbers, satisfying 0 < q / p < 1, q / p is the terminal attractor; the outer - loop composite reaching law is: ； In the formula, Let be the first-order derivative of the sliding mode surface of the outer loop voltage, and define it as the outer loop composite reaching law. , , , , and All are constants. for saturation function; Step A2: Substitute the voltage loop state equation of the unmatched power winding outer loop voltage subsystem into the designed outer loop voltage sliding surface, differentiate it, and then combine it with the outer loop composite reaching law to determine the overall control law of the outer loop full-order terminal sliding controller.

2. The full-order terminal sliding mode control method for a brushless doubly-fed induction generator independent power generation system according to claim 1, characterized in that, The overall control law of the outer loop full-order terminal sliding mode controller in step A2 is: ； in, This is the overall control law for the outer-loop full-order terminal sliding mode controller. and These are the equivalent control law and the actual control law of the outer-loop full-order terminal sliding mode controller, respectively. Let q and p be constants, both of which are odd numbers, satisfying 0 This represents the error between the output voltage amplitude of the power winding and the given value. This is the upper bound of the integral. For sliding surface parameters, , , , , and All are constants. It is the sliding surface of the outer loop voltage.​ 3. The full-order terminal sliding mode control method for a brushless doubly-fed induction generator independent power generation system according to claim 1 or 2, characterized in that, Where Δ is the boundary value of the boundary layer, and both k and Δ are constants.

4. The full-order terminal sliding mode control method for a brushless doubly-fed induction generator independent power generation system according to claim 1, characterized in that, The specific process for designing the inner-loop full-order terminal sliding mode controller is as follows: Step B1: Based on the current loop state equation of the inner loop current subsystem of the matched control winding, design the inner loop current sliding surface and the inner loop composite reaching law. Step B2: Substitute the current loop state equation of the matching control winding inner loop current subsystem into the designed inner loop current sliding surface, differentiate it, and then combine it with the inner loop composite reaching law to determine the overall control law of the inner loop full-order terminal sliding controller.

5. The full-order terminal sliding mode control method for a brushless doubly-fed induction generator independent power generation system according to claim 4, characterized in that, In step B1, the inner loop current sliding surface is: ； Where: is the inner - loop current sliding surface, and e i is the error between the output current of the control winding and the current reference value, \(C_2=\text{diag}(c 21 , c 22 )\), \(C_2\) represents a positive - definite diagonal matrix of the parameters of the designed inner - loop current sliding surface, \(c 21 and \(c 22 are the first parameter and the second parameter in \(C_2\) respectively, both \(q\) and \(p\) are odd numbers, satisfying \(0\lt q / p\lt1\), \(q / p\) is the terminal attractor; the inner - loop composite reaching law is: In the formula, It is the first-order derivative of the sliding mode surface of the inner loop current, and is defined as the inner loop composite reaching law. , , , , and All are constants. for The saturation function.

6. The full-order terminal sliding mode control method for a brushless doubly-fed induction generator independent power generation system according to claim 4, characterized in that, The overall control law of the inner-loop full-order terminal sliding mode controller in step B2 is: in, This is the overall control law for the inner-loop full-order terminal sliding mode controller. and These are the equivalent control law and the actual control law of the inner-loop full-order terminal sliding mode controller, respectively. , This is the correlation matrix of the inner loop with respect to the dq-axis current of the control winding. To control the autocorrelation matrix of the winding dq-axis current, where R 20 To control the initial value of the winding resistance, To control the combined inductance parameters of the rotor and rotor windings, L 20 To control the initial value of the winding inductance, To control the dq axis current of the winding, As an intermediate variable, , To control the electrical angular frequency on the winding side, This represents the original value of the mutual inductance between the power winding and the rotor winding. To control the initial value of the mutual inductance between the rotor winding and the rotor winding, This represents the amplitude of the control winding current at the equilibrium point. This is the initial value of the self-inductance of the power winding. This represents the initial self-inductance of the rotor winding. This represents the d-axis current of the power winding. The initial resistance value of the power winding. This represents the q-axis current of the power winding. To control the error of the winding d-axis current reference value, This is the upper bound of the integral. For sliding surface parameters, , , , , , , and All are constants. for The saturation function, It is the sliding surface of the inner loop current.

7. The full-order terminal sliding mode control method for a brushless doubly-fed induction generator independent power generation system according to claim 5 or 6, characterized in that, Where Δ is the boundary value of the boundary layer, and both k and Δ are constants.

8. The full-order terminal sliding mode control method for a brushless doubly-fed induction generator independent power generation system according to claim 1, characterized in that, The voltage loop state equation is: ； in, This represents the error between the output voltage amplitude of the power winding and the given value. To control the d-axis current of the winding, This serves as the overall control law for the outer-loop full-order terminal sliding mode controller. For custom parameters Parameter perturbation, For bounded values, As an intermediate variable, , To control the electrical angular frequency on the winding side, This represents the original value of the mutual inductance between the power winding and the rotor winding. To control the initial value of the mutual inductance between the rotor winding and the rotor winding, This represents the amplitude of the control winding current at the equilibrium point. This is the initial value of the self-inductance of the power winding. This represents the initial self-inductance of the rotor winding. This represents the d-axis current of the power winding. The initial resistance value of the power winding. This represents the q-axis current of the power winding. To control the error between the winding d-axis current and the reference value, The gain is uncertain, and it satisfies the following boundary conditions: ; The current loop state equation is: ； in, To control the dq axis current of the winding, for The differential, To control the autocorrelation matrix of the winding dq-axis current, It is a constant. , To control the self-inductance of the winding, For the rotor winding self-inductance, , To control the d-axis component of the winding current and q-axis components Cross-disturbance vectors between To control the dq axis voltage of the winding, It also serves as the overall control law for the inner loop full-order terminal sliding mode controller.

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