Metal detector with sampling architecture

Abstract

This invention relates to the field of magnetic detection technology. In particular, this invention relates to a metal detector (100) for detecting at least one metallic object exposed to a magnetic detection field. The invention provides a metal detector using a receiver with a sampling architecture. Specifically, in this type of architecture, the sampling operation is separated from the quantization and encoding operations. In contrast, in conventional architectures, these three operations are typically performed in parallel on an analog-to-digital converter (ADC). In the context of the sampling architecture, sampling is performed directly on the received signal, and most signal processing is performed in discrete time (e.g., by using a switched capacitor system with MOS switches for switching purposes).

Description

Technical Field

[0001] This invention relates to the technical field of magnetic detection. In particular, this invention relates to a metal detector for detecting at least one metallic object exposed to a detection magnetic field. Background Technology

[0002] Metal detectors are primarily used in security fields, such as airports to detect weapons hidden on airplane passengers, military fields for mine clearance, recreational fields for searching for various buried objects, and archaeological fields for searching for antiques.

[0003] Among the most extensive range of metal detectors, there are so-called "passive" metal detectors and so-called "active" metal detectors.

[0004] Passive metal detectors sense and assess deformation of the Earth's magnetic field, while active metal detectors generate an electromagnetic field that induces eddy currents in metals during their transmission phase.

[0005] Before describing examples of active detectors, which are the subject of this invention, we will briefly describe examples of passive metal detectors.

[0006] A magnetometer is a passive detector that does not transmit any signal. Specifically, given that the Earth's magnetic field is inherently homogeneous, it detects breaks in this homogeneity. Therefore, any metallic particle magnetized by the Earth's magnetic field, depending on its size, location, and metallic properties, more or less strongly disrupts this homogeneity. The basic principle of a magnetometer is based on two coils sensitive to the magnetic field and installed in opposite phases to provide a zero-electrical measurement of the magnetic field. If one of these coils encounters a breakdown field, the difference stops at zero, generating a voltage that makes it possible to locate and identify metallic objects.

[0007] Active detectors specifically include beat frequency oscillators (BFO), inductive balance (IB), and pulse induction (PI) detectors.

[0008] In a BFO detector, an oscillator, which is a resonant circuit, is used, with the sensor's coil acting as an inductor. The presence of a metallic object changes the inductance and thus the oscillator's frequency. Analyzing this frequency by mixing it with adjacent frequencies makes it possible to indicate the presence of metal. In this case, the sensor's coil simultaneously functions as both a transmitter and a receiver.

[0009] In detector 1B, at least two coils are used. One is called the transmitter and is powered by a repetitive generator, while the other is the receiver. These two coils are arranged such that their mutual coupling (their mutual inductance) is zero (or as low as possible). Unlike a power transformer, which is arranged such that the coupling between the two coils (primary and secondary) is at its maximum, the receiving coil ideally does not transmit a signal when the transmitting coil is operating. The inductive balance is constructed based on the symmetry property of the magnetic field induced by the coils. Once a metallic object is immersed in this magnetic field, the balance is disrupted, and the receiving coil transmits a signal. This signal is out of phase with respect to the excitation field. Analysis of its phase and / or its gain makes it possible to detect the presence and properties of the metallic object. This detector is widely used due to its advantageous characteristics. It is commonly found among precious metal seekers, beachcombers, and in the construction industry when searching for buried pipes or cable tunnels.

[0010] Finally, the PI detector operates twice. In the first stage, a pulsed magnetic field is induced. This is typically an approximation of a real pulse (Dirac). The metallic portion surrounded by the field is the trajectory of eddy currents. Once the excitation ends, the eddy currents relax (decreasing over time to reach zero). These eddy currents induce a magnetic field, a secondary or reverse induced field, which is measured by the detector during a second stage called the listening stage. During this stage, the voltage induced across the terminals of the coil is extremely low, typically a few mV to less than 1 μV. Typically, the detector implements an averaging mechanism, which accumulates the signal over several pulses to improve the signal-to-noise ratio. The detector transcribes this average value from the audio signal. This detector is characterized by very low sensitivity to ferromagnetic materials (as long as they are poor conductors, such as iron oxide) and better sensitivity to others. A single coil can be used as both transmitter and receiver, or they can be separate. The increased complexity of the peripheral circuitry used to read and interpret the signal makes this sensor more expensive. Applications of this detector include gold mining and mine sweeping.

[0011] In other words, in order to locate and identify metallic objects in a given monitored area, an active metal detector uses a transmitter-receiver system (with one or more coils) that utilizes the physical phenomenon of magnetic induction.

[0012] In practice, the transmitter is configured to emit a time-dependent primary magnetic field in the monitored area via a transmitter coil. The receiver is then configured to measure a secondary magnetic field via a receiver coil, which is generated by eddy currents induced in any metallic object present in the region of influence of the primary magnetic field (also referred to as "remanent magnetization").

[0013] Because of this system, the detection of secondary magnetic fields within the monitored area indicates the presence of at least one metallic object within the monitored area.

[0014] In practice, the detection operation involves filtering, amplifying, demodulating, and identifying the actual received signal at the receiver, which varies as a function of time, and the receiver coil receives the actual received signal.

[0015] Typically, demodulation can be performed in the frequency domain (then using the terms frequency domain or FD detector) or in the time domain (then using the terms time domain or TD detector) for identification.

[0016] In the special case of an FD detector, the transmitter is typically powered by one or more sinusoidal alternating currents to power the transmitter coil, thereby establishing one or more magnetic fields transmitted at one or more transmission frequencies (then referred to as a sinusoidal detector or CW (continuous wave) detector). Therefore, the FD detector receiver performs demodulation by identifying the amplitude and phase of one or more detection frequencies that are substantially equal to the transmission frequency.

[0017] CW receivers can receive signals on one or more detection channels.

[0018] The term "detection channel" should be understood as referring to the frequency range of electromagnetic waves.

[0019] In a known manner, each detection channel has a specific detection frequency f. D (Also known as the carrier frequency), which corresponds to the center frequency of the frequency range of the detection channel.

[0020] In the first example, one or more detection frequencies are selected from the VLF (Very Low Frequency) band.

[0021] The term "VLF signal" refers to the so-called very low frequency band, which is approximately between 3 kHz and 30 kHz.

[0022] In the second example, one or more detection frequencies are selected from the eVLF (extended very low frequency) band.

[0023] The term “eVLF signal” should be understood to refer to a frequency known as extended very low frequency, which is approximately between 20 Hz and 200 kHz.

[0024] Typically, the “bandwidth” of a detection channel refers to the width of the frequency range of the detection channel encompassed between the minimum and maximum frequencies.

[0025] In practice, the bandwidth of a detection channel is called "narrow" because its center frequency is very high compared to the bandwidth of the received signal. For example, in the field of metal detectors, the bandwidth of the detection channel is determined by the relative displacement velocity of the target relative to the detector and rarely exceeds 100 Hz.

[0026] Figure 1The received spectrum of the actual received signal s(t) received by the receiver coil of the metal detector is shown. Figure 1 In the example, the received signal s(t) is received through ten channels CH1 to CH10. Specifically, each channel CH1 to CH10 has a first corresponding detection frequency f. d1 to f d10 and corresponding bandwidth B1 to B 10 In fact, the bandwidth from B1 to B... 10 equal.

[0027] In practice, to facilitate demodulation, the signal is arranged to obtain a received signal with two separate channels (then referred to as an "IQ signal," where I is the in-phase component and Q is the quadrature component). One advantage of this signal form is that it represents the entire spectrum of the received signal without information loss. Furthermore, in a detector, the I and Q components can be used to determine the reactive and resistive portions of a metallic object, and in a way that allows for the identification of the metal that constitutes the object.

[0028] In the example, the two channels of the IQ signal are orthogonal in phase. In practice, the IQ signal is a complex signal represented by the following mathematical form:

[0029] [Mathematical Expression 1]

[0030] IQ signal = A * exp(j * α)

[0031] A is its amplitude and α is its phase.

[0032] Then formula [Mathematical Expression 1] can be decomposed as follows:

[0033] [Mathematical Expression 2]

[0034] IQ signal=A*cos(α)+j*A*sin(α)

[0035] Then, formula [Mathematical Expression 2] can be simplified as follows:

[0036] [Mathematical Expression 3]

[0037] IQ signal=I+jQ

[0038] Where I = A*cos(α) and Q = A*cos(α).

[0039] Finally, the amplitude and phase can be easily recovered using the IQ signal represented by formula [Mathematical Formula 3].

[0040] Specifically, the amplitude is obtained using the following formula:

[0041] [Mathematical Expression 4]

[0042]

[0043] Alternatively, the phase can be obtained using one of the following formulas:

[0044] [Mathematical Expression 5]

[0045]

[0046] Where [·] is the modular function;

[0047] [Mathematical Expression 6]

[0048] α=arg(I+j*Q)=atani2(I,Q)

[0049] Where atan2(·) is a known function that returns the arctangent of the ratio of two variables x and y, expressed in radians, between -π and π (inclusive).

[0050] Of course, it is also conceivable to use a bijective basis different from orthogonal separation, especially one different from... Phase separation.

[0051] In examples of existing CW detectors, the single-frequency real received signal s(t) takes the following mathematical form:

[0052] [Mathematical Expression 7]

[0053] s(t)=A(t)*cos(2×π×f D ×t+α(t)),

[0054] Where A is its amplitude, f D α is its detection frequency, t is time, and α is its phase.

[0055] In the case of multi-frequency received signals, this corresponds to the sum of several single-frequency received signals s(t).

[0056] To simplify the formula [Mathematical Expression 7], 2×π×f is usually used. D Compressed to ω, ω corresponds to the angular pulse of the signal, which is almost time-invariant. In this case, formula [Mathematical Equation 7] can be rewritten as follows:

[0057] [Mathematical Expression 8]

[0058] s(t)=A(t)*cos(ω*t+α(t))

[0059] Next, we extract the I and Q channels of s(t) separately.

[0060] To extract channel I, perform the following operations:

[0061] [Mathematical Expression 9]

[0062]

[0063] Then, applying a low-pass filter to the result of Equation [Mathematical Formula 9] suppresses the frequency 2×f D Signal at the location to obtain

[0064]

[0065] To extract channel Q, perform the following operations:

[0066] [Mathematical Expression 10]

[0067]

[0068] Then, a low-pass filter is applied to the result of formula [Mathematical Equation 10], which suppresses the frequency 2×f. D The signal at the location is obtained:

[0069]

[0070] In a known manner, to simplify computation in digital systems, the different values ​​that the functions cos(·) and sin(·) can take are stored in memory. Multiplication is then performed using a mixer. However, such an architecture consumes a significant amount of memory and computational power.

[0071] Therefore, there is a need for an alternative demodulation solution that does not consume large amounts of memory and computing power. Summary of the Invention

[0072] This invention aims to at least partially satisfy these needs.

[0073] The present invention particularly relates to a metal detector for detecting at least one metallic object exposed to a detection magnetic field.

[0074] Specifically, the metal detector includes:

[0075] - At least one receiving coil, the at least one receiving coil being arranged to receive a magnetic field signal referred to as a received signal s(t), the received signal s(t) representing at least one modification of the detected magnetic field under the influence of the metallic object, the received signal s(t) comprising at least one detection channel of a given bandwidth, each detection channel being associated with a separate detection frequency f. D Related, and

[0076] - At least one sampling architecture receiver coupled to a receiving coil.

[0077] in,

[0078] The sampling architecture receiver includes a first sampling device of a non-quantized type, referred to as the first sampler, which is configured to sample the received signal s(t) or the signals |s(t)| and |s(t)| that are derivatives of the received signal at a first sampling frequency f1 that obeys the Nyquist-Shannon bandwidth sampling theorem. * |、|s'[t]|、|s′[t] * |、|s′ + [t]| is sampled, so that

[0079] - Transmit a discrete-time signal called the first sampled signal s[t], which corresponds in the frequency domain to a copy of the received signal s(t) or a copy of the signal s′[t], which is the derivative of the received signal, and

[0080] - Convert the frequency of the received signal s(t) to a frequency lower than the detection frequency f. D The first intermediate frequency.

[0081] In a first embodiment, the sampling architecture receiver further includes a real-to-complex conversion device coupled downstream of the first sampler, and the real-to-complex conversion device is configured to convert the first sampled signals s[t] and s”[t] into a complex signal s[t] having an in-phase component I[t] and a quadrature component Q[t]. * 、s”[t] * .

[0082] In this first embodiment, the first sampling frequency f1 follows the following relationship:

[0083]

[0084] Where f D is the detection frequency of the detection channel for the received signal s(t), n is a natural integer and f1 is the first sampling frequency.

[0085] Furthermore, in this first embodiment, the real-to-complex conversion device includes a first discrete-time mixer, a second discrete-time mixer, and a discrete-time low-pass filter, wherein the discrete-time low-pass filter is coupled as follows:

[0086] - Downstream of the first discrete-time mixer, so as to transmit the in-phase component I[t], and

[0087] - Downstream of the second discrete-time mixer, so as to transmit the quadrature component Q[t].

[0088] In a second embodiment, the sampling architecture receiver further includes a continuous-time anti-aliasing filter coupled upstream of the first sampler, the continuous-time anti-aliasing filter having a bandwidth configured to be centered around at least one of the detection frequencies associated with the received signal s(t), so as to limit the bandwidth of the received signal s(t) around at least one of the detection frequencies and pass the filtered received signal |s(t)|.

[0089] In the third embodiment, the sampling architecture receiver further includes,

[0090] - A non-quantized second sampling device, coupled upstream of the first sampler, and

[0091] - A discrete-time anti-aliasing filter, which is coupled between the first sampler and the second sampler.

[0092] Furthermore, in this third embodiment,

[0093] The second sampler is configured to sample the received signal s(t) at a second sampling frequency f2 following the Nyquist-Shannon bandwidth sampling theorem, so as to transmit a signal called the second sampled signal s′[t], which is a derivative of the received signal, and the second sampled signal corresponds in the spectral domain to a copy of the received signal s(t).

[0094] - The discrete-time anti-aliasing filter has a bandwidth configured to be centered on at least one of the detection frequencies associated with the received signal s(t) so as to limit the bandwidth of the second sampled signal s′[t] around at least one of the detection frequencies and to pass the sampled and filtered received signal |s'[t]|.

[0095] Finally, in this third embodiment, the second sampling frequency f2 is further configured to transform the frequency of the received signal s(t) to a value greater than the first intermediate frequency and less than or equal to the detection frequency f. D The second intermediate frequency.

[0096] In an embodiment of the third embodiment, the metal detector further includes a non-quantization type oversampling device, referred to as the oversampler, coupled between the second sampler and the discrete-time anti-aliasing filter.

[0097] In specific embodiments of the second and third embodiments, the continuous-time anti-aliasing filter and / or discrete-time anti-aliasing filter are complex bandpass type filters configured to convert the received signal s(t) or the signal s′[t], which is the derivative of the received signal, into a filtered complex signal |s(t) having an in-phase component I[t] and a quadrature component Q[t].* |、|s′[t] * |

[0098] In the fourth embodiment, the sampling architecture receiver further includes a quantizer and an encoder, the quantizer quantizing the first sampled signal s[t] and / or the second sampled signal s′[t] in such a way as to transmit a quantized signal, and the encoder being configured to encode the quantized signal in such a way as to transmit a digital signal.

[0099] In the fifth embodiment, one or more detection frequencies are selected from the VLF band and the eVLF band.

[0100] In a sixth embodiment, the metal detector further includes at least one transmitting coil arranged to generate the detection magnetic field.

[0101] In the seventh embodiment, the metal detector further includes a plurality of sampling architecture receivers, wherein each sampling architecture receiver is configured to operate at a specific detection frequency f. D Process the received signal s(t).

[0102] In the eighth embodiment, the metal detector further includes a component for automatically or under user command to select one or more detection frequencies.

[0103] In the ninth embodiment, the metal detector further includes,

[0104] - Support rod,

[0105] - A detection head, disposed at one end of the support rod and configured to accommodate the receiving coil and / or the transmitting coil, and

[0106] - At least one processor, coupled to the detection head and configured to manage the operation of the detection head and utilize information from the detection head. Attached Figure Description

[0107] Other features and advantages of the invention will be better understood by reading the following description and referring to the accompanying drawings, which are given by way of illustration rather than limitation.

[0108] [ Figure 1 ] Figure 1 The spectrum of the actual received signal at two receiving frequencies is shown.

[0109] [ Figure 2 ] Figure 2 A metal detector is shown.

[0110] [ Figure 3 ] Figure 3 The invention is shown Figure 2The receiver of the metal detector.

[0111] [ Figure 4 ] Figure 4 A diagram illustrating the Nyquist-Shannon bandwidth sampling theorem is shown.

[0112] [ Figure 5 ] Figure 5 It shows Figure 4 The receiver is a real-to-complex number conversion device.

[0113] [ Figure 6 ] Figure 6 It shows Figure 2 An example of a receiver for a metal detector.

[0114] [ Figure 7 ] Figure 7 It shows Figure 6 Variations of the embodiments.

[0115] [ Figure 8 ] Figure 8 It shows Figure 2 Another embodiment of the receiver of the metal detector.

[0116] [ Figure 9 ] Figure 9 It shows Figure 8 The first variation of the embodiment.

[0117] [ Figure 10 ] Figure 10 It shows Figure 8 The second variation of the embodiment.

[0118] [ Figure 11 ] Figure 11 It shows Figure 8 The third variation of the embodiment.

[0119] In the various accompanying drawings, dashed lines and arrows indicate optional elements, steps, and sequences. Detailed Implementation

[0120] One of the objectives of this invention is to provide a metal detector that offers an alternative demodulation solution that does not consume too much memory and computing power.

[0121] To this end, the inventors provide a metal detector that uses a sampling architecture receiver in which frequency transformation is performed by subsampling, which uses the Nyquist-Shannon bandwidth sampling theorem to controllably utilize spectral aliasing (meaning it does not seek to avoid spectral aliasing).

[0122] Therefore, unlike existing technologies, it is not necessary to use a mixer (i.e., to perform the multiplication of the received signal with the local oscillator) to perform frequency conversion.

[0123] However, it is known that using mixers in receiver architectures consumes a significant amount of memory and / or computing power.

[0124] In particular, in the sampling architecture receiver, the sampling operation is separated from the quantization and encoding operations. In contrast, in conventional architectures, these three operations are typically performed in parallel within the analog-to-digital converter (or ADC).

[0125] In the context of a sampling architecture, sampling is performed directly on the received signal, and most of the signal processing is done in discrete time (e.g., using a switched capacitor system with MOS switches for switching).

[0126] The term "discrete time" refers to a field that represents a value of time defined by a countable set within a finite time window. For the sake of simplicity in this document, the discrete time period will be assumed to be fixed.

[0127] In this specification, the metal detector according to the present invention will be considered as an FD detector.

[0128] Therefore, the present invention relates to a metal detector for detecting at least one metallic object.

[0129] In this invention, the metal object is exposed to a detection magnetic field, such as a detection magnetic field generated by the transmitting coil of a metal detector.

[0130] In particular, the detection magnetic field can magnetize metallic objects.

[0131] In this invention, the metal detector includes at least one receiving coil and at least one receiver coupled to the receiving coil.

[0132] Therefore, the present invention also covers a metal detector comprising two or more receiving coils and / or two or more receivers.

[0133] Of course, the present invention also includes a multi-channel system (i.e., multi-frequency) that includes several receivers according to the present invention.

[0134] Therefore, this invention covers multi-channel metal detectors using two, three, four, five or more detection frequencies.

[0135] Figure 2 A metal detector 100 is shown, which includes a receiving coil 110 and a sampling architecture receiver 120.

[0136] Specifically, a receiving coil 110 of a known type is arranged to receive magnetic field signals, referred to as receiving signals.

[0137] Specifically, the receiving coil 110 is configured to sense the residual magnetization of the metal object after the transmitting coil (not shown) has magnetized the metal object.

[0138] Figure 2 A receiving coil 110 is shown, which receives the received signal s(t), where t represents time.

[0139] In fact, the received signal s(t) is a continuous-time real analog signal.

[0140] The term "continuous time" should be understood as a field representing the value of time as defined by an uncountable set within a finite time window.

[0141] Furthermore, the received signal s(t) represents at least one modification of the detected magnetic field under the influence of the metallic object.

[0142] In other words, the metallic object modifies the detected magnetic field, and the received signal s(t) modulates this modification of the magnetic field.

[0143] In a known manner, the received signal s(t) is amplified by a low-noise amplifier (LNA) before being passed to the sampling architecture receiver 120.

[0144] In this invention, the sampling architecture receiver 120 includes a first sampling device of non-quantizer type, referred to as the first sampler.

[0145] Figure 3 A sampling architecture receiver 120 including a first sampler 121 is shown.

[0146] In fact, the first sampler 121 is a sampling device of a known type, but it does not necessarily include quantization or encoding operations.

[0147] For example, the first sampler 121 can be a sample and hold circuit without quantization and encoding operations.

[0148] In this invention, the first sampler 121 is configured to sample the received signal s(t) at a first sampling frequency that follows the Nyquist-Shannon bandwidth sampling theorem (also known as bandpass sampling, undersampling, subsampling, downsampling, sub-Nyquist, super-Nyquist, or harmonic sampling).

[0149] As a reminder, the Nyquist-Shannon bandwidth sampling theorem is an extension of the "classical" Nyquist-Shannon theorem applied to narrowband signals. That is, sampling is performed as a function of bandwidth, rather than as a function of the maximum frequency of the received signal s(t).

[0150] Figure 4 It shows the definition used in f L and f HThis theorem relates to the bandwidth of signals between [the two sides].

[0151] In fact, this theorem shows that if the sampling frequency f of the signal... S At least twice the width of the signal bandwidth (i.e., f) L with f H If the frequency band between the baseband and intermediate frequency (IF) is used, the signal can be reconstructed. Therefore, according to this theorem, a signal can be intentionally aliased so that at least one of its aliased frequencies is then transformed to the intermediate frequency or baseband. In practice, the useful signal is transformed to the baseband and half the sampling frequency (i.e., ...). The intermediate frequency F between ) i .

[0152] In other words, the Nyquist-Shannon bandwidth sampling theorem makes it possible to transform or demodulate a narrowband signal around a given frequency by sampling at a frequency lower than the given frequency, provided that the sampling frequency is at least twice the bandwidth of the real signal and at least once the bandwidth of the complex signal. This is because the spectrum of such a sampled signal is found to repeat without superposition around multiples of the sampling frequency. Among these aliasings, aliasing located in or near the baseband makes it possible to demodulate a bandwidth signal without using a mixer.

[0153] By extension, note that in the case of this invention using the Nyquist-Shannon bandwidth sampling theorem, the sampling frequency f S It must be less than the detection frequency f D Used for replicas to undergo frequency transformations.

[0154] In the first specific implementation, the following conditions are met: f S ≤2×f D .

[0155] In a first aspect of the first specific implementation, the following conditions are met: f S ≤f D .

[0156] In a second aspect of the first specific implementation, the following conditions are met: f S <f D

[0157] In the second specific implementation, the following conditions are met: quantity The value is not an integer.

[0158] In one aspect of the second specific implementation, the following conditions are met: quantity The value is not an integer.

[0159] Therefore, in practice, the Nyquist-Shannon bandwidth sampling theorem must be obeyed in this invention, but most importantly, the "classical" Nyquist-Shannon sampling theorem cannot be obeyed. This is because if the "classical" Nyquist-Shannon sampling theorem were obeyed, frequency transformation would be impossible.

[0160] In the context of this invention, Figure 3 A first sampler 121 using a sampling frequency f1 is shown, which is selected to comply with the Nyquist-Shannon bandwidth sampling theorem for real signals as described above.

[0161] In particular, the selection of the first sampling frequency f1 allows the first sampler 121 to transmit a discrete-time signal, called the first sampled signal, which corresponds to a copy of the received signal s(t) in the frequency domain.

[0162] Therefore, in this invention, the sampling frequency f1 is strictly less than the detection frequency f D .

[0163] In fact, the copied signal includes the spectral copy of the received signal s(t).

[0164] Figure 3 A first sampler 121 is shown, which transmits the first sampled signal s[t] in response to the reception of the received signal s(t) and as a function of the first sampling frequency f1.

[0165] In fact, the first sampled signal s[t] is a discrete-time analog sampled signal. As mentioned above, such a signal has not been quantized but only sampled. In contrast, in conventional architectures, analog signals are maintained on a continuous-time basis until they are quantized in the analog-to-digital converter.

[0166] Furthermore, for each detection channel, the selection of the first sampling frequency f1 allows the first sampler 121 to transform the frequency of the received signal s(t) to a frequency lower than the detection frequency f of the detection channel. D The first intermediate frequency.

[0167] In fact, this first intermediate frequency can be found in the baseband.

[0168] In a first specific embodiment of the invention, signal processing techniques using a flexible and configurable first sampled signal s[t] may be advantageous.

[0169] For this purpose, the sampling architecture receiver 120 also includes a processing unit (not shown) for the first sampled signal s[t] using discrete-time analog signal processing techniques.

[0170] For example, a switched capacitor system with a MOS switch for switching can be used.

[0171] In a second specific embodiment of the invention, it may be advantageous, for example, to digitally process the first sampled signal s[t] using a processor.

[0172] For this purpose, the sampling architecture receiver 120 also includes a quantizer (not shown) and an encoder (not shown).

[0173] Specifically, the quantizer coupled downstream of the first sampler 121 is configured to quantize the first sampled signal s[t] in such a way as to transmit the quantized signal. Furthermore, the encoder coupled downstream of the quantizer is configured to encode the quantized signal in such a way as to transmit a digital signal.

[0174] Of course, it is conceivable to combine the quantizer and encoder in a single device, and doing so would not require any substantial modification to the invention.

[0175] In a third specific embodiment of the invention, it is advantageous to convert the first sampled signal s[t] into a complex signal of type IQ signal, and this is done to simplify its processing. In this case, the sampled signal s[t] will preferably be at an intermediate frequency.

[0176] Therefore, such as Figure 3 As shown, the sampling architecture receiver 120 also includes a real-to-complex conversion device 122 coupled downstream of the first sampler 121.

[0177] Specifically, the real-to-complex conversion device 122 is configured to convert the first sampled signal s[t], which is a real signal, into a complex signal of the type IQ signal as described above.

[0178] exist Figure 3 In this process, the real-to-complex conversion device 122 transmits the complex signal s[t] in response to the reception of the first sampled signal s[t]. * Specifically, the complex signal s[t] * It includes the in-phase component I[t] and the quadrature component Q[t].

[0179] Of course, it can be specified that the real-to-complex conversion device 122 only transmits the in-phase component I[t] and omits the quadrature component Q[t].

[0180] In a third specific embodiment of the invention, it may be advantageous to use a simple and efficient structure for the real-to-complex conversion device 122.

[0181] Therefore, such as Figure 5 As shown, the real-to-complex conversion device 122 includes a first discrete-time mixer 1221, a second discrete-time mixer 1222, and at least one discrete-time low-pass filter 1223.

[0182] In practice, the first discrete-time mixer 1221 and the discrete-time low-pass filter 1223 are configured to transmit the complex signal s[t] for each sample of the first sampled signal s[t]. * The in-phase component I[t]. The second discrete-time mixer 1222 and the discrete-time low-pass filter 1223 are configured to transmit the complex signal s[t] for each sample of the first sampled signal s[t]. * The orthogonal component Q[t].

[0183] Of course, it is possible to specify the use of two low-pass filters instead of a single low-pass filter. In this case, the first discrete-time low-pass filter can be coupled downstream of the first discrete-time mixer 1221 to obtain the complex signal s[t]. * The in-phase component I[t]. Furthermore, a second discrete-time low-pass filter can be coupled downstream of the second discrete-time mixer 1222 to obtain the complex signal s[t]. * The orthogonal component Q[t].

[0184] Furthermore, in an embodiment of the third specific embodiment of the invention, in order to reduce the resources required at the processor level, it may be advantageous to optimize the ratio of the first sampling frequency f1 to the bandwidth so that the lowest possible first sampling frequency f1 can be selected.

[0185] Therefore, it is preferable to choose a sampling frequency f1 such that it conforms to the following relationship:

[0186] [Mathematical Expression 11]

[0187]

[0188] f D f1 is the detection frequency of the detection channel for the received signal s(t), n' is a natural integer, f1 is the first sampling frequency, and n is a natural integer.

[0189] In the first example of formula [Mathematical Expression 11], the following relationship can be used:

[0190] [Mathematical Expression 12]

[0191]

[0192] In this case, the term "quarter-band" configuration can be used.

[0193] In the second example of formula [Mathematical Expression 11], the following relationship can be used:

[0194] [Mathematical Expression 13]

[0195]

[0196] In this case, the term "three-quarters band" configuration can be used.

[0197] By selecting the sampling frequency f1 according to the arrangement in Formula [Mathematical Equation 11], a first discrete-time mixer and a second discrete-time mixer can be used, utilizing only the multiplication coefficients in the following factors, to obtain the complex signal s[t]. * The in-phase component I[t] and the quadrature component Q[t]:

[0198] -1, 0, and 1.

[0199] Furthermore, one can envision performing additional optimizations by utilizing, for example, the property of zero multiplicative factors of two samples. This is especially true for certain finite response filter (FIR) topologies.

[0200] Furthermore, other real-to-complex conversion techniques, such as the Hilbert transform, can be used without requiring substantial modifications to this invention.

[0201] In a fourth specific embodiment of the invention, it may be advantageous to reduce the amount of aliasing noise during sampling performed by the first sampler 121. Furthermore, if the transmitter of the metal detector 100 transmits several frequencies simultaneously, it may be advantageous to isolate the signal of the detection channel from other detection channels.

[0202] For this purpose, the sampling architecture receiver 120 also includes a continuous-time anti-aliasing filter.

[0203] Figure 6 A continuous-time anti-aliasing filter 123 coupled upstream of the first sampler 121 is shown.

[0204] Specifically, the continuous-time anti-aliasing filter 123 is configured to transmit a filtered received signal |s(t)| in response to the reception of the received signal s(t). In this case, the first sampler 121 transmits a first sampled signal s[t] in response to the reception of the filtered received signal |s(t)|, and the filtered received signal |s(t)| is therefore the derivative of the received signal s(t).

[0205] Furthermore, the continuous-time anti-aliasing filter 123 has a bandwidth that is configured to be centered on at least one of the detection frequencies associated with the received signal s(t) so as to limit the bandwidth of the received signal s(t) to around at least one of the detection frequencies.

[0206] In the first specific embodiment of the fourth specific embodiment of the present invention ( Figure 6In (not shown), it may be specified to add a real-to-complex conversion device 122 downstream of the first sampler 121 to convert the first sampled signal s[t] into a complex signal of type IQ signal, and to do so in order to facilitate its processing.

[0207] In a second specific embodiment of the fourth specific embodiment of the present invention, the above-described real-to-complex conversion device 122 can be advantageously omitted.

[0208] For this purpose, a continuous-time anti-aliasing filter 123 can be selected, making it a complex bandpass type. Therefore, in addition to performing filtering operations, such a filter is configured to filter out negative frequencies. The signal transformed by the first sampler 121 can then be transformed into a complex signal of the IQ signal type as described above. Specifically, there will be no aliasing of the inherent negative frequencies of the real signal.

[0209] Figure 7 A continuous-time anti-aliasing filter 123 of the complex bandpass type is shown, which transmits a filtered complex received signal |s(t) in response to the reception of the received signal s(t). * The filtered complex received signal has an in-phase component I(t) and a quadrature component Q[t].

[0210] In addition, Figure 7 In this configuration, the first sampler 121 is configured to process the filtered complex received signal |s(t) * |Sampling is performed to transmit a complex sampled signal |s(t) that includes the in-phase component I(t) and the quadrature component Q[t]. * |

[0211] In a fifth specific embodiment of the invention, simplifying the configuration of the bandwidth filter may be advantageous, particularly when having a configurable detection frequency f. D In systems and / or in multi-channel (i.e., multi-frequency) systems.

[0212] To this end, the sampling architecture receiver 120 also includes a non-quantized type second sampling device, a second sampler, and a discrete-time anti-aliasing filter.

[0213] Figure 8 A discrete-time anti-aliasing filter 125 coupled between a first sampler 121 and a second sampler 124 is shown.

[0214] In fact, the second sampler 124 is configured to sample the received signal s(t) at a second sampling frequency that obeys the Nyquist-Shannon bandwidth sampling theorem, as described above with reference to the first sampling frequency f1.

[0215] In a particular embodiment, in a multi-channel (i.e., multi-frequency) system, the second sampler 124 is shared by two receiving channels.

[0216] Figure 8 A second sampler 124 using a sampling frequency f2 is shown, which is selected to comply with the Nyquist-Shannon bandwidth sampling theorem as described above.

[0217] In particular, the selection of the second sampling frequency f2 allows the second sampler 124 to transmit a discrete-time signal, called the second sampled signal, which corresponds to a copy of the received signal s(t) in the frequency domain.

[0218] Therefore, in this invention, the sampling frequency f2 satisfies the following condition: f2 < 2 × f D Strictly less than the detection frequency f D .

[0219] In fact, the copied signal includes the spectral copy of the received signal s(t).

[0220] Figure 8 A second sampler 124 is shown, which responds to the reception of the received signal s(t) and transmits the second sampled signal s′[t] as a function of the second sampling frequency f2.

[0221] In fact, the second sampled signal s′[t] is a discrete-time analog sampled signal, which is the derivative of the received signal s(t). As mentioned above, such a signal has not been quantized but only sampled. In contrast, in conventional architectures, analog signals are maintained on a continuous-time basis until quantization in the analog-to-digital converter.

[0222] Furthermore, for each detection channel, the selection of the second sampling frequency f2 allows the second sampler 124 to transform the frequency of the received signal s(t) to a value greater than the first intermediate frequency and less than or equal to the detection frequency f. D The second intermediate frequency.

[0223] Furthermore, the discrete-time anti-aliasing filter 125 is configured to transmit the sampled and filtered received signal |s'[t]| in response to the reception of the second sampled signal s'[t]. Therefore, the first sampler 121 transmits the first sampled signal s”[t] in response to the reception of the second sampled and filtered signal |s'[t]|, where the second sampled and filtered signal |s'[t]| is the derivative of the received signal s(t).

[0224] Furthermore, the discrete-time anti-aliasing filter 125 has a bandwidth that is configured to be centered on at least one of the detection frequencies associated with the received signal s(t) so as to limit the bandwidth of the second sampled signal s′[t] around said at least one detection frequency.

[0225] In a first embodiment of the fifth specific embodiment of the invention, it may be advantageous to convert the first sampled signal s”[t] into a complex signal of the IQ signal type in order to simplify its processing.

[0226] Therefore, such as Figure 9 As shown, the sampling architecture receiver 120 also includes the above-mentioned... Figure 2 The real-to-complex conversion device 122 is coupled downstream of the first sampler 121.

[0227] Specifically, the real-to-complex conversion device 122 is configured to convert the first sampled signal s”[t], which is a real signal, into a complex signal of the type IQ signal as described above.

[0228] exist Figure 9 In this process, the real-to-complex conversion device 122 transmits the complex signal s″[t] in response to the reception of the first sampled signal s”[t]. * Specifically, the complex signal s″[t] * It includes the in-phase component I[t] and the quadrature component Q[t].

[0229] Of course, it can be specified that the real-to-complex conversion device 122 only transmits the in-phase component I(t) and omits the quadrature component Q[t].

[0230] In the second embodiment of the fifth specific embodiment of the present invention, omitting the above-described real-to-complex conversion device 122 may be advantageous.

[0231] For this purpose, a discrete-time anti-aliasing filter 125 can be selected, making it a complex bandpass type. Therefore, in addition to performing filtering operations, such a filter is configured to filter out negative frequencies. The signal transformed by the second sampler 124 can then be transformed into a complex signal of the IQ signal type as described above. Specifically, aliasing of the inherent negative frequencies of the real signal will be eliminated.

[0232] Figure 10 Including with Figure 8 The same components are used, and a discrete-time anti-aliasing filter 125 of the complex bandpass type is shown, which, in response to the reception of the second sampled signal s′[t], transmits the sampled and filtered complex received signal |s′[t]. * The complex received signal has an in-phase component I[t] and a quadrature component Q[t].

[0233] In addition, Figure 10 In this configuration, the first sampler 121 is configured to process the sampled and filtered complex received signal |s′[t]. * Sampling is performed to transmit a complex sampled signal s″[t] comprising the in-phase component I[t] and the quadrature component Q[t]. * .

[0234] In a third embodiment of the fifth specific embodiment of the invention, it may be advantageous to perform frequency conversion by a fraction of the first sampling frequency f1 rather than a multiple of f1.

[0235] For this purpose, the sampling architecture receiver 120 also includes a non-quantized type of oversampling device, referred to as an oversampler.

[0236] Figure 11 Including with Figure 8 The same components are used, and an oversampler 126 coupled between the second sampler 124 and the anti-aliasing filter 125 is shown.

[0237] The oversampler 126 makes it possible to boost the second sampled signal s′[t] to a third sampling frequency by interpolation and decimation, based on a given oversampling factor.

[0238] In practice, for an oversampling factor N, the oversampler 126 inserts (N-1) new samples with a value of zero between each sample of the second sampled signal s′[t]. Therefore, the oversampler 126 outputs a signal with a frequency N times higher than that of the second sampled signal s′[t] and a Fourier transform consisting of N aliases of the Fourier transform of the input signal.

[0239] In particular, Figure 11 In the process, oversampler 126 (which has an oversampling factor N) receives the second sampled signal s′[t] and transmits an oversampled version s′ of the second sampled signal s′[t]. + [t].

[0240] Furthermore, the discrete-time anti-aliasing filter 125 is configured to transmit the sampled and filtered received signal |s′ in response to the reception of the second sampled signal s′[t]. + [t]|. Subsequently, the first sampler 121 responds to the second sampled and filtered signal |s′ + [t]| is received, while the first sampled signal s″ is transmitted. + [t], the second sampled and filtered signal |s′ + [t]| is the derivative of the received signal s(t).

[0241] The present invention has been described and illustrated. However, the invention is not limited to the forms of the described embodiments. Therefore, those skilled in the art can deduce other variations and embodiments upon reading the specification and drawings.

[0242] In the first embodiment, the metal detector 100 further includes at least one transmitting coil arranged to generate a detection magnetic field.

[0243] In the second embodiment, the metal detector includes a plurality of sampling architecture receivers 120, wherein each sampling architecture receiver 120 is configured to operate at a specific detection frequency f. D Process the received signal s(t).

[0244] In an example of the second embodiment, the metal detector also includes components for automatically or under user command to select one or more detection frequencies.

[0245] In the third embodiment, the metal detector further includes,

[0246] - Support rod,

[0247] - A detection head, disposed at one end of the support rod and configured to accommodate the receiving coil and / or the transmitting coil, and

[0248] - At least one processor, coupled to the detection head and configured to manage the operation of the detection head and utilize information from the detection head.

[0249] In the first example of the third embodiment, the metal detector includes one or more processors.

[0250] In a second example of the third embodiment, the metal detector also includes an electronic control unit to house the processor and is positioned at the other end of the support rod.

[0251] In a third example of the third embodiment, the metal detector further includes an operator headset coupled to the processor and configured to send a detection audio signal to the operator indicating that a metal object has been detected.

[0252] The present invention can be the subject of many variations and applications beyond those described above. In particular, unless otherwise stated, the different structural and functional features of each of the above embodiments should not be considered as combinations and / or tightly and / or inseparably linked together, but rather as simple juxtapositions. Furthermore, the structural and / or functional features of the different embodiments described above can be wholly or partially the subject of any different juxtaposition or any different combination.

Claims

1. A metal detector (100) for detecting at least one metallic object exposed to a detection magnetic field, said metal detector (100) comprising: - At least one receiving coil (110) is arranged to receive a magnetic field signal called a received signal s(t), the received signal s(t) representing at least one modification of the detected magnetic field under the influence of the metallic object, the received signal s(t) comprising at least one detection channel of a given bandwidth, each detection channel being associated with a separate detection frequency f. D Related, and - At least one sampling architecture receiver (120) coupled to the receiving coil (110), in, The sampling architecture receiver (120) includes a non-quantized type first sampling device called a first sampler (121), which is configured to sample the received signal s(t) or the signal |s(t)|, |s(t) being the derivative of the received signal at a first sampling frequency f1 that obeys the Nyquist-Shannon bandwidth sampling theorem. * |、|s′[t]|、|s′[t] * |、|s′ + [t]| Sample, s′ + [t] is an oversampled version of the signal s′[t], making - Transmit a discrete-time signal called the first sampled signal s[t], which corresponds in the frequency domain to a copy of the received signal s(t) or a copy of the signal s′[t] that is the derivative of the received signal, and - Convert the frequency of the received signal s(t) to a frequency lower than the detection frequency f. D The first intermediate frequency.

2. The metal detector (100) according to claim 1, wherein, The sampling architecture receiver (120) further includes a real-to-complex conversion device (122) coupled downstream of the first sampler (121), and configured to convert the first sampled signals s[t] and s″[t] into a complex signal s[t] having an in-phase component I[t] and a quadrature component Q[t]. * 、s″[t] * .

3. The metal detector (100) according to claim 2, wherein the first sampling frequency f1 follows the following relationship: Where f D f1 is the detection frequency of the detection channel for the received signal s(t), where n is a natural integer and f1 is the first sampling frequency. And among them, The real-to-complex conversion device (122) includes a first discrete-time mixer (1221), a second discrete-time mixer (1222), and a discrete-time low-pass filter (1223), the discrete-time low-pass filter (1223) being coupled as follows: Downstream of the first discrete-time mixer (1221), in order to transmit the in-phase component I[t], and - Downstream of the second discrete-time mixer (1222), so as to transmit the quadrature component Q[t].

4. The metal detector (100) according to claim 1, wherein, The sampling architecture receiver (120) also includes a continuous-time anti-aliasing filter (123) coupled upstream of the first sampler (121), the continuous-time anti-aliasing filter (123) having a bandwidth configured to center around at least one of the detection frequencies associated with the received signal s(t), so as to limit the bandwidth of the received signal s(t) around at least one of the detection frequencies and pass the filtered received signal |s(t)|.

5. The metal detector (100) according to claim 1, wherein, The sampling architecture receiver (120) also includes, - A non-quantized second sampling device, the second sampler (124) being coupled upstream of the first sampler (121), and - A discrete-time anti-aliasing filter (125) is coupled between the first sampler (121) and the second sampler (124). in, - The second sampler (124) is configured to sample the received signal s(t) at a second sampling frequency f2 following the Nyquist-Shannon bandwidth sampling theorem, so as to transmit a signal called the second sampled signal s′[t], which is a derivative of the received signal, and the second sampled signal s′[t] corresponds in the spectral domain to a copy of the received signal s(t), and The discrete-time anti-aliasing filter (125) has a bandwidth configured to be centered on at least one of the detection frequencies associated with the received signal s(t), so as to limit the bandwidth of the second sampled signal s′[t] around at least one of the detection frequencies and to transmit the sampled and filtered received signal |s′[t]|. Furthermore, the second sampling frequency f2 is configured to convert the frequency of the received signal s(t) to a value greater than the first intermediate frequency and less than or equal to the detection frequency f. D The second intermediate frequency.

6. The metal detector (100) according to claim 5 further includes a non-quantized type oversampling device called an oversampler (126), said oversampler (126) being coupled between the second sampler (124) and the discrete-time anti-aliasing filter (125).

7. The metal detector (100) according to any one of claims 4 to 6, wherein, The continuous-time anti-aliasing filter (123) and / or the discrete-time anti-aliasing filter (125) are complex bandpass type filters configured to convert the received signal s(t) or the signal s′[t] which is the derivative of the received signal into a filtered complex signal |s(t) with in-phase component I[t] and quadrature component Q[t]. * |、|s′[t] * | 8. The metal detector (100) according to any one of claims 1 to 6, wherein, The sampling architecture receiver (120) further includes a quantizer and an encoder, the quantizer quantizing the first sampled signal s[t] and / or the second sampled signal s′[t] in such a way as to transmit the quantized signal, and the encoder being configured to encode the quantized signal in such a way as to transmit the digital signal.

9. The metal detector (100) according to any one of claims 1 to 6, wherein, The one or more detection frequencies are selected from the VLF band and the eVLF band.

10. The metal detector (100) according to any one of claims 1 to 6 further includes at least one transmitting coil arranged to generate the detection magnetic field.

11. The metal detector (100) according to any one of claims 1 to 6, comprising a plurality of sampling architecture receivers (120), wherein each sampling architecture receiver (120) is configured to operate at a specific detection frequency f D The received signal s(t) is processed.

12. The metal detector (100) of claim 11 further includes a component for automatically or under user command selecting one or more detection frequencies.

13. The metal detector (100) according to any one of claims 1 to 6 further comprises: - Support rod, - A detection head, disposed at one end of the support rod and configured to accommodate the receiving coil and / or transmitting coil, and - At least one processor, coupled to the detection head and configured to manage the operation of the detection head and utilize information from the detection head.

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