Overlap band calibration method for broadband interleaved acquisition system with fixed phase compensation position

Abstract

The present invention belongs to the field of digital communication, and specifically relates to an overlap band calibration method for a broadband interleaved acquisition system adapted to a fixed phase compensation position. The specific technical scheme is: constructing a linear phase compensation module and a nonlinear phase compensation module, and building and solving a fitting coefficient optimization function of the optimized overlap band compensation module. The present invention changes the value of the optimization objective function so that there is no need to consider the overall trend of the fitting phase difference, thereby avoiding the defect that the traditional method has an unsatisfactory compensation effect in some cases, and can achieve effective phase compensation of the overlap band phase difference under the condition of a fixed compensation position.

Description

Technical Field

[0001] The invention belongs to the field of digital communication, and in particular relates to an overlap band calibration method for a broadband interleaved acquisition system with an adaptive fixed phase compensation position. Background Art

[0002] High-speed data acquisition systems play an indispensable role in modern electronic information systems and are essential in fields such as wireless communication measurement, aerospace, and industrial medical instruments. However, due to device process limitations, the highest indicators of single-chip ADCs in mature commercial products are 10.4GSa / s sampling rate, 8GHz bandwidth, and 12-bit effective bits, which are difficult to meet the indicator requirements of high sampling rate and large bandwidth acquisition systems. Wideband interleaved sampling technology can break through the limitations of single-chip ADCs and achieve higher sampling rates and larger bandwidths. This technology decomposes the acquisition system into multiple sub-bands, performs frequency division on different sub-bands, and then down-converts them to low frequency bands. Low-performance ADC chips are used for sampling, and then interpolation filtering and up-mixing are used to achieve distortion-free sampling of the signal.

[0003] Although broadband interleaved sampling technology can improve bandwidth and sampling rate at the same time, the non-ideal characteristics of the hardware design of the sub-band decomposition filter in the frequency domain interleaving process will cause the same-frequency signal to enter the two adjacent sub-bands at the same time, which is called the overlapping band. At the same time, the difference in hardware characteristics of different sub-bands will also introduce phase differences of the same-frequency signals. The introduction of this phase difference will cause the amplitude of the reconstructed and spliced ​​signals in the overlapping band of the two sub-bands to cancel each other, resulting in a decrease in system performance. In order to correct the phase difference caused by the inconsistency of the frequency response between sub-bands, the traditional compensation algorithm solves the coefficient optimization equation and constructs a cascade structure of integer delay, fractional delay and all-pass filters to fit the overlapping band phase difference for compensation. However, the phase lead / lag relationship between adjacent sub-bands in the overlapping band estimated by this method is unknown, resulting in the overall trend of the phase difference of multiple frequency points in the overlapping band being uncertain upward / downward, while the stable all-pass second-order section filter group delay is always positive, which makes the traditional algorithm fixed to a certain sub-band for overlapping band compensation likely to have the problem of failure of optimization equation fitting.

[0004] Based on the defects of traditional algorithms in this aspect, this paper proposes an overlapping band calibration method for a broadband interleaved acquisition system that adapts to a fixed phase compensation position. By improving the original objective function, the overlapping band adaptive phase compensation effect is achieved. Summary of the invention

[0005] In order to solve the above technical problems, the present invention provides an overlapping band calibration method for a broadband interleaved acquisition system with an adaptive fixed phase compensation position, which can be used in a broadband interleaved sampling system that simulates the phase cancellation phenomenon of overlapping band signals after splicing caused by the non-ideality of the transition band of the analog filter, and can perform delay and filtering on any sub-band, thereby achieving the purpose of rapid compensation of the phase difference of the overlapping band signal.

[0006] To achieve the above-mentioned purpose, the technical solution adopted by the present invention is: an overlap band calibration method of a broadband interleaved acquisition system with an adaptive fixed phase compensation position, comprising the following steps:

[0007] S1, determining the overlapping band frequency interval according to the amplitude-frequency response characteristics of the transition bands of two adjacent sub-bands;

[0008] S2, estimate the overlapping band sweep phase difference;

[0009] S3, constructing an overlapping band compensation module: constructing a linear phase compensation module and a nonlinear phase compensation module, and building and solving a fitting coefficient optimization function of the overlapping band compensation module; the fitting coefficient optimization function is as follows:

[0010]

[0011] Where E represents the sum of squares of the difference between the correction structure and the system phase-frequency response; δ D Indicates linear delay; M p represents the modulus length of the pth second-order node pole; θ p represents the argument of the pth second-order node pole; k is a positive integer, indicating that the phase-frequency response of the compensation module differs from the overlap band by k cycles 2π; P / 2 means that the all-pass filter is composed of P / 2 second-order cascaded IIR filters; p represents the pth second-order node pole;

[0012] R represents the weighted error function value; W = [w(ω1), w(ω2), …, w(ω N )] T , W represents the weight value, Indicates the amplitude difference of the same frequency point between sub-bands in the overlapping band; e(ω n )=φ c (ω n )-Δφ(ω n ), represents the error between the phase difference of the compensation module and the phase difference to be fitted, φ c (ω) is the phase difference of the compensation module after adding 2πk, Δφ(ω n ) is the calculated overlap band phase difference; ω represents the digital angular frequency; represents a dot product operation; the superscript T represents a transpose operation; and st represents a constraint condition.

[0013] S4. Construct a sub-band phase difference compensation module.

[0014] Preferably, the linear phase compensation module is constructed as follows:

[0015]

[0016] δ Drepresents linear delay; It is the integer drop point module; is the fractional filter module; j represents the imaginary unit; δ Id represents integer delay; δ Fd Indicates fractional delay;

[0017] The phase-frequency response of the linear phase compensation module is D (ω) is: φ D (ω)=-δ D ω.

[0018] Preferably, a nonlinear phase compensation module is constructed: the nonlinear phase compensation module is composed of a cascade of all-pass filters, and its transfer function is:

[0019]

[0020] The pole position of the pth second-order node is and p=1,2,…P / 2; z represents a complex variable; Re{·} represents a real part operation.

[0021] The phase-frequency response of the nonlinear phase compensation module is p (ω) is:

[0022]

[0023] Preferably, in step S1, the maximum amplitude frequency difference M of the same frequency points of two sub-bands at the overlapping band boundary is preset according to engineering experience. th , in dB; the amplitude-frequency response of each sub-band transition band is calculated by sine sweep, and the amplitude difference M between two sub-bands at the same frequency point diff (ω) is:

[0024]

[0025] M0(ω) and M1(ω) are the signal amplitude values ​​when the digital angular frequency is ω. At this time, the overlapping band frequency interval range should satisfy: |M diff (ω)|≤M th ; where ω is the sweep frequency sequence.

[0026] Preferably, in step S2, the frequencies within the overlapping band are divided into N equal parts, and the corresponding frequency points are [ω1, ω2, ...ω N ], the sweep step frequency is ω step =ω2-ω1;

[0027] The signal source inputs a sine sweep signal with different frequencies, and uses the sine three-parameter or Fourier transform method to estimate the phase φ1(ω i ) and φ2(ωi ), calculate the phase difference Δφ(ω i )for:

[0028] Δφ(ω i )=φ1(ω i )-φ2(ω i ),i=1,2,…N.

[0029] Preferably, in step S3, the method for solving the fitting coefficient optimization function includes a particle swarm algorithm, a LM algorithm, and a hybrid particle swarm algorithm.

[0030] Preferably, a particle swarm algorithm is used for solving: Set the maximum number of iterations K of the particle swarm algorithm hpslom , the particle swarm population size S, the number of second-order nodes of the nonlinear compensation module cascade P / 2; the optimal individual of the population is calculated [M p ,θ p ,k], p=1,2,…P / 2.

[0031] Preferably, in step S4, both sub-band 1 and sub-band 2 have an integer delay module. and Its value is calculated as follows:

[0032]

[0033] When performing overlapping band compensation, subband 2 first undergoes a delay δ Fd =δ D -floor(δ D ) fractional filter module, where floor(·) means rounding down; then passing through the all-pass filter module Perform nonlinear compensation; finally, the integer delay values ​​of subband 1 and subband 2 are obtained according to the formula and Perform integer delay to complete overlapping band phase difference compensation.

[0034] The present invention has the following beneficial effects:

[0035] 1. Compared with the optimization calculation method of the traditional compensation coefficient, the present invention changes the optimization objective function value so that there is no need to consider the overall trend of the fitted phase difference, thus avoiding the defect of the traditional method that the compensation effect is not ideal in some cases, and can achieve effective phase compensation of the overlapping band phase difference under the condition of fixed compensation position. It can be used in a broadband interleaved sampling system that simulates the phase cancellation phenomenon of overlapping band signals after splicing caused by the non-ideality of the transition band of the analog filter, and can delay and filter on any sub-band, so as to achieve the purpose of fast compensation of the overlapping band signal phase difference.

[0036] 2. Compared with the compensation module of the traditional algorithm, the present invention only adds an integer delay module on another sub-band, and the system resource overhead of the compensation module is basically the same as that of the traditional algorithm. BRIEF DESCRIPTION OF THE DRAWINGS

[0037] Figure 1 It is a flowchart of the calibration method of the present invention;

[0038] Figure 2 It is a schematic diagram of the amplitude-frequency response of each sub-band transition band calculated by the sine frequency sweep method of the present invention;

[0039] Figure 3 The phase-frequency response φ of a single second-order section of the present invention is p (ω) in θ p Schematic diagram of the change situation;

[0040] Figure 4 A schematic diagram showing the comparison of the phase difference after fitting in the method of the present invention and after fitting in the traditional method;

[0041] Figure 5 A schematic diagram of compensation in sub-band 2 of the present invention;

[0042] Figure 6 A schematic diagram of the phase difference between two sub-bands in an uncompensated state in an embodiment of the present invention;

[0043] Figure 7 Schematic diagram showing the comparison of amplitude-frequency responses after calibration between the method of the present invention and the traditional method in an embodiment of the present invention. DETAILED DESCRIPTION

[0044] The following will be combined with the drawings in the embodiments of the present invention to clearly and completely describe the technical solutions in the embodiments of the present invention. Obviously, the described embodiments are only part of the embodiments of the present invention, not all of the embodiments. Unless otherwise specified, the technical means used in the embodiments are conventional means well known to those skilled in the art.

[0045] like Figure 1 As shown, the present invention discloses an overlap band calibration method for a broadband interleaved acquisition system with an adaptive fixed phase compensation position, comprising the following steps:

[0046] S1. Determine the overlapping band frequency interval according to the amplitude-frequency response characteristics of the transition band between two adjacent sub-bands.

[0047] Specifically, based on engineering experience, the maximum amplitude frequency difference M of the same frequency points of two sub-bands at the overlapping band boundary is preset. th , in dB. Figure 2 , calculate the amplitude-frequency response of each sub-band transition band by sine sweep, and define the amplitude difference M between two sub-bands at the same frequency point diff (ω) is:

[0048]

[0049] Among them, M0(ω) and M1(ω) are the signal amplitude values ​​when the digital angular frequency is ω. At this time, the overlapping band frequency interval range should satisfy:

[0050] |M diff (ω)|≤M th (2)

[0051] Where ω is the sweep frequency sequence. This formula means that when the absolute value of the amplitude difference between the same frequency points of the two sub-bands is less than M th , the frequency point is within the overlapping band.

[0052] S2. Estimate the overlapping band sweep phase difference.

[0053] Divide the frequency in the overlapping band into N equal parts, and the corresponding frequency points are [ω1,ω2,…ω N ], the sweep step frequency is ω step =ω2-ω1. The signal source inputs a sine sweep signal with different frequencies, and uses the sine three-parameter or Fourier transform method to estimate the phase φ1(ω i ) and φ2(ω i ), calculate the phase difference Δφ(ω i )for:

[0054] Δφ(ω i )=φ1(ω i )-φ2(ω i ),i=1,2,…N (3)

[0055] S3. Construct an overlapping band compensation module.

[0056] The phase difference of the overlapping band usually includes a linear part and a nonlinear part. The cascade method of the linear delay structure and the nonlinear phase correction structure can simultaneously compensate for the linear and nonlinear phase differences of the overlapping band. At the same time, the calculation method of the phase difference in formula (3) will affect the compensation position of the overlapping band phase compensation module. For example, using the phase φ1(ω i ) minus the phase φ2(ω i ), then the all-pass filter module and the fractional filter module need to be performed in sub-band two.

[0057] S31, construct the linear phase compensation module as follows:

[0058]

[0059] δ D represents linear delay; It is the integer drop point module; is the fractional filter module; j represents the imaginary unit; δ Id represents integer delay; δ Fd Indicates fractional delay.

[0060] The phase-frequency response of the linear phase compensation module is D (ω) is: φ D (ω)=-δ D ω.

[0061] S32, construct a nonlinear phase compensation module: The nonlinear phase compensation module is composed of a cascade of all-pass filters, and its transfer function is:

[0062]

[0063] Formula (4) indicates that the all-pass filter consists of P / 2 second-order cascaded IIR filters, where the pole position of the pth second-order section is and p=1,2,…P / 2; z represents a complex variable; Re{·} represents a real part operation.

[0064] The phase-frequency response of the nonlinear phase compensation module is p (ω) is:

[0065]

[0066] refer to Figure 3 , the phase-frequency response of a single second-order section φ p (ω) in θ p The most dramatic change is at M p The larger the p The more drastic the change of (ω), the more its integral slope is less than zero (the group delay is always positive).

[0067] S33. The phase-frequency response of the compensation module system built by fixing the integer delay module in a certain sub-band compensation method is also the optimization objective function for solving the fitting coefficient as follows:

[0068]

[0069] It should be noted that the method of fixing the integer delay module in a certain sub-band compensation mentioned in step S33 is specifically: application number 202010448355.2, patent name "A frequency overlapping band phase compensation method for a bandwidth alternating sampling system"; application number 202010952125.X, patent name "A digital all-pass filter design method based on a hybrid particle swarm algorithm".

[0070] If equation (6) is used to fit the overlapping band phase difference, when the overall slope of Δφ(ω) calculated according to equation (3) is greater than zero, that is, sub-band 1 lags behind sub-band 2 at the same frequency point, the fitting will fail. Figure 4 The simulated data of the overlap band phase difference is calculated by delaying sub-band 2 by 100 points. At this time, the traditional algorithm performs phase difference compensation in the second sub-band, and the fitted φ c There is a big gap between (ω) and Δφ(ω). There are two main reasons:

[0071] 1. The overlapping band frequency sweep does not start from zero frequency. There is a periodic winding problem when calculating the phase difference, that is, Δφ(ω i ) differs from the actual value by an integer multiple of 2π. Figure 4 , the actual phase difference is between [124, 128] rad / s, while the phase difference obtained by overlapping band sweep is between [-2, 2.5] rad / s. If the phase-frequency compensation φ of the linear delay module is used D (ω),ω∈[ω1,ω2,...ω N ] to fit the frequency sweep phase difference and solve the linear delay δ D There are great optimization constraints in the process: even if there is a δ D The feasible solution is -100, which means that the subband is delayed by 100 points. D The calculated value of (ω) is the actual phase difference, which is quite different from the frequency sweep phase difference. This feasible solution will also be discarded. D The optimization calculation of (ω) is greatly limited. Therefore, it is difficult to perform effective phase difference compensation by fixing the loss point in the second sub-band.

[0072] 2. Reference Figure 4 , when the sub-band 2 is delayed by 100 points, the overlap band phase difference Δφ(ω i ) The overall trend is upward, and the group delay of the phase difference between sub-bands is negative. In the analysis of step S32, the stable physically achievable second-order section group delay is always positive. At this time, the second-order section of the traditional algorithm fixed in the second sub-band for compensation will fail, that is, the phase difference after fitting is approximately a straight line.

[0073] In summary, both the traditional algorithm’s approach of fixing two sub-band compensations and the optimization objective function have limitations. The present invention first considers the design defects of the traditional algorithm’s phase compensation and optimizes the phase-frequency response of the compensation module. When the amplitude-frequency response of the compensation module differs from the phase difference of the overlapping band by k times 2π (k is a positive integer), it will not affect the amplitude change after the sub-bands are spliced. Therefore, k can be added as an optimization variable to the optimization function of the fitting coefficient of the compensation module. This method allows the phase-frequency response of the compensation module to differ from the overlapping band by k periods of 2π. This can effectively solve the above problems by broadening the feasible solution space of the original optimization problem. See reference. Figure 4Secondly, the compensation module structure was updated, which is described in detail in S4.

[0074] The optimized function of the fitting coefficient of the compensation module after optimization is as follows:

[0075]

[0076] Where k is a positive integer; θ p represents the argument of the pth second-order node pole; M p represents the modulus length of the pth second-order node pole; then the objective optimization function is:

[0077]

[0078] Where E represents the sum of squares of the difference between the correction structure and the system phase-frequency response; δ D Indicates linear delay; M p represents the modulus length of the pth second-order node pole; θ p represents the angle of the pth second-order node pole; k is a positive integer, indicating that the phase-frequency response of the compensation module differs from the overlap band by k cycles 2π; P / 2 means that the all-pass filter is composed of P / 2 second-order cascaded IIR filters; p represents the pth second-order node pole.

[0079] R represents the weighted error function value; W = [w(ω1), w(ω2), …, w(ω N )] T , W represents the weight value, Indicates the amplitude difference of the same frequency point between sub-bands in the overlapping band; e(ω n )=φ c (ω n )-Δφ(ω n ), represents the error between the phase difference of the compensation module and the phase difference to be fitted, φ c (ω) is the phase difference of the compensation module after adding 2πk, Δφ(ω n ) is the calculated overlap band phase difference; ω represents the digital angular frequency; represents a dot product operation; the superscript T represents a transpose operation; and st represents a constraint condition.

[0080] Solving equation (8) yields the linear delay δ D , the pole modulus M of each second-order section of the all-pass filter p , phase angle θ p The compensation module can be determined.

[0081] S34, solving the fitting coefficient optimization function.

[0082] Compared with the traditional method, the optimization equation of formula (8) has only one more variable k, and after k is added as an additional term, it does not affect the gradient calculation results of other parameters. Therefore, various optimization algorithms such as but not limited to particle swarm algorithm, LM algorithm, hybrid particle swarm algorithm, etc. can be used for calculation. For example, the steps of using particle swarm algorithm to solve the above formula are as follows:

[0083] S341. Initial parameter setting: Setting the maximum number of iterations K of the particle swarm algorithm hpslom , the particle swarm population size S, the number of second-order sections of the nonlinear compensation module cascade P / 2;

[0084] S342, Particle Swarm Optimization: Use the particle swarm optimization algorithm to optimize the calculation and obtain the optimal individual of the population [M p ,θ p ,k], p=1,2,…P / 2.

[0085] S4. Sub-band phase difference compensation module design.

[0086] Figure 5 A schematic diagram of compensation in sub-band 2 is given. Compared with the traditional method, the present invention has an integer delay module δ in both sub-band 1 and sub-band 2. Id1 and δ Id2 , whose value is calculated as follows:

[0087]

[0088] When performing overlapping band compensation, subband 2 first undergoes a delay δ Fd =δ D -floor(δ D ) fractional filter module, where floor(·) means rounding down; then passing through the all-pass filter module Perform nonlinear compensation; Finally, the integer delay values ​​of subband 1 and subband 2 are obtained according to formula (9): and Perform integer delay to complete overlapping band phase difference compensation.

[0089] Example

[0090] The following is a detailed description of the technical effects of the present invention in combination with actual engineering tests:

[0091] 1. Test conditions and contents

[0092] The experimental equipment is a high-speed digital storage oscilloscope with a sampling rate of 80GSa / s and a bandwidth of 18GHz, which is built based on broadband interleaved sampling technology, and a signal source that can generate a sinusoidal signal with a frequency range of 20GHz. The signal source generates a sinusoidal swept frequency signal and inputs it into the oscilloscope. The oscilloscope collects the time domain signals of the two sub-bands without phase compensation at the overlapping band for analysis and calculation compensation.

[0093] 2. Test result analysis

[0094] Reference Figure 6 , the phase difference between the two sub-bands in the uncompensated state is calculated based on the collected data, as shown with a star curve, and the phase difference span is between [0,8]rad / s. At this time, the overall trend of the phase difference calculated according to formula (3) is upward, that is, the sampling of sub-band two lags behind that of sub-band one. The traditional optimization method performs compensation in sub-band two, so it is difficult to fit the overall upward phase difference trend, which makes the phase-frequency response of the compensation module far from the overlapping band phase difference. However, the optimization method of the present invention changes the target optimization function and the structure of the compensation module, so that the fitting curve is closer to the phase difference between the two sub-bands in the uncompensated state, and the compensation effect is also better. The overlapping band phase difference after compensation by the method proposed in this article is closer to zero.

[0095] Reference Figure 7 It can be seen that if compensation is performed on sub-band two, but the relative sampling relationship of sub-band two lags behind that of sub-band one, the phase difference fitting fails due to the defects of the original optimization function, so the compensation effect is not very good. Only when the overlapping band amplitude is close to some frequency points, the compensation effect is good, and the amplitude after compensation is not smooth in the overlapping band. The overlapping band amplitude-frequency response based on the compensation method of the present invention presents an ideal phenomenon without amplitude cancellation between sub-bands, and the overlapping band amplitude-frequency response after compensation of the present invention is smoother.

[0096] The embodiments described above are only descriptions of the preferred modes of the present invention and are not intended to limit the scope of the present invention. Without departing from the design spirit of the present invention, various deformations, modifications, and substitutions made to the technical solutions of the present invention by ordinary technicians in this field should all fall within the protection scope determined by the claims of the present invention.

Claims

1. An overlap band calibration method for a broadband interleaved acquisition system with a fixed phase compensation position is characterized by: The steps include: S1, determining the overlapping band frequency interval according to the amplitude-frequency response characteristics of the transition bands of two adjacent sub-bands; S2, estimate the overlapping band sweep phase difference; S3, constructing an overlapping band compensation module: constructing a linear phase compensation module and a nonlinear phase compensation module, and building and solving a fitting coefficient optimization function of the overlapping band compensation module; The optimized function of the compensation module's fitting coefficient is as follows: Where k is a positive integer; θ p represents the argument of the pth second-order node pole; M p represents the modulus length of the pole of the pth second-order node; The objective optimization function is as follows: Where E represents the sum of squares of the difference between the correction structure and the system phase-frequency response; δ D Indicates linear delay; M p represents the modulus length of the pth second-order node pole; θ p represents the argument of the pth second-order node pole; k is a positive integer, indicating that the phase-frequency response of the compensation module differs from the overlap band by k cycles 2π; P / 2 means that the all-pass filter is composed of P / 2 second-order cascaded IIR filters; p represents the pth second-order node pole; n=1,2,…N, R represents the weighted error function value; W=[w(ω1),w(ω2),…,w(ω N )] T , W represents the weight value, Indicates the amplitude difference of the same frequency point between sub-bands in the overlapping band; e(ω n )=φ c (ω n )-Δφ(ω n ), represents the error between the phase difference of the compensation module and the phase difference to be fitted, φ c (ω) is the phase difference of the compensation module after adding 2πk, Δφ(ω n ) is the calculated overlap band phase difference; ω represents the digital angular frequency; represents the dot product operation; the superscript T represents the transposition operation; st represents the constraint condition; S4. Construct a sub-band phase difference compensation module.

2. The overlapping band calibration method of the broadband interleaved acquisition system with an adaptive fixed phase compensation position according to claim 1 is characterized in that: The linear phase compensation module is constructed as: δ D represents linear delay; It is the integer drop point module; is the fractional filter module; j represents the imaginary unit; δ Id represents integer delay; δ Fd Indicates fractional delay; The phase-frequency response of the linear phase compensation module is D (ω) is: φ D (ω)=-δ D ω.

3. The overlapping band calibration method of the broadband interleaved acquisition system with an adaptive fixed phase compensation position according to claim 2 is characterized in that: Constructing nonlinear phase compensation module: The nonlinear phase compensation module consists of a cascade of all-pass filters, and its transfer function is: The pole position of the pth second-order node is and p=1,2,…P / 2;z represents a complex variable;Re{} represents the real part operation; The phase-frequency response of the nonlinear phase compensation module is p (ω) is:

4. The overlapping band calibration method of the broadband interleaved acquisition system with an adaptive fixed phase compensation position according to claim 1 is characterized in that: In step S1, the maximum amplitude frequency difference M of the same frequency points of two sub-bands at the overlapping band boundary is preset according to engineering experience. th , in dB; the amplitude-frequency response of each sub-band transition band is calculated by sine sweep, and the amplitude difference M between two sub-bands at the same frequency point diff (ω) is: M0(ω) and M1(ω) are the signal amplitude values ​​when the digital angular frequency is ω. At this time, the overlapping band frequency interval range should satisfy: |M diff (ω)|≤M th ; where ω is the sweep frequency sequence.

5. The overlapping band calibration method of the broadband interleaved acquisition system with an adaptive fixed phase compensation position according to claim 1 is characterized in that: In step S2, the frequency in the overlapping band is divided into N equal parts, and the corresponding frequency points are [ω1, ω2, ...ω N ], the sweep step frequency is ω step =ω2-ω1; The signal source inputs a sine sweep signal with different frequencies, and uses the sine three-parameter or Fourier transform method to estimate the phase φ1(ω i ) and φ2(ω i ), calculate the phase difference Δφ(ω i )for: Df(ω i )=φ1(ω i )-φ2(ω i ),i=1,2,…N.

6. The overlapping band calibration method of the broadband interleaved acquisition system with an adaptive fixed phase compensation position according to claim 3 is characterized in that: In step S3, the solution method of the fitting coefficient optimization function includes a particle swarm algorithm, a LM algorithm, and a hybrid particle swarm algorithm.

7. The overlapping band calibration method of the broadband interleaved acquisition system with an adaptive fixed phase compensation position according to claim 6 is characterized in that: Use particle swarm algorithm to solve: Set the maximum number of iterations K of particle swarm algorithm hpslom , the particle swarm population size S, the number of second-order nodes of the nonlinear compensation module cascade P / 2; the optimal individual of the population is calculated [M p ,θ p ,k], p=1,2,…P / 2.

8. The overlapping band calibration method of the adaptive fixed phase compensation position broadband interleaved acquisition system according to claim 3 is characterized by: In step S4, both sub-band 1 and sub-band 2 have an integer delay module. and Its value is calculated as follows: When performing overlapping band compensation, subband 2 first undergoes a delay δ Fd =δ D -floor(δ D ) fractional filter module, where floor(·) means rounding down; then passing through the all-pass filter module Perform nonlinear compensation; finally, the integer delay values ​​of subband 1 and subband 2 are obtained according to the formula and Perform integer delay to complete overlapping band phase difference compensation.

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