An Optimized Control Method for a Single-Phase High-Frequency Chain AC-DC Electrolytic Capacitor-Free Converter

Abstract

The application relates to a single-phase high-frequency chain AC-DC electrolytic capacitor-free converter optimization control method, and belongs to the technical field of power electronics. In order to realize power factor correction and soft switching without a closed-loop control circuit, the application comprises a converter and a controller, and the converter and the controller are connected through signals; the controller collects signals of the converter as feedback signals, generates driving pulse signals of the converter in an SPWM mode, and adopts a phase-shifting control method and a dead zone adaptive control method to optimize control of the frequency and pulse width of a first series-connected switch tube, a second series-connected switch tube, a third series-connected switch tube and a fourth series-connected switch tube in a primary side matrix converter and a secondary side first switch tube, a secondary side second switch tube, a secondary side third switch tube and a secondary side fourth switch tube of a secondary side rectifier. The application generates driving pulse signals of the front-stage matrix converter, eliminates a DC link energy storage capacitor of a traditional two-stage rectifier circuit, reduces the volume, and saves the hardware cost.

Description

Technical Field

[0001] This invention belongs to the field of power electronics technology, specifically relating to an optimized control method for a single-phase high-frequency AC-DC electrolytic capacitor-free converter. Background Technology

[0002] With the large-scale application of new energy power systems, higher demands are being placed on power electronics optimization technology. Among these, advanced rectification technology is currently a key research focus. The high-frequency chain AC-DC converter, a type of bidirectional isolated converter, is an extension of the matrix rectifier. It achieves the mutual conversion of energy between AC and DC by employing a bidirectional switch array. Furthermore, due to its advantages such as high power density, low harmonic content, minimal power conversion, and a wide input voltage range, it has become a hot research topic.

[0003] The simplest control method for high-frequency chain AC-DC converters is single-phase-shift control, which adjusts power transmission by controlling the phase shift angle of the high-frequency AC voltage across the full-bridge transformer. However, single-phase-shift control has low controllability and high current stress, leading to significant power backflow when there is a mismatch between the preceding and following stage voltages. Dual-phase-shift control expands the control freedom, thereby reducing power backflow and current stress. For high-frequency chain AC-DC converter control, the dead time is often fixed, making it impossible to guarantee sufficient time for the charging and discharging process to complete. Furthermore, traditional two-stage transmission converter topologies contain large-volume voltage regulator capacitors in the intermediate stage, resulting in low efficiency, high complexity, complex boost inductor design, and large-volume electrolytic capacitors on the bus side. While existing control schemes can achieve single-stage operation, they still require large electrolytic capacitors. Therefore, research on control strategies for resonant high-frequency chain AC-DC capacitor-free converters is highly significant. Summary of the Invention

[0004] The problem this invention aims to solve is to achieve power factor correction and soft switching without the need for closed-loop control circuitry, and proposes an optimized control method for a single-phase high-frequency AC-DC electrolytic capacitor-free converter.

[0005] To achieve the above objectives, the present invention provides the following technical solution:

[0006] An optimized control method for a single-phase high-frequency AC-DC capacitor-free converter includes a converter and a controller, which are connected by a signal.

[0007] The controller collects the converter's signal as a feedback signal, generates the converter's drive pulse signal using SPWM, and optimizes the frequency and pulse width of the first series switch S1, the second series switch S2, the third series switch S3, and the fourth series switch S4 in the primary matrix converter, as well as the second-side first switch Q1, the second-side second switch Q2, the third-side third switch Q3, and the fourth-side fourth switch Q4 in the secondary rectifier using phase-shift control and dead-time adaptive control methods.

[0008] The converter includes an input AC power supply, an AC filter, a primary-side matrix converter, a high-frequency transformer, an inductor, a secondary-side rectifier, and a DC power supply.

[0009] The AC filter includes an AC-side filter inductor L1 and a filter capacitor C1; the primary-side matrix converter is composed of the first primary-side bridge arm and the second primary-side bridge arm connected in parallel, the first primary-side bridge arm is composed of the first series-connected switch S1 and the second series-connected switch S2 connected in series, and the second primary-side bridge arm is composed of the third series-connected switch S3 and the fourth series-connected switch S4 connected in series.

[0010] One end of the AC power supply is connected to the AC side filter inductor L1. The AC side filter inductor L1 is connected to the filter capacitor C1 and one end of the primary side matrix converter. The other end of the filter capacitor C1 and the primary side matrix converter is connected to the other end of the AC power supply.

[0011] The secondary-side rectifier is composed of a first secondary-side bridge arm and a second secondary-side bridge arm connected in parallel. The first secondary-side bridge arm is composed of a first secondary-side switch Q1 and a second secondary-side switch Q2 connected in series. The second secondary-side bridge arm is composed of a third secondary-side switch Q3 and a fourth secondary-side switch Q4 connected in series. The two ends of the DC power supply are respectively connected to the two ends of the secondary-side rectifier.

[0012] One end of the primary side of the high-frequency transformer is connected to the midpoint of the first bridge arm of the primary-side matrix converter, and the other end of the primary side of the high-frequency transformer is connected to the midpoint of the second bridge arm of the primary-side matrix converter; one end of the secondary side of the high-frequency transformer is connected to an inductor L, and the other end of the inductor L is connected to the midpoint of the first bridge arm of the secondary-side rectifier, and the other end of the secondary side of the high-frequency transformer is connected to the midpoint of the second bridge arm of the secondary-side rectifier.

[0013] The controller uses the converter's signal as feedback signal and employs phase-shift control and dead-time adaptive control strategies to optimize the frequency and pulse width of the first series switch S1, the second series switch S2, the third series switch S3, and the fourth series switch S4 in the primary matrix converter, as well as the second-side first switch Q1, the second-side second switch Q2, the third-side third switch Q3, and the fourth-side fourth switch Q4 in the secondary rectifier.

[0014] Furthermore, the first series-connected switch S1, the second series-connected switch S2, the third series-connected switch S3, and the fourth series-connected switch S4 are all connected in a common-source configuration. The first series-connected switch includes switch S... 11 S 12 The second series-connected switching transistor includes switching transistor S. 21 S 22 The third series-connected switching transistor includes switching transistor S. 31 S 32 The fourth series-connected switch includes switch S. 41 S 42 .

[0015] Furthermore, the phase-shift control method includes the following steps:

[0016] In AC-side control, modulation is achieved using a triangular carrier wave u0. In DC-side control, modulation is achieved using phase-shifted triangular carrier waves u1 and u2 derived from the triangular carrier wave u0. The converter switching pulses are adjusted via phase-shift control, and the adjustment range of the phase-shift angle φ is as follows:

[0017] 0≤φ≤π (1);

[0018] The controller first uses numerical calculations to calculate the peak current i of the primary-side matrix converter. p The calculation expression is:

[0019]

[0020] Where p is the transmission power and k is the voltage conversion ratio;

[0021] Solve for partial derivatives The obtained optimal control coordinates φ op , where i p This represents the peak current of the primary-side matrix converter.

[0022] Due to the limitations of the boundary conditions in equation (1), when φ op When within the operating area, φ = φ op When φ op When the value is outside the operating region, the closest extreme point on the boundary of the operating region is selected as the optimization result;

[0023] The formula for calculating the voltage conversion ratio k is:

[0024] k = v dc / nv ac

[0025] Among them, v dc DC power supply voltage, v ac AC power supply voltage;

[0026] Then calculate the phase shift φ between the DC and AC sides of the converter. op The calculation expression is:

[0027]

[0028] Furthermore, the SPWM control method also includes modulation via a triangular carrier u0 in the AC side control. When u0 is greater than 0, the first and fourth series switches of the primary matrix converter are turned on, and when u0 is less than 0, the second and third series switches of the primary matrix converter are turned on.

[0029] In DC-side control, the triangular carriers u1 and u2 are modulated by phase-shifted triangular carrier u0, with u1 phase-shifted by φ relative to u0 and u2 phase-shifted by 180° relative to u1.

[0030] DC side through modulation signal u control Modulation duty cycle, defining the modulation signal u control The expression is:

[0031] u control =-|sin(ω) s t)|+1 (4)

[0032] Where, ω s t is the switching angular frequency, and t is time.

[0033] Furthermore, the dead-time adaptive control strategy is set such that the dead time interval between two switches on a bridge arm should be longer than the time required for the parallel capacitor to fully discharge. In order to avoid increasing switching losses due to excessively long dead time, adaptive dead-time control is adopted. The relationship between dead time and drain-source current is as follows:

[0034]

[0035] Among them, t d V is the dead time interval. DS i is the drain-source voltage. Lrms C is the effective value of the inductor current. oss It is a parallel capacitor; the dead time interval is calculated in real time based on the sampling inductor current.

[0036] Furthermore, the controller controls the converter to achieve six operating states within one high-frequency cycle:

[0037] Mode 1: Time interval t0 to t1, S 11 S 12 S 41 and S 42 When the circuit is turned on, a loop is formed with the input inductor, and Q1 and Q4 operate in synchronous rectification mode.

[0038] Mode 2: During the time interval t1 to t2, the flow through S 21 and S 22 As the current rises, Q1 turns off, and Q2 turns on at zero voltage. The current flows through the anti-parallel diode Q2 to form a closed circuit.

[0039] Mode 3: During the time interval t2 to t3, the current begins to decrease, Q4 is turned off, and Q3 is turned on. During the dead time, the current flows through the anti-parallel diode of Q3 to form a closed circuit, so Q3 achieves zero-voltage turn-on. Q2 and Q3 are in synchronous rectification mode.

[0040] Mode 4: Time interval t3 to t4, S 21 S 22 S 31 S 32 When the converter is turned on, it is in rectification mode, the DC-side switch state remains unchanged, and the input inductor current i g As energy is transferred to the DC side, Q2 and Q3 operate in synchronous rectification mode to reduce conduction losses.

[0041] Mode 5: During the time period t4 to t5, the inductor current i in the converter circuit L As the current continues to rise, during the dead time, the current flows through the anti-parallel diode Q4 to form a closed circuit, and Q4 achieves zero-voltage turn-on.

[0042] Mode 6: Time interval t5 to t6, S 21 S 22 S 31 S 32 As the current in the transformer begins to rise, the DC current flows through Q1 and Q4 to form a closed circuit, and Q1 achieves zero-voltage turn-on.

[0043] The beneficial effects of this invention are:

[0044] The present invention discloses an optimized control method for a single-phase high-frequency AC-DC capacitor-free converter. This method uses SPWM to generate the drive pulse signal for the front-stage matrix converter, eliminating the DC link energy storage capacitor of the traditional two-stage rectifier circuit, reducing the size, and saving hardware costs.

[0045] The present invention discloses an optimized control method for a single-phase high-frequency chain AC-DC capacitor-free converter. In specific operation, the controller collects the voltage signal across the load as a feedback signal, adopts a phase-shift modulation strategy to control the power transmission in the primary-side matrix converter and rectifier circuit, and uses a current peak optimization method to calculate the optimized control variable to reduce switching losses.

[0046] The present invention discloses an optimized control method for a single-phase high-frequency AC-DC capacitor-free converter, which includes adaptive dead-time control to prevent the parallel capacitor from being fully discharged during the dead time interval between two switches on a line, while also avoiding switch losses caused by excessively long dead time. Attached Figure Description

[0047] Figure 1 This is a schematic diagram of the topology of a single-phase high-frequency chain AC-DC electrolytic capacitor-free converter according to the present invention;

[0048] Figure 2 This is an equivalent topology diagram of a single-phase high-frequency chain AC-DC electrolytic capacitor-free converter according to the present invention;

[0049] Figure 3 A control principle diagram of a single-phase high-frequency chain AC-DC electrolytic capacitor-free converter provided by the present invention;

[0050] Figure 4 A waveform diagram illustrating the SPWM modulation principle of a single-phase high-frequency AC-DC capacitor-free converter provided by this invention;

[0051] Figure 5 The principle waveform diagram of a single-phase high-frequency chain AC-DC electrolytic capacitor-free converter optimization control method provided by the present invention within one switching cycle;

[0052] Figure 6 This invention provides current path diagrams for different modes within one switching cycle of an optimized control method for a single-phase high-frequency chain AC-DC electrolytic capacitor-free converter, where (a) represents mode 1, (b) represents mode 2, (c) represents mode 3, (d) represents mode 4, (e) represents mode 5, and (f) represents mode 6.

[0053] Figure 7 The figures show experimental waveforms of the optimized control method and the traditional single-phase phase-shift control method when power is transferred from the DC side to the AC side in a single-phase high-frequency AC-DC capacitor-free converter with k=1.33 according to the present invention. (a) is the optimized control strategy of the present invention, and (b) is the traditional control strategy. Detailed Implementation

[0054] To make the objectives, technical solutions, and advantages of this invention clearer, the invention will be further described in detail below with reference to the accompanying drawings and specific embodiments. It should be understood that the specific embodiments described herein are only for explaining the invention and are not intended to limit the invention; that is, the described specific embodiments are merely a part of the embodiments of the invention, and not all of them. The components of the specific embodiments of the invention described and shown in the accompanying drawings can generally be arranged and designed in various different configurations, and the invention may also have other embodiments.

[0055] Therefore, the following detailed description of specific embodiments of the invention provided in the accompanying drawings is not intended to limit the scope of the claimed invention, but merely to illustrate selected specific embodiments of the invention. All other specific embodiments obtained by those skilled in the art based on these specific embodiments without inventive effort are within the scope of protection of this invention.

[0056] To further understand the invention's content, features, and effects, the following specific embodiments are provided, along with accompanying drawings. Figure 1 -Appendix Figure 7 Detailed explanation is as follows:

[0057] Example 1:

[0058] An optimized control method for a single-phase high-frequency AC-DC capacitor-free converter includes a converter and a controller, which are connected by a signal.

[0059] The controller collects the converter's signal as a feedback signal, generates the converter's drive pulse signal using SPWM, and optimizes the frequency and pulse width of the first series switch S1, the second series switch S2, the third series switch S3, and the fourth series switch S4 in the primary matrix converter, as well as the second-side first switch Q1, the second-side second switch Q2, the third-side third switch Q3, and the fourth-side fourth switch Q4 in the secondary rectifier using phase-shift control and dead-time adaptive control methods.

[0060] The converter includes an input AC power supply, an AC filter, a primary-side matrix converter, a high-frequency transformer, an inductor, a secondary-side rectifier, and a DC power supply.

[0061] The AC filter includes an AC-side filter inductor L1 and a filter capacitor C1; the primary-side matrix converter is composed of the first primary-side bridge arm and the second primary-side bridge arm connected in parallel, the first primary-side bridge arm is composed of the first series-connected switch S1 and the second series-connected switch S2 connected in series, and the second primary-side bridge arm is composed of the third series-connected switch S3 and the fourth series-connected switch S4 connected in series.

[0062] One end of the AC power supply is connected to the AC side filter inductor L1. The AC side filter inductor L1 is connected to the filter capacitor C1 and one end of the primary side matrix converter. The other end of the filter capacitor C1 and the primary side matrix converter is connected to the other end of the AC power supply.

[0063] The secondary-side rectifier is composed of a first secondary-side bridge arm and a second secondary-side bridge arm connected in parallel. The first secondary-side bridge arm is composed of a first secondary-side switch Q1 and a second secondary-side switch Q2 connected in series. The second secondary-side bridge arm is composed of a third secondary-side switch Q3 and a fourth secondary-side switch Q4 connected in series. The two ends of the DC power supply are respectively connected to the two ends of the secondary-side rectifier.

[0064] One end of the primary side of the high-frequency transformer is connected to the midpoint of the first bridge arm of the primary-side matrix converter, and the other end of the primary side of the high-frequency transformer is connected to the midpoint of the second bridge arm of the primary-side matrix converter; one end of the secondary side of the high-frequency transformer is connected to an inductor L, and the other end of the inductor L is connected to the midpoint of the first bridge arm of the secondary-side rectifier, and the other end of the secondary side of the high-frequency transformer is connected to the midpoint of the second bridge arm of the secondary-side rectifier.

[0065] The controller uses the converter's signal as feedback signal and employs phase-shift control and dead-time adaptive control strategies to optimize the frequency and pulse width of the first series switch S1, the second series switch S2, the third series switch S3, and the fourth series switch S4 in the primary matrix converter, as well as the second-side first switch Q1, the second-side second switch Q2, the third-side third switch Q3, and the fourth-side fourth switch Q4 in the secondary rectifier.

[0066] Furthermore, the first series-connected switch S1, the second series-connected switch S2, the third series-connected switch S3, and the fourth series-connected switch S4 are all connected in a common-source configuration. The first series-connected switch includes switch S... 11 S 12 The second series-connected switching transistor includes switching transistor S. 21 S 22 The third series-connected switching transistor includes switching transistor S. 31 S 32 The fourth series-connected switch includes switch S. 41 S 42 .

[0067] Furthermore, the topology includes an AC filter, which comprises an AC-side filter inductor L1 and a filter capacitor C1. The voltage across the capacitor is v1, and the voltage difference at the midpoint of the AC-side bridge arm is v. p1 The secondary voltage of the high-frequency transformer is v p2 The primary current of the transformer is i p1 The voltage difference at the midpoint of the DC side bridge arm is v s The DC power supply is V. dc The AC power supply is V. ac The AC input current is i g High-frequency transformer T r The ratio is 1:n. L It is the current flowing through inductor L.

[0068] Furthermore, all switching transistors are SiC MOSFETs.

[0069] Furthermore, the phase-shift control method includes the following steps:

[0070] In AC-side control, modulation is achieved using a triangular carrier wave u0. In DC-side control, modulation is achieved using phase-shifted triangular carrier waves u1 and u2 derived from the triangular carrier wave u0. The converter switching pulses are adjusted via phase-shift control, and the adjustment range of the phase-shift angle φ is as follows:

[0071] 0≤φ≤π (1);

[0072] The controller first uses numerical calculations to calculate the peak current i of the primary-side matrix converter. p The calculation expression is:

[0073]

[0074] Where p is the transmission power and k is the voltage conversion ratio;

[0075] Solve for partial derivatives The obtained optimal control coordinates φ op , where i p This represents the peak current of the primary-side matrix converter.

[0076] Due to the limitations of the boundary conditions in equation (1), when φ op When within the operating area, φ = φ op When φ op When the value is outside the operating region, the closest extreme point on the boundary of the operating region is selected as the optimization result;

[0077] The formula for calculating the voltage conversion ratio k is:

[0078] k = v dc / nv ac

[0079] Among them, v dc DC power supply voltage, v ac AC power supply voltage;

[0080] Then calculate the phase shift φ between the DC and AC sides of the converter. op The calculation expression is:

[0081]

[0082] Furthermore, the SPWM control method also includes modulation via a triangular carrier u0 in the AC side control. When u0 is greater than 0, the first and fourth series switches of the primary matrix converter are turned on, and when u0 is less than 0, the second and third series switches of the primary matrix converter are turned on.

[0083] In DC-side control, the triangular carriers u1 and u2 are modulated by phase-shifted triangular carrier u0, with u1 phase-shifted by φ relative to u0 and u2 phase-shifted by 180° relative to u1.

[0084] DC side through modulation signal u control Modulation duty cycle, defining the modulation signal u control The expression is:

[0085] u control =-|sin(ω) s t)|+1 (4)

[0086] Where, ω s t is the switching angular frequency, and t is time.

[0087] Furthermore, the dead-time adaptive control strategy is set such that the dead time interval between two switches on a bridge arm should be longer than the time required for the parallel capacitor to fully discharge. In order to avoid increasing switching losses due to excessively long dead time, adaptive dead-time control is adopted. The relationship between dead time and drain-source current is as follows:

[0088]

[0089] Among them, t d V is the dead time interval. DS i is the drain-source voltage. Lrms C is the effective value of the inductor current. oss It is a parallel capacitor; the dead time interval is calculated in real time based on the sampling inductor current.

[0090] Figure 6 This embodiment presents the current path diagram for different modes within one switching cycle of an optimized control method for a single-phase high-frequency AC-DC capacitor-free converter. Assuming all components in the converter are ideal, and based on the working principle, there are six operating states within one high-frequency cycle. Furthermore, the controller controls the converter to achieve these six operating states within one high-frequency cycle:

[0091] Mode 1: Time interval t0 to t1, S 11 S 12 S 41 and S 42 When the circuit is turned on, a loop is formed with the input inductor, and Q1 and Q4 operate in synchronous rectification mode.

[0092] Mode 2: During the time interval t1 to t2, the flow through S 21 and S 22 As the current rises, Q1 turns off, and Q2 turns on at zero voltage. The current flows through the anti-parallel diode Q2 to form a closed circuit.

[0093] Mode 3: During the time interval t2 to t3, the current begins to decrease, Q4 is turned off, and Q3 is turned on. During the dead time, the current flows through the anti-parallel diode of Q3 to form a closed circuit, so Q3 achieves zero-voltage turn-on. Q2 and Q3 are in synchronous rectification mode.

[0094] Mode 4: Time interval t3 to t4, S 21 S 22 S 31 S 32 When the converter is turned on, it is in rectification mode, the DC-side switch state remains unchanged, and the input inductor current i g As energy is transferred to the DC side, Q2 and Q3 operate in synchronous rectification mode to reduce conduction losses.

[0095] Mode 5: During the time period t4 to t5, the inductor current i in the converter circuit L As the current continues to rise, during the dead time, the current flows through the anti-parallel diode Q4 to form a closed circuit, and Q4 achieves zero-voltage turn-on.

[0096] Mode 6: Time interval t5 to t6, S 21 S 22 S 31 S 32 As the current in the transformer begins to rise, the DC current flows through Q1 and Q4 to form a closed circuit, and Q1 achieves zero-voltage turn-on.

[0097] Figure 2 This is the equivalent topology diagram of the single-phase high-frequency chain AC-DC electrolytic capacitor-free converter in this embodiment; v p1 and v p2 It refers to the primary and secondary voltages of the transformer, v. s This is the midpoint voltage of the subsequent rectifier. p1 It is the primary current of the transformer, i L It is the current flowing through inductor L.

[0098] Figure 4 This embodiment provides a waveform diagram illustrating the SPWM modulation principle of a single-phase high-frequency AC-DC capacitor-free converter. s This is the midpoint voltage of the subsequent rectifier. The two bridge arm drive signals of the subsequent rectifier have a 180° phase shift, u control U1 is the DC-side modulated signal, u2 and u1 are the DC-side carrier signals, u0 is the AC-side carrier signal, and D is the DC-side duty cycle.

[0099] Figure 5 The schematic diagram of the principle waveform of a single-phase high-frequency chain AC-DC electrolytic capacitor-free converter optimization control method provided in this embodiment within one switching cycle is shown. Figure 5 For the corresponding Figure 4 Current path diagrams for different modes within a single switching cycle. L V is the inductor current of the converter. p2 It is the secondary voltage of the transformer. p2 and v s The phase shift angle at the midpoint is φ, and D is the DC duty cycle.

[0100] To verify the effectiveness of the method in this embodiment, the optimization method was experimentally verified and compared using the constructed experimental platform.

[0101] Figure 7 The figures show the voltage and current waveforms of the proposed optimized control method and the traditional single-phase-shift control method for high-frequency chain capacitorless converters when transmitting power from the DC side to the AC side on a microsecond timescale, with k=1.33 and n=1. It can be observed from the figures that the proposed control strategy significantly reduces the peak-to-peak value of the inductor current.

[0102] The above analysis shows that this circuit reduces current stress and improves power transmission efficiency by using a current peak optimization formula to obtain the optimal duty cycle. Simultaneously, it achieves single-stage power transmission without relying on additional buffers or clamping circuits, eliminating the need for large-volume electrolytic capacitors in intermediate stages. Adaptive dead-time control prevents parallel capacitors from failing to fully discharge during the dead time interval between two switches on a line, while also avoiding switching losses caused by excessively long dead times. The converter has a simple structure, significant advantages, and broad application prospects.

[0103] It should be noted that relational terms such as "first" and "second" are used merely to distinguish one entity or operation from another, and do not necessarily require or imply any such actual relationship or order between these entities or operations. Furthermore, the terms "comprising," "including," or any other variations thereof are intended to cover non-exclusive inclusion, such that a process, method, article, or apparatus that comprises a list of elements includes not only those elements but also other elements not expressly listed, or elements inherent to such a process, method, article, or apparatus. Without further limitations, an element defined by the phrase "comprising one..." does not exclude the presence of other identical elements in the process, method, article, or apparatus that includes said element.

[0104] Although this application has been described above with reference to specific embodiments, various modifications can be made and components can be replaced with equivalents without departing from the scope of this application. In particular, as long as there is no structural conflict, the features in the specific embodiments disclosed in this application can be combined with each other in any way. The lack of an exhaustive description of these combinations in this specification is merely for the sake of brevity and resource conservation. Therefore, this application is not limited to the specific embodiments disclosed herein, but includes all technical solutions falling within the scope of the claims.

Claims

1. A single-phase high-frequency link AC-DC electrolytic capacitorless converter optimal control method, characterized in that, The converter and the controller are connected by signals; The controller generates driving pulse signals of the converter by collecting signals of the converter as feedback signals in an SPWM mode, and optimizes the frequency and pulse width of the first series switch S1, the second series switch S2, the third series switch S3 and the fourth series switch S4 in the primary matrix converter and the first secondary switch Q1, the second secondary switch Q2, the third secondary switch Q3 and the fourth secondary switch Q4 in the secondary rectifier by a phase shift control method and a dead zone adaptive control method; The phase shift control method comprises the following steps: In the AC side control, modulation is performed by a triangular carrier u0, and in the DC side control, modulation is performed by triangular carriers u1 and u2 which are phase shifted from the triangular carrier u0, and the switching pulse of the converter is adjusted by phase shift control, and the adjustment range of the phase shift angle φ is as follows: 0≤φ≤π (1); The controller first calculates the peak current i of the primary matrix converter by numerical operation p The calculation expression is: Wherein, p is the transmission power, and k is the voltage conversion ratio; Solving partial derivatives Optimal control coordinates φ obtained op where i p is the peak current of the line matrix converter; Due to the restriction of the boundary condition in formula (1), when φ op is located in the operating region, φ = φ op ; when φ op is located outside the operating region, the extreme point closest to the boundary of the operating region is selected as the optimization result; The calculation formula of the voltage conversion ratio k is as follows: k = v dc / nv ac wherein v dc is the DC supply voltage, v ac is the AC supply voltage; The phase shift φ between the AC side and the DC side of the converter is then calculated op The expression is calculated as The method of SPWM control further comprises that in the AC side control, modulation is performed by a triangular carrier u0, when u0 is greater than 0, the first series switch and the fourth series switch of the primary matrix converter are turned on, and when u0 is less than 0, the second series switch and the third series switch of the primary matrix converter are turned on; In the DC side control, modulation is performed by triangular carriers u1 and u2 which are phase shifted from the triangular carrier u0, u1 is phase shifted from u0 by φ, and u2 is phase shifted from u1 by 180°; The DC side is modulated by the modulation signal u control The modulation duty cycle defines the modulation signal u control The expression for u is u control = -|sin(ω s t)| + 1 (4) where ω s is the switching angular frequency and t is time.

2. The single-phase high-frequency link AC-DC electrolytic capacitorless converter optimal control method according to claim 1, characterized in that, The converter comprises an input AC power supply, an AC filter, a primary matrix converter, a high-frequency transformer, an inductor, a secondary rectifier and a DC power supply; The AC filter comprises an AC side filter inductor L1 and a filter capacitor C1; the primary matrix converter is composed of a primary first bridge arm and a primary second bridge arm in parallel, the primary first bridge arm is composed of a first series switch S1 and a second series switch S2 in series, and the primary second bridge arm is composed of a third series switch S3 and a fourth series switch S4 in series; One end of the AC power supply is connected to the AC side filter inductor L1, the AC side filter inductor L1 is connected to the filter capacitor C1 and one end of the primary matrix converter respectively, and the other end of the filter capacitor C1 and the primary matrix converter is connected to the other end of the AC power supply; The secondary rectifier is composed of a secondary first bridge arm and a secondary second bridge arm in parallel, the secondary first bridge arm is composed of a first secondary switch Q1 and a second secondary switch Q2 in series, and the secondary second bridge arm is composed of a third secondary switch Q3 and a fourth secondary switch Q4 in series; the DC power supply is connected to the secondary rectifier at both ends; One end of the primary side of the high-frequency transformer is connected to the midpoint of the first bridge arm of the primary matrix converter, the other end of the primary side of the high-frequency transformer is connected to the midpoint of the second bridge arm of the primary matrix converter; one end of the secondary side of the high-frequency transformer is connected to the inductor L, the other end of the inductor L is connected to the midpoint of the first bridge arm of the secondary rectifier, and the other end of the secondary side of the high-frequency transformer is connected to the midpoint of the second bridge arm of the secondary rectifier.

3. The single-phase high-frequency link AC-DC electrolytic capacitorless converter optimal control method of claim 2, wherein, The first, second, third and fourth series switch tubes S1, S2, S3 and S4 are common source connection, the first series switch tube includes switch tube S 11 , 12 ; the second series switch tube includes switch tube S 21 , 22 ; the third series switch tube includes switch tube S 31 , 32 ; the fourth series switch tube includes switch tube S 41 , 42 .

4. The single-phase high-frequency link AC-DC electrolytic capacitorless converter optimal control method of claim 3, wherein, The dead-time adaptive control strategy is set to be longer than the time required for the parallel capacitor to be fully discharged, and in order to avoid the increase of switch loss caused by too long dead-time, the adaptive dead-time control is adopted, and the relationship between the dead-time and the drain-source current is as follows: where t d is the dead time interval, V DS is the drain-source voltage, i Lrms is the inductor current root mean square, C oss is the parallel capacitance; the dead time interval is calculated in real time according to the sampled inductor current.

5. The single-phase high-frequency link AC-DC electrolytic capacitorless converter optimal control method of claim 4, wherein, The controller controls the converter to realize 6 working states in a high-frequency period: Mode 1: t0~t1 period, S 11 , S 12 , S 41 and S 42 Open, with the input inductance to form a circuit, Q1 and Q4 in synchronous rectification mode Mode 2: t1~t2 period, current flows through S 21 and S 22 rises, Q1 is off, Q2 realizes zero-voltage turn-on, and the current flows through the antiparallel diode of Q2 to form a closed circuit; Mode 3: In the time period t2~t3, the current begins to drop, Q4 is turned off, Q3 is turned on, and the current flows through the anti-parallel diode of Q3 to form a closed circuit in the dead-time, so that Q3 realizes zero-voltage turn-on, and Q2 and Q3 are in the synchronous rectification mode; Mode 4: t3~t4 period, S 21 , S 22 , S 31 , S 32 , S g , S g , S g , S g , S g , S g , S g , S g , S g , S g , S g , S g , S g , S g , S g , S g , S g , S g , S g , S g , S g , S < Mode 5: Time period t4~t5, inductor current i in the transformer circuit L Continuing to rise, the current flows through the Q4 antiparallel diode in the dead time to form a closed circuit, and Q4 realizes zero voltage turn-on; Mode 6: t5~t6 period, S 21 , S 22 , S 31 , S 32 And the current of transformer begins to rise, the current of DC side flows through Q1 and Q4 to form a closed circuit, and Q1 realizes zero-voltage turn-on.

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