Auxiliary coil type dynamic tuning IPT system and dynamic tuning method thereof

Abstract

The invention provides an auxiliary coil type dynamic tuning IPT system and a dynamic tuning method thereof, and is used for solving the problems that higher harmonics contained in inversion output current in an energy transfer channel are not considered in tuning of an existing IPT system, and the offset range of a receiving mechanism is limited. The system comprises a transmitting circuit, a receiving circuit and a tuning circuit, the transmitting circuit comprises a direct-current power supply, an inverter circuit I, a transmitting end compensation circuit and a transmitting coil, and the receiving circuit comprises a receiving coil, a receiving end compensation circuit, a rectification filter circuit, an impedance matching circuit and a load. The tuning circuit comprises a DC-DC conversion circuit, an inverter circuit II, an auxiliary compensation circuit and an auxiliary coil which are connected in sequence, the input end of the DC-DC conversion circuit is connected with the DC power supply, and the auxiliary coil is arranged on the transmitting coil. The auxiliary coil serves as a tuning channel, and it can be ensured that the system is always in a resonant state through linear adjustment of the input voltage of the tuning channel.

Description

Technical Field

[0001] The present invention relates to the field of wireless power transmission, and in particular to an auxiliary coil type dynamic tuning IPT system and a dynamic tuning method thereof. Background Art

[0002] Inductively coupled wireless power transfer (IPT) technology uses the principle of electromagnetic induction to transfer energy from the transmitter to the receiver in a non-contact manner. It can solve the problems of mechanical wear and safety hazards caused by traditional plug-in power extraction. It has the advantages of reliability, convenience and flexibility, and can meet the power transmission requirements of various harsh environments. In recent years, this technology has become a hot research topic and has good application prospects in electric vehicles, medical and other fields. The receiving coil of the IPT system will shift and deflect during the energy transmission process. When the coil position changes, due to the presence of the transmitting and receiving iron cores, the system self-inductance and mutual inductance parameters will change with the position, resulting in the IPT system transmitter being unable to operate in a resonant state. For the system detuning problem caused by the position offset of the receiving coil, the existing tuning methods can be divided into four types: using variable capacitors or inductors; adjusting the system operating frequency; switching capacitor arrays; and using coil structure self-tuning.

[0003] The first type uses variable capacitors or inductors, which requires that the switching frequency of the variable capacitor or inductor control circuit is consistent with the operating frequency of the IPT system. However, due to the influence of sampling accuracy and inherent errors of components, there is inevitably a deviation between the actual switching frequency and the system frequency, which aggravates the negative effect of high-order harmonics on the resonant state of the system. The second type uses the method of adjusting the system operating frequency to make the IPT system always in a resonant state by adjusting the operating frequency. However, within the offset range of the receiving end and the load value range, the resonant frequency curve may have a frequency bifurcation phenomenon; the identification and tracking of the frequency bifurcation operating point increases the complexity of the control strategy of this method. The third type uses the method of switching capacitor arrays. Due to the discreteness of the capacitor capacitance, the switching capacitor array method cannot achieve continuous tuning of the IPT system. The fourth type uses the method of self-tuning using the coil structure. Based on the coupling effect of multiple coils, it can achieve self-tuning of the IPT system. However, due to the invariance of the coil parameters and the circulating current effect between the multiple coils, not only the offset range of the receiving coil is limited, but also this method is mainly suitable for low-power applications. Therefore, the existing tuning method does not take into account the higher harmonics contained in the inverter output current in the energy transmission channel, and the offset range of the receiving mechanism is limited. Summary of the invention

[0004] The purpose of the present invention is to provide an auxiliary coil type dynamic tuning IPT system and a dynamic tuning method thereof, which is used to solve the technical problem that the existing IPT system tuning method does not consider the high-order harmonics contained in the inverter output current in the energy transmission channel, and the offset range of the receiving mechanism is limited.

[0005] An auxiliary coil type dynamic tuning IPT system includes a transmitting circuit and a receiving circuit, wherein the transmitting circuit includes a DC power supply, an inverter circuit I, a transmitting end compensation circuit and a transmitting coil, and the receiving circuit includes a receiving coil, a receiving end compensation circuit, a rectifier filter circuit and a load, and also includes a tuning circuit and an impedance matching circuit;

[0006] The impedance matching circuit is arranged between the rectification and filtering circuit and the load, the tuning circuit includes a DC-DC conversion circuit, an inverter circuit II, an auxiliary compensation circuit and an auxiliary coil which are connected in sequence, the input end of the DC-DC conversion circuit is connected to the DC power supply, and the auxiliary coil is arranged on the transmitting coil.

[0007] Optionally, the transmitting coil and the receiving coil are both DD-type coils, the auxiliary coil is a flat spiral coil, and the axial direction of the flat spiral coil is arranged parallel to the arrangement direction of the DD-type coils.

[0008] Optionally, the impedance matching circuit is a Boost conversion circuit, and the DC-DC conversion circuit is a Sepic conversion circuit.

[0009] Optionally, the transmitting end compensation circuit and the receiving end compensation circuit constitute an LCC-S resonant circuit, and the auxiliary compensation circuit is an LC resonant circuit.

[0010] Optionally, it also includes a dynamic tuning control circuit, a harmonic suppression control circuit and a data acquisition circuit;

[0011] The data acquisition circuit is used to collect the inverter output current and voltage of the inverter circuit I to obtain the collected data. The harmonic suppression control circuit is used to calculate the duty cycle D of the impedance matching circuit according to the collected data, and generate a control signal to drive the impedance matching circuit. The dynamic tuning control circuit is used to calculate the duty cycle α of the driving DC-DC conversion circuit according to the collected data. s , and generate a control signal to drive the DC-DC conversion circuit.

[0012] A dynamic tuning method of an auxiliary coil type dynamic tuning IPT system adopts the above-mentioned auxiliary coil type dynamic tuning IPT system, and the specific steps are as follows:

[0013] S1: Set the initial system parameters;

[0014] S2: Collect the inverter output current of inverter circuit I and perform Fourier analysis to obtain the harmonic suppression ratio r a , according to the harmonic suppression ratio r a Calculating a duty cycle D of the impedance matching circuit and generating a control signal to drive the impedance matching circuit;

[0015] S3: Collect the inverter output voltage of inverter circuit I Calculate the output phase difference θ and the output phase difference θ and the expected phase value θ ref The difference is used as input to calculate the duty cycle α of the DC-DC converter circuit. s , and generate a control signal to drive the DC-DC conversion circuit.

[0016] Optionally, the initial system parameters in step S1 include the expected phase value θ ref、 Duty cycle α s The preset range and duty cycle α s The step value of

[0017] And the current control cycle phase value θ at the initial moment cur , the phase value of the upper wheel control cycle at the initial moment θ pre , the initial duty cycle α s , the intermediate quantity k of the control process at the initial moment and the scanning direction mark quantity d of the control process at the initial moment.

[0018] Optionally, the specific steps in step S3 are:

[0019] S3.1 collects the inverter output voltage of inverter circuit I The measurement signal is conditioned into a square wave by using the zero-crossing detection module, and the output phase difference θ between the output voltage and current of the inverter circuit I is output by the phase detection module;

[0020] S3.2 Determine that the output phase difference θ is less than the expected phase value θ ref Is it true? If so, the system meets the resonance condition and returns to step S3.1. If not, the system does not meet the resonance condition and returns to step S3.3.

[0021] S3.3 Determine the current control cycle phase value θ cur The absolute value of the control cycle is less than the phase value θ of the previous round pre Is it true? If so, go to step S3.4. If not, invert the scanning direction flag d and go to step S3.4.

[0022] S3.4 Determine the duty cycle α s Is it within the preset range? If so, go to step S3.5. If not, determine that the receiving circuit exceeds the preset range and cannot perform tuning control;

[0023] S3.5 Let K = K + 1, and set the current duty cycle α s With duty cycle α s The sum of the step values ​​is used as input to generate a control signal to drive the DC-DC conversion circuit, and then returns to step S3.1.

[0024] Due to the adoption of the above technical solution, the present invention has the following advantages:

[0025] 1. This application uses the auxiliary coil as a tuning channel. Within the set offset range, the linear adjustment of the input voltage of the tuning channel can ensure that the system is always in a resonant state. At the same time, the impedance matching of the equivalent load is used to suppress the proportion of high-order harmonics.

[0026] 2. The auxiliary coil type dynamic tuning system constructed in the present application realizes stepless and continuous tuning of the energy transfer channel without changing the topology of the energy transfer channel and having no influence on the pickup power.

[0027] Other advantages, objectives and features of the present invention will be described in the following description to some extent, and to some extent, will be obvious to those skilled in the art based on the following examination and study, or can be taught from the practice of the present invention. The objectives and other advantages of the present invention can be realized and obtained through the following description. BRIEF DESCRIPTION OF THE DRAWINGS

[0028] The accompanying drawings of the present invention are as follows.

[0029] Figure 1 It is a structural schematic diagram of the auxiliary coil type dynamic tuning IPT system of the present invention.

[0030] Figure 2 It is a schematic diagram of the three-dimensional structure of the coupling mechanism of the present invention.

[0031] Figure 3 It is a front view of the coupling mechanism of the present invention.

[0032] Figure 4 The figure is a schematic diagram of the circuit winding of the transmitting coil and the auxiliary coil of the present invention.

[0033] Figure 5 This is a flow chart of the method for dynamic tuning of the IPT system of the present invention.

[0034] Figure 6 This is a control system diagram for dynamic tuning of the IPT system of the present invention.

[0035] Figure 7 This is an equivalent circuit diagram of the present invention based on auxiliary coil tuning.

[0036] Figure 8 This is an impedance matching equivalent diagram based on the Boost conversion circuit of the present invention.

[0037] Fig. 9 It is the envelope diagram of various parameters of the IPT system under dynamic tuning of the present invention.

[0038] Fig.10It is the transient waveform diagram of the IPT system under dynamic tuning of the present invention.

[0039] Fig.11 θ and r before and after dynamic tuning under XOY plane offset of the present invention a 's change graph.

[0040] Fig.12 This is the voltage and current waveform diagram of the dynamic tuning of the experimental prototype of the present invention.

[0041] Fig.13 It is a dynamic waveform diagram of the operation process of the impedance matching circuit and the tuning channel of the present invention.

[0042] In the figure: t x is the transmitting coil, r x is the receiving coil, c x is the auxiliary coil, B1 is the magnetic core, and B2 is the shielding aluminum plate. DETAILED DESCRIPTION

[0043] The present invention will be further described below in conjunction with the accompanying drawings and embodiments.

[0044] Embodiment 1:

[0045] like Figure 1 The auxiliary coil type dynamic tuning IPT system shown in the figure includes a transmitting circuit, a receiving circuit and a tuning circuit, wherein the transmitting circuit includes a DC power supply U connected in sequence dc , inverter circuit I, transmitter compensation circuit and transmitter coil t x The receiving circuit includes a receiving coil r x , receiving end compensation circuit, rectification and filtering circuit, impedance matching circuit and load.

[0046] The tuning circuit includes a DC-DC conversion circuit, an inverter circuit II, an auxiliary compensation circuit and an auxiliary coil C connected in sequence. x The input end of the DC-DC conversion circuit is connected to the DC power supply U dc connection, the auxiliary coil c x Set in the transmitting coil t x superior.

[0047] In this embodiment, if Figure 1 As shown, the transmitting circuit and the receiving circuit form an energy transmission channel, the tuning circuit is a tuning channel, the inverter circuit I and the inverter circuit II shown are both full-bridge high-frequency inverters, and the transmitter compensation circuit includes a compensation inductor L f1 , compensation capacitor C1 and compensation capacitor C f1 The receiving end compensation circuit includes a compensation capacitor C2, and the rectifier filter circuit is composed of a diode and a filter capacitor C d . Transmitting coil t xThe self-sensitivity is L t , receiving coil r x The self-sensitivity is L r , auxiliary coil c x The self-sensitivity is L c , transmitting coil t x With receiving coil r x The mutual inductance is M e , transmitting coil t x With auxiliary coil c x The mutual inductance is M c . Figure 1 In the figure, i2 is the transmitting coil t x The excitation current, R L is the equivalent resistance of the load, R eq and R eqo are the input equivalent resistances of the impedance matching circuit and the rectifier filter circuit respectively. i5 is the auxiliary tuning coil c x The excitation current, u et is the alternating voltage output by inverter circuit I, u c It is the alternating voltage output by inverter circuit II.

[0048] In this embodiment, the alternating voltage u output by the inverter circuit I of the energy transfer channel et , after passing through the LCC resonant circuit, the transmitting coil is excited t x Generates an alternating magnetic field, receiving coil r x The alternating voltage is induced from the coupled magnetic field and then supplied to the rectifier circuit and the equivalent load of the subsequent circuit through the compensation capacitor C2. Among them, the impedance matching circuit adjusts the equivalent resistance R in real time according to the position of the transmitting and receiving coils. eq , so as to achieve the effect of suppressing the output harmonics of the inverter circuit I.

[0049] The operating frequency of the inverter circuit II of the tuning channel is consistent with that of the inverter circuit I. The alternating voltage u generated by the inverter circuit II is c Through the LC resonant circuit, the auxiliary coil c x Generates an excitation current i5, which flows through the transmitting coil t x The induced voltage is generated, and then the LCC circuit is superimposed on the inverter circuit I of the energy transmission channel. With the help of the DC-DC conversion circuit, u is adjusted in real time. c Amplitude, to ensure that the input voltage u of the energy transfer inverter circuit I e and current i 1t Same phase.

[0050] As an embodiment of the present application, the impedance matching circuit selects a Boost conversion circuit, and the DC-DC conversion circuit selects a Sepic conversion circuit.

[0051] like Figure 2 , Figure 3 and Figure 4 The magnetic coupling mechanism shown, the transmitting coil t x and receiving coil r x They are all DD type coils, the auxiliary coil is a flat spiral coil, and the axial direction of the flat spiral coil is arranged parallel to the arrangement direction of the DD type coils.

[0052] In this embodiment, the transmitting coil t x The bottom and receiving coil r x A magnetic conductive core B1 and a shielding aluminum plate B2 are provided in sequence, and a transmitting coil t x and receiving coil r x It constitutes an energy transmission channel. Figure 5 As shown, the auxiliary coil c x The hollow flat winding method and relative position make c x With receiving coil r x The effect of mutual inductance can be ignored.

[0053] like Figure 6 As shown, the system also includes a dynamic tuning control circuit, a harmonic suppression control circuit and a data acquisition circuit; the data acquisition circuit is used to collect the inverter output current and voltage of the inverter circuit I to obtain the collected data. The harmonic suppression control circuit is used to calculate the duty cycle D according to the collected data and generate a control signal to drive the impedance matching circuit. The dynamic tuning control circuit is used to calculate the duty cycle α of the DC-DC conversion circuit according to the collected data. s , and generate a control signal to drive the DC-DC conversion circuit.

[0054] In this embodiment, the data acquisition circuit includes a voltage sensor and a current sensor, which are respectively used to collect the inverter output voltage and current of the inverter circuit I; the dynamic tuning control circuit includes a zero-crossing detection module, a phase detection module, a controller II and a drive circuit, the zero detection module is used to condition the measured output voltage signal into a square wave, the phase detection module is used to output the conditioned square wave to the output voltage of the inverter circuit I and the output phase difference θ of the current, and the controller II is used to linearly scan the output to drive the duty cycle α of the DC-DC conversion circuit s .

[0055] The harmonic suppression control circuit includes a Fourier analysis module, a controller I and a drive circuit; the Fourier analysis module is used to collect the inverter output current of the inverter circuit I and perform Fourier analysis to obtain the harmonic suppression ratio r a The controller I is used to control the harmonic suppression ratio r a The duty cycle D of the impedance matching circuit is linearly scanned.

[0056] In this embodiment, the auxiliary coil type dynamic tuning IPT system is analyzed:

[0057] 1. Analysis of dynamic tuning principle based on auxiliary coil:

[0058] like Figure 7 The fundamental wave equivalent circuit of the auxiliary coil type dynamic tuning IPT system is shown in FIG. is the reference phasor of the equivalent circuit, that is and Indicates that the excitation current passes through the mutual inductance M e and M c The induced voltage generated in the corresponding coil:

[0059]

[0060] In the formula, and Using associative reference direction, and A non-associative reference direction is used, so The sign is negative, The sign is positive.

[0061] L t0 is the self-inductance of the transmitting coil when the coupling mechanism is aligned, ΔL t is the change in the self-inductance of the transmitting coil compared to the positive alignment position, L r0 and ΔL r Then the corresponding self-inductance and self-inductance change of the receiving coil at the positive alignment position, the mutual inductance of the transmitting and receiving coils at the positive alignment position is expressed as M e0 , the change is ΔM e , from which we can get:

[0062]

[0063] according to Figure 7 Writing the KVL equation, we can get:

[0064]

[0065] Where Z L , Z C Respectively represent the impedance of the resonant inductor and capacitor, and the specific expression of the parameters is:

[0066]

[0067] The parameter configuration conditions of the LCC-S and LC resonant circuits are shown in equation (5):

[0068]

[0069] When the energy transmission channel is running alone and the tuning channel is not working, that is, When , the current of each circuit can be obtained by equations (1)-(5):

[0070]

[0071] In the equation, the subscript "e" of the current symbol in the equation group indicates that the current is supplied by the voltage source. Produced. Where A1 and X 1e The meaning of is shown in formula (7):

[0072]

[0073] When only the horizontal movement of the receiving mechanism is considered, R eqo Usually, equation (8) is satisfied, that is, A1≈R eqo 2.

[0074] R eqo 2 >>ω 2 △L r 2 (8)

[0075] The loop currents included in the energy transfer channel are shown in (a), (b) and (c) of equation (6). As can be seen from (c) of equation (6), the receiving coil current There is a real part and an imaginary part, the imaginary part of which depends only on ΔL r ; Energy transfer channel input current The imaginary part of ΔL t and ΔL r Related; Transmitter coil current The amplitude of is only related to the energy transfer voltage U and the inductance L f1 and angular frequency ω, but not ΔL t and ΔL r impact.

[0076] Equations (d) and (e) of (6) reflect the effect of the energy transfer channel on the tuning channel. Equation (6) (e) shows that the transmitting coil current No induced current will be generated in the auxiliary coil, because L f2 and C f2 The equivalent open circuit effect of the parallel resonant circuit formed. Equation (6) (d) shows that The corresponding induced voltage source A current will be formed on the input side of the tuning channel.

[0077] The input impedance angle of the energy transfer channel can be obtained by solving (a) of equation (6):

[0078]

[0079] Combining equation (2) and equation (6), equation (9) can be expressed as:

[0080]

[0081] From formula (10), we can see that the input impedance angle θ e At the same time, ΔL t , ΔL r and ΔM e The influence of these three factors means that when the relative positions of the transmitting and receiving coils change, the input impedance angle of the energy transfer channel based on the LCC-S resonant circuit changes accordingly.

[0082] According to formula (10), we can deduce ΔL under single action t , ΔL r and ΔM e The reactive power P caused in the energy transfer channel qe0 As shown in formula (11):

[0083]

[0084] Combining equations (10) and (11), we can see that ΔL t , ΔL r and ΔM e The three factors make the input impedance of the energy transfer channel unable to be in a zero phase angle state, and the input apparent power of the energy transfer channel contains reactive power and is thus in a detuned state.

[0085] When the tuning channel is running alone and the energy transmission channel is not working, that is, When , the current expressions of each loop can be deduced from equations (1)-(5):

[0086]

[0087] In the equation, the current subscript "c" represents the voltage source The loop current of the tuning channel is generated as shown in (d) and (e) of equation (12).

[0088] It can be seen that the input current of the tuning channel and auxiliary coil current Not affected by the energy transfer channel ΔL t , ΔL r and ΔM e The influence of the three, and only with the tuned voltage source Inductance L f2 , L c and angular frequency ω.

[0089] Equations (b) and (c) in (12) show that the tuning channel has no effect on the transmitting and receiving coil currents of the energy transfer channel. This means that the output power and transmission efficiency of the energy transfer channel are affected by the tuning voltage. The original value is always maintained during the real-time adjustment process, that is, the tuning channel will not interfere with the output. Similar to the situation of working alone, the auxiliary coil will form current on the input side of the energy transfer channel.

[0090] According to formula (12), the voltage source can be obtained The reactive power P injected into the tuned channel LCL network qc0 for:

[0091]

[0092] Combining equations (6) and (12), the currents of each branch of the energy transfer channel superimposed tuning effect can be obtained as:

[0093]

[0094] Under the superposition of the tuning channels, the input impedance angle θ of the energy transfer channel is:

[0095]

[0096] From formula (15), we can deduce that θ=0 corresponds to for:

[0097]

[0098] This means that when ΔL t , ΔL r and ΔM e When changes occur, adjust accordingly Satisfying equation (16) can ensure that the energy transfer channel is in a resonant state.

[0099] When the energy transfer channel is in the resonant state, the voltage source of the tuning channel can be obtained by combining equations (16) and (d) in (12): Only reactive power is injected and its expression P qc for:

[0100]

[0101] Among them, P qe0 The component is used to offset the reactive power inside the energy transfer channel, P qc0 The component provides reactive power to the tuning channel LCL network. qc0 =P qe0 , that is, the total reactive power P injected by the tuning channel qcis zero, and the parameter configuration conditions of the LCL network are:

[0102]

[0103] The specific expressions of the parameters are shown in formula (19), and γ p =1.

[0104]

[0105] This means that under this parameter condition, there is no reactive component in the input power of the energy transfer channel and the tuning channel. Combining equations (16) and (17), the input impedance Z of the tuning channel can be obtained: c for:

[0106]

[0107] Substituting equation (20) into equation (18), we can see that when γ p =1 when input reactance value X zc tends to infinity, at which point the tuning channel operates in an equivalent open-circuit state; when γ p >1, X zc >0, the input impedance of the tuning channel is inductive; otherwise, γ p <1X zc <0, the input impedance is capacitive. In order to ensure that the energy transfer channel is maintained in a resonant state within the offset range of the receiving mechanism, and the inverter circuit II of the tuning channel works in ZVS mode, the parameter configuration of the LCL network must meet γ p >1.

[0108] 2. Harmonic suppression analysis based on impedance matching:

[0109] 2.1 Analysis of the influence of high-order harmonics on the resonance state of the system: Within the offset range of the receiving mechanism, equations (5) and (16) give the resonance conditions of the system under the fundamental component. In addition, the output current of the energy transfer channel inverter circuit I It also contains triple and quintuple harmonic components, so The expression is:

[0110]

[0111] The KVL equation under high-order harmonics is thus written as equation (22), where the impedance of the inductor and capacitor under high-order harmonics, the mutual inductance M e and M c The generated induced voltage source is shown in equation (23). Substituting equations (5), (16) and (23) into equation (22), we can get the inverter output current Medium and high harmonic components. Since the third and fifth harmonic components are the dominant components, the influence of the third and fifth harmonic components on the resonant state of the system is mainly analyzed.

[0112]

[0113] When n = 1, the triple harmonic component can be obtained for:

[0114]

[0115] The real part R 1(3ω) , imaginary part X 1(3ω) The expression is:

[0116]

[0117] The parameter A in formula (25) (3ω) , B (3ω) , C (3ω) , D (3ω) 、E (3ω) and a (3ω) , b (3ω) 、c (3ω) d (3ω) As shown in equations (26) and (27).

[0118]

[0119] It can be seen from formula (25) that the triple harmonic component Contains a non-zero imaginary part X 1(3ω) , this non-zero imaginary part X 1(3ω) Will cause the output current to reverse With voltage This shows that the IPT system cannot operate in a resonant state under the action of the triple harmonic component.

[0120] When n = 2, the fifth harmonic component can be obtained for:

[0121]

[0122] The imaginary part X 1(5ω) The expression is:

[0123]

[0124] Parameter A in formula (29) (5ω) , B (5ω) , C (5ω) , D (5ω) 、E (5ω) and a (5ω) , b (5ω)、c (5ω) d (5ω) As shown in equations (30) and (31):

[0125]

[0126] Combining equations (25) and (29), it can be seen that the inverter output current The third and fifth harmonic components in the equation contain both real and imaginary parts, and the imaginary part X 1(3ω) , X 1(5ω) The non-zero imaginary parts of the triple and quintuple harmonic components will not only cause the inverter output current With voltage The phase difference makes the system unable to operate in a resonant state, and the excessively high harmonic current will also cause serious distortion of the inverter output current.

[0127] 2.2 Analysis of harmonic suppression principle based on impedance matching: In order to suppress the influence of current high-order harmonics on the system resonance state, this paper proposes a harmonic suppression method based on impedance matching. This method matches the equivalent load R eqo To reduce the harmonic content of the inverter current, so that the fundamental component of the inverter current dominates.

[0128] 2.2.1 Analysis of equivalent resistance R eqo And harmonic suppression ratio r a Relationship: Due to the tuning channel input voltage The imaginary part of the fundamental current is zero. According to the fundamental current (a) in equation (14): Expression to find the current Valid values ​​are:

[0129]

[0130] Combining equations (25), (26) and (27), we can obtain the triple harmonic current The effective value of I 1(3ω)RMS for:

[0131]

[0132] The parameter K i(3ω) (i=1,2,3,4) and the specific expressions of α, β, γ, λ are as shown in equations (34) and (35):

[0133]

[0134] In order to quantify R eqoThe suppression effect of harmonics is defined as the effective value ratio of the fundamental wave to the third harmonic, G, as shown in formula (36). The third harmonic is selected as the comparison object because when the proportion of the fundamental current is high enough compared to the harmonic current, the effect of the harmonic current on the input impedance angle of the system can be ignored; on the other hand, when the excitation voltage source of the LCC resonant network is in the form of a square wave, the amplitude of the third harmonic current of the LCC input current is higher than that of the fifth harmonic.

[0135]

[0136] To simplify the ratio G, set the parameters H and J to:

[0137]

[0138] For IPT systems, the transmitting coil's self-inductance usually satisfies:

[0139]

[0140] At this time, (d) of equation (38) can be approximated as λ = 8α, and parameter H = J. On this basis, let H > 10, and equation (37) can be obtained:

[0141]

[0142] When the constraint condition of formula (39) is satisfied, formula (36) can be simplified as:

[0143]

[0144] According to this, R eqo For r a The functional relationship is:

[0145]

[0146] When equation (41) satisfies the constraint condition:

[0147]

[0148] Then we can solve R eqo R a The derivative of is:

[0149]

[0150] From formula (43), we can see that R eqo R a The derivative result is less than 0, indicating that the equivalent resistance R eqo And harmonic suppression ratio r a It is a linear inverse proportional relationship. This shows that R eqo A smaller value of corresponds to a lower harmonic proportion.

[0151] 2.1.1 Equivalent impedance matching analysis: Based on the equivalent resistance R eqo And harmonic suppression ratio r a The mapping relationship between the rectifier circuit and the load end of the Boost conversion circuit can achieve equivalent impedance matching, such as Figure 8 shown.

[0152] Figure 8 R eqo With R L The equivalent relationship between them is:

[0153]

[0154] Where D is the duty cycle of the Boost converter circuit, and the value of D is between (0,1). eqo The equivalent relationship is:

[0155]

[0156] Substituting equation (41) into equation (45), we can get the duty cycle D and harmonic suppression ratio r a The relationship is:

[0157]

[0158] Formula (45) shows that R eqo Negatively linearly correlated with D, R eqo Depends on three variables M e , ΔL r and r a , M e and ΔL r The value range of is determined by the offset range of the receiving mechanism, and the harmonic suppression ratio r a is the set value. According to formula (41), R eqo The upper and lower limits of the value are obtained, and then the theoretical value range of the duty cycle D is obtained by equation (46). In addition, the upper limit of the duty cycle D is also limited by the saturation characteristics of the inductor magnetic ring, and its lower limit must comprehensively consider the disconnection delay time of the power switch tube. The offset range of the prototype in this paper is ±30% on the X axis and ±50% on the Y axis. Combined with the characteristics of the power devices and magnetic components used, the duty cycle D range of the impedance matching circuit is set to [0.1, 0.9].

[0159] Embodiment 2:

[0160] like Figure 5 The dynamic tuning method of an auxiliary coil type dynamic tuning IPT system shown in the embodiment 1 is an auxiliary coil type dynamic tuning IPT system, and the specific steps are:

[0161] S1: Set the initial system parameters; including the expected phase value θ ref、 Duty cycle α s The preset range and duty cycle α s The step value of

[0162] And the current control cycle phase value θ at the initial moment cur , the phase value of the upper wheel control cycle at the initial moment θ pre , the initial duty cycle α s , the intermediate quantity k of the control process at the initial moment and the scanning direction mark quantity d of the control process at the initial moment.

[0163] In this embodiment, the desired phase value θ is set ref =3°, duty cycle α s The preset range is: 0<α s <1, duty cycle α s The step value is 0.05. At the initial moment θ cur =θ pre =0, set the initial duty cycle α s =0.1, the intermediate quantity k of the control process at the initial moment is 0, and the scanning direction mark quantity d of the control process at the initial moment is 1.

[0164] S2: Collect the inverter output current of inverter circuit I and perform Fourier analysis to obtain the harmonic suppression ratio r a , according to the harmonic suppression ratio r a The duty cycle D of the impedance matching circuit is calculated, and a control signal is generated to drive the impedance matching circuit.

[0165] S3: Collect the inverter output voltage of inverter circuit I Calculate the output phase difference θ and the output phase difference θ and the expected phase value θ ref The difference is used as input to calculate the duty cycle α of the DC-DC converter circuit. s , and generate a control signal to drive the DC-DC conversion circuit. The specific steps are:

[0166] S3.1 collects the inverter output voltage of inverter circuit I The measurement signal is conditioned into a square wave by using the zero-crossing detection module, and the output phase difference θ between the output voltage and current of the inverter circuit I is output by the phase detection module;

[0167] S3.2 Determine that the output phase difference θ is less than the expected phase value θ ref Is it true? If so, the system meets the resonance condition and returns to step S3.1. If not, the system does not meet the resonance condition and returns to step S3.3.

[0168] S3.3 Determine the current control cycle phase value θ curThe absolute value of the control cycle is less than the phase value θ of the previous round pre Is it true? If so, go to step S3.4. If not, invert the scanning direction flag d and go to step S3.4.

[0169] S3.4 Determine the duty cycle α s Is it within the preset range? If so, go to step S3.5. If not, determine that the receiving circuit exceeds the preset range and cannot perform tuning control;

[0170] In this embodiment, when the receiving circuit exceeds the preset range and cannot be tuned and controlled, it means that the receiving mechanism is in a non-tunable position and the control process ends.

[0171] S3.5 Let K = K + 1, and set the current duty cycle α s With duty cycle α s The sum of the step values ​​is used as input to generate a control signal to drive the DC-DC conversion circuit, and then returns to step S3.1.

[0172] S4 simulation and experimental verification:

[0173] S4.1 parameter settings:

[0174] To verify the effectiveness of the proposed dynamic tuning method, a system model was built on the Ansys finite element simulation platform and Simulink circuit simulation tool. The physical dimensions of the coupling mechanism refer to the national standard GB / T 38775, and the relative position parameters of the tuning coil are optimized with the help of the Maxwell toolbox. The specific parameter values ​​are shown in Table 1.

[0175] Table 1 Dimensional parameters of magnetic coupling mechanism

[0176]

[0177] The dynamic offset process of the receiving end is simulated using Simplorer and Simulink simulation platforms. During the offset process, the simulation model takes the positive alignment position as the starting point, takes the length l2 of the energy transmission coil as the reference, and uses 5% of l2 (i.e. 15mm) as the step value. Due to the symmetry of the magnetic coupling mechanism, the simulation model selects five positions of positive alignment (0, 0), (45, 75), (90, 0), (0, 150) and (90, 150), and the unit of the position points is mm. The measured parameters of each position point are shown in Table 2. The parameters Lt (X, Y), Lr (X, Y) and Me (X, Y) in Table 2 represent the self-inductance and mutual inductance of the transmitting and receiving coils at different positions. The resonant circuit parameters of the simulation model are obtained according to the constraint formula (5), and the main component parameters of the energy transmission channel and the tuning channel are shown in Table 3.

[0178] Table 2 Self-inductance and mutual inductance parameters of magnetic coupling mechanism

[0179]

[0180]

[0181] Table 3 System resonant network parameters

[0182]

[0183] S4.2 Simulation verification:

[0184] The simulation results of the constructed system during the dynamic tuning process are shown in Fig. 9 shown. Fig. 9 (a) to (d) are phase difference θ, harmonic suppression ratio r a , equivalent resistance R eqo And tuning voltage U c Envelope diagram of the changes of four parameters. Fig. 9 In the example, the T1, T2 and T3 moments respectively represent the position of the receiving end shifting from (0, 150) to (90, 150), the impedance matching circuit performing linearization scanning, and adjusting the duty cycle α of the Sepic circuit. s .

[0185] In the period from 0 to T1, Fig. 9 (a) and (b) show that the system phase difference θ is 7.37° and the harmonic suppression ratio r a It is 10.04, that is, the ratio of the fundamental wave to the higher harmonic content is ten times the order of magnitude, the harmonics are effectively suppressed, and the system is in a resonant state. Fig. 9 (c) and (d) give the equivalent resistance R during this period. eqo is 4.202Ω, the tuning channel input voltage U c It is 79.56 V, which is consistent with the calculation results of equations (16) and (44).

[0186] When t = T1, the receiving end is offset to (90, 150). Fig. 9 (a) and (b) show that the system phase difference θ increases from 7.37° to 46.66°. At this time, the system is in a detuned state and the harmonic suppression ratio r a It decreased from 10.04 to 3.25, and the proportion of high-order harmonic content increased.

[0187] When t = T2, the receiving end Boost impedance matching circuit starts linearization scanning to obtain the required equivalent resistance R eqo .Depend on Fig. 9 It can be seen that after impedance matching, the equivalent resistance R eqo From 4.202Ω to 1.106Ω, the harmonic suppression ratio r aIt recovered from 3.25 to 10.54, and the phase difference θ decreased from 46.66° to 11.97°.

[0188] When t = T3, the Sepic circuit at the transmitter starts to adjust α s .Depend on Fig. 9 (d) It can be seen that the tuning channel U c From 79.56V to 40.24V, the harmonic suppression ratio is r a It is further reduced to 10.04, and the phase difference θ is reduced from 11.97° to 7.63°. At this time, the system is in a resonant state.

[0189] Fig.10 (a1), (b1) and (c1) are respectively the position change before and after T1, the impedance matching before and after T2 and the adjustment of α at T3. s The output voltage U of the inverter circuit before and after et and output current I 1t Waveform diagram of Fig.10 (a2) and (b2) are the input voltage U of the receiving end rectifier circuit before and after the position change at time T1 and before and after the impedance matching at time T2. rec and input current I rec Waveform diagram of Fig.10 (c2) is the adjustment of α at time T3 s U before and after c And the change diagram of I4 waveform.

[0190] Depend on Fig.10 (a1) It can be seen that before time T1, U et with I 1t The waveforms remain in phase, indicating that the system is in a resonant state; after time T1, the receiving end shifts to (90, 150), U et and I 1t The waveform begins to have a phase difference, and U rec and I rec The waveforms always remain in phase.

[0191] Depend on Fig.10 (b1) shows that the system is in a detuned state before time T2, and I 1t The waveform is distorted; after time T2, the receiving end performs impedance matching, and the high-order harmonics are suppressed. 1t The waveform distortion is alleviated, but the voltage U et and current I 1t There is still a phase difference.

[0192] Depend on Fig.10 (c1) It can be seen that the voltage U before T3 et and current I 1tThere is a phase difference of about 12°; after time T3, the Sepic circuit of the tuning channel starts to adjust the duty cycle α s , after which the phase difference decreases to about 7°, at which point the system can be considered to be in a resonant state.

[0193] Depend on Fig.10 (a2) and Fig.10 (b2) It can be seen that the input voltage U of the receiving end rectifier circuit rec and current I rec are all in phase, indicating that the receiving end always picks up active power before and after impedance matching; Fig.10 (c2) It can be seen that the tuning channel input I4 is relatively large compared to the voltage U c 90° ahead, and in U c The positive and negative half-cycles are symmetrical, indicating that the tuning channel only injects reactive power to compensate the energy transfer channel.

[0194] Fig.11 The simulation results of the receiving end before and after tuning in the range of (-90, 90) on the X-axis and Y-axis. Fig.11 (a) is the phase difference θ before and after tuning corresponding to the position of the receiver in the XOY plane. When the receiver is in an offset position, the phase difference θ before tuning is above 15°, and θ increases with the increase of the offset distance. The maximum θ value is about 27°, which is at the four diagonal positions of (-90, -90), (-90, 90), (90, -90), and (90, 90). At this time, the system is in a detuned state. After the dynamic tuning method, the θ value in the XOY plane is reduced to about 7°.

[0195] Fig.11 (b) is the harmonic suppression ratio r a Changes before and after tuning on the XOY plane. When the receiving end is offset, the r a are all below 4 and decrease with the increase of the offset degree. In this state, the proportion of high-order harmonics increases. a The value is about 1, corresponding to the four diagonal positions of the receiving position area. At this time, the inverter output current is seriously distorted. After dynamic tuning, all r a When it rises to about 10, the proportion of high-order harmonics is effectively suppressed.

[0196] From the above analysis, it can be seen that the dynamic tuning method proposed in the present application can achieve high-order harmonic suppression and ensure that the system is in a resonant state within the offset position range of the receiving end.

[0197] S4.3 Experimental verification:

[0198] Comprehensive parameters in Table 1-Table 3, based on Figure 6The dynamic tuning control system shown is based on a DD-type auxiliary coil IPT system.

[0199] like Fig.12 The voltage and current waveforms of the inverter output side and the rectifier bridge input side of the energy transfer channel and tuning channel are measured, and the inverter voltage U et With current I 1t The phase difference θ and total harmonic distortion THD are measured. The experimental results are obtained with the help of a DC power supply (IT7805), an electronic load (IT8817), an oscilloscope (DS7054) and an impedance meter (E4980AL). During the experiment, the receiving mechanism is offset by ±30% and ±50% of the coil length on the X-axis and Y-axis respectively. Considering the symmetry of the magnetic coupling mechanism, the experimental waveforms at the four positions of (0, 0), (0, 150), (45, 75) and (90, 150) are measured.

[0200] Fig.12 (a0)-(a3) are the voltage and current waveforms of the inverter output side and the rectifier input side of the energy transfer channel at four positions without impedance matching. The equivalent resistance R eqo Both are 20Ω. Fig.12 (a0) shows that the system phase difference θ at the positive alignment position is 13°, THD = 0.158, and the current I 1t There is distortion. Fig.12 From (a1) to (a3), it can be seen that as the offset distance of the receiving end increases, the THD of the current waveform increases accordingly. The THD at the position (90, 150) reaches 0.3076.

[0201] Fig.12 (a0)-(a3) are the U values ​​at four locations after impedance matching. et ,I 1t , U rec with I rec Waveform. Fig.12 (b0)-(b3) and Fig.12 From the horizontal comparison between (a0) and (a3), it can be seen that after impedance matching, the I 1t The current THD is reduced by 0.058, 0.1041, 0.1045 and 0.2079 respectively, and the corresponding Boost circuit duty cycle D is 0.22, 0.52, 0.48 and 0.76 respectively. The experimental value is consistent with the calculated value of formula (46), which verifies the effectiveness of the harmonic suppression method based on impedance matching. In addition, at the positive alignment position, the phase difference θ is reduced to 2°, while the inverter output phase difference θ at the three positions of (0, 150), (45, 75) and (90, 150) is still about 12°.

[0202] Fig.12 (c1)-(c3) are the inverter output voltage and current waveforms of the energy transfer and tuning channels after the auxiliary coil is running. Fig.12 (c1)-(c3) and Fig.12 From the comparison between (b1) and (b3), we can see that by tuning the auxiliary coil on the basis of impedance matching, the current phase difference θ at the four positions is reduced from 12° to 2°, and the system can be regarded as operating in a resonant state. The corresponding Sepic circuit duty cycle α s 0.53, 0.34 and 0.16 respectively. The experimental values ​​are consistent with the simulation values, which further verifies the accuracy of the dynamic tuning method based on the auxiliary coil. In addition, the voltage and current of the energy transfer channel after dynamic tuning of the auxiliary coil still maintain a phase difference of 2°, which is to make the resonant network weakly inductive to achieve zero voltage turn-on of the power device of the transmitter inverter circuit.

[0203] Fig.13 The dynamic waveform of the impedance matching circuit and the tuning channel operation process at the position (90, 150). Fig.13 (a) It can be seen that the adjustment time of the impedance matching circuit is 5ms, and during the adjustment process, the output of the inverter and the input of the rectifier circuit do not overshoot. This is because the duty cycle D of the impedance matching circuit is obtained through a linear scanning method. Fig.13 (b) shows that the operation time of the tuning channel is 40ms, and there is no overshoot during the tuning process, which proves the feasibility of the tuning control loop. Fig.13 (a) and (b) show that the time it takes for the tuning channel to reach stability is 8 times the impedance matching adjustment time. This is because the tuning process includes zero-crossing detection of voltage and current and phase detection, and the signal conditioning and processing time in the detection process are several milliseconds.

[0204] It should be noted that the receiving coil L r With auxiliary coil L c The mutual inductance simulation value between the two coils is always lower than 1nH within the offset range, and the experimental measurement value is lower than 0.7μH. e and auxiliary coil mutual inductance M c , L r and L c The mutual inductance is one order of magnitude lower; in addition, the input of the tuning circuit does not affect L c The output voltage amplitude of the receiving end is affected; the magnitude of the comprehensive inductance and the experimental results show that L r With L c The effect of mutual inductance can be ignored.

[0205] In summary, this application proposes an auxiliary coil type dynamic tuning IPT system and its dynamic tuning method, reveals the power distribution mechanism of the two energy channels where the auxiliary coil and the energy transmission coil are located, derives the load equivalent resistance corresponding to the high-order harmonic suppression ratio of the energy transmission channel, and provides a tuning channel input voltage control method that can both meet the resonance conditions of the energy transmission channel and achieve high-order harmonic suppression. The experimental results show that within the set offset range, the proposed dynamic tuning method can ensure that the IPT system is always in a resonant state, and the proportion of high-order harmonics is lower than the required value. The constructed auxiliary coil type dynamic tuning method can realize stepless continuous tuning of the energy transmission channel without changing the topological form of the energy transmission channel and has no effect on the pickup power. The experimental process found that the asynchronous driving of the two channels and the manufacturing error of the resonant element will cause the energy transmission channel to generate an anti-phase circulation at the excitation source of the tuning channel, and this circulation increases the power device loss of the tuning channel. In order to eliminate the influence of the anti-phase circulation, the driving phase of the two-channel inverter can be corrected accordingly. This part of the research content will be carried out in the subsequent work of this article.

[0206] Finally, it should be noted that the above embodiments are only used to illustrate the technical solutions of the present invention rather than to limit it. Although the present invention has been described in detail with reference to the above embodiments, ordinary technicians in the relevant field should understand that the specific implementation methods of the present invention can still be modified or replaced by equivalents. Any modification or equivalent replacement that does not depart from the spirit and scope of the present invention should be covered within the scope of protection of the claims of the present invention.

Claims

1. An auxiliary coil type dynamic tuning IPT system, comprising a transmitting circuit and a receiving circuit, wherein the transmitting circuit comprises a DC power supply, an inverter circuit I, a transmitting end compensation circuit and a transmitting coil, and the receiving circuit comprises a receiving coil, a receiving end compensation circuit, a rectifier filter circuit and a load, characterized in that: Also includes tuning circuits and impedance matching circuits; The impedance matching circuit is arranged between the rectification and filtering circuit and the load, the tuning circuit includes a DC-DC conversion circuit, an inverter circuit II, an auxiliary compensation circuit and an auxiliary coil which are connected in sequence, the input end of the DC-DC conversion circuit is connected to the DC power supply, and the auxiliary coil is arranged on the transmitting coil.

2. The auxiliary coil type dynamic tuning IPT system according to claim 1, characterized in that: The transmitting coil and the receiving coil are both DD-type coils, the auxiliary coil is a flat spiral coil, and the axial direction of the flat spiral coil is arranged in parallel with the arrangement direction of the DD-type coils.

3. The auxiliary coil type dynamic tuning IPT system according to claim 2, characterized in that: The impedance matching circuit is a Boost conversion circuit, and the DC-DC conversion circuit is a Sepic conversion circuit.

4. The auxiliary coil type dynamic tuning IPT system according to claim 2, characterized in that: The transmitting end compensation circuit and the receiving end compensation circuit constitute an LCC-S resonant circuit, and the auxiliary compensation circuit is an LC resonant circuit.

5. The auxiliary coil type dynamic tuning IPT system according to claim 1, characterized in that: It also includes a dynamic tuning control circuit, a harmonic suppression control circuit and a data acquisition circuit; The data acquisition circuit is used to collect the inverter output current and voltage of the inverter circuit I to obtain the collected data. The harmonic suppression control circuit is used to calculate the duty cycle D of the impedance matching circuit according to the collected data and generate a control signal to drive the impedance matching circuit. The dynamic tuning control circuit is used to calculate the duty cycle α according to the collected data. s , and generate a control signal to drive the DC-DC conversion circuit.

6. A dynamic tuning method for an auxiliary coil type dynamic tuning IPT system, characterized in that: The auxiliary coil type dynamic tuning IPT system according to any one of claims 1 to 5 is adopted, and the specific steps are as follows: S1: Set the initial system parameters; S2: Collect the inverter output current of inverter circuit I and perform Fourier analysis to obtain the harmonic suppression ratio r a , according to the harmonic suppression ratio r a Calculating a duty cycle D of the impedance matching circuit and generating a control signal to drive the impedance matching circuit; S3: Collect the inverter output voltage of inverter circuit I Calculate the output phase difference θ and the output phase difference θ and the expected phase value θ ref The difference is used as input to calculate the duty cycle α of the DC-DC converter circuit. s , and generate a control signal to drive the DC-DC conversion circuit.

7. The dynamic tuning method of the auxiliary coil type dynamic tuning IPT system according to claim 6, characterized in that: The initial system parameters in step S1 include the expected phase value θ ref、 Duty cycle α s The preset range and duty cycle α s The step value of And the current control cycle phase value θ at the initial moment cur , the phase value of the upper wheel control cycle at the initial moment θ pre , the initial duty cycle α s , the intermediate quantity k of the control process at the initial moment and the scanning direction mark quantity d of the control process at the initial moment.

8. The dynamic tuning method of the auxiliary coil type dynamic tuning IPT system according to claim 6, characterized in that: The specific steps in step S3 are: S3.1 collects the inverter output voltage of inverter circuit I The measurement signal is conditioned into a square wave by using the zero-crossing detection module, and the output phase difference θ between the output voltage and current of the inverter circuit I is output by the phase detection module; S3.2 Determine that the output phase difference θ is less than the expected phase value θ ref Is it true? If so, the system meets the resonance condition and returns to step S3.

1. If not, the system does not meet the resonance condition and returns to step S3.

3. S3.3 Determine the current control cycle phase value θ cur The absolute value of the control cycle is less than the phase value θ of the previous round pre Is it true? If so, go to step S3.

4. If not, invert the scanning direction flag d and go to step S3.

4. S3.4 Determine the duty cycle α s Is it within the preset range? If so, go to step S3.

5. If not, determine that the receiving circuit exceeds the preset range and cannot perform tuning control; S3.5 Let K = K + 1, and set the current duty cycle α s With duty cycle α s The sum of the step values ​​is used as input to generate a control signal to drive the DC-DC conversion circuit, and then returns to step S3.1.

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