A method for determining switching waveforms of two-level PWM converters for electromagnetic interference prediction

Abstract

The present invention discloses a method for determining a switching waveform of a two-level PWM converter for electromagnetic interference prediction, which belongs to the field of electromagnetic compatibility of power electronics. The method is based on solving steady-state and transient quantities of a state equation group, and can efficiently obtain a switching waveform containing nonlinear transients. After conversion and detection processing of an equivalent interference source, a wideband, high-precision switching waveform spectrum for electromagnetic interference prediction can be obtained. Thanks to the specified initial values ​​of state variables, nonlinear device models, simplified calculation of steady-state values ​​of state variables, and parallel calculation of transient values, the method has the advantages of good convergence, high precision, and high computational efficiency. It realizes the calculation of the switching waveform spectrum for electromagnetic interference prediction from the physical-level parameters of the switching device, with a spectrum range covering 0-1 GHz and a calculation time of only one hundred seconds. This can improve the speed of electromagnetic interference modeling, enhance prediction accuracy, and ultimately accurately guide the electromagnetic interference suppression design of the two-level PWM converter.

Description

Technical Field

[0001] The present invention belongs to the field of power electronic electromagnetic compatibility, and more particularly relates to a method for determining a switching waveform of a two-level PWM converter for electromagnetic interference prediction. Background Art

[0002] In recent years, two-level PWM converters have been widely used in modern power systems, rail transit, and consumer electronics. However, the inherent switching operation of two-level PWM converters can cause serious electromagnetic interference (EMI) issues, potentially endangering the proper operation of surrounding electronic equipment or affecting the reliability and lifespan of electronic systems. Before deployment, two-level PWM converters must pass stringent EMC certification. Effective EMI suppression design is essential for successful EMC certification. Compared to prototype testing-based suppression design, model-based suppression design requires less effort and is more efficient, offering a wider range of feasible solutions and ultimately improving product performance. Consequently, EMI modeling for two-level PWM converters is currently attracting widespread attention and application.

[0003] Electromagnetic interference modeling for two-level PWM converters uses the physical parameters of active and passive components, utilizing field / path simulation or code programming, to predict the electromagnetic interference spectrum measured at the LISN or antenna during operation of the proposed two-level PWM converter prototype. Electromagnetic interference modeling primarily involves modeling interference sources and interference paths. Compared to the well-established modeling of interference paths, modeling of interference sources remains understudied. Interference source modeling aims to obtain the spectral amplitude of the switching waveform. Switching waveform modeling for electromagnetic interference prediction, also known as interference source modeling, establishes a functional model linking the physical parameters of the switching devices and associated auxiliary circuits with the spectral amplitude of the switching waveform during operation.

[0004] Switching waveform modeling is crucial for electromagnetic interference (EMI) modeling of two-level PWM converters. It not only impacts the design of converter port EMI filters but is also closely related to the design of driver and snubber circuits, which significantly impact converter efficiency. Switching waveform modeling for EMI prediction is primarily categorized into three methods: time domain, frequency domain, and time-frequency domain. Time domain modeling methods based on nonlinear switching waveforms offer slightly higher accuracy, but suffer from extremely long simulation times and poor convergence. Frequency domain modeling methods based on ideal linear switching waveforms offer shorter simulation times but lower accuracy. There is also a class of time-frequency domain modeling methods that combine the advantages of both methods, but these methods are complex, have poor reproducibility, fail to account for the nonlinear effects of switching devices, and still exhibit low accuracy. Furthermore, these methods are limited to a very low frequency range, achieving EMI predictions up to approximately 10 MHz. However, the required frequency range for EMI modeling of two-level PWM converters is 0 Hz to 1 GHz, and no model yet exists that can achieve such a wide frequency range. In summary, current switching waveform models for EMI prediction still suffer from low accuracy and efficiency. Summary of the Invention

[0005] In response to the above defects or improvement needs of the prior art, the present invention provides a method for determining a two-level PWM converter switching waveform for electromagnetic interference prediction, thereby solving the problems of low accuracy and low efficiency of the existing two-level PWM converter switching waveform modeling methods.

[0006] To achieve the above object, according to a first aspect of the present invention, a method for determining a switching waveform of a two-level PWM converter for electromagnetic interference prediction is provided, comprising:

[0007] S1, compare the modulated wave of the converter with the carrier wave to obtain the steady-state value V of the gate voltage of the first and second switching tubes in the converter gs1_sta 、V gs2_sta , and according to V gs1_sta 、V gs2_sta and load current I L Determine the steady-state value V of the voltage across the first and second switching tubes ds1_sta 、V ds2_sta and the steady-state value of the output current I s1_sta , I s2_sta ;

[0008] Among them, at a certain moment, if the modulation wave amplitude is greater than the carrier amplitude, then at that moment, V gs1_sta =V H , V gs2_sta =V L , I s1_sta =-I L , I s2_sta =0, when I L ≤0, V ds1_sta=V moson 、V ds2_sta =V DC -V moson , when I L >0, V ds1_sta =-V Don 、V ds2_sta =V DC +V Don If the modulation wave amplitude is less than or equal to the carrier amplitude, then at this moment, V gs1_sta =V L , V gs2_sta =V H , I s1_sta =0, I s2_sta =I L , when I L When V ds1_sta =V DC +V Don 、V ds2_sta =-V Don , when I L When it is negative, V ds1_sta =V DC -V moson 、V ds2_sta =V moson ; V H 、V L They are driving voltage high and low level respectively, V DC 、V Don 、V moson They are input voltage, diode conduction voltage, and MOS tube conduction voltage respectively.

[0009] S2, taking the voltage across both ends, gate voltage and output current of each switch tube as its state variables, and determining the transient initial value and driving voltage of the state variables of each switch tube;

[0010] Among them, the transient initial values ​​of the voltage across the two ends and the output current of each switch tube are the steady-state values ​​of the voltage across the two ends and the output current at the transient start time n respectively; if I L If it is positive, the initial transient values ​​of the gate voltage and drive voltage of the first switch tube are both V L , the transient initial values ​​of the gate voltage and driving voltage of the second switch tube are the steady-state values ​​of the gate voltage at time n and time n+1 respectively; if I L If negative, the transient initial values ​​of the gate voltage and drive voltage of the first switch tube are the steady-state values ​​of the gate voltage at time n and time n+1 respectively, and the transient initial values ​​of the gate voltage and drive voltage of the second switch tube are both V L ;

[0011] S3, establishing a differential equation group of the state variables of each switch tube with respect to the transient moment based on the switch tube model, the diode model, and the voltage equation of the junction capacitance, and solving the differential equation group in combination with the transient initial value of the state variable of each switch tube and the driving voltage to obtain the transient value of each state variable;

[0012] S4, splicing the steady-state value and transient value of each state variable of each switching tube to obtain the complete time domain waveform of each state variable, so as to calculate the common-mode and differential-mode interference source switching waveforms of the converter, and performing detection processing on them to obtain the two-level PWM converter switching waveform for electromagnetic interference prediction.

[0013] According to a second aspect of the present invention, there is provided an electronic device comprising: a computer-readable storage medium and a processor;

[0014] The computer-readable storage medium is used to store executable instructions;

[0015] The processor is configured to read the executable instructions stored in the computer-readable storage medium and execute the method according to the first aspect.

[0016] According to a third aspect of the present invention, a computer-readable storage medium is provided, wherein the computer-readable storage medium stores computer instructions, and the computer instructions are used to enable a processor to execute the method according to the first aspect.

[0017] According to a fourth aspect of the present invention, there is provided a computer program product comprising a computer program or instructions, which implement the method according to the first aspect when executed by a processor.

[0018] In general, the above technical solutions conceived by the present invention can achieve the following beneficial effects compared with the prior art:

[0019] 1. The method provided by the present invention calculates the initial value of the state variable of the switch tube in a specified manner rather than iteratively, ensuring that the I / V characteristics of the nonlinear device (switching devices such as IGBT or MOS and diode) are in the correct on or off range, thereby ensuring the computational convergence of the switching waveform model and avoiding the computational errors of the existing time-domain modeling method that do not converge. The method provided by the present invention can realize the switching waveform calculation with both computing power selectivity and support for parallel processing. In the steady state without switching, the state equation group is not solved, and the state equation group is solved only in the switching transient state, focusing the limited computing power on the switching transient state that requires high precision, thereby improving the computational efficiency. Moreover, the calculations of different switching transients are independent of each other, and parallel computing can be used to further improve the computational efficiency. In addition, the method provided by the present invention takes into account the voltage-dependent characteristics of the device junction capacitance when establishing the differential equation group of the state variable, which can further improve the computational accuracy. Simulation verification shows that the computational efficiency (minute level) and convergence of this method far exceed those of existing foreign commercial software such as Pspice (hour level).

[0020] 2. Furthermore, the method provided by the present invention takes into account that the existing transconductance model of the switching device only considers the device drain current I d With the gate voltage V gs The change of drain-source voltage V ds The effect can only be seen on the drain-source voltage V ds It is used in a larger amplification area and is not applicable in the device conduction stage or the initial stage of shutdown. The transconductance model is improved and the improved model also takes V ds and V gs Drain current I d It is applicable to all stages of the operation of switching devices and has higher accuracy and greater versatility.

[0021] 3. The method provided by the present invention uses the Runge-Kutta algorithm to solve the state equation group to obtain the transient value of the state variable, which has higher accuracy and is not prone to divergence.

[0022] 4. The method provided by the present invention takes into account the skin effect of ringing when solving the state equations. When the switching waveform enters the oscillation stage, the skin resistance is made effective, which can further improve the calculation accuracy. BRIEF DESCRIPTION OF THE DRAWINGS

[0023] Figure 1 A flow chart of a method for determining a switching waveform of a two-level PWM converter for electromagnetic interference prediction provided by an embodiment of the present invention;

[0024] Figure 2 (a) and (b) are schematic diagrams of function curves when the switching device model adopts a transconductance model and an improved model provided by an embodiment of the present invention, respectively;

[0025] Figure 3 (a) and (b) are Figure 2 Schematic diagram of the input, output and reverse transfer capacitance voltage-variation curves and the converted junction capacitance voltage-variation curves provided in the datasheet of the voltage-variation junction capacitance model of the IGBT or MOS device;

[0026] Figure 4 (a) and (b) are schematic diagrams of a diode model using equivalent junction capacitance to characterize reverse recovery and a schematic diagram of a diode forward characteristic curve, respectively;

[0027] Figure 5 (a) and (b) are Figure 4 Schematic diagram of the switching device and the combined model of the anti-parallel diode junction capacitance;

[0028] Figure 6 Schematic diagram of the switching transient circuit model of a two-level PWM converter;

[0029] Figure 7 (a) and (b) are typical topology diagrams of single-phase two-level PWM converter and three-phase two-level PWM converter respectively.

[0030] Figure 8 The state variable V provided in the embodiment of the present invention ds Waveform diagram;

[0031] Figure 9 The state variable I provided in the embodiment of the present invention s Waveform diagram;

[0032] Figure 10 A schematic diagram of the spectrum of a conducted common-mode interference source provided by an embodiment of the present invention;

[0033] Figure 11 A schematic diagram of the spectrum of a conducted differential mode interference source provided by an embodiment of the present invention;

[0034] Figure 12 A schematic diagram of the source frequency of radiated common-mode interference provided by an embodiment of the present invention;

[0035] Figure 13 A schematic diagram of a spectrum of a radiated differential mode interference source is provided for an embodiment of the present invention. DETAILED DESCRIPTION

[0036] In order to make the objectives, technical solutions and advantages of the present invention more clearly understood, the present invention is further described in detail below with reference to the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are merely for the purpose of explaining the present invention and are not intended to limit the present invention. In addition, the technical features involved in the various embodiments of the present invention described below may be combined with each other as long as they do not conflict with each other.

[0037] Due to the inherent switching operation of two-level PWM converters, they generate strong electromagnetic interference during operation. Improperly suppressed electromagnetic interference can severely pollute the surrounding electromagnetic environment and power supply system through conduction and radiation coupling. The electromagnetic interference suppression design of two-level PWM converters relies heavily on accurate switching waveform models. Currently, frequency-domain models based on ideal switching waveforms lack consideration of device nonlinearities and ringing effects, resulting in low accuracy. Existing time-domain models based on nonlinear switching waveforms suffer from poor computational convergence and inefficiency. This is especially true for radiated electromagnetic interference switching waveforms that require extremely high prediction frequencies, making it impossible to calculate correct results. Therefore, existing switching waveform determination methods for electromagnetic interference prediction suffer from low accuracy and computational efficiency.

[0038] An embodiment of the present invention provides a method for determining a switching waveform of a two-level PWM converter for electromagnetic interference prediction, comprising:

[0039] S1, compare the modulated wave of the converter with the carrier wave to obtain the steady-state value V of the gate voltage of the first and second switching tubes in the converter gs1_sta 、V gs2_sta , and according to V gs1_sta 、V gs2_sta and load current I L Determine the steady-state value V of the voltage across the first and second switching tubes ds1_sta 、V ds2_sta and the steady-state value of the output current I s1_sta , I s2_sta ;

[0040] At a certain moment, if the modulation wave amplitude is greater than the carrier amplitude, then at that moment, V gs1_sta =V H , V gs2_sta =V L , I s1_sta =-I L , I s2_sta =0, when I L ≤0, V ds1_sta =V moson 、V ds2_sta =V DC -V moson , when I L >0, V ds1_sta =-V Don 、V ds2_sta =V DC +V Don If the modulation wave amplitude is less than or equal to the carrier amplitude, then at this moment, V gs1_sta =V L , V gs2_sta=V H , I s1_sta =0, I s2_sta =I L , when I L When V ds1_sta =V DC +V Don 、V ds2_sta =-V Don , when I L When it is negative, V ds1_sta =V DC -V moson 、V ds2_sta =V moson ; V H 、V L They are driving voltage high and low level respectively, V DC 、V Don 、V moson They are input voltage, diode voltage, and MOS tube conduction voltage respectively.

[0041] Specifically, in step S1, a steady-state value of a state variable is generated by coding according to a modulation strategy.

[0042] In the present invention, the state variable refers to the voltage V across each switch tube. ds , gate voltage V gs , the output current of the switch tube I s ; Among them, V gs For calculating V ds and I s The intermediate variable of the two-level PWM converter; Since the two-level PWM converter has two switching tubes (MOS tube or IGBT), the switching waveform of the two-level PWM converter refers to the V ds and I s waveform.

[0043] Since the forward voltage V of the diode and MOS tube is calculated Don and V moson The diode model and the switch device model are needed when constructing the differential equation group in step S3, and the diode model, the switch device model and the junction capacitance voltage change model of the switch tube are needed. Therefore, the above models are first described below. It is worth noting that the above models can all adopt existing models (as shown in equations (1), (3) and (5)). Preferably, the switch device model can adopt the improved model of the present invention (as shown in equation (2)).

[0044] Extract physical parameters from the data sheet of the switching device. Extract the switching tube model based on the output and transfer characteristic curves. d =g(V ds ,V gs), extract the diode model I according to the diode forward characteristics D =f(V D ) According to the curve of junction capacitance and voltage across the switch tube, the voltage-dependent characteristic curve of junction capacitance C=p(V ds ). And based on the diode model and the switch tube model, calculate the steady-state voltage and current of each switch tube when there is no transient switching process and it is only in the on or off state.

[0045] 1. Output / transfer characteristic model of IGBT or MOS device (i.e. switching device model)

[0046] The transconductance model widely used in the existing switching waveform modeling method is shown in formula (1). The present invention takes into account that the transconductance model (1) only considers the device drain current I d With the gate voltage V gs The change of drain-source voltage V ds The effect can only be seen on the drain-source voltage V ds Therefore, preferably, the switching device model used in the present invention is an improved model, which takes into account V ds and V gs Drain current I d The model is applicable to all stages of the operation of switching devices, has higher accuracy and greater versatility, and improves the model accuracy of the device in the saturation region.

[0047] The improved model is shown in formula (2). In these two expressions, V gsth , V dsth , K is a parameter that needs to be fitted according to the device data sheet. Figure 2 (a) and (b) compare the two model function curves and point out the key parameter V of the model. gsth , V dsth The remaining parameter K needs to be calculated by reading a point in the data sheet.

[0048] I mos =g(V gs -V gsth ) (1)

[0049]

[0050] Among them, I mos is the channel current of the switch tube, V gs 、V ds They are the gate voltage and the voltage across the switch tube, V gsth 、V dsth , K are fitting parameters, V this the gate threshold voltage, ∨ and ∧ are the calculation symbols of or and respectively.

[0051] 2. Junction capacitance voltage change model of IGBT or MOS device

[0052] The junction capacitance of the device is closely related to the rising and falling slopes of the switching transient. The voltage-dependent effect of the junction capacitance leads to the variable slope of the switching transient. The voltage-dependent effect is characterized by formula (3). The two waveform parameters C gd0 and m gd Need to be extracted from the data sheet. Figure 3 (a) in the figure plots the measured capacitance versus drain-source voltage curve provided in the data sheet. Figure 3 (b) in the figure plots the Figure 3 The junction capacitance of the device after the conversion of the data in (a). The intersection of the curve and the y-axis is C gd0 and C ds0 , and one more data point needs to be read to complete the modeling of the voltage-variation effect. Since the present invention is applicable to both IGBT and MOS, and the modeling method is the same, all the following discussions of the present invention will only introduce MOS tubes.

[0053]

[0054] 3. Device model of anti-parallel diode

[0055] The diodes in the two-level PWM converter are anti-parallel diodes. The forward conduction model of the diodes is selected from equations (4) and Figure 4 The broken line model shown in (a) in Figure 1 is also shown. It should also be noted that the diode also has junction capacitance, which is the direct cause of the reverse recovery characteristics. The modeling of the junction capacitance can be based on the recovery charge Q provided in the data sheet. R and the reverse voltage V when measuring R Determine, as shown in formula (5). Due to the junction capacitance C of the diode D and MOS tube junction capacitance C ds Since they are in parallel, it is not necessary to calculate the diode junction capacitance C separately. D The current is combined to calculate C D +C ds The current can be Figure 5 As shown in (a) and (b) in .

[0056] This capacitance is hereinafter referred to as C' ds

[0057]

[0058] C d ' s =C D +C ds (6)

[0059] In step S1, the voltage V ds , the encoding generates the known modulation wave and carrier sequence, and the two sequences are compared to obtain V ds The ideal square wave of the square wave sequence is generated in parallel, which is much more efficient than the traditional time domain step simulation. ds The amplitude of the square wave depends on whether the MOS tube is in forward conduction or the diode is in freewheeling. gs The method of obtaining V ds The acquisition method is similar to that of the modulation wave and carrier wave, which are generated by coding. The two sequences are compared and driven by the high level V H and driving low level V L , we get V gs For the ideal square wave of the switch tube output current I s The output current waveform amplitude of the switch tube is determined according to the current on or off state of the switch tube and the positive direction of the device current. This waveform is also generated in parallel, with high calculation efficiency.

[0060] The ideal switching waveform is generated by coding rather than circuit simulation, which makes it easy to modify the waveform parameters. The pseudo code of the coding algorithm is as follows.

[0061] Algorithm: Calculation of steady-state values ​​of state variables.

[0062] Input: Input voltage V DC , output voltage V o , load current effective value I Load , power factor PF, switching frequency F s , modulation wave frequency F u .

[0063] Output: The steady-state matrix V of the state variables of the two switching tubes ds_sta =[V ds1_sta ,V ds2_sta ],V gs_sta =[V gs1_sta ,V gs2_sta ], I s_sta =[I s1_sta ,I s2_sta ],V ds_sta ,V gs_sta ,I s_sta Each element is a row with 1 and column with f s / F u Array of .

[0064] 1. Initialize waveform duration t max 1 / F u , sampling frequency f sFor 2GHz, the calculation time vector t=[1 / f s :1 / f s :t max ], length is N. According to the input voltage V DC Output voltage V o Calculate the modulation ratio m = 1.414 × V o / V DC .

[0065] 2. According to the power factor PF, modulation wave frequency F u and load current effective value I Load Generate load sinusoidal current time domain waveform I L Vector, the length of the vector is N.

[0066] 3. Generate switching frequency F s Triangular carrier, with amplitude [0,1]. Carrier is a vector of N.

[0067] 4. The single-phase two-level PWM converter generates a phase with an amplitude of m and a frequency of F according to the sinusoidal modulation (SPWM) strategy. u The sinusoidal modulation wave D, that is, D = sin(2×F u t). If it is a three-phase two-level PWM converter, the amplitude of one phase is m and the frequency is F according to the space vector modulation (SVPWM) measurement. u SVPWM sinusoidal modulation wave D ra , the formula is:

[0068]

[0069] 5. Compare the amplitudes of the modulation wave D and the carrier at each moment one by one, and combine the high level of the driving voltage V H and low level V L Get V gs =[V gs1 ,V gs2 ]. That is: when D is greater than Carrier, V gs1 =V H , V gs2 =V L , when D is less than or equal to Carrier, V gs1_sta =V L , V gs2_sta =V H , as shown in Table 1.

[0070] 6. According to the load current I L , diode model V D =f -1 (I D) and switch device models (e.g., MOS device model V ds =g -1 (I mos ,V gs )) Calculate the forward voltage V of the diode and MOS tube Don and V moson . Both are vectors of length N.

[0071] 7. According to V gs High or low level and load current I L Is it positive or negative? Determine the conduction device at this time and calculate the voltage V across each switch tube. ds and output current I s .Right now:

[0072] Table 1. Rules for determining steady-state values ​​of state variables

[0073] D <![CDATA[I L ]]> <![CDATA[V gs1_sta ]]> <![CDATA[V gs2_sta ]]> <![CDATA[V ds1_sta ]]> <![CDATA[V ds2_sta ]]> <![CDATA[I s1_sta ]]> <![CDATA[I s2_sta ]]> >Carrier <＝0 <![CDATA[V H ]]> <![CDATA[V L ]]> <![CDATA[V moson ]]> <![CDATA[V DC -V moson ]]> <![CDATA[-I L ]]> 0 >Carrier >0 <![CDATA[V H ]]> <![CDATA[V L ]]> <![CDATA[-V Don ]]> <![CDATA[V DC +V Don ]]> <![CDATA[-I L ]]> 0 <=Carrier <＝0 <![CDATA[V L ]]> <![CDATA[V H ]]> <![CDATA[V DC +V Don ]]> <![CDATA[-V Don ]]> 0 <![CDATA[I L <!-- 6 -->]]> <=Carrier >0 <![CDATA[V L ]]> <![CDATA[V H ]]> <![CDATA[V DC -V moson ]]> <![CDATA[V moson ]]> 0 <![CDATA[I L ]]>

[0074] At this point, the steady-state values ​​V of the three state variables are ds_sta , I s_sta and V gs_sta All are obtained.

[0075] S2, taking the voltage across both ends, gate voltage and output current of each switch tube as its state variables, and determining the transient initial value and driving voltage of the state variables of each switch tube;

[0076] Among them, the transient initial values ​​of the voltage across the two ends and the output current of each switch tube are the steady-state values ​​of the voltage across the two ends and the output current at the transient start time n respectively; if I L If it is positive, the initial transient values ​​of the gate voltage and drive voltage of the first switch tube are both V L , the transient initial values ​​of the gate voltage and driving voltage of the second switch tube are the steady-state values ​​of the gate voltage at time n and time n+1 respectively; if I L If negative, the transient initial values ​​of the gate voltage and drive voltage of the first switch tube are the steady-state values ​​of the gate voltage at time n and time n+1 respectively, and the transient initial values ​​of the gate voltage and drive voltage of the second switch tube are both V L .

[0077] Specifically, the initial value of the state variable and the driving voltage of the switch transient state are determined according to the steady-state value of the state variable. gs The waveform of the switching transient is used to determine the start time of the switching transient. Then, the forward transistor of the two IGBT / MOS switching transistors is determined based on the direction of the output current. For the forward transistor and the non-forward transistor, specific initial values ​​of the state variables and the driving voltage V are specified based on the current switching waveform. drive =[V drive1 ,V drive2 ].

[0078] Calculate V gs_sta The moment when the amplitude starts to jump (that is, the sequence number of the array) is stored in the transient start time matrix T _pul =[t _pul1 ,t _pul2 ,…,t _pulm ], where m is the number of transient states.

[0079] State variable I s and V ds The initial value is equal to the steady-state value of the state variable at the beginning of the switching transient, and the steady-state value has been calculated in the previous step.

[0080] State variable V gs Initial value and driving voltage V drive The determination of is based on the opening or closing of the forward transistor. The forward transistor refers to the MOS transistor whose device current is in the same direction (from drain to source) as the current. At the beginning of each switching transient, the forward transistor must be switched, and the non-forward transistor is already in steady state. The gate voltage V gs The corresponding steady-state value at the start of the transient should be selected, and the driving voltage V drive The corresponding steady-state value at the end of the transient should be selected. gs and the driving voltage V drive The shutdown level V L This is because in order to avoid the bridge arm from being directly connected, when the forward tube is turned on or off, the non-forward tube must be in the off steady state.

[0081] Based on this, Table 2 lists the rules for determining the initial values ​​of state variables and the driving voltage. n is the time at which the transient begins. This rule applies to the initial values ​​of state variables at the beginning of different subtransients.

[0082] Table 2. Rules for determining the initial values ​​of state variables and driving voltages for a two-level PWM converter

[0083] <![CDATA[V ds10 ]]> <![CDATA[V gs10 ]]> <![CDATA[I s10 ]]> <![CDATA[V drive1 ]]> <![CDATA[V ds20 ]]> <![CDATA[V gs20 ]]> <![CDATA[I s20 ]]> <![CDATA[V drive2 ]]> <![CDATA[I L >0]]> <![CDATA[V ds1 (n)]]> <![CDATA[V L ]]> <![CDATA[I s1 (n)]]> <![CDATA[V L ]]> <![CDATA[V ds2 (n)]]> <![CDATA[V gs2_sta (n)]]> <![CDATA[I s2 (n)]]> <![CDATA[V gs2_sta (n+1)]]> <![CDATA[I L ≤0]]> <![CDATA[V ds1 (n)]]> <![CDATA[V gs1_sta (n)]]> <![CDATA[I s1 (n)]]> <![CDATA[V gs1_sta (n+1)]]> <![CDATA[V ds2 (n)]]> <![CDATA[V L ]]> <![CDATA[I s2 (n)]]> <![CDATA[V L ]]>

[0084] Determine the driving voltage V drive Finally, you should also follow V drive Select the drive resistor R g , when V drive When the voltage is high, the driving resistor R g Select R gon , when V drive When it is low, the driving resistor R g Select R goff .

[0085] S3. Based on the switch tube model, diode model and junction capacitance voltage equation, a differential equation group of the state variables of each switch tube with respect to the transient moment is established, and the transient initial value and driving voltage of the state variables of each switch tube are combined to solve the differential equation group to obtain the transient value of each state variable.

[0086] Specifically, the state equations are written and encoded according to the two-level PWM converter topology. The gate voltage V gs , output voltage V ds and the device output current I s The state equations of these three state variables give rise to a function model for calculating the differentials of the state variables at different times.

[0087] The switching transient model of the two-level PWM converter is as follows: Figure 6 As shown, nonlinear switching devices, voltage-dependent junction capacitance, and skin effect are all included. Below, the circuit equations are written based on Kirchhoff's current law (KCL) and voltage law (KVL) and converted into a system of state equations.

[0088] for Figure 6 The red node in the figure is the gate or g-pole of the MOS transistor. The following equations can be written using KCL. Since there are two MOS transistors, i = 1 and 2, and there are two equations.

[0089]

[0090] for Figure 6 The blue node in the graph is the drain or d-pole of the MOS transistor. The following equations can be written using KCL. Since there are two MOS transistors, i = 1 and 2, and there are two equations.

[0091]

[0092] for Figure 6 The loop formed by the two MOS transistors, parasitic inductance and skin resistance, and DC power supply can be written as the following equations based on KVL. There are two MOS transistors here, so i = 1, 2, and there are two equations. It should also be noted that the current differentials of the two MOS transistors are the same, because the difference between the two currents is I L is a constant value. Here I ring It refers to the oscillating current flowing through the skin resistance, and its determination is introduced in the next section.

[0093]

[0094] In addition, the three capacitor voltages of each switch tube also satisfy the KVL theorem, that is,

[0095] V gsi +V gdi -Vdsi =0, i=1,2 (11)

[0096] After transforming the equations (7)-(10) together, we can obtain the following set of state equations.

[0097]

[0098] Current I d By MOS current I mos and the diode current I D Synthesis, that is:

[0099] I di =I mosi -I Di , i=1,2 (13)

[0100] According to formula (2), the MOS current I mosi The state variable V dsi and V gsi Calculation. According to formula (4), the diode current I D The state variable V dsi Calculation, this is because the forward voltage of the diode is opposite to the voltage of the MOS tube V D =-V dsi According to equations (3) and (6), the voltage variable capacitor can be expressed by the state variable V dsi Calculation. So far, the right side of the equal sign in the equation group (11) can be represented by the state variable V ds 、V gs and I s Solving the equations converts them into a system of state equations for the state variables and their differentials. To facilitate the following explanation, the three state variable equations can be written together in the following form. x represents the package of three state variables, and y is the functional relationship between the state variable x and its differential, which is Equation (11).

[0101]

[0102] It can be understood that Equation (12) assumes that the converter modulation strategy is SVPWM, and the diode model, switch device model, and switch tube junction capacitance pressure change model are respectively based on Equations (2), (3), and (5). The state variable differential equations for the switch tube established by this system are merely examples, and the present invention does not limit this system to any uniqueness. For other modulation strategies and diode models, switch device models, and switch tube junction capacitance pressure change models, those skilled in the art may select them according to actual conditions and derive the state variable differential equations according to common knowledge.

[0103] The derivation of the state equation group adopts an existing algorithm, such as the improved Euler method, etc. Considering that the step size required for the improved Euler method and other solution methods to calculate the equation without divergence is too small, the present invention preferably selects the Runge-Kutta algorithm with higher accuracy and less divergence to solve and obtain the transient value of the state variable.

[0104] The pseudo code of the solution process is as follows. For the transient start time vector T pul Each element t in pul By repeating this process, the transient values ​​of all state variables can be calculated.

[0105] Algorithm: Calculation of transient values ​​of state variables.

[0106] Input: state equation model y, state variable transient initial value V ds0 V gs0 I s0 , transient time t pul .

[0107] Output: transient value of state variable.

[0108] 1. Set the maximum calculation time to 20us and the calculation step h to 0.25ns. To ensure accuracy, the calculation step of transient state variables is smaller than that of steady-state state variables. The initial value of skin resistance is set to zero. According to the load current I L and transient time t pul Calculate the transient load current i L .

[0109] 2. According to the two state variables I s1 I s2 The initial value of determines the ringing current term I ring The reason for this choice is that the state variable with a larger absolute amplitude will only have a ringing current during the ringing period at the end of the transient process, and no AC power current. The ringing current at this time is I ring .

[0110] Table 3. Ringing current term I ring Selection rules

[0111] Criteria <![CDATA[I ring ]]> <![CDATA[|I s1 |>|I s2 |]]> <![CDATA[I s1 ]]> <![CDATA[|I s1 |<=|I s2 |]]> <![CDATA[I s2 ]]>

[0112] 3. Use the Runge-Kutta method to calculate the state variables at the next moment n+1.

[0113]

[0114] 4. Determine whether the diode is operating at each step h of each transient process. If it starts to operate, the switching waveform enters the oscillation stage, and the skin resistance R qIt should take effect. It can be understood that this step is a preferred step.

[0115] When solving the state equations, the transient end criterion used is: Taking N = 50, M = J = 10000 as an example, calculate the variance of all state variables from n-49 to n, these 50 moments. If the state variable V ds The variance is less than the input voltage V DC One ten-thousandth of, and V gs The variance is less than the high and low level difference V H -V L One ten-thousandth of s The variance of the load current i L , then it can be considered that the state variable has reached a steady state, that is, the transient process is over, and the transient value V of the state variable obtained at this time is stored. ds_pul and I s_pul , and the transient occurrence time t pul If not, go back to step 3 and use the calculated x(n+1) to continue calculating x(n+2) according to the Runge-Kutta algorithm and repeat the above process.

[0116] In summary, in S3, the state equations are solved by combining the initial state value and the driving voltage. During the solution process, it is determined whether the transient state has ended and the skin resistance is updated. The following process is calculated in parallel for each transient state of the switching waveform obtained in S1. Set the sampling frequency f to 4 GHz. s , get the time series. At each moment, based on the state variable value of the previous moment, first calculate the voltage-variable junction capacitance of the switch tube according to the capacitance-voltage-variable model, calculate the working current of the current switch device according to the device I / V curve model, then call the function model in S3 to calculate the state variable differential, and determine the state variable value at the current moment according to the state variable differential, and iterate the calculation. When the calculation reaches the time length (about N / f s =12.5ns), the variance of the last N values ​​of the state variable is used to determine whether the state variable waveform has exited the transient state. Furthermore, the diode current calculated from the state variable is used to determine whether the waveform has entered the ringing phase. At this point, the skin resistance is corrected to allow the skin effect to take effect.

[0117] S4, splicing the steady-state value and transient value of each state variable of each switching tube to obtain the complete time domain waveform of each state variable, so as to calculate the common-mode and differential-mode interference source switching waveforms of the converter, and performing detection processing on them to obtain the two-level PWM converter switching waveform for electromagnetic interference prediction.

[0118] Specifically, step S4 includes:

[0119] S41, synthesize the complete state variable time domain waveform using the steady-state value and transient value of the state variable.

[0120] The steady-state value of the state variable obtained by S3 is replaced by the transient state variable value of the switch obtained in S4 during the switching transient time, and the device state variable time domain waveform is synthesized taking into account the device nonlinear I / V characteristics, device junction capacitance pressure change effect and skin effect.

[0121] For m transients, repeat the following process: set the steady-state value of the state variable V ds_sta and I s_sta Starting from the transient state t_ pul The next section of the waveform is replaced by the transient value V ds_pul and I s_pul , the complete state variables can be synthesized. The pseudo code is as follows:

[0122] Algorithm: Synthesize the complete state variable V ds , I s .

[0123] Input: Steady-state value of state variable V ds_sta and I s_sta , transient start time t _pul , transient state variable V ds_pul and I s_pul .

[0124] Output: Contains transient and steady-state state variables V ds , I s .

[0125] 1. Calculate the duration p of the transient state variable.

[0126] 2. If the transient duration p is equal to the transient start time t _pul The sum does not exceed the longest time t of the steady-state state variable end , then the steady-state value of the state variable is changed from the transient start time t _pul to p+t _pul All steady-state variables are replaced by transient values.

[0127] 3. If the transient duration p is equal to the transient start time t _pul The sum has exceeded the longest time t of the steady-state state variable end , then only the steady-state variables are changed from the transient start time t _pul To the last time t end The steady-state state variables are replaced by transient values.

[0128] S42, according to V ds and I s The state variables calculate the switching waveforms of common-mode and differential-mode interference sources.

[0129] In view of the symmetry of different phase bridge arms in the inverter, based on the calculated Vds and I s The state variables generate the state variables of other bridge arms. According to the functional relationship between the common-differential mode interference source and the bridge arm state variables, the switching waveform of the common-differential mode interference source is calculated.

[0130] Two-level PWM converters are divided into three-phase and single-phase, which need to be discussed separately.

[0131] For a single-phase PWM converter, such as Figure 7 In (a), unipolar frequency multiplication modulation technology is generally used. At this time, the switch voltage of the other bridge arm can be calculated as follows: V ds3 Equal to V ds2 Phase shift half a cycle, V ds4 Equal to V ds1 Phase shifted by half a cycle. s3 Equal to I s2 Phase shifted by half a cycle, I s4 Equal to I s1 The phase shift is half a cycle. This is because, in unipolar frequency modulation, the carriers of the two bridge arms are 180° apart.

[0132] The switching waveform of the common-mode interference source of the single-phase inverter is equal to the average value of the voltage waveform of the lower tube of the two bridge arms and the sum of the current of the lower tube of the two bridge arms, that is:

[0133]

[0134] For a three-phase PWM converter, such as Figure 7 In (b), space vector modulation technology is generally used. At this time, the switch voltage of the other bridge arm can be calculated as follows: V ds3 Equal to V ds1 Phase shift 1 / 3 cycle, V ds5 Equal to V ds1 Phase shift 2 / 3 cycle. V ds4 Equal to V ds2 Phase shift 1 / 3 cycle, V ds6 Equal to V ds2 Phase shift 2 / 3 cycle. s3 Equal to I s1 Phase shift 1 / 3 cycle, I s5 Equal to I s1 Phase shift 2 / 3 cycle. s4 Equal to I s2 Phase shift 1 / 3 cycle, I s6 Equal to I s2 The phase shift is 2 / 3 of a cycle. This is because, in space vector modulation, the carriers of the two bridge arms are 120° apart.

[0135] The switching waveform of the common-mode interference source of the three-phase inverter is equal to the average value of the voltage waveform of the three bridge arms and the sum of the current of the three bridge arms, that is:

[0136]

[0137] S43 , detecting the switching waveforms of the common-mode and differential-mode electromagnetic interference sources to obtain quasi-peak and average switching waveform spectra for electromagnetic interference prediction.

[0138] For example, an algorithm that simulates an EMI receiver can be used to convert input time-domain electromagnetic interference into a spectrum similar to that used in EMI receiver testing, based on civilian or military EMI standards. This spectrum can be used to predict EMI emissions. Specifically, the function input is the interference time-domain waveform, and the output is the quasi-peak and average values ​​obtained after testing in accordance with simulated EMI standards.

[0139] The physical layer parameters of the known single-phase two-level PWM converter are shown in Table 4. The switching waveform spectrum calculation time of the common-mode and differential-mode interference sources based on the model proposed in the present invention is only 80 seconds.

[0140] Table 4 Physical layer parameters of single-phase two-level PWM converter

[0141]

[0142]

[0143] After model calculation, the drain-source voltage V ds Can be predicted, the time domain waveform is as follows Figure 8 As shown in FIG, it can be seen from the switching transient waveform that the slope change characteristics when the voltage rises or falls can be simulated, which shows the accuracy of the model.

[0144] The output current of the switching device I s It can also be predicted that the time domain waveform is as follows Figure 9 As shown in Figure 3, the reverse recovery current peak when the switch is turned on can be accurately simulated.

[0145] For the switching waveform spectrum of the common-differential mode interference source in the conduction frequency band, the results are as follows: Figure 10 and Figure 11 Both the quasi-peak value and the average value can be modeled and predicted, and the ringing effect around 18 MHz can be predicted.

[0146] The proposed model can also predict the common and differential mode interference sources in the radiation frequency band up to 1 GHz, such as Figure 12 and Figure 13 shown.

[0147] In summary, the present invention addresses the pain points of low accuracy and low efficiency of the existing switching waveform model used for electromagnetic interference prediction of two-level PWM converters, and proposes a broadband, efficient, and high-precision switching waveform determination method based on a state equation group. The switching transient is solved based on the state equation group, and each switching transient is solved jointly according to the specified initial value of the state variable, the equivalent state equation group, the nonlinear device transfer function, and the voltage-dependent characteristics of the junction capacitance; the device switching waveform is synthesized with the steady-state value and transient value of the state variable, and then the device switching waveform and the common-differential mode interference source switching waveform are converted to obtain the switching waveform for electromagnetic interference prediction. Finally, the switching waveform spectrum is obtained after detection processing of the analog EMI receiver.

[0148] An embodiment of the present invention provides an electronic device, comprising: a computer-readable storage medium and a processor;

[0149] The computer-readable storage medium is used to store executable instructions;

[0150] The processor is configured to read the executable instructions stored in the computer-readable storage medium and execute the method described in any one of the above embodiments.

[0151] An embodiment of the present invention provides a computer-readable storage medium, wherein the computer-readable storage medium stores computer instructions, and the computer instructions are used to enable a processor to execute the method described in any embodiment.

[0152] An embodiment of the present invention provides a computer program product, including a computer program or instructions, which implements the method described in any embodiment when executed by a processor.

[0153] It will be easily understood by those skilled in the art that the above description is merely a preferred embodiment of the present invention and is not intended to limit the present invention. Any modifications, equivalent substitutions, and improvements made within the spirit and principles of the present invention should be included in the scope of protection of the present invention.

Claims

1. A method for determining a switching waveform of a two-level PWM converter for electromagnetic interference prediction, characterized in that: include: S1, compare the modulated wave of the converter with the carrier wave to obtain the steady-state value V of the gate voltage of the first and second switching tubes in the converter gs1_sta 、V gs2_sta , and according to V gs1_sta 、V gs2_sta and load current I L Determine the steady-state value V of the voltage across the first and second switching tubes ds1_sta 、V ds2_sta and the steady-state value of the output current I s1_sta , I s2_sta ; Among them, at a certain moment, if the modulation wave amplitude is greater than the carrier amplitude, then at that moment, V gs1_sta =V H , V gs2_sta =V L , I s1_sta =-I L , I s2_sta =0, when I L ≤0, V ds1_sta =V moson 、V ds2_sta =V DC -V moson , when I L >0, V ds1_sta =-V Don 、V ds2_sta =V DC +V Don If the modulation wave amplitude is less than or equal to the carrier amplitude, then at this moment, V gs1_sta =V L , V gs2_sta =V H , I s1_sta =0, I s2_sta =I L , when I L When V ds1_sta =V DC +V Don 、V ds2_sta =-V Don , when I L When it is negative, V ds1_sta =V DC -V moson 、V ds2_sta =V moson ; V H 、V L They are driving voltage high and low level respectively, V DC 、V Don 、V moson They are input voltage, diode conduction voltage, and MOS tube conduction voltage respectively. S2, taking the voltage across both ends, gate voltage and output current of each switch tube as its state variables, and determining the transient initial value and driving voltage of the state variables of each switch tube; Among them, the transient initial values ​​of the voltage across the two ends and the output current of each switch tube are the steady-state values ​​of the voltage across the two ends and the output current at the transient start time n respectively; if I L If it is positive, the initial transient values ​​of the gate voltage and drive voltage of the first switch tube are both V L , the transient initial values ​​of the gate voltage and driving voltage of the second switch tube are the steady-state values ​​of the gate voltage at time n and time n+1 respectively; if I L If negative, the transient initial values ​​of the gate voltage and drive voltage of the first switch tube are the steady-state values ​​of the gate voltage at time n and time n+1 respectively, and the transient initial values ​​of the gate voltage and drive voltage of the second switch tube are both V L ; S3, establishing a differential equation group of the state variables of each switch tube with respect to the transient moment based on the switch tube model, the diode model, and the voltage equation of the junction capacitance, and solving the differential equation group in combination with the transient initial value of the state variable of each switch tube and the driving voltage to obtain the transient value of each state variable; S4, splicing the steady-state value and transient value of each state variable of each switching tube to obtain the complete time domain waveform of each state variable, so as to calculate the common-mode and differential-mode interference source switching waveforms of the converter, and performing detection processing on them to obtain the two-level PWM converter switching waveform for electromagnetic interference prediction.

2. The method according to claim 1, wherein The switch tube model is: Among them, I mos is the channel current of the switch tube, V gs 、V ds They are the gate voltage and the voltage across the switch tube, V gsth 、V dsth , K are fitting parameters, V th is the gate threshold voltage, ∨ and ∧ are the calculation symbols of OR and AND respectively.

3. The method according to claim 2, wherein The differential equations are: Where i=1,2, L is the parasitic inductance, C gdi 、C gsi 、C d ' si They are the drain-gate capacitance, gate capacitance, and parasitic capacitance of the anti-parallel diode C of the i-th switch tube. D The equivalent capacitance between the drain and source of a MOS tube or the emitter and collector of an IGBT, V drivei 、V gsi , I si , I di 、V dsi They are the driving voltage, gate voltage, output current, drain current, and voltage across the i-th switch tube, V DC is the input voltage, I ring is the oscillating current of the skin resistance, R q is the skin resistance, R g is the driving resistor.

4. The method according to claim 3, wherein In step S3, the Runge-Kutta algorithm is used to solve the differential equations.

5. The method according to claim 1 or 4, wherein: In step S3, when solving the differential equations, the variance of the state variables at N adjacent transient moments is calculated. If the voltage V ds The variance is less than the input voltage V DC One-Mth of the gate voltage V gs The variance of the driving voltage is less than one-J of the high and low levels, and the output current I s If the variance of is less than the load current IL, the state variable has reached a steady state; where N, M, and J are all positive numbers.

6. The method according to claim 1 or 4, wherein: In step S3, when solving the differential equations, it is determined whether the diode is operating at each step h of each transient process. If so, the skin resistance R q Not zero, otherwise R q zero.

7. An electronic device, characterized in that: include: Computer-readable storage medium and processor; The computer-readable storage medium is used to store executable instructions; The processor is configured to read the executable instructions stored in the computer-readable storage medium and execute the method according to any one of claims 1 to 6.

8. A computer-readable storage medium, characterized in that The computer-readable storage medium stores computer instructions, and the computer instructions are used to enable a processor to execute the method according to any one of claims 1 to 6.

9. A computer program product comprising a computer program or instructions, characterized in that When the computer program or instruction is executed by a processor, the method according to any one of claims 1 to 6 is implemented.

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