Method for operating an electric machine

Abstract

The invention relates to a method for operating an electric machine (2) having a brushless electric motor (4) with a stator and a rotor and having a regulating unit (64) for sensorless field-oriented control and / or regulation of the electric motor (4) in a d-q reference frame which has a d-axis and a q-axis and is fixed relative to the rotor, the electric motor (4) is controlled and / or regulated on the basis of the estimated motor position, a test signal (UdTest) is generated for determining a deviation between the real motor position and the estimated motor position, the test signal (UdTest) is fed into the d-axis, a compensation signal (UqComp) is determined on the basis of the test signal (UdTest), and the compensation signal (UqComp) is controlled and / or regulated on the basis of the compensation signal (UqComp). The compensation signal (UqComp) is fed into the q-axis such that a torque change of the electric motor (4) caused by the test signal (UdTest) is reduced when the compensation signal (UqComp) is fed into the q-axis, and wherein the compensation signal (UqComp) is fed into the q-axis.

Description

Technical Field

[0001] The invention relates to a method for operating an electric machine having a brushless electric motor with a stator and a rotor and having a control unit for sensorless field-oriented control and / or regulation of the electric motor in a d / q coordinate system fixed relative to the rotor, an electric machine for carrying out the method, and software on a data carrier. Background Art

[0002] Motor vehicle components such as window lifters, seat adjustments, door and sliding roof drives or radiator fan drives as well as pumps and interior blowers that are driven or operated by electric motors typically have an electric drive with a controlled electric motor. So-called brushless electric motors (brushless direct current motors, BLDC motors) are increasingly being used for such electric motor drives, in which the wear-prone brush elements of a rigid (mechanical) commutator are replaced by electronic commutation of the motor current.

[0003] Electric motor drives for motor vehicles are usually fed by a (high-voltage) battery as an energy storage device inside the vehicle, from which the electric motor is supplied with electrical energy in the form of direct current (direct voltage). In order to convert the direct current into motor current, a current converter (converter, inverter) is interconnected between the battery and the electric motor in a suitable manner. The current converter has a bridge circuit, which is supplied with direct current or direct voltage from the energy storage device via an electrical intermediate circuit. The motor current is generated as a multi-phase output current by pulse-width modulated (PWM) control or regulation of the semiconductor switches of the bridge circuit. Due to the pulses of the PWM signal, the semiconductor switches are switched between the conducting state and the blocked state in a clocked manner.

[0004] During operation, the bridge circuit feeds electrical motor current (rotating current) into the stator coils of the electric motor, which then generate a rotating magnetic field that rotates relative to the stator. The rotor of the electric motor has a number of permanent magnets in a suitable manner, wherein the interaction of the permanent magnets with the rotating field generates a resultant torque that causes the rotor to rotate.

[0005] The phases of the generated rotating current and the associated rotating field are referred to as (motor) phases. In a paraphrase sense, this is also understood to be the stator coils (phase windings) respectively assigned to such phases with the associated connecting lines (phase ends). In this case, the phases are interconnected with one another, for example in the star point of a star circuit.

[0006] For efficient operation, it is necessary to supply the phases with current at the right time. For this purpose, for example, vector control can be performed, which is also called field-oriented control (English: Field Oriented Control, FOC). In such field-oriented control or FOC, the rotating current is identified into two orthogonal components, which can be visualized using the current space vector. One component (the direct-axis component) defines the magnetic flux of the motor, the other component defines the torque (the quadrature-axis current).

[0007] Field-oriented regulation regulates the rotating current in the dq reference frame (reference frame) of the electric motor. Ideally, the current space vector is fixed in value and direction (orthogonal) with respect to the rotor, i.e., independent of the rotation. Since the current space vector is static in the dq reference frame, the current regulation is based on a direct current signal. This isolates the regulator from the time-varying winding currents and winding voltages and thus eliminates the limitations of the regulator frequency response and phase shift on the motor torque and speed.

[0008] The electric motor has an associated motor control unit, which determines the corresponding current component target values ​​as a function of the flux target values ​​and the torque target values ​​specified by the speed control. The motor current or the phase current is converted into the dq reference system.

[0009] In order to ensure that the phases are supplied with current at the correct point in time, it is necessary to accurately determine the relative positioning of the rotor and stator.

[0010] For example, for position determination, the rotor position (rotor position) is determined by means of an additional rotation sensor, such as a Hall sensor. However, such rotation sensors or encoders are expensive, so the position determination should preferably be performed sensorlessly.

[0011] Sensorless position determination is based, for example, on the detection of induced current and / or voltage signals due to the back electromotive force (back-EMF), which is induced in the phase winding by the rotating permanent magnet. The induced back-EMF signal is proportional to the rotor speed, so that at low speeds or when the electric motor is at a standstill, little or no information is available to the motor control unit for position determination in a disadvantageous manner. In particular, at low speeds, the signal-to-noise ratio decreases. This limitation also exists for sensorless measurement methods based on flux. As a result, position determination or position recognition is generally not possible below a threshold speed, which makes safe and reliable operation of the electric motor disadvantageously difficult, especially during startup from a standstill or during operation at low speeds.

[0012] In this case, it is possible, for example, to estimate the rotor position near a (motor) standstill and to control the electric motor based on this estimate. In order to determine the deviation between the estimated position and the actual position and then to compensate for this deviation, for example, a test signal or test pulse is fed into the imaginary d-axis. However, this test signal leads to torque ripples in the electric motor, which can cause unnecessary acoustic damage.

[0013] These acoustic impairments usually only exist briefly during the start-up period or at low motor speeds of the electric motor, so that these acoustic disturbances are usually acceptable. Alternatively, the acoustic influence of the test signal can be reduced by mechanical measures. However, this is accompanied by additional costs. Summary of the invention

[0014] The object of the present invention is to specify a particularly suitable method for moving an electric motor. In particular, reliable sensorless control and / or regulation of the motor operation should be possible even at low rotational speeds. Furthermore, motor operation with minimal noise should be possible. The object of the present invention is also to specify a particularly suitable electric motor and particularly suitable software on a data carrier.

[0015] The advantages and configurations listed with respect to the method can also be transferred to the motor and / or the software, and vice versa. The conjunction "and / or" is to be understood here and below to mean that the features connected by means of this conjunction can be formed both together and as alternatives to each other.

[0016] The method according to the invention is provided and suitable and designed for operating an electric machine. With regard to the method steps described below, advantageous embodiments of the electric machine are obtained in particular in that the electric machine is designed to carry out one or more of these method steps.

[0017] According to the method, a regulating unit for generating control variables for an electric motor of an electric machine is provided. The electric motor is implemented as a brushless electric motor, which has a multi-phase, especially at least three-phase rotating field winding. The regulating unit generates control variables for a motor control unit based on current actual values ​​(input currents U, V, W) and positioning actual values ​​(rotor positioning). The current actual value is understood here as an input current for a motor phase, wherein the positioning actual value particularly describes the mechanical positioning or electrical positioning of the rotor of the electric motor. The mechanical positioning (mechanical angle) particularly describes the absolute mechanical position of the rotor relative to the stator, wherein the electrical positioning (electrical angle) particularly describes the position value that plays a decisive role in the commutation of the motor current. The electrical positioning particularly describes the phase of the current vector used to commutate the electric motor. Preferably, the positioning actual value is equivalent to the electrical rotor positioning.

[0018] The operation of the electric motor is controlled and / or regulated by means of a field-oriented control (FOC) of a control unit. In this case, the motor current or phase current of the electric motor is regulated as a current actual value in a dq reference system (d / q coordinate system) with a DC voltage component along the d-axis and a quadrature-axis current along the q-axis by means of current regulation. The electric machine or electric motor is implemented sensorlessly, which means that no positioning sensor is provided for directly or directly detecting the rotor position. Therefore, the positioning actual value is determined in a sensorless manner, for example based on the back electromotive force of the electric motor.

[0019] At the start of operation, i.e., for example, when the electric motor is at a standstill, in which in particular there is not yet enough back electromotive force for determining the actual positioning value, the method controls and / or regulates the electric motor based on the estimated motor position. In other words, the estimated actual positioning value is initially used for FOC. Here and below, "estimation" is understood to mean the approximate determination of the motor position or the actual positioning value by evaluating the back electromotive force, for example by visual inspection, pre-characterized measurements, stored tables or characteristic curves, or by means of statistical mathematical measures. The estimated actual positioning value can also be a stored starting value in this case.

[0020] In order to check and adjust the position estimate, a test signal or test pulse is generated, by means of which a deviation between the real motor position (real actual position value) and the estimated motor position (estimated actual position value) is generated. The test signal is fed into the (estimated) d-axis of the electric motor.

[0021] According to the invention, a compensation signal is generated in addition to the test signal and fed into the (estimated) q-axis of the electric motor. The compensation signal is determined based on the test signal so that when the compensation signal is fed into the q-axis, the torque change of the electric motor caused by the test signal is reduced or compensated. As a result, the torque ripple caused by the test signal is compensated via the other drive signal (compensation signal). As a result, the acoustic performance of the electric motor is improved, especially at startup. A particularly suitable method for operating an electric motor is thus achieved.

[0022] In contrast to conventional control methods, in the present invention, not only the DC voltage component but also the quadrature axis component is controlled. These values ​​or the motor response to these values ​​can be measured very well, for example, via the phase currents.

[0023] Preferably, the torque change of the electric motor caused by the test signal is reduced as completely as possible by the compensation signal, so that a particularly smooth starting of the electric motor can be achieved. This improves user comfort, in particular when the electric motor is used as an adjustment drive in the vehicle interior of a motor vehicle.

[0024] In a suitable improvement, the motor response to the test signal and the compensation signal is detected, for example in the form of a current in the q-axis, and the motor response is used to determine the motor positioning. The test signal is used to check the positioning estimate. The current in the q direction depends on the deviation from the estimated motor positioning, and when there is no error between the estimated positioning and the actual positioning, that is, when there is no deviation between the real motor positioning and the estimated motor positioning, no current is generated, for example, in the stator winding of the electric motor. The induced current can be detected as a motor response, for example, by means of an ammeter, in particular by means of a shunt resistor. Therefore, even at low motor speeds, the estimated motor positioning can be controlled and / or adjusted by the test signal, wherein the compensation signal ensures that no torque ripple occurs here. Therefore, the estimated motor positioning can be iteratively or gradually matched to the real motor positioning. The method is preferably performed until another sensorless method (especially based on electromotive force evaluation) reliably takes over the positioning determination. The estimated speed can be used as an alternation or switching criterion between methods.

[0025] The invention is based on the following formula for the torque of an electric motor: , Where p is the number of pole pairs of the electric motor, Ψ is the magnetic flux, i q and i d is the current along the q-axis or d-axis, and L d and L q is the inductance of the stator (i.e. the stator coil / winding) along the q-axis or d-axis.

[0026] In this case, the operating point applies

[0027] In order to reduce the disturbing influence of the test signal, i.e. the torque ripple, it is therefore necessary to q and Δi d When the torque changes, the torque change ΔM is zero:

[0028] Therefore, the current change Δi is obtained q / Δi d The ratio between Where k is the compensation coefficient.

[0029] For FOC, the corresponding voltage u is typically adjustedd and u q Current i d and i q Here with u d and u q Therefore, the Laplace transform method is used to obtain the following equation: Where s is the Laplace transform parameter, F N is the normalization coefficient, F dd 、F qd 、F dq 、F qq are the transfer matrix components.

[0030] For boundary conditions Conclusion

[0031] Normalization coefficient F N and the transfer matrix components F dd 、F qd 、F dq 、F qq It can be identified based on the motor model. Which motor model is used here is not important at first. It is possible to use different motor models for different electric motors. For example, a dq motor model is used here, which is described below in the figure ( Figure 5 This is explained in detail in . In such a motor model,

[0032] In a suitable embodiment, the compensation signal Δu q Based on the following formula Here, Δu d is the test signal, ω el is the rotor (rotation) frequency, L d and L q is the inductance of the stator along the d-axis or q-axis, R S is the ohmic resistance of the stator (stator coil / stator winding), s is the Laplace transform parameter, and k is the compensation coefficient.

[0033] In a suitable improvement, in order to reduce the computational workload, the compensation signal Δu is used. q Simplified calculation of .

[0034] For low rotor frequencies (ω el~ 0 kHz), for example when starting from standstill, the above formula can be used In addition, if R S Can also be ignored, then we get:

[0035] If R S can be ignored, but ω el Cannot be ignored, then: Among them, Δu q is the compensation signal, Δu d is the test signal, ω el is the rotor frequency, L d and L q is the inductance value of the stator along the d-axis or q-axis, s is the Laplace transformation parameter, and k is the compensation coefficient.

[0036] In an advantageous embodiment, the compensation factor k is based on the following formula To simplify the calculation and thus to reduce the computational effort, a simplified proportional calculation can be used. The above ratio calculation can be used Perform approximate calculations.

[0037] In order to determine the compensation signal in a particularly resource-saving manner and with reduced computational effort, the compensation signal Δu q In a suitable embodiment, for While ignoring R S and ω el The impact applies when: Among them, Δu q is the compensation signal, Δu d is the test signal, Ψ is the magnetic flux, L d / L q is the inductance of the stator along the d / q axis.

[0038] In a preferred improvement, a sinusoidal voltage with a constant frequency is used as the test signal. Therefore, when calculating the compensation signal, to replace the Laplace transform parameter s, where j is the imaginary unit and ω ud is the frequency of the test signal. So we have

[0039] As before, to reduce the computational effort we can use the simplification and / or ignore R S and ω el .

[0040] If the test signal is a sinusoidal voltage, the compensation signal is also a sinusoidal voltage. The amplitude and phase values ​​can also be stored in a table and used here, for example. These values ​​can be known via the equations shown. Alternatively, however, they can also be known empirically (for example by means of acoustic measurements). The table values ​​can be created and used, for example, depending on the temperature, the estimated rotational speed or the estimated load.

[0041] The electric machine according to the invention is implemented, for example, as a control drive in a motor vehicle. The electric machine has a brushless electric motor with a stator and a rotor. In addition, the electric machine has a control unit for sensorless field-oriented control and / or regulation of the electric motor. The control unit is, for example, coupled to a controller (that is, a control unit) or integrated into the controller.

[0042] In general, the controller is hereby set up (in terms of programming and / or circuit technology) to carry out the method according to the invention described above. Thus, the controller is specifically set up to estimate the initial motor position, to generate a test signal and feed the test signal into the estimated d-axis, and to determine a compensation signal based on the test signal and feed the compensation signal into the q-axis.

[0043] In a preferred embodiment, the controller is formed at least in its core by a microcontroller having a processor and a data memory, in which the functions for executing the method according to the invention are implemented in programming in the form of operating software (firmware), so that when the operating software is executed in the microcontroller, the method is automatically executed (if necessary in interaction with the user of the installation). Alternatively, however, within the scope of the invention, the controller can also be formed by a non-programmable electronic component, such as an application-specific integrated circuit (ASIC) or by an FPGA (field programmable gate array), in which the functions for executing the method according to the invention are implemented using circuit components.

[0044] An additional or further aspect of the invention provides a software on a medium or data carrier for implementing or executing the above method. This means that the software is stored on a data carrier and is configured and suitable and designed to perform the above method. The software is in particular a computer program product, which includes instructions that, when the program is executed by a computer, cause the computer to perform the above method.

[0045] This results in a particularly suitable software for operating the electric machine, with which the functions for carrying out the method according to the invention are implemented in terms of programming technology. The software is therefore in particular operating software (firmware), wherein the data carrier is, for example, a data memory of a controller. BRIEF DESCRIPTION OF THE DRAWINGS

[0046] The following is a detailed explanation of the embodiments of the present invention based on the diagrams.

[0047] Figure 1 An electric machine having a current source and an electric motor and a converter interconnected therebetween is shown;

[0048] Figure 2 The phase windings of a three-phase electric motor of a machine are shown in a star circuit;

[0049] Figure 3 A bridge module of a bridge circuit of a converter for driving a phase winding of an electric motor is shown;

[0050] Figure 4 An equivalent circuit diagram for a current source is shown; and

[0051] Figure 5 A block diagram for field-oriented control of an electric motor is shown.

[0052] The present invention will be explained below based on a driving device with a B6 circuit by way of example. However, the present invention can also be applied to other arrangements.

[0053] In all the figures, parts and variables that correspond to one another are always provided with the same reference symbols. DETAILED DESCRIPTION

[0054] Figure 1 The electric machine 2 for an electric motor-driven drive of a vehicle (not shown in detail), for example a motor vehicle or an electrically driven or electrically drivable bicycle (e-bike) is shown. The machine 2 comprises a three-phase brushless electric motor 4, which is connected to a current source (voltage supply) 8 by means of a current converter (converter, inverter) 6. In the present exemplary embodiment, the current source 8 comprises a vehicle-internal energy storage device in the form of a (motor vehicle) battery 10 and a (DC voltage) intermediate circuit 12 connected to the energy storage device, which intermediate circuit extends at least partially into the current converter 6.

[0055] The intermediate circuit 12 is essentially formed by an outgoing line 12a and a return line 12b, by means of which the current converter 6 is connected to the battery 10. The lines 12a and 12b are at least partially led into the current converter 6, in which an intermediate circuit capacitor 14 and a bridge circuit 16 are interconnected between these lines.

[0056] When the machine 2 is in operation, the input current I supplied to the bridge circuit 16 is E The three-phase output current (motor current, rotation current) I converted into the three phases U, V, and W for the electric motor 4 U ,I V ,I W The output current I is also referred to as the phase current in the following text. U ,I V ,I W are led to the corresponding phases (windings) U, V, W of a stator (not shown in detail) Figure 2 ).

[0057] exist Figure 2 1 shows a star circuit 18 of three-phase windings U, V, W. The phase windings U, V and W are respectively connected to respective bridge modules 20 ( 20 ) of the bridge circuit 16 with (phase) ends 22 , 24 , 26 . Figure 3 ), and are interconnected with each other at opposite ends in a star point 28 as a common connection terminal. Figure 2 In the illustration of FIG, the phase windings U, V and W are each shown with the aid of an equivalent circuit diagram in the form of an inductance 30 and an ohmic resistor 32 and respective voltage drops 34, 36, 38. The voltages 34, 36, 38 dropped across the phase windings U, V, W are schematically represented by arrows and result from the sum of the voltage drops across the inductance 30 and the ohmic resistor 32 and the induced voltage 40. Figure 2 The voltage 40 induced by the movement of the rotor of the electric motor 4 is shown in FIG. 4 by means of circles (electromagnetic force, electromotive force, EMF).

[0058] The star circuit 18 is driven by means of a bridge circuit 16. The bridge circuit 16 is implemented in particular as a B6 circuit using a bridge module 20. In this design, during operation, a switch is made between a higher (DC) voltage level of the power supply line 12a and a lower voltage level of the return line 12b at a higher switching frequency on each phase winding U, V, W. The higher voltage level is in particular the intermediate circuit voltage U of the intermediate circuit 12. ZK , where the lower voltage level is preferably ground potential U G The clock-based control mode is implemented by the controller 42 as (in Figure 1 The PWM control (indicated by arrows) is implemented by means of which the speed, power and rotation direction of the electric motor 4 can be controlled and / or regulated.

[0059] The bridge module 20 includes two semiconductor switches 44 and 46, which are connected to each other in the Figure 2In FIG. 1 , only the W phase is shown schematically and by way of example. On the one hand, the bridge module 20 is connected to the power supply line 12 a and thus to the intermediate circuit voltage U 2 via a potential terminal 48 . ZK On the other hand, the bridge module 20 contacts the return line 12b with the second potential terminal 50, and then with the ground potential U G The respective phase ends 22 , 24 , 26 of the phases U, V, W can be connected to the intermediate circuit voltage U via semiconductor switches 44 , 46 either ZK Connection, either to ground potential U G If the semiconductor switch 44 is closed (conducting) and the semiconductor switch 46 is open (non-conducting, blocking), the phase ends 22, 24, 26 are connected to the intermediate circuit voltage U ZK Accordingly, when the semiconductor switch 44 is opened and the semiconductor switch 46 is closed, the phases U, V, W are connected to the ground potential U G Thus, it is possible to apply two different voltage levels to each phase winding U, V, W using PWM control.

[0060] exist Figure 3 2 shows a simplified representation of a single bridge module 20. In the present exemplary embodiment, semiconductor switches 44 and 46 are implemented as MOSFETs (metal oxide semiconductor field effect transistors), which are each switched between a switched state and a blocked state in a clocked manner by means of PWM control. For this purpose, the respective gate terminals are connected to corresponding control voltage inputs 52, 54, via which the signals of the PWM control of the controller 42 are transmitted.

[0061] Figure 4 An equivalent circuit diagram for the current source 8 is shown. In operation, the battery 10 generates a battery power P Bat ( Figure 5 )、Battery voltage U Bat And the corresponding battery current I Bat , for operating the current converter 6. Figure 4 In FIG. 5 , the internal resistance of the battery 10 is shown as an ohmic resistance 56, and the self-inductance of the battery 10 is shown as an inductance 58. A shunt resistor 60 is connected in the loop 12b.

[0062] Depending on the switching state of the (power) semiconductor switches 44 , 46 , the phase current I U ,I V ,I W The phase current I flows through the shunt resistor 60. The voltage drop across the shunt resistor 60 is amplified and evaluated. The controller 42 reconstructs the phase current I by measuring and detecting the switching state of the semiconductor switches 44 and 46. U ,I V ,I WOther measurement measures can also be used to determine the motor current (such as direct phase current measurement). U , U V , U W ) together, the phase voltage (U U , U V , U W ) and phase current I U ,I V ,I W For use by controller 42.

[0063] exist Figure 1 In the embodiment of the present invention, the motor current is detected by means of an ammeter 62, for example, by means of a shunt resistor 60, and is directed to the controller 42. The controller 42 detects the motor current based on the motor parameters, in particular the detected phase current I U ,I V ,I W And the calculated phase voltage U U , U V , U W And based on other parameters (such as motor resistance, motor inductance, PWM voltage duty cycle), the rotation parameters θ, ω, namely the motor position (rotor position) θ and / or (rotor) speed ω, are calculated or estimated. In particular, the electric motor position θ is calculated or estimated. el Or electrical frequency / speed ω el .

[0064] Figure 5 A block diagram is shown for operating the electric machine 2 according to the method, wherein the electric motor 4 is controlled and / or regulated in a dq reference system having a d-axis and a q-axis.

[0065] According to the method, the regulating unit 64 of the controller 42 predetermines the current target value I for the target current along the d-axis and the q-axis. dsoll and I qsoll The control unit 64 determines a corresponding actuation signal U for the voltage used to drive the electric motor 4 by means of field-oriented control. dFOC and U qFOC .

[0066] At the beginning of operation, for example, when the electric motor 4 is started from a stationary state, the back electromotive force is not sufficient to generate enough voltage 40, so that the current signal measured by the ammeter 62 is not sufficient to determine the motor position θ el Therefore, the target current value I for adjusting unit 64 is dsoll and I qsoll cannot be determined based on the measured current signal. Therefore, at the beginning of the method or run, the controller estimates the motor position θ el, and accordingly determine the target current value I for regulating unit 64 dsoll and I qsoll .

[0067] In order to check and adjust the position estimate, a test signal or test pulse U is generated by the compensation unit 66. dTest By means of a test signal or a test pulse, the deviation between the actual motor position and the estimated motor position can be determined. dTest is fed into the d-axis of the electric motor 4; in particular, the test signal U dTest With the control signal U dFOC Add.

[0068] In addition to the test signal U dTest In addition, the compensation unit 66 also generates a compensation signal U qComp , which is fed into the q axis of the electric motor 4. In particular, the compensation signal U qComp With the control signal U qFOC Add. Compensation signal U qComp Based on the test signal U dTest Determined by calculation 68, so that the compensation signal U qComp When fed into the q axis, reduce or compensate for the test signal U dTest The resulting torque change of the electric motor 4.

[0069] Test signal U dTest and compensation signal U qComp Corrected control signal U d '、U q ', is converted into a corresponding PWM signal for driving the bridge circuit 16 by a PWM driver not shown in detail.

[0070] Preferably, the compensation signal U qComp As far as possible, the test signal U dTest The resulting change in the torque of the electric motor 4 allows a particularly smooth starting of the electric motor 2 .

[0071] Used to determine the compensation signal U qComp The calculation 68 of is based on a motor model for the electric motor 4. Preferably, the calculation 68 is based on Figure 5 The purpose of the modified actuation signal U shown in FIG. d '、U q 'Converted into motor current I q and I d The dq motor model of the electric motor 4 is used as the basis.

[0072] The following will be based on Figure 5 Explain the dq motor model in detail.

[0073] Used to correct the control signal U d 'Converted into motor current I d The d-model includes the coil component and the ohmic component on the one hand and the reactance component on the other hand.

[0074] The coil share is 1 / L d Coefficient (where L d An amplifier 70 having an inductance value (wherein d is the inductance value of the inductor 30 along the d-axis) and modeled by an integrator 72 .

[0075] The ohmic component corresponds to the voltage loss caused by the ohmic resistance 32 . The ohmic component is implemented as a negative feedback of an amplifier 74 with a factor R (R being the ohmic resistance 32 of the stator or of the electric motor 4 ).

[0076] The reactance or reactive impedance is based on the motor current I q The reactance component is determined (e.g. estimated) by the controller for the motor speed or rotor frequency ω el . Motor speed ω el and the motor current I determined by the q model explained below q Multiplied by the coefficient L through amplifier 76 q , where L q is the inductance value of the inductor 30 along the q axis. The reactance component is added to the adjustment value U d 'superior.

[0077] Used to correct the control signal U q 'Converted into motor current I q The q-model includes the coil component and the ohmic component on the one hand, and the reactance component and the inductive component on the other hand.

[0078] The coil share is 1 / L q The coefficient amplifier 78 is modeled by an integrator 80.

[0079] The ohmic component is implemented using a negative feedback of an amplifier 82 having a resistance value R.

[0080] For reactance share or reactive impedance share, motor speed ω el The motor current I determined by the d model d Multiply by the coefficient L through amplifier 84 d From the adjustment value U q 'Minus the reactance share.

[0081] The induction share is used as the motor speed ω elThe product of the magnetic flux Ψ is modeled, wherein the product is implemented as an amplifier 86 with a coefficient PSiM and wherein PSiM corresponds to the magnetic flux Ψ. q 'Minus the induction share.

[0082] For example, the compensation signal U qComp By calculation based on the following formula 68 Here, Δu q is the compensation signal U qComp , Δu d is the test signal U dTest ,ω el is the motor speed, L d and L q is the inductance value of the inductor 30 along the d-axis or q-axis, R S is the resistance value of the ohmic resistor 32, s is the Laplace transformation parameter, and k is the compensation coefficient.

[0083] In order to reduce the computational effort, a simplified calculation 68 can also be used in which at least one approximation is performed.

[0084] For low rotor frequencies (ω el ~ 0 kHz), for example when starting from standstill, the above formula can be used In addition, if R S Can also be ignored, then we get:

[0085] If R S can be ignored, but el Cannot be ignored, then: Among them, Δu q is the compensation signal U qComp , Δu d is the test signal U dTest ,ω el is the motor speed, L d and L q is the inductance value of the inductor 30 along the d-axis or the q-axis, s is the Laplace transformation parameter, and k is the compensation coefficient.

[0086] The compensation factor k is based in particular on the following formula Determine, where Ψ is the magnetic flux, i q and i dis the motor current I q and I d , and Δi q and Δi d Because the test signal U dTest and compensation signal U qComp The resulting current changes along the d-axis and q-axis. In order to simplify the calculation 68 and thus to reduce the computational effort, a simplified proportional calculation can be used. For large magnetic flux The above ratio calculation can be used Perform approximate calculations.

[0087] In order to determine the compensation signal in a particularly resource-saving manner and with reduced computational effort, the above-mentioned approximations can be combined. While ignoring R S and ω el The impact applies when: Among them, Δu q is the compensation signal U qComp , Δu d is the test signal U dTest , Ψ is the magnetic flux, and L d / L q is the inductance value of the inductor 30 along the d / q axis.

[0088] For example, a sinusoidal voltage with a constant frequency is used as the test signal U dTest Therefore, in calculation 68, we can use to replace the Laplace transform parameter s, where j is the imaginary unit and ω ud is the frequency of the test signal. So we have

[0089] In order to reduce the computational workload, a simplified and / or ignore R S and ω el .

[0090] For the test signal U dTest and compensation signal U qComp The motor response, that is, the motor current I q and I d , detected by current measurement 62, and used to determine the motor position or to determine the deviation between the estimated motor position and the actual motor position. The deviation is adjusted by the controller 42 so that it is as small as possible, so that the estimated motor position and the actual motor position are as consistent as possible.

[0091] The claimed invention is not limited to the above-described embodiments. On the contrary, a person skilled in the art may also derive other variants of the invention within the scope of the disclosed claims without departing from the subject matter of the claimed invention. In particular, all individual features described in conjunction with the different embodiments may also be combined in other ways within the scope of the disclosed claims without departing from the subject matter of the claimed invention.

[0092] Reference numerals list

[0093] 2 Machines

[0094] 4 Electric motor

[0095] 6 Current Converter

[0096] 8 Current Source

[0097] 10. Battery

[0098] 12 Intermediate circuit

[0099] 12a Outgoing line

[0100] 12b loop line

[0101] 14 Intermediate circuit capacitor

[0102] 16 Bridge Circuit

[0103] 18 Star Circuit

[0104] 20 Bridge Module

[0105] 22, 24, 26 phase ends

[0106] 28 star points

[0107] 30 Inductor

[0108] 32 Resistors

[0109] 34, 36, 38 Voltage drop

[0110] 40 Voltage

[0111] 42 Controller

[0112] 44, 46 Semiconductor switch

[0113] 48, 50 potential terminals

[0114] 52, 54 Control voltage input terminal

[0115] 56 Resistors

[0116] 58 Inductor

[0117] 60 shunt resistor

[0118] 62 Ammeter

[0119] 64 Adjustment units

[0120] 66 Compensation Unit

[0121] 70 Amplifier

[0122] 72 Integrator

[0123] 74, 76, 78 amplifiers

[0124] 80 Integrator

[0125] 82, 84, 86 amplifiers

[0126] U, V, W phase / phase winding

[0127] I U ,I V ,I W Phase current / output current

[0128] I E Input Current

[0129] U ZK Intermediate circuit voltage

[0130] U G Ground potential

[0131] I Bat Battery Current

[0132] U Bat Battery voltage

[0133] ω el (Electric) frequency / motor speed

[0134] I dsoll ,I qsoll Current target value

[0135] U dFOC , U qFOC Manipulating signals

[0136] U dTest Test signal

[0137] U qComp Compensation signal

[0138] I q ,I d Motor current

[0139] U d '、Uq 'Manipulation signal

Claims

1. A method for operating an electric machine (2) comprising a brushless electric motor (4) with a stator and a rotor and comprising a control unit (64) for sensorless field-oriented control and / or regulation of the electric motor (4) in a dq reference system fixed relative to the rotor with a d-axis and a q-axis, -in, At the start of operation, controlling and / or regulating the electric motor (4) based on the estimated motor position, wherein a test signal (U) is generated for determining the deviation between the actual motor position and the estimated motor position dTest ), - wherein the test signal (U dTest ) is fed into the d-axis, - wherein, based on the test signal (U dTest ) determines the compensation signal (U qComp ), so that the compensation signal (U qComp ) is fed into the q axis to reduce the test signal (U dTest ) caused by the torque change of the electric motor (4), and - wherein the compensation signal (U qComp ) is fed into the q-axis.

2. The method according to claim 1, It is characterized in that Through the compensation signal (U qComp ) is completely reduced due to the test signal (U dTest ) caused by the torque change of the electric motor (4).

3. The method according to claim 1 or 2, It is characterized in that Detection of the test signal (U dTest ) and the compensation signal (U qComp ) and use the motor response to determine motor positioning.

4. The method according to any one of claims 1 to 3, It is characterized in that The compensation signal (U qComp ) is based on the following formula To determine, among them, -Δu q is the compensation signal (U qComp ), -Δu d is the test signal (U dTest ), -ω el is the frequency of the rotor, -L d is the inductance of the stator along the d-axis, -L q is the inductance of the stator along the q axis, -R S is the ohmic resistance of the stator, -s is the Laplace transform parameter, and -k is the compensation factor.

5. The method according to any one of claims 1 to 4, It is characterized in that The compensation signal (U qComp ) is based on the following formula To determine, among them, -Δu q is the compensation signal (U qComp ), -Δu d is the test signal (U dTest ), -R S is the ohmic resistance of the stator, -L d is the inductance of the stator along the d-axis, -L q is the inductance of the stator along the q axis, -s is the Laplace transform parameter, and -k is the compensation factor.

6. The method according to any one of claims 1 to 5, It is characterized in that The compensation signal (U qComp ) is based on the following formula To determine, among them, -Δu q is the compensation signal (U qComp ) -Δu d is the test signal (U dTest ), -Ψ is the magnetic flux, -L d is the inductance of the stator along the d-axis, and -L q is the inductance of the stator along the q-axis.

7. The method according to any one of claims 4 to 6, It is characterized in that The compensation coefficient k is based on the following formula To determine, among them, -i q is the motor current along the q-axis, -i d is the motor current along the d-axis, -Δi q is the current change along the q-axis caused by the compensation signal, -Δi d is the current change along the d-axis caused by the test signal, -Ψ is the magnetic flux, -L d is the inductance of the stator along the d-axis, and -L q is the inductance of the stator along the q-axis.

8. The method according to any one of claims 4 to 7, It is characterized in that The compensation coefficient k is based on the following formula To determine, among them, -i q is the motor current along the q-axis, -Ψ is the magnetic flux, -L d is the inductance of the stator along the d-axis, and -L q is the inductance of the stator along the q-axis.

9. The method according to any one of claims 1 to 8, It is characterized in that A sinusoidal voltage with a constant frequency is used as the test signal (U dTest ).

10. A motor (2), comprising: - a brushless electric motor (4) with a rotor and a stator, - a control unit (64) for sensorless, field-oriented control and / or regulation of the electric motor (4), and - A controller (42) for performing the method according to any one of claims 1 to 9.

11. Software on a data carrier for carrying out the method according to any one of claims 1 to 9 when the software is run on a computer.

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