Single-phase open-circuit fault-tolerant control system and method for dual three-phase permanent magnet synchronous motors

Abstract

The present invention discloses a single-phase open-circuit fault-tolerant control system and method for a dual three-phase permanent magnet synchronous motor. The control system includes a PI controller, a PIR controller, an SVPWM module, a phase current collector, a second coordinate transformation module, and a z1-z2 current loop fault-tolerant controller. The z1-z2 current loop fault-tolerant controller includes a notch filter, a harmonic QPR controller, and an adaptive resonant controller. The adaptive resonant controller is a fundamental frequency QPR controller connected in series with a switch, and the switch is controlled on and off by a z1-z2 axis reference current signal. The harmonic QPR controller includes parallel third, fifth, and / or seventh harmonic QPR controllers, the notch filter and the harmonic QPR controller are connected in series and then in parallel with the adaptive resonant controller. The z1-z2 current loop fault-tolerant controller inputs the z1-z2 axis actual current signal and outputs the z1-z2 axis reference voltage signal to the SVPWM module. The present invention achieves smooth transient fault transition and high torque output during fault-tolerant steady-state operation.

Description

Technical Field

[0001] The present invention relates to a single-phase open-circuit fault-tolerant control system for a synchronous motor, and in particular to a single-phase open-circuit fault-tolerant control system and method for a dual three-phase permanent magnet synchronous motor. Background Art

[0002] Dual-three-phase permanent magnet synchronous motors (DT-PMSMs) offer advantages such as high power density, multiple degrees of freedom, and strong fault tolerance, and hold great promise for application in fields such as ship propulsion, electric vehicles, and aerospace. Open-circuit winding failures caused by mechanical, electrical, or environmental stresses are inevitable in motor systems, and high-performance fault-tolerant control for these open-circuit faults has become a research hotspot.

[0003] Currently, most conventional fault-tolerant control strategies for multiphase motors are based on full-order vector space decoupling transformations. Since the motor's decoupling transformation matrix remains unchanged after a phase loss, the voltage, flux, and torque equations remain unaffected. Only additional current constraints are required: coupling the fundamental and harmonic plane currents. The essence of fault-tolerant control is to modify the harmonic plane reference current to avoid control conflicts with the fundamental plane current. This new residual healthy phase current is then derived to ensure that the resultant rotating magnetomotive force remains unchanged, allowing the motor to continue operating undisturbed.

[0004] After a single-phase open-circuit fault, the multi-phase motor still has additional control degrees of freedom. The solution to the fault-tolerant reference current is not unique, and the reference current can be modified according to different current optimization objectives. Due to the lack of open-circuit phase winding current, regardless of the current optimization objective, the effective value of the current of some phase windings will be greater than the normal value after the fault to output the same torque. In order to avoid winding overheating and secondary faults caused by the effective value of the phase current exceeding the rated value under fault-tolerant operation, different current optimization objectives have a torque output range to avoid the corresponding maximum phase current effective value exceeding the rated value after the torque output exceeds the limit. Accordingly, the torque output capacity can be defined as the ratio of the effective value of the phase current during normal operation to the effective value of the maximum phase current during fault-tolerant operation.

[0005] The optimization goal of the minimum copper loss (ML) strategy is to provide the lowest stator copper loss at each torque. However, some phase currents have large amplitudes, reaching the rated current quickly after the torque reference is increased, resulting in lower torque output capacity. The optimization goal of the maximum torque (MT) strategy is to minimize the maximum amplitude of the phase current. This allows MT to achieve the maximum torque output range after a fault. An improved MT technique, based on the relationship between torque output capacity and single-phase copper loss under open-circuit faults, simultaneously optimizes the copper loss of the remaining healthy five-phase windings without considering constraints such as the sinusoidal current pattern, achieving higher torque output capacity than the traditional MT strategy.

[0006] These traditional fault-tolerance methods require pre-fault detection before modifying the current reference to switch to fault-tolerant operation. However, fault detection inherently involves a detection delay. During this detection time, the current reference remains constant, and conflicting control commands between the fundamental and harmonic planes can cause significant fluctuations in speed and torque.

[0007] To improve transient performance during the transition from normal mode to fault-tolerant mode, conflict-free natural fault-tolerant control strategies have attracted widespread attention. One fault-tolerant self-healing scheme iteratively replaces the harmonic plane current reference with the actual value, gradually shifting the harmonic plane current toward its ideal value. However, this strategy still requires a transition period from healthy to fault-tolerant operation. Another approach evaluated the natural fault-tolerant capability of utilizing a virtual voltage vector. The core concept is to synthesize the reference voltage of the harmonic plane to zero to avoid conflicts with the fundamental plane controller after a fault. Researchers at home and abroad maintain closed-loop control of the harmonic plane current. After an open-circuit fault occurs, the harmonic plane reference voltage reaches a set low saturation threshold, automatically deactivating the PI controller, thereby achieving control compatibility between the fundamental and harmonic planes. Furthermore, some researchers have further clarified that the coupling component of the harmonic plane is the fundamental frequency current and proposed corresponding fault-tolerant strategies. The basic idea is to embed a filter in the harmonic plane to filter out the fundamental component, releasing control of the fundamental frequency current in the harmonic plane to achieve natural fault tolerance.

[0008] Although the natural fault-tolerant control strategy improves the transient performance of the DT-PMSM before and after an open-circuit fault, the current target of the natural fault-tolerant control approaches the ML current mode when the harmonic plane open-loop control and the fundamental plane closed-loop control are combined. Previous natural fault-tolerant control methods failed to further improve the torque output range because they did not control the fundamental frequency current in the harmonic plane to meet the MT current mode. Furthermore, the conventional MT current mode requires the injection of the third, fifth, and seventh harmonics of the phase current, which requires complex controller design and makes it difficult to accurately track the high-frequency current. Summary of the Invention

[0009] This paper proposes a single-phase open-circuit fault-tolerant control system and method for a dual three-phase permanent magnet synchronous motor based on an adaptive resonant controller. This method improves the torque output capability of the dual three-phase permanent magnet synchronous motor under fault conditions while taking into account fault transient performance.

[0010] The technical solution adopted by the present invention to solve the technical problems existing in the known technology is:

[0011] A single-phase open-circuit fault-tolerant control system for a dual three-phase permanent magnet synchronous motor. The dual three-phase permanent magnet synchronous motor is a surface-mount DT-PMSM with two sets of windings with isolated neutral points and a 30° phase shift. The driver of the surface-mount DT-PMSM includes a first converter and a second converter. The two converters are controlled by the same control system. The control system includes a PI controller, a PIR controller, a first coordinate transformation module, and an SVPWM module connected in sequence, as well as a phase current collector, a second coordinate transformation module, and a z1-z2 current loop fault-tolerant controller.

[0012] The phase current collector is used to collect the phase current of each phase motor winding;

[0013] The second coordinate transformation module is used to map the variables in the natural coordinate system to the fundamental wave dq plane and the harmonic z1-z2 plane. It inputs the current signals of each phase collected by the phase current collector; it outputs the actual current signals of the dq axes to the PIR controller, and it outputs the actual current signals of the z1-z2 axes;

[0014] The PI controller inputs the rotor reference angular velocity signal and the rotor actual angular velocity signal respectively; it outputs the q-axis reference current signal;

[0015] The PIR controller inputs a q-axis reference current signal, a d-axis reference current signal, and a dq-axis actual current signal; it outputs a dq-axis reference voltage signal to the first coordinate transformation module;

[0016] The first coordinate transformation module is used to transform the variables in the dq coordinate system into variables in the α-β coordinate system; it outputs the α-β reference voltage signal to the SVPWM module;

[0017] The z1-z2 current loop fault-tolerant controller includes a notch filter, a harmonic QPR controller and an adaptive resonant controller; the adaptive resonant controller is a fundamental frequency QPR controller connected in series with a switching switch, and the switching switch is controlled by the z1-z2 axis reference current signal; the notch filter is used to filter out the fundamental frequency current component mapped from the α-β plane to the z1-z2 plane; the harmonic QPR controller includes parallel third, fifth and / or seventh harmonic QPR controllers, the notch filter is connected in series with the harmonic QPR controller and then connected in parallel with the adaptive resonant controller; the z1-z2 current loop fault-tolerant controller, which inputs the z1-z2 axis actual current signal from the second coordinate transformation module, and outputs the z1-z2 axis reference voltage signal to the SVPWM module;

[0018] The SVPWM module inputs the α-β reference voltage signal and the z1-z2 axis reference voltage signal, and outputs PWM signals to the first converter and the second converter respectively.

[0019] Furthermore, the system also includes a current vector trajectory module and an MT optimization controller; the current vector trajectory module is used to track the harmonic plane current trajectory, and its input is the actual current signal of the z1-z2 axis from the second coordinate transformation module, and it outputs the current vector angle of the z1-z2 plane; the MT optimization controller optimizes the control parameters through an optimization tool, and its optimization goal is to minimize the maximum single-phase copper loss. It inputs the q-axis reference current signal from the PI controller and the current vector angle of the z1-z2 plane from the current vector trajectory module, and it outputs the z1-z2 axis reference current signal to the z1-z2 current loop fault-tolerant controller.

[0020] Furthermore, assume that one of the two sets of windings includes A, B, and C phase windings, and the other set of windings includes D, E, and F phase windings; assume that x is any one of the A, B, C, D, E, or F phase windings;

[0021] The optimization function of the MT optimization controller is as follows:

[0022]

[0023] p max =max{p A ,p B ,p C ,p D ,p E ,p F};

[0024] The copper loss calculation formula of the x-phase winding is as follows:

[0025]

[0026] The coefficients in the formula are

[0027]

[0028] Where:

[0029] is the spatial phase angle of the x-phase winding; x∈{A, B, C, D, E, F}; when x is A, B, C, D, E, F phase respectively,

[0030] k d is the amplitude coefficient of the d-axis current;

[0031] p x is the copper loss of x-phase winding;

[0032] p A is the copper loss of phase A winding;

[0033] p B is the copper loss of phase B winding;

[0034] p C is the copper loss of phase C winding;

[0035] p D is the copper loss of D-phase winding;

[0036] p E is the copper loss of E-phase winding;

[0037] p F is the copper loss of F phase winding;

[0038] R s is the phase winding resistance of the motor stator;

[0039] i x is the x-phase current;

[0040] i d is the d-axis current in the dq plane;

[0041] i q is the q-axis current in the dq plane;

[0042] I dc is the DC current coefficient in the phase winding copper loss;

[0043] α d is the phase deviation angle of the d-axis current;

[0044] θ e is the electrical angle of the motor rotor position;

[0045] I1 is the cosine coefficient of phase winding copper loss;

[0046] I2 is the sinusoidal coefficient of phase winding copper loss;

[0047] k1, k2, k3, and k4 are coefficients to be optimized;

[0048] i α is the α-axis current in the α-β plane;

[0049] i β is the β-axis current in the α-β plane;

[0050] i z1 is the z1-axis current in the z1-z2 plane;

[0051] i z2 is the z2-axis current in the z1-z2 plane;

[0052] p max is the maximum single-phase copper loss;

[0053] min p max Indicates minimizing the maximum single-phase copper loss of the motor;

[0054] The optimization function of the MT optimization controller is solved by the MATLAB optimization toolbox, and the optimal solution of the following coefficients is obtained: α d 、k d , k1, k2, k3, k4.

[0055] Furthermore, a limiting module is included between the PI controller and the PIR controller. The limiting module performs limiting processing on the q-axis reference current signal output by the PI controller, and outputs the q-axis reference current signal after the limiting processing to the PIR controller.

[0056] The present invention also provides a single-phase open-circuit fault-tolerant control method for a dual three-phase permanent magnet synchronous motor. The dual three-phase permanent magnet synchronous motor is a surface-mounted DT-PMSM with two sets of winding neutral points isolated and a phase shift of 30 degrees. The driver of the surface-mounted DT-PMSM includes a first converter and a second converter. The two converters are controlled by the same control system. The control system is provided with a PI controller, a PIR controller, a first coordinate transformation module, and an SVPWM module connected in sequence, as well as a phase current collector, a second coordinate transformation module, and a z1-z2 current loop fault-tolerant controller.

[0057] The phase current collector is used to collect the phase current of each phase motor winding;

[0058] The second coordinate transformation module is used to map the variables in the natural coordinate system to the fundamental wave dq plane and the harmonic z1-z2 plane, so that it inputs the current signals of each phase collected by the phase current collector; it outputs the actual current signals of the dq axes to the PIR controller, which outputs the actual current signals of the z1-z2 axes;

[0059] The PI controller is input with the rotor reference angular velocity signal and the rotor actual angular velocity signal respectively; and is output with the q-axis reference current signal;

[0060] The PIR controller is configured to input a q-axis reference current signal, a d-axis reference current signal, and a dq-axis actual current signal; and to output a dq-axis reference voltage signal to the first coordinate transformation module;

[0061] The first coordinate transformation module is used to transform the variables in the dq coordinate system into variables in the α-β coordinate system, so that it outputs the α-β reference voltage signal to the SVPWM module;

[0062] The z1-z2 current loop fault-tolerant controller is provided with a notch filter, a harmonic QPR controller and an adaptive resonant controller; the adaptive resonant controller is a fundamental frequency QPR controller connected in series with a switching switch, and the switching switch is controlled by the z1-z2 axis reference current signal; the notch filter is used to filter out the fundamental frequency current component mapped from the α-β plane to the z1-z2 plane; the harmonic QPR controller is provided with parallel third, fifth and / or seventh harmonic QPR controllers, so that the notch filter and the harmonic QPR controller are connected in series and then in parallel with the adaptive resonant controller; the z1-z2 current loop fault-tolerant controller is input with the z1-z2 axis actual current signal from the second coordinate transformation module, so that it outputs the z1-z2 axis reference voltage signal to the SVPWM module;

[0063] The SVPWM module is input with the α-β reference voltage signal and the z1-z2 axis reference voltage signal, so that the module outputs the PWM signal to the first converter and the second converter respectively;

[0064] During the normal operation and natural fault-tolerant operation stages of the motor, the z1-z2 axis reference current is zero, the switching switch is disconnected, and the adaptive resonant controller is in the fundamental frequency current open-loop control mode; during the current optimization operation stage, when the z1-z2 axis reference current is not zero, the switching switch is turned on, and the adaptive resonant controller is in the fundamental frequency current closed-loop control mode.

[0065] Furthermore, the method also provides a current vector trajectory module and an MT optimization controller; the current vector trajectory module is used to track the harmonic plane current trajectory, so that it inputs the actual current signal of the z1-z2 axis from the second coordinate transformation module, and outputs the current vector angle of the z1-z2 plane; the MT optimization controller optimizes the control parameters through an optimization tool, so that its optimization target is to minimize the maximum single-phase copper loss, so that it inputs the q-axis reference current signal from the PI controller and the current vector angle of the z1-z2 plane from the current vector trajectory module respectively, and outputs the z1-z2 axis reference current signal to the z1-z2 current loop fault-tolerant controller.

[0066] At the moment of fault, the adaptive resonant controller is disabled, and the fundamental frequency filtering effect of the notch filter allows the fundamental frequency component related to the torque to flow through the harmonic plane without interference; after the MT optimization controller outputs a new z1-z2 axis reference current based on the current vector angle of the z1-z2 plane, the adaptive resonant controller is reactivated to follow the fundamental frequency current reference value of the harmonic plane, switching from the natural fault-tolerant operation stage to the current control optimization operation stage.

[0067] Furthermore, assume that one of the two sets of windings includes A, B, and C phase windings, and the other set of windings includes D, E, and F phase windings; assume that x is any one of the A, B, C, D, E, or F phase windings;

[0068] The optimization function of the MT optimization controller is as follows:

[0069]

[0070] p max =max{p A ,p B ,p C ,p D ,p E ,p F};

[0071] The copper loss calculation formula of the x-phase winding is as follows:

[0072]

[0073] The coefficients in the formula are

[0074]

[0075] Where:

[0076] is the spatial phase angle of the x-phase winding; x∈{A, B, C, D, E, F}; when x is A, B, C, D, E, F phase respectively,

[0077] k d is the amplitude coefficient of the d-axis current;

[0078] p x is the copper loss of x-phase winding;

[0079] p A is the copper loss of phase A winding;

[0080] p B is the copper loss of phase B winding;

[0081] p C is the copper loss of phase C winding;

[0082] p D is the copper loss of D-phase winding;

[0083] p E is the copper loss of E-phase winding;

[0084] p F is the copper loss of F phase winding;

[0085] R s is the phase winding resistance of the motor stator;

[0086] i x is the x-phase current;

[0087] i d is the d-axis current in the dq plane;

[0088] i q is the q-axis current in the dq plane;

[0089] I dc is the DC current coefficient in the phase winding copper loss;

[0090] α d is the phase deviation angle of the d-axis current;

[0091] θ e is the electrical angle of the motor rotor position;

[0092] I1 is the cosine coefficient of phase winding copper loss;

[0093] I2 is the sinusoidal coefficient of phase winding copper loss;

[0094] k1, k2, k3, and k4 are coefficients to be optimized;

[0095] i α is the α-axis current in the α-β plane;

[0096] i β is the β-axis current in the α-β plane;

[0097] i z1 is the z1-axis current in the z1-z2 plane;

[0098] i z2 is the z2-axis current in the z1-z2 plane;

[0099] p max is the maximum single-phase copper loss;

[0100] min p max Indicates minimizing the maximum single-phase copper loss of the motor;

[0101] The optimization function of the MT optimization controller is solved by the MATLAB optimization toolbox, and the optimal solution of the following coefficients is obtained: α d 、k d , k1, k2, k3, k4.

[0102] Furthermore, after a phase failure occurs, the MT optimization controller determines the fault phase based on the current vector trajectory direction on the harmonic plane using the following method:

[0103] When the current in the z1-z2 plane moves along the 0 / 180° direction, it is determined that phase A is faulty;

[0104] When the current in the z1-z2 plane moves along the -120° / 60° direction, it is determined that the B phase is faulty;

[0105] When the current in the z1-z2 plane moves along the 120° / -60° direction, it is determined that the C phase is faulty;

[0106] When the current in the z1-z2 plane moves along the 150° / -30° direction, it is determined that the D phase is faulty;

[0107] When the current in the z1-z2 plane moves along the 30° / -150° direction, it is determined that the E phase is faulty;

[0108] When the current in the z1-z2 plane moves along the -90° / 90° direction, it is determined that the F phase is faulty;

[0109] Based on the fault phase information, the MT optimization controller obtains the optimal coefficient in the maximum torque current mode and calculates the output z1-z2 axis reference current signal.

[0110] Furthermore, a limiting module is provided between the PI controller and the PIR controller. The limiting module performs limiting processing on the q-axis reference current signal output by the PI controller, and outputs the q-axis reference current signal after limiting processing to the PIR controller. After a fault occurs, the limiting value of the limiting module is changed according to the z1-z2 current reference value so that the motor can operate safely without exceeding the limit.

[0111] The advantages and positive effects of the present invention are as follows: In response to the inherent problems of natural fault-tolerant control technology, the present invention takes the surface-mounted DT-PMSM as the research object and proposes a single-phase open-circuit fault-tolerant control strategy based on an adaptive resonant controller. During the transition period from when the motor fails to when it enters the current optimization operation stage, that is, during the moment of the fault, the fundamental frequency current of the harmonic plane is open-loop controlled, so that the motor operates in the natural fault-tolerant operation stage and inherits the same good fault transient characteristics as the natural fault-tolerant control. Furthermore, the harmonic plane current vector trajectory under the natural fault-tolerant operation state is used to obtain a unified new MT reference current. The adaptive resonant controller then resumes control of the fundamental frequency current of the harmonic plane to follow the reference value, smoothly switching from the natural fault-tolerant operation stage to the current optimization operation stage, and only injecting the third harmonic current can obtain the maximum torque output capacity.

[0112] The proposed fault-tolerant strategy, along with traditional natural fault-tolerant control, can both achieve a smooth transition from a healthy state to a natural fault-tolerant state at the moment of a fault. The proposed fault-tolerant strategy is superior in that it quickly and smoothly switches from the ML mode inherent in the natural fault-tolerant state to the MT mode after a fault, effectively improving torque output capacity.

[0113] Conventional MT fault-tolerant modes either offer limited torque boost or require the controller to handle multiple harmonics and high-frequency currents, leading to complex controller design and poor control accuracy. In contrast, the novel MT fault-tolerant mode proposed in this paper eliminates the zero-value constraint on the d-axis current and achieves the desired high torque output capability by injecting only the second harmonic component. This allows for convenient calculation of optimal coefficients and simplified controller design.

[0114] The fault-tolerant control strategy proposed in the present invention combines the advantages of natural fault-tolerant control and MT fault-tolerant mode. While ensuring low algorithm complexity, the present invention achieves smooth transition of fault transients and a high torque output range for fault-tolerant steady-state operation. BRIEF DESCRIPTION OF THE DRAWINGS

[0115] Figure 1 is the topology of the converter and DT-PMSM winding;

[0116] Figure 2 is the natural fault-tolerant control phase current waveform under phase A fault;

[0117] Figure 3 The DT-PMSM single-phase open-circuit fault-tolerant control scheme proposed by the present invention;

[0118] Figure 4 This is the block diagram of the z1-z2 current loop fault-tolerant controller;

[0119] Figure 5 Bode diagram of the z1-z2 current loop open-loop transfer function under open-loop control of the baseband QPR controller;

[0120] Figure 6 Bode diagram of the open-loop transfer function of the z1-z2 current loop under closed-loop control of the baseband QPR controller;

[0121] Figure 7 This is the phase current waveform of the new MT fault-tolerant mode under phase A fault.

[0122] In the picture:

[0123] Indicates: the reference electrical angular velocity of the motor.

[0124] ω e (k) represents: the actual electrical angular velocity of the motor.

[0125] Indicates: q-axis reference current.

[0126] Indicates: d-axis reference current.

[0127] Represents: the reference current vector in the z1-z2 plane.

[0128] i s (k) represents the actual current vector in the dq plane.

[0129] i z (k) represents: the actual current vector in the z1-z2 plane.

[0130] Represents: the reference voltage vector of the dq plane.

[0131] Represents: the reference voltage vector of the z1-z2 plane.

[0132] Represents: the reference voltage vector of the α-β plane.

[0133] i A,B,C (k) represents the current sampling value of the A, B, and C phase windings.

[0134] i A,B,C (t) represents the actual current value of the A, B, and C phase windings.

[0135] i D,E,F (k) represents the current sampling value of the D, E, and F phase windings.

[0136] i D,E,F (t) represents the actual current value of the D, E, and F phase windings.

[0137] d / dt represents the mathematical derivative symbol.

[0138] PIR stands for Proportional-Integral-Resonant controller.

[0139] PI stands for Proportional Integral controller.

[0140] DT-PMSM stands for Dual Three-Phase Permanent Magnet Synchronous Motor.

[0141] VSC stands for Voltage Source Converter Module.

[0142] SVPWM stands for Vector Space Pulse Width Modulation module.

[0143] G pz (z) is the reference voltage on the z1-z2 plane to i z (z) transfer function;

[0144] i z (z) is the discretized current vector in the z1-z2 plane;

[0145] is the discretized reference current vector in the z1-z2 plane;

[0146] is the discretized reference voltage vector in the z1-z2 plane;

[0147] δ z (z) is the discretized periodic perturbation in the z1-z2 plane;

[0148] G QPR-1 (z) is the transfer function of the first-order QPR controller in the discrete domain;

[0149] G QPR-n (z) is the transfer function of the n-th order QPR controller in the discrete domain;

[0150] G NF (z) is the transfer function of the fundamental frequency notch filter in the discrete domain. DETAILED DESCRIPTION

[0151] The present invention will be described in detail below with reference to the accompanying drawings and in combination with embodiments. It should be understood that the preferred embodiments described herein are only used to illustrate and explain the present invention and are not used to limit the present invention.

[0152] The Chinese meanings of the following English words, phrases and abbreviations, as well as English-Chinese combinations, are as follows:

[0153] PI controller: Proportional-Integral controller.

[0154] PIR controller: Proportional-Integral-Resonant controller.

[0155] SVPWM module: vector space pulse width modulation module.

[0156] PWM signal: pulse width modulation signal.

[0157] VSC module: voltage source inverter module.

[0158] ML: Minimum copper loss mode.

[0159] MT: Maximum torque mode.

[0160] DT-PMSM: Dual three-phase permanent magnet synchronous motor.

[0161] VSD: Vector Space Decoupling.

[0162] Bode diagram: Bode diagram.

[0163] QPR controller: Quasi-proportional resonant controller.

[0164] Matlab: Commercial mathematical software produced by MathWorks, an American company, used in data analysis, wireless communications, deep learning, image processing and computer vision, signal processing, quantitative finance and risk management, robotics, control systems and other fields.

[0165] See Figures 1 to 7 A single-phase open-circuit fault-tolerant control system for a dual three-phase permanent magnet synchronous motor. The dual three-phase permanent magnet synchronous motor is a surface-mount DT-PMSM with two sets of windings with isolated neutral points and a 30° phase shift. The driver of the surface-mount DT-PMSM includes a first converter and a second converter. The two converters are controlled by the same control system. The control system includes a PI controller, a PIR controller, a first coordinate transformation module, and an SVPWM module connected in sequence, as well as a phase current collector, a second coordinate transformation module, and a z1-z2 current loop fault-tolerant controller.

[0166] The phase current collector is used to collect the phase current of each phase motor winding;

[0167] The second coordinate transformation module is used to map the variables in the natural coordinate system to the fundamental wave dq plane and the harmonic z1-z2 plane. It inputs the current signals of each phase collected by the phase current collector; it outputs the actual current signals of the dq axes to the PIR controller, and it outputs the actual current signals of the z1-z2 axes;

[0168] The PI controller inputs the rotor reference angular velocity signal and the rotor actual angular velocity signal respectively; it outputs the q-axis reference current signal;

[0169] The PIR controller inputs a q-axis reference current signal, a d-axis reference current signal, and a dq-axis actual current signal; it outputs a dq-axis reference voltage signal to the first coordinate transformation module;

[0170] The first coordinate transformation module is used to transform the variables in the dq coordinate system into variables in the α-β coordinate system; it outputs the α-β reference voltage signal to the SVPWM module;

[0171] The z1-z2 current loop fault-tolerant controller includes a notch filter, a harmonic QPR controller and an adaptive resonant controller; the adaptive resonant controller is a fundamental frequency QPR controller connected in series with a switching switch, and the switching switch is controlled by the z1-z2 axis reference current signal; the notch filter is used to filter out the fundamental frequency current component mapped from the α-β plane to the z1-z2 plane; the harmonic QPR controller includes parallel third, fifth and / or seventh harmonic QPR controllers, the notch filter is connected in series with the harmonic QPR controller and then connected in parallel with the adaptive resonant controller; the z1-z2 current loop fault-tolerant controller, which inputs the z1-z2 axis actual current signal from the second coordinate transformation module, and outputs the z1-z2 axis reference voltage signal to the SVPWM module;

[0172] The SVPWM module inputs the α-β reference voltage signal and the z1-z2 axis reference voltage signal, and outputs PWM signals to the first converter and the second converter respectively.

[0173] It may also include a rotor position electrical angle collector and a differentiator; the rotor position electrical angle collector is used to collect the electrical angle of the motor rotor; the differentiator is used to convert the rotor position electrical angle signal into a rotor angular velocity signal; it inputs the electrical angle signal collected by the rotor position electrical angle collector and outputs the actual rotor angular velocity signal to the PI controller.

[0174] Preferably, the system may also include a current vector trajectory module and an MT optimization controller; the current vector trajectory module can be used to track the harmonic plane current trajectory, which can input the actual current signal of the z1-z2 axis from the second coordinate transformation module, and can output the current vector angle of the z1-z2 plane; the MT optimization controller can optimize the control parameters through the optimization tool, and its optimization goal can be to minimize the maximum single-phase copper loss, which can respectively input the q-axis reference current signal from the PI controller and the current vector angle of the z1-z2 plane from the current vector trajectory module, and can output the z1-z2 axis reference current signal to the z1-z2 current loop fault-tolerant controller.

[0175] Preferably, one of the two sets of windings may include A, B, and C phase windings, and the other set of windings may include D, E, and F phase windings; and x may be any one of the A, B, C, D, E, and F phase windings.

[0176] The optimization function of the MT optimization controller can be as follows:

[0177]

[0178] p max=max{p A ,p B ,p C ,p D ,p E ,p F};

[0179] The copper loss calculation formula of the x-phase winding is as follows:

[0180]

[0181] The coefficients in the formula are

[0182]

[0183] Where:

[0184] is the spatial phase angle of the x-phase winding; x∈{A, B, C, D, E, F}; when x is A, B, C, D, E, F phase respectively,

[0185] k d is the amplitude coefficient of the d-axis current;

[0186] p x is the copper loss of x-phase winding;

[0187] p A is the copper loss of phase A winding;

[0188] p B is the copper loss of phase B winding;

[0189] p C is the copper loss of phase C winding;

[0190] p D is the copper loss of D-phase winding;

[0191] p E is the copper loss of E-phase winding;

[0192] p F is the copper loss of F phase winding;

[0193] R s is the phase winding resistance of the motor stator;

[0194] i x is the x-phase current;

[0195] i d is the d-axis current in the dq plane;

[0196] i q is the q-axis current in the dq plane;

[0197] Idc is the DC current coefficient in the phase winding copper loss;

[0198] α d is the phase deviation angle of the d-axis current;

[0199] θ e is the electrical angle of the motor rotor position;

[0200] I1 is the cosine coefficient of phase winding copper loss;

[0201] I2 is the sinusoidal coefficient of phase winding copper loss;

[0202] k1, k2, k3, and k4 are coefficients to be optimized;

[0203] i α is the α-axis current in the α-β plane;

[0204] i β is the β-axis current in the α-β plane;

[0205] i z1 is the z1-axis current in the z1-z2 plane;

[0206] i z2 is the z2-axis current in the z1-z2 plane;

[0207] p max is the maximum single-phase copper loss;

[0208] min p max Indicates minimizing the maximum single-phase copper loss of the motor;

[0209] max{} represents the maximum value function.

[0210] The optimization function of the MT optimization controller can be solved by the Matlab optimization toolbox to obtain the optimal solution of the following coefficients: α d 、k d , k1, k2, k3, k4.

[0211] The present invention also provides a single-phase open-circuit fault-tolerant control method for a dual three-phase permanent magnet synchronous motor. The dual three-phase permanent magnet synchronous motor is a surface-mounted DT-PMSM with two sets of winding neutral points isolated and a phase shift of 30 degrees. The driver of the surface-mounted DT-PMSM includes a first converter and a second converter. The two converters are controlled by the same control system. The control system is provided with a PI controller, a PIR controller, a first coordinate transformation module, and an SVPWM module connected in sequence, as well as a phase current collector, a second coordinate transformation module, and a z1-z2 current loop fault-tolerant controller.

[0212] The phase current collector is used to collect the phase current of each phase motor winding;

[0213] The second coordinate transformation module is used to map the variables in the natural coordinate system to the fundamental wave dq plane and the harmonic z1-z2 plane, so that it inputs the current signals of each phase collected by the phase current collector; it outputs the actual current signals of the dq axes to the PIR controller, which outputs the actual current signals of the z1-z2 axes;

[0214] The PI controller is input with the rotor reference angular velocity signal and the rotor actual angular velocity signal respectively; and is output with the q-axis reference current signal;

[0215] The PIR controller is configured to input a q-axis reference current signal, a d-axis reference current signal, and a dq-axis actual current signal; and to output a dq-axis reference voltage signal to the first coordinate transformation module;

[0216] The first coordinate transformation module is used to transform the variables in the dq coordinate system into variables in the α-β coordinate system, so that it outputs the α-β reference voltage signal to the SVPWM module;

[0217] The z1-z2 current loop fault-tolerant controller is provided with a notch filter, a harmonic QPR controller and an adaptive resonant controller; the adaptive resonant controller is a fundamental frequency QPR controller connected in series with a switching switch, and the switching switch is controlled by the z1-z2 axis reference current signal; the notch filter is used to filter out the fundamental frequency current component mapped from the α-β plane to the z1-z2 plane; the harmonic QPR controller is provided with parallel third, fifth and / or seventh harmonic QPR controllers, so that the notch filter and the harmonic QPR controller are connected in series and then in parallel with the adaptive resonant controller; the z1-z2 current loop fault-tolerant controller is input with the z1-z2 axis actual current signal from the second coordinate transformation module, so that it outputs the z1-z2 axis reference voltage signal to the SVPWM module;

[0218] The SVPWM module is input with the α-β reference voltage signal and the z1-z2 axis reference voltage signal, so that the module outputs the PWM signal to the first converter and the second converter respectively;

[0219] During the normal operation and natural fault-tolerant operation stages of the motor, the z1-z2 axis reference current is zero, the switching switch is disconnected, and the adaptive resonant controller is in the fundamental frequency current open-loop control mode; during the current optimization operation stage, when the z1-z2 axis reference current is not zero, the switching switch is turned on, and the adaptive resonant controller is in the fundamental frequency current closed-loop control mode.

[0220] Preferably, a limiting module can also be set between the PI controller and the PIR controller. The limiting module performs limiting processing on the q-axis reference current signal output by the PI controller, and outputs the q-axis reference current signal after limiting processing to the PIR controller; after a fault, the limiting value of the limiting module is changed according to the z1-z2 current reference value to ensure that the motor runs safely without exceeding the limit.

[0221] Preferably, the method may further include a current vector trajectory module and an MT optimization controller; the current vector trajectory module may be used to track the harmonic plane current trajectory, may input the actual current signal of the z1-z2 axis from the second coordinate transformation module, and may output the current vector angle of the z1-z2 plane; the MT optimization controller may optimize the control parameters through an optimization tool, may set its optimization target as minimizing the maximum single-phase copper loss, may input the q-axis reference current signal from the PI controller and the current vector angle of the z1-z2 plane from the current vector trajectory module respectively, and may output the z1-z2 axis reference current signal to the z1-z2 current loop fault-tolerant controller.

[0222] At the moment of fault, the adaptive resonant controller is disabled, and the fundamental frequency filtering effect of the notch filter allows the fundamental frequency component related to the torque to flow through the harmonic plane without interference; after the MT optimization controller outputs a new z1-z2 axis reference current based on the current vector angle of the z1-z2 plane, the adaptive resonant controller is reactivated to follow the fundamental frequency current reference value of the harmonic plane, switching from the natural fault-tolerant operation stage to the current control optimization operation stage.

[0223] Preferably, one of the two sets of windings includes A, B, and C phase windings, and the other set of windings includes D, E, and F phase windings; let x be any one of the A, B, C, D, E, or F phase windings;

[0224] The optimization function of the MT optimization controller is as follows:

[0225]

[0226] p max =max{p A ,p B ,p C ,p D ,p E ,p F};

[0227] The copper loss calculation formula of the x-phase winding is as follows:

[0228]

[0229] The coefficients in the formula are

[0230]

[0231] Where:

[0232] is the spatial phase angle of the x-phase winding; x∈{A, B, C, D, E, F}; when x is A, B, C, D, E, F phase respectively,

[0233] k d is the amplitude coefficient of the d-axis current;

[0234] p x is the copper loss of x-phase winding;

[0235] p A is the copper loss of phase A winding;

[0236] p B is the copper loss of phase B winding;

[0237] p C is the copper loss of phase C winding;

[0238] p D is the copper loss of D-phase winding;

[0239] p E is the copper loss of E-phase winding;

[0240] p F is the copper loss of F phase winding;

[0241] R s is the phase winding resistance of the motor stator;

[0242] i x is the x-phase current;

[0243] i d is the d-axis current in the dq plane;

[0244] i q is the q-axis current in the dq plane;

[0245] I dc is the DC current coefficient in the phase winding copper loss;

[0246] α d is the phase deviation angle of the d-axis current;

[0247] θ e is the electrical angle of the motor rotor position;

[0248] I1 is the cosine coefficient of phase winding copper loss;

[0249] I2 is the sinusoidal coefficient of phase winding copper loss;

[0250] k1, k2, k3, and k4 are coefficients to be optimized;

[0251] i α is the α-axis current in the α-β plane;

[0252] i β is the β-axis current in the α-β plane;

[0253] i z1 is the z1-axis current in the z1-z2 plane;

[0254] i z2 is the z2-axis current in the z1-z2 plane;

[0255] p max is the maximum single-phase copper loss;

[0256] min p max Indicates minimizing the maximum single-phase copper loss of the motor;

[0257] The optimization function of the MT optimization controller is solved by the MATLAB optimization toolbox, and the optimal solution of the following coefficients is obtained: α d 、k d , k1, k2, k3, k4.

[0258] Preferably, after a phase failure occurs, the MT optimization controller can determine the fault phase based on the current vector trajectory direction on the harmonic plane by the following method:

[0259] When the current in the z1-z2 plane moves along the 0 / 180° direction, it is determined that phase A is faulty;

[0260] When the current in the z1-z2 plane moves along the -120° / 60° direction, it is determined that the B phase is faulty;

[0261] When the current in the z1-z2 plane moves along the 120° / -60° direction, it is determined that the C phase is faulty;

[0262] When the current in the z1-z2 plane moves along the 150° / -30° direction, it is determined that the D phase is faulty;

[0263] When the current in the z1-z2 plane moves along the 30° / -150° direction, it is determined that the E phase is faulty;

[0264] When the current in the z1-z2 plane moves along the -90° / 90° direction, it is determined that the F phase is faulty;

[0265] Based on the fault phase information, the MT optimization controller can obtain the optimal coefficient in the maximum torque current mode and calculate the output z1-z2 axis reference current signal.

[0266] The structure, workflow and working principle of the present invention are further described below with reference to a preferred embodiment of the present invention:

[0267] The present invention mainly studies a surface-mounted DT-PMSM with two sets of winding neutral points isolated and a phase shift of 30°. Figure 1 The topology of the DT-PMSM driver is shown in the figure. dc is the DC bus voltage.

[0268] The vector space decoupling transformation matrix T can be used VSD , mapping the variables in the natural coordinate system to the fundamental wave α-β plane, harmonic z1-z2 plane and zero sequence o1-o2 plane. Vector space decoupling transformation matrix T VSD It can be as follows:

[0269]

[0270] Since the neutral points of the two windings are isolated, the zero-sequence plane current i o1 and i o2 It cannot flow and is therefore omitted in subsequent analysis. r , the α-β plane variables are mapped to the dq plane of the synchronously rotating coordinate system. r It can be written as:

[0271]

[0272] Where: θ e is the electrical angle of the rotor position.

[0273] The voltage equations in the dq plane and the z1-z2 plane are:

[0274]

[0275] u s =u d +ju q ,u z =u z1 +ju z2 ;i s =i d +ji q ,i z =i z1 +ji z2 ;L s =3L m +L z , e z =e z1 +je z2 ;

[0276] In formula 1 to formula 3:

[0277] T VSD is the vector space decoupling transformation matrix;

[0278] u s is the voltage vector in the dq plane;

[0279] i s is the current vector in the dq plane;

[0280] ud is the d-axis voltage;

[0281] u q is the q-axis voltage;

[0282] i d is the d-axis current;

[0283] i q is the q-axis current;

[0284] R s is the stator resistance;

[0285] ω e is the rotor electrical angular frequency;

[0286] L s is the inductance of the motor phase winding;

[0287] L m Feel for the Lord;

[0288] L z is leakage inductance;

[0289] ψ f is the permanent magnet flux amplitude;

[0290] u z is the voltage vector in the z1-z2 plane;

[0291] i z is the current vector in the z1-z2 plane;

[0292] u z1 is the z1 axis voltage;

[0293] u z2 is the z2 axis voltage;

[0294] i z1 is the z1 axis current;

[0295] i z2 is the z2 axis current;

[0296] e z is the back electromotive force induced by the permanent magnet flux harmonic in the z1-z2 plane;

[0297] e z1 is the back electromotive force of z1 axis;

[0298] e z2 is the z2-axis back electromotive force;

[0299] L z is the motor inductance in the z1-z2 plane.

[0300] When the motor is running healthily, the electromagnetic torque T e for:

[0301] T e =3n p ψ f i q (4)

[0302] In formula 4: n p is the number of motor pole pairs. e is the electromagnetic torque.

[0303] The discretized current equation in the z1-z2 plane is expressed as:

[0304] i z (k+1)=ζ z i z (k)+κ z u z (k)-k z e z (k) (5)

[0305] In formula 5, each parameter is expressed as:

[0306]

[0307] The reference voltage of the z1-z2 plane to i z The transfer function of (z) is:

[0308]

[0309] In Formula 5 and Formula 6:

[0310] i z (k+1) is the current vector in the z1-z2 plane at time k+1;

[0311] ζ z is an intermediate variable;

[0312] i z (k) is the current vector in the z1-z2 plane at time k;

[0313] κ z is an intermediate variable;

[0314] u z (k) is the voltage vector on the z1-z2 plane at time k;

[0315] e z (k) is the back EMF vector in the z1-z2 plane at time k;

[0316] T s To control the cycle;

[0317] τ is a variable representing time and is used for mathematical integration operations;

[0318] G pz (z) is the reference voltage on the z1-z2 plane to i z (z) transfer function;

[0319] i z (z) is the discretized current vector in the z1-z2 plane;

[0320] is the discretized reference voltage vector in the z1-z2 plane;

[0321] z is a variable representing the discrete z transform.

[0322] ML current mode under natural fault-tolerant control:

[0323] When an open-circuit fault occurs in a motor winding, no current can flow through the damaged phase winding. The current in the x-phase where the fault occurs is limited to zero.

[0324]

[0325] Take the open circuit fault of phase A as an example. Substituting into equation (7), we get the current constraint in the system:

[0326] i z1 =-i α (8)

[0327] In formula 7 and formula 8:

[0328] i x is the current of phase x; subscript x∈{A,B,C,D,E,F}; the corresponding phase angle

[0329] The α-β plane and the z1-z2 plane are no longer independent of each other. The z1-z2 controller attempts to control the z1-z2 current to zero, while the dq controller aims to track flux and torque to maintain pre-fault torque. This creates a conflict between the control objectives of the dq and z1-z2 current controllers. For traditional fault-tolerant strategies, this control conflict during fault detection is unavoidable, impacting torque and current control performance at the moment of fault.

[0330] The essence of the natural fault-tolerant control strategy is to not control the fundamental frequency component of the current coupled between the z1-z2 plane and the α-β plane, which essentially overcomes the control conflict of the traditional fault-tolerant method during fault detection. d = 0 control architecture, the VSD currents after natural fault-tolerant control are:

[0331]

[0332] In formula 9:

[0333] i o1 is the o1 axis current;

[0334] i o2 is the o2 axis current.

[0335] However, if we pass T through equation (9), VSD The inverse transformation of the phase current after natural fault-tolerant control is (10), and the corresponding fault-tolerant phase current waveform is as follows: Figure 2 Therefore, under the strong control force of the speed loop and dq current loop PI controllers, the natural fault-tolerant control is consistent with the current target of the ML mode.

[0336]

[0337] In formula 10, θ T is the initial phase angle; I s is the phase current amplitude during normal operation.

[0338] i B is the phase current of phase B;

[0339] i C is the phase current of phase C;

[0340] i D is the phase current of phase D;

[0341] i E is the phase current of phase E;

[0342] i F is the phase current of phase F.

[0343] Since the current target under natural fault-tolerant control is ML, it is impossible to achieve other current targets, the torque output capacity is only 55.4%, and the torque output range cannot be further expanded.

[0344] The fault-tolerant control strategy proposed by the present invention:

[0345] To address the inherent problems of natural fault-tolerant control strategies, the present invention proposes a fault-tolerant control strategy based on an adaptive resonant controller. The fault-tolerant process after a single-phase open-circuit fault is divided into a natural fault-tolerant operation phase and a current optimization operation phase. At the moment of the fault, the adaptive resonant controller in the z1-z2 current loop fault-tolerant controller is disabled. Combined with the fundamental frequency filtering effect of the notch filter, the fundamental frequency component related to torque generation will flow through the harmonic plane without interference, allowing a smooth transition to the natural fault-tolerant operation phase after the fault. After obtaining a new MT current reference based on the harmonic plane current trajectory, the adaptive resonant controller is reactivated to follow the fundamental frequency current reference value of the harmonic plane, smoothly switching from the natural fault-tolerant operation phase to the current optimization operation phase, during which the motor operates in MT current mode.

[0346] The MT current mode refers to a fault-tolerant operating current mode with maximum torque output capability after a motor fault.

[0347] The present invention proposes a single-phase open-circuit fault-tolerant control system for a dual three-phase permanent magnet synchronous motor. Figure 3 shown.

[0348] After a single-phase open circuit fault, there is a coupling effect between the current components in the α-β plane and the z1-z2 plane. A notch filter (NF) is used to filter out the fundamental frequency current component mapped from the α-β plane to the z1-z2 plane, avoiding the conflict between the current control targets of the α-β plane and the z1-z2 plane. This allows a smooth transition to the natural fault-tolerant operation stage after a fault occurs without the need for a fault detection link. Using the Tustin approximation method, the transfer function of the fundamental frequency notch filter in the discrete domain is:

[0349]

[0350] Where,

[0351]

[0352] Formula 11 and coefficient calculation formula:

[0353] G NF (z) is the transfer function of the fundamental frequency notch filter in the discrete domain;

[0354] p0, p1, p2, q0, q1, q2 are all intermediate variables;

[0355] ω e is the rotor electrical angular velocity of the motor;

[0356] k b k is the adjustment parameter of the notch filter bandwidth. b Used to adjust the bandwidth of the notch filter.

[0357] The quasi-proportional resonance (QPR) controller is set in the z1-z2 plane to suppress the e z The third, fifth, and seventh current harmonics caused by non-ideal factors such as hysteresis and oscillation can be controlled, and the impact on the fundamental plane current can be avoided. The tertiary current QPR controller also has the ability to track and control the third component of the z1-z2 current reference value in the new MT fault-tolerant mode. Using the Tustin approximation method, the transfer function of the nth order QPR controller in the discrete domain is:

[0358]

[0359] Where,

[0360]

[0361] Formula 12 and coefficient calculation formula:

[0362] k p and k q are proportional coefficient and resonant gain respectively; (nω e ) is the resonant frequency; ω c is the controller bandwidth.

[0363] G QPR-n (z) is the transfer function of the n-th order QPR controller in the discrete domain;

[0364] b n 、a 0-n 、a 1-n 、a 2-n All are intermediate variables;

[0365] k q-n is the resonant gain of the n-th QPR controller;

[0366] ω c is the controller bandwidth

[0367] n represents the current control frequency of the QPR controller;

[0368] k p is the resonant gain.

[0369] Figure 4 The structure of the current loop fault-tolerant controller is given. In the figure, δ z (z) is the periodic disturbance caused by factors such as back EMF harmonics.

[0370] Based on the notch filter series three, five, and seven-order QPR controllers, an adaptive resonant controller with the z1-z2 axis current reference as input is connected in parallel. The adaptive resonant controller is a baseband QPR controller with an input switching switch. The switching signal F TIt is set according to the z1-z2 axis current reference, as shown in equation (13).

[0371]

[0372] In formula 13:

[0373] F T is the switching signal;

[0374] is the reference current of z1 axis;

[0375] is the z2-axis reference current.

[0376] The "adaptive" technical solution of the adaptive resonant controller provided by the present invention is to change the state of the baseband QPR controller according to the magnitude of the z1-z2 axis current reference. According to formula (13), the adaptive resonant controller can be divided into two states: the baseband QPR controller open-loop control and the closed-loop control, corresponding to the natural fault-tolerant operation stage and the current optimization operation stage, respectively. 1) When the motor is operating normally, the z1-z2 axis current reference is zero. At this time, the input of the baseband QPR controller is cut off, which is equivalent to the baseband current open-loop control. At the same time, considering the baseband filtering effect of the notch filter, after the fault occurs, the control target conflict between the α-β plane and the z1-z2 plane is avoided at the moment of the fault, and the motor smoothly transitions to the natural fault-tolerant operation stage; 2) The MT current reference is obtained based on the harmonic plane current vector trajectory under the natural fault-tolerant operation state. That is, when the z1-z2 axis current reference is not zero, the input of the baseband QPR control is put in, and the baseband QPR closed-loop control is used to restore the tracking of the z1-z2 axis baseband current reference. The baseband and cubic QPR controllers jointly achieve the predetermined MT current optimization target.

[0377] based on Figure 4 And formula (13) is drawn as Figure 5 、 Figure 6 The Bode diagram of the z1-z2 current loop open-loop transfer function is shown, and the corresponding fundamental frequency ω e =20πrad / s, controller control period T s =0.1ms. When the baseband QPR controller is in open loop control, Figure 5 The gain of the open-loop transfer function in the fundamental frequency is very small, which effectively filters out the fundamental frequency current coupled with the α-β plane. When the fundamental frequency QPR controller is closed-loop controlled, Figure 6 The open-loop transfer function in has a large gain at the fundamental frequency, which can effectively track the fundamental current reference and cooperate with the triple QPR to achieve the new MT current target. Figure 5 、 Figure 6 The open-loop transfer function has large gains at three, five, and seven times the fundamental frequency, which can effectively suppress the corresponding frequency disturbances.

[0378] The steps to achieve the MT current mode optimization goal are as follows:

[0379] According to formula (4), the electromagnetic torque of the surface-mounted DT-PMSM is only related to the q-axis current i q The d-axis current i d Typically controlled to zero, the present invention injects a second harmonic component into the d-axis current after a phase failure occurs. This is then mapped to the first and third harmonic components of the phase current through coordinate transformation. This third harmonic component in the phase current removes the inherent constraints of the sinusoidal distribution of the phase current, granting the DT-PMSM greater control freedom and thus further improving post-fault torque output capability.

[0380] The new MT fault-tolerant current mode, which injects second-harmonic current on the d-axis, only requires the injection of low-order harmonics and a simple controller design. This avoids the problems of complex controller design and difficult control accuracy caused by multiple harmonics, high-frequency current injection, or online real-time calculation in the conventional MT fault-tolerant mode. It also has similar copper loss and torque output capabilities to the conventional MT fault-tolerant mode.

[0381] The expression of the d-axis current after injecting the second harmonic component is:

[0382] i d =i q k d sin(2θ e +α d ) (14)

[0383] Where:

[0384] k d is the amplitude coefficient of the d-axis current;

[0385] α d is the phase angle of the d-axis current.

[0386] The q-axis current comes from the speed loop output and remains unchanged in order to maintain the electromagnetic torque. r The inverse transformation of α-β axis current is:

[0387]

[0388] According to the post-fault phase current constraint shown in equation (7), the z1-z2 axis current can be expressed as

[0389]

[0390] Transform Equation (15) and Equation (16) through T VSD The inverse transformation of is used to obtain the expression of each phase current. Combined with formula (7), the copper loss of each phase winding is calculated as:

[0391]

[0392] The coefficients in the formula are

[0393]

[0394] Wherein, the subscript x can be any phase among A, B, C, D, E or F.

[0395] Therefore, the maximum single-phase copper loss in the six-phase winding can be expressed as

[0396] p max =max{p A ,p B ,p C ,p D ,p E ,p F} (19)

[0397] In Equations 15 to 19:

[0398] Where:

[0399] is the spatial phase angle of the x-phase winding; x∈{A, B, C, D, E, F}; when x is A, B, C, D, E, F phase respectively,

[0400] k d is the amplitude coefficient of the d-axis current;

[0401] p x is the copper loss of x-phase winding;

[0402] p A is the copper loss of phase A winding;

[0403] p B is the copper loss of phase B winding;

[0404] p C is the copper loss of phase C winding;

[0405] p D is the copper loss of D-phase winding;

[0406] p E is the copper loss of E-phase winding;

[0407] p F is the copper loss of F phase winding;

[0408] R s is the phase winding resistance of the motor stator;

[0409] i x is the x-phase current;

[0410] i d is the d-axis current in the dq plane;

[0411] i q is the q-axis current in the dq plane;

[0412] I dc is the DC current coefficient in the phase winding copper loss;

[0413] α d is the phase deviation angle of the d-axis current;

[0414] θ e is the electrical angle of the motor rotor position;

[0415] I1 is the cosine coefficient of phase winding copper loss;

[0416] I2 is the sinusoidal coefficient of phase winding copper loss;

[0417] k1, k2, k3, and k4 are coefficients to be optimized;

[0418] i α is the α-axis current in the α-β plane;

[0419] i β is the β-axis current in the α-β plane;

[0420] i z1 is the z1-axis current in the z1-z2 plane;

[0421] i z2 is the z2-axis current in the z1-z2 plane;

[0422] p max is the maximum single-phase copper loss.

[0423] During normal operation, the six-phase currents of a DT-PMSM all exhibit the same sinusoidal waveform, and rated torque is achieved when they simultaneously reach their rated RMS values. However, as previously analyzed, the motor's natural fault-tolerance strategy operates in the ML current mode after a single-phase winding failure. The RMS values ​​of the six-phase currents vary, and when the output torque reaches the rated value, localized overheating may occur, with some winding phase currents exceeding their rated RMS values. To ensure safe motor operation, the maximum single-phase copper loss must not exceed the rated single-phase copper loss.

[0424] The copper loss of a single-phase winding is directly related to the effective value of the current in each phase. In other words, the torque output capacity η T It can be expressed as the effective phase current I during normal operation N The maximum effective value of the six-phase current during fault-tolerant operation I max Therefore, the torque output capacity η T for:

[0425]

[0426] Where: η T is the torque output capacity.

[0427] From formula (20), we can see that by minimizing p max To achieve torque output capacity η T The solution of the DT-PMSM maximum torque MT current mode can be transformed into an objective function optimization problem with multiple variables under constraints. The optimization function is expressed as follows:

[0428]

[0429] min p max Indicates minimizing the maximum single-phase copper loss of the motor.

[0430] The constraint in the optimization process is the restriction condition that the fault phase current is zero. The optimization function of the MT optimization controller is solved by the MATLAB optimization toolbox to obtain the optimal solution of the following coefficients: α d 、k d , k1, k2, k3, k4.

[0431] The optimization process is not affected by changes in motor parameters and operating conditions and can be optimized offline, thus reducing the computational burden of the controller. Table 1 lists the optimal coefficients of the MT current mode when faults occur in different phases. Figure 6 The phase current waveform of MT current mode when phase A is faulty is given.

[0432] Table 1 Optimal coefficients of MT current mode when faults occur in different phases

[0433]

[0434] To demonstrate the advantages of the proposed fault-tolerance scheme over traditional natural fault-tolerance methods, a comparison of torque output capacity is presented in Table 2. For ease of understanding, all performance data in Table 2 is compared in units (pu) under normal operating conditions. It can be seen that after a single-phase open-circuit fault in the DT-PMSM occurs, the proposed fault-tolerance scheme achieves a maximum torque output capacity of 71.4% by injecting only the optimal second harmonic current into the d-axis without the need for additional complex controller design. Compared to traditional natural fault-tolerance control methods, the proposed fault-tolerance scheme significantly improves torque output capacity by 28.6%.

[0435] Table 2 Performance comparison between traditional natural fault tolerance method and proposed fault tolerance method

[0436]

[0437] Furthermore, a new MT fault-tolerant strategy is implemented based on the current vector trajectory of the harmonic plane when the remaining five phases are faulted. As shown in Equation (9), the z1-z2 plane current moves in the 0 / 180° direction when phase A fails. In order to obtain the z1-z2 plane current vector trajectory after an open-circuit fault in any phase during the natural fault-tolerant operation phase, further analysis is carried out using phase B as an example. According to Equation (7), the current constraint when phase B fails is:

[0438]

[0439] In formula (9), i α and i β Substituting into the above formula and further simplifying it, we get:

[0440]

[0441] From formula (23), we can see that i z1 and i z2 With the same frequency and phase, i can be set separately z1 =m1i q sin(θ e -2π / 3), i z2 =m2i q sin(θ e -2π / 3). At the same time, from the fault analysis of phase A, it can be known that the current vector amplitude in the z1-z2 plane is i q , the following relationship holds true for the simultaneous equation (23):

[0442]

[0443] Where: m1 and m2 are intermediate variables.

[0444] Therefore, the current vector in the z1-z2 plane is along the -120° / 60° direction. Similar to the analysis process of the phase B fault, the current vector trajectory in the z1-z2 plane after the fault of different phases has its own corresponding angle. The current vector trajectory in the z1-z2 plane after the fault is listed in Table 3. v,off (v=A, B, C, D, E, F) is the current vector angle in the z1-z2 plane after the fault; Φ v is the angle corresponding to the fault phase v in the α-β plane.

[0445] Table 3 Current vector trajectory in the z1-z2 plane

[0446]

[0447] In the natural fault-tolerant operation phase, through θ v,offAfter the fault phase is known, the optimal coefficient of the new MT current mode is further obtained according to Table 1. The d-z1-z2 axis current reference is switched from zero to the MT current reference value, and the fundamental frequency PR controller of the z1-z2 current loop is enabled synchronously.

[0448] In addition, after a fault occurs, the speed loop output limit value is changed according to the z1-z2 current reference value to ensure that the motor runs safely without exceeding the limit.

[0449]

[0450] Where,

[0451]

[0452] In Equation 25 and Equation 26:

[0453] is the reference value of the q-axis current after the limiting link.

[0454] I m is the rated phase current amplitude.

[0455] I L is an intermediate variable;

[0456] It is the q-axis current reference value output by the speed loop PI controller.

[0457] The above-mentioned PI controller, PIR controller, first coordinate transformation module, SVPWM module, phase current collector, second coordinate transformation module, z1-z2 current loop fault-tolerant controller, notch filter, harmonic QPR controller, adaptive resonant controller, switching switch, fundamental frequency QPR controller, current vector trajectory module and MT optimization controller can all adopt applicable functional modules in the existing technology, or adopt functional modules and software in the existing technology and adopt conventional technical means to construct them.

[0458] The embodiments described above are only used to illustrate the technical ideas and features of the present invention. Their purpose is to enable those skilled in the art to understand the contents of the present invention and implement them accordingly. The scope of the patent of the present invention cannot be limited by these embodiments alone. That is, any equivalent changes or modifications made to the spirit disclosed by the present invention still fall within the scope of the patent of the present invention.

Claims

1. A single-phase open-circuit fault-tolerant control system for a dual three-phase permanent magnet synchronous motor, wherein the dual three-phase permanent magnet synchronous motor is a surface-mount DT-PMSM with two sets of windings having neutral points isolated and a phase shift of 30°; the driver of the surface-mount DT-PMSM includes a first converter and a second converter; the two converters are controlled by the same control system; and the following features are provided: The control system includes a PI controller, a PIR controller, a first coordinate transformation module and an SVPWM module, which are connected in sequence, as well as a phase current collector, a second coordinate transformation module and a z1-z2 current loop fault-tolerant controller; The phase current collector is used to collect the phase current of each phase motor winding; The second coordinate transformation module is used to map the variables in the natural coordinate system to the fundamental wave dq plane and the harmonic z1-z2 plane. It inputs the current signals of each phase collected by the phase current collector; it outputs the actual current signals of the dq axes to the PIR controller, and it outputs the actual current signals of the z1-z2 axes; The PI controller inputs the rotor reference angular velocity signal and the rotor actual angular velocity signal respectively; It outputs the q-axis reference current signal; The PIR controller inputs the q-axis reference current signal, the d-axis reference current signal, and the dq-axis actual current signal; It outputs dq axis reference voltage signals to the first coordinate transformation module; The first coordinate transformation module is used to transform the variables in the dq coordinate system into variables in the α-β coordinate system; it outputs the α-β reference voltage signal to the SVPWM module; The z1-z2 current loop fault-tolerant controller includes a notch filter, a harmonic QPR controller, and an adaptive resonant controller; The adaptive resonant controller is a baseband QPR controller connected in series with a switch, and the on and off of the switch is controlled by the z1-z2 axis reference current signal; The notch filter is used to filter out the fundamental frequency current component mapped from the α-β plane to the z1-z2 plane; the harmonic QPR controller includes parallel third, fifth and / or seventh harmonic QPR controllers, the notch filter is connected in series with the harmonic QPR controller and then connected in parallel with the adaptive resonant controller; the z1-z2 current loop fault-tolerant controller inputs the z1-z2 axis actual current signal from the second coordinate transformation module and outputs the z1-z2 axis reference voltage signal to the SVPWM module; The SVPWM module inputs the α-β reference voltage signal and the z1-z2 axis reference voltage signal, and outputs PWM signals to the first converter and the second converter respectively.

2. The single-phase open-circuit fault-tolerant control system of a dual three-phase permanent magnet synchronous motor according to claim 1, characterized in that: The system also includes a current vector trajectory module and an MT optimization controller; the current vector trajectory module is used to track the harmonic plane current trajectory, and its input is the actual current signal of the z1-z2 axis from the second coordinate transformation module, and it outputs the current vector angle of the z1-z2 plane; the MT optimization controller optimizes the control coefficient through an optimization tool, and its optimization goal is to minimize the maximum single-phase copper loss. It inputs the q-axis reference current signal from the PI controller and the current vector angle of the z1-z2 plane from the current vector trajectory module, and it outputs the z1-z2 axis reference current signal to the z1-z2 current loop fault-tolerant controller.

3. The single-phase open-circuit fault-tolerant control system of a dual three-phase permanent magnet synchronous motor according to claim 2, characterized in that: Assume that one of the two windings includes phases A, B, and C, and the other includes phases D, E, and F. Assume that x is any phase winding among phases A, B, C, D, E, or F. The optimization function of the MT optimization controller is as follows: p max =max{p A ,p B ,p C ,p D ,p E ,p F }; The copper loss calculation formula of the x-phase winding is as follows: Where: Where: is the spatial phase angle of the x-phase winding; x∈{A, B, C, D, E, F}; when x is A, B, C, D, E, F phase respectively, k d is the amplitude coefficient of the d-axis current; p x is the copper loss of x-phase winding; p A is the copper loss of phase A winding; p B is the copper loss of phase B winding; p C is the copper loss of phase C winding; p D is the copper loss of D-phase winding; p E is the copper loss of E-phase winding; p F is the copper loss of F phase winding; R s is the phase winding resistance of the motor stator; i x is the x-phase current; i d is the d-axis current in the dq plane; i q is the q-axis current in the dq plane; I dc is the DC current coefficient in the phase winding copper loss; α d is the phase deviation angle of the d-axis current; θ e is the electrical angle of the motor rotor position; I1 is the cosine coefficient of phase winding copper loss; I2 is the sinusoidal coefficient of phase winding copper loss; k1, k2, k3, and k4 are coefficients to be optimized; i α is the α-axis current in the α-β plane; i β is the β-axis current in the α-β plane; i z1 is the z1-axis current in the z1-z2 plane; i z2 is the z2-axis current in the z1-z2 plane; p max is the maximum single-phase copper loss; min p max Indicates minimizing the maximum single-phase copper loss of the motor; The optimization function of the MT optimization controller is solved by the MATLAB optimization toolbox, and the optimal solution of the following coefficients is obtained: α d 、k d , k1, k2, k3, k4.

4. The single-phase open-circuit fault-tolerant control system of a dual three-phase permanent magnet synchronous motor according to claim 1, characterized in that: A limiting module is also included between the PI controller and the PIR controller. The limiting module performs limiting processing on the q-axis reference current signal output by the PI controller, and outputs the q-axis reference current signal after the limiting processing to the PIR controller.

5. A single-phase open-circuit fault-tolerant control method for a dual three-phase permanent magnet synchronous motor, wherein the dual three-phase permanent magnet synchronous motor is a surface-mount DT-PMSM with two sets of windings having neutral points isolated and a phase shift of 30°; the driver of the surface-mount DT-PMSM includes a first converter and a second converter; the two converters are controlled by the same control system; characterized in that: The control system is provided with a PI controller, a PIR controller, a first coordinate transformation module and an SVPWM module connected in sequence, as well as a phase current collector, a second coordinate transformation module and a z1-z2 current loop fault-tolerant controller; The phase current collector is used to collect the phase current of each phase motor winding; The second coordinate transformation module is used to map the variables in the natural coordinate system to the fundamental wave dq plane and the harmonic z1-z2 plane, so that it inputs the current signals of each phase collected by the phase current collector; it outputs the actual current signals of the dq axes to the PIR controller, which outputs the actual current signals of the z1-z2 axes; The PI controller is input with the rotor reference angular velocity signal and the rotor actual angular velocity signal respectively; and is output with the q-axis reference current signal; The PIR controller is configured to input a q-axis reference current signal, a d-axis reference current signal, and a dq-axis actual current signal; and to output a dq-axis reference voltage signal to the first coordinate transformation module; The first coordinate transformation module is used to transform the variables in the dq coordinate system into variables in the α-β coordinate system, so that it outputs the α-β reference voltage signal to the SVPWM module; The z1-z2 current loop fault-tolerant controller sets a notch filter, a harmonic QPR controller and an adaptive resonant controller; The adaptive resonant controller is a baseband QPR controller connected in series with a switch, and the on and off of the switch is controlled by the z1-z2 axis reference current signal; The notch filter is used to filter out the fundamental frequency current component mapped from the α-β plane to the z1-z2 plane; the harmonic QPR controller is provided with a third, fifth, and / or seventh harmonic QPR controller in parallel, so that the notch filter and the harmonic QPR controller are connected in series and then in parallel with the adaptive resonant controller; the z1-z2 current loop fault-tolerant controller is input with the z1-z2 axis actual current signal from the second coordinate transformation module, so that it outputs the z1-z2 axis reference voltage signal to the SVPWM module; The SVPWM module is input with the α-β reference voltage signal and the z1-z2 axis reference voltage signal, so that the module outputs the PWM signal to the first converter and the second converter respectively; During the normal operation and natural fault-tolerant operation of the motor, the reference current of the z1-z2 axis is zero, the switch is disconnected, and the adaptive resonant controller is in the fundamental frequency current open-loop control mode; In the current optimization operation stage, when the z1-z2 axis reference current is not zero, the switch is turned on and the adaptive resonant controller is in the fundamental frequency current closed-loop control mode.

6. The single-phase open-circuit fault-tolerant control method for dual three-phase permanent magnet synchronous motors according to claim 5, characterized in that: The method further provides a current vector trajectory module and an MT optimization controller; the current vector trajectory module is used to track the harmonic plane current trajectory, so that it inputs the actual current signal of the z1-z2 axis from the second coordinate transformation module and outputs the current vector angle of the z1-z2 plane; the MT optimization controller optimizes the control coefficient through an optimization tool, so that its optimization goal is to minimize the maximum single-phase copper loss, and inputs the q-axis reference current signal from the PI controller and the current vector angle of the z1-z2 plane from the current vector trajectory module respectively, so that it outputs the z1-z2 axis reference current signal to the z1-z2 current loop fault-tolerant controller; At the moment of fault, the adaptive resonant controller is deactivated and the fundamental frequency component related to the torque flows through the harmonic plane without being disturbed by the fundamental frequency filtering action of the notch filter; After the MT optimization controller outputs a new z1-z2 axis reference current according to the current vector angle of the z1-z2 plane, it re-enables the adaptive resonant controller to follow the fundamental frequency current reference value of the harmonic plane, switching from the natural fault-tolerant operation stage to the current control optimization operation stage.

7. The single-phase open-circuit fault-tolerant control method for dual three-phase permanent magnet synchronous motors according to claim 6, characterized in that: Assume that one of the two windings includes phases A, B, and C, and the other includes phases D, E, and F. Assume that x is any phase winding among phases A, B, C, D, E, or F. The optimization function of the MT optimization controller is as follows: p max =max{p A ,p B ,p C ,p D ,p E ,p F }; The copper loss calculation formula of the x-phase winding is as follows: The coefficients in the formula are Where: is the spatial phase angle of the x-phase winding; x∈{A, B, C, D, E, F}; when x is A, B, C, D, E, F phase respectively, k d is the amplitude coefficient of the d-axis current; p x is the copper loss of x-phase winding; p A is the copper loss of phase A winding; p B is the copper loss of phase B winding; p C is the copper loss of phase C winding; p D is the copper loss of D-phase winding; p E is the copper loss of E-phase winding; p F is the copper loss of F phase winding; R s is the phase winding resistance of the motor stator; i x is the x-phase current; i d is the d-axis current in the dq plane; i q is the q-axis current in the dq plane; I dc is the DC current coefficient in the phase winding copper loss; α d is the phase deviation angle of the d-axis current; θ e is the electrical angle of the motor rotor position; I1 is the cosine coefficient of phase winding copper loss; I2 is the sinusoidal coefficient of phase winding copper loss; k1, k2, k3, and k4 are coefficients to be optimized; i α is the α-axis current in the α-β plane; i β is the β-axis current in the α-β plane; i z1 is the z1-axis current in the z1-z2 plane; i z2 is the z2-axis current in the z1-z2 plane; p max is the maximum single-phase copper loss; min p max Indicates minimizing the maximum single-phase copper loss of the motor; The optimization function of the MT optimization controller is solved by the MATLAB optimization toolbox, and the optimal solution of the following coefficients is obtained: α d 、k d , k1, k2, k3, k4.

8. The single-phase open-circuit fault-tolerant control method for dual three-phase permanent magnet synchronous motors according to claim 7, characterized in that: After a phase failure occurs, the MT optimization controller determines the fault phase based on the current vector trajectory direction on the harmonic plane using the following method: When the current in the z1-z2 plane moves along the 0 / 180° direction, it is determined that phase A is faulty; When the current in the z1-z2 plane moves along the -120° / 60° direction, it is determined that the B phase is faulty; When the current in the z1-z2 plane moves along the 120° / -60° direction, it is determined that the C phase is faulty; When the current in the z1-z2 plane moves along the 150° / -30° direction, it is determined that the D phase is faulty; When the current in the z1-z2 plane moves along the 30° / -150° direction, it is determined that the E phase is faulty; When the current in the z1-z2 plane moves along the -90° / 90° direction, it is determined that the F phase is faulty; Based on the fault phase information, the MT optimization controller obtains the optimal coefficient in the maximum torque current mode and calculates the output z1-z2 axis reference current signal.

9. The single-phase open-circuit fault-tolerant control method for dual three-phase permanent magnet synchronous motors according to claim 5, characterized in that: A limiting module is also set between the PI controller and the PIR controller. The limiting module limits the q-axis reference current signal output by the PI controller and outputs the limited q-axis reference current signal to the PIR controller. After a fault occurs, the limiting value of the limiting module is changed according to the z1-z2 current reference value to ensure that the motor runs safely without exceeding the limit.

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