DFT-s-OFDM SIGNAL WITH SPECTRAL EXTENSION

Abstract

Embodiments of the present invention relate to a DFT-s-OFDM signal with spectral expansion. A first communication device (100) acquires Ndata Fourier coefficients on the basis of Ndata data symbols, and repeats Ne Fourier coefficients among the Ndata Fourier coefficients to acquire Nsc Fourier coefficients, where Ne is determined on the basis of at least one of a modulation symbol constellation of the Ndata data symbols and an FDSS window of the first communication device (100); the Nsc Fourier coefficients are multiplied by an FDSS window of a size of Nsc to obtain Nsc frequency shaping Fourier coefficients mapped onto Nsc subcarriers to obtain a transmitted DFT-s-OFDM signal (510). Therefore, the DFT-s-OFDM signal can be transmitted at a lower PAPR (Peak to Average Power Ratio). Furthermore, embodiments of the invention also relate to a corresponding second communication device (300), a corresponding method and a computer program.

Description

Technical Field

[0001] Embodiments of the present invention relate to a first communication device and a second communication device for a DFT-s-OFDM signal with spectrum spreading. Furthermore, embodiments of the present invention also relate to a corresponding method and a computer program. Background Art

[0002] Due to the limited transmission power of user equipment (UE), uplink transmission is considered to be the coverage bottleneck of modern wireless systems. Therefore, uplink coverage enhancement is an important topic in the current 3GPP new radio (NR) standardization. In the case of limited coverage, the envelope fluctuation of the transmitted signal (usually measured by its cubic metric (CM) or peak-to-average power ratio (PAPR)) should be as small as possible. This is because signals with higher CM / PAPR require the power amplifier to operate with a larger power back-off to avoid signal distortion caused by its nonlinear amplification range, which directly affects the link budget of the transmission.

[0003] Orthogonal frequency division multiplexing (OFDM) is the main modulation waveform of wireless communication systems to date, and is also the basic waveform of the 3GPP standard in uplink and downlink. Discrete Fourier transform spread OFDM (DFT-s-OFDM) is a variant of OFDM, mainly used to achieve lower PAPR transmission compared to standard OFDM. In DFT-s-OFDM, data symbols are first DFT precoded before OFDM modulation, forming a single carrier waveform because the energy of the modulation symbol is then distributed across all subcarriers in the allocated spectrum. In contrast, in the time domain, most of the symbol energy is concentrated in shorter intervals, resulting in time domain multiplexing of symbols.

[0004] Although the use of DFT-s-OFDM can improve PAPR, the effect may still be unsatisfactory, for example in scenarios with limited indoor deep coverage. Therefore, additional low PAPR techniques have been considered in 3GPP for uplink transmission. For example, in addition to the traditional LTE constellation, 5G NR also supports π / 2-BPSK in the uplink, aiming to further reduce PAPR and thus improve the power efficiency of the radio frequency (RF) amplifier at lower data rates. In the current 5G NR, π / 2-BPSK can also be used in conjunction with frequency domain spectrum shaping (FDSS). FDSS can also further reduce PAPR at the cost of manageable self-interference. In fact, no specific FDSS window is defined in the NR specification, but the use of the FDSS window is indirectly allowed through the looser RAN4 spectrum flatness requirements for NR uplinks using π / 2-BPSK.

[0005] Spectrum extension (SE) with FDSS is another PAPR reduction technique that is not currently specified in NR but is listed as a potential solution. The main goal is to achieve lower PAPR also for higher order modulations (such as QPSK), thereby improving coverage at higher data rates. Summary of the invention

[0006] An object of embodiments of the present invention is to provide a solution to reduce or solve the disadvantages and problems of conventional solutions.

[0007] Another object of an embodiment of the present invention is to provide a solution that can reduce PAPR compared to conventional solutions.

[0008] The above objects and others are achieved by the subject-matter of the independent claims. Further embodiments of the invention can be found in the dependent claims.

[0009] According to a first aspect of the present invention, the above and other objects are achieved by a first communication device for a communication system, wherein the first communication device is configured to:

[0010] Based on N data Data symbols are obtained N data Fourier coefficients, where N data is a positive integer;

[0011] Repeat N data N of the Fourier coefficients e Fourier coefficients to obtain N sc Fourier coefficients, where N e and N scis a positive integer, and N e is determined based on at least one of the following: data a modulation symbol constellation of data symbols, and a frequency domain spectrum shaping (FDSS) window of the first communication device;

[0012] The N sc Fourier coefficients and size N sc The FDSS window is multiplied to obtain N sc frequency-shaping Fourier coefficients;

[0013] The N sc The frequency-shaping Fourier coefficients are mapped to N sc subcarriers to obtain discrete Fourier transform spread orthogonal frequency division multiplexing (DFT-s-OFDM) signals;

[0014] The DFT-s-OFDM signal is transmitted.

[0015] An advantage of the first communication device according to the first aspect is that the first communication device is able to reduce the PAPR of the DFT-s-OFDM signal. This is because the optimal reduction in PAPR and the N e The value of depends largely on the modulation symbol constellation and the FDSS window used for the data symbols. Therefore, the first communication device as a transmitter device can use a lower power back-off and thereby increase the coverage of the DFT-s-OFDM signal.

[0016] In an implementation manner of the first communication device according to the first aspect, further based on N sc or N data To determine N e .

[0017] The advantage of this implementation is that the PAPR of the DFT-s-OFDM signal is further reduced according to the bandwidth allocation. This is because the N e The values ​​increase roughly in proportion to the bandwidth allocation, but vary slightly.

[0018] In an implementation manner of the first communication device according to the first aspect, the N data The modulation symbol constellation of the data symbols is a π / 2-BPSK constellation or a QAM constellation.

[0019] The advantage of this implementation is that the constellation used in the 3GPP standard is that π / 2-BPSK is specifically designed to achieve very low PAPR transmission and can be achieved with a very different N than the QAM constellation. e value for further improvement.

[0020] In an implementation manner of the first communication device according to the first aspect,

[0021] N e is predetermined; or

[0022] The first communication device is used to receive an indication N e The first control signal.

[0023] The advantage of this implementation is that it has a predetermined N e It is easy to minimize the signaling overhead; or if N is indicated by signaling e , then it is possible to better reduce the PAPR based on other transmission parameters or device-specific characteristics.

[0024] In an implementation manner of the first communication device according to the first aspect, the N e The Fourier coefficients are repeated:

[0025] included in resources allocated for transmission of the DFT-s-OFDM signal; or

[0026] added to the resources allocated for transmitting the DFT-s-OFDM signal.

[0027] The advantage of this implementation is that for adjacent multiple user bandwidths, if repeated Fourier coefficients are included in the allocated resources for each user, interference is avoided and repeated symbols can be used to improve demodulation performance; or if repeated Fourier coefficients are added to the allocated resources, the spectrum efficiency is improved.

[0028] In an implementation manner of the first communication device according to the first aspect, the first communication device is configured to:

[0029] Send a second control signal, wherein the second control signal instructs the first communication device to repeat the N e Fourier coefficients.

[0030] The advantage of this implementation is that the N indicated by the first control signal can be determined based on the capability of the first communication device indicated by the second control signal. e This enables optimized PAPR reduction based on several configuration parameters and implementation aspects of the first communication device.

[0031] In an implementation manner of the first communication device according to the first aspect, when the N sc Before multiplying the Fourier coefficients by the FDSS window, the first communication device is used to:

[0032] The L Fourier coefficients are used to represent the N sc Fourier coefficients are cyclically shifted to obtain N sc cyclically shifted Fourier coefficients, where L is a positive integer;

[0033] The N sc The cyclically shifted Fourier coefficients are related to the size N sc The FDSS window is multiplied to obtain N sc Fourier coefficients of frequency shaping and cyclic shifting;

[0034] The N sc The frequency-shaped and cyclically shifted Fourier coefficients are mapped to the N sc subcarriers to obtain the DFT-s-OFDM signal.

[0035] The advantage of this implementation is that when the PAPR varies as a function of the cyclic shift coefficient L, the PAPR can be further reduced.

[0036] In an implementation manner of the first communication device according to the first aspect,

[0037] L is predetermined; or

[0038] The first communication device is used to receive a first control signal indicating L.

[0039] The advantage of this implementation is that having a predetermined L can easily minimize the signaling overhead; or if L is indicated by signaling, it enables better PAPR reduction based on other transmission parameters or device-specific characteristics.

[0040] In an implementation manner of the first communication device according to the first aspect, L is determined based on any one of the following:

[0041] N e 、N data or N sc ;

[0042] The N data A modulation symbol constellation for data symbols;

[0043] formula or or or or or Where k is a positive integer, round(x) gives the integer closest to x. and They are the ceiling and floor operators for x respectively.

[0044] The advantage of this implementation is that the PAPR can be optimally reduced by selecting the optimal shift function.

[0045] In an implementation manner of the first communication device according to the first aspect, Less than the maximum allowed Fourier coefficient repetition capability of the first communication device (100) and is based on the FDSS window, L, and N sc and N data Any one of the items is determined.

[0046] The advantage of this implementation is that a smaller N e To improve spectral efficiency and further avoid choosing too large N which may increase PAPR e .

[0047] In an implementation manner of the first communication device according to the first aspect, the first communication device is configured to:

[0048] Based on the FDSS window, the N data The modulation symbol constellation of the data symbols, L, and N sc or N data Any one of the So as to minimize the peak-to-average-power ratio (PAPR) of transmitting the DFT-s-OFDM signal.

[0049] The advantage of this implementation is that N can be selected to optimize PAPR performance based on transmission parameters or device-specific characteristics. e .

[0050] In an implementation manner of the first communication device according to the first aspect, the first communication device is configured to:

[0051] Send Instructions The second control signal.

[0052] The advantage of this implementation is that the selected N is ensured according to the specific first communication device capabilities and implementation. e There will be no adverse effect on PAPR.

[0053] According to a second aspect of the present invention, the above and other objects are achieved by a second communication device for a communication system, wherein the second communication device is configured to:

[0054] A DFT-s-OFDM signal is received from a first communication device, wherein the DFT-s-OFDM signal includes a sc N subcarriers sc Fourier coefficients, where N sc is a positive integer;

[0055] The N is obtained based on the DFT-s-OFDM signal sc Fourier coefficients, the N sc Fourier coefficients include N data Fourier coefficients and the N data N of the Fourier coefficients e Repeated Fourier coefficients, where N data and N e is a positive integer; data The Fourier coefficients are based on N data data symbols are obtained, and N e is determined based on at least one of the following: data a modulation symbol constellation of data symbols, and a FDSS window of the first communication device;

[0056] Based on the N sc Fourier coefficients to obtain the N data Fourier coefficients;

[0057] For the N data Fourier coefficients are decoded to obtain the N data data symbol.

[0058] An advantage of the second communication device according to the second aspect is that the first communication device is able to reduce the PAPR of the DFT-s-OFDM signal. This is because the optimal reduction in PAPR and the N required to achieve this goal are e The value of depends largely on the modulation symbol constellation and the FDSS window used for the data symbols. Therefore, the first communication device as a transmitter device can use a lower power back-off and thereby increase the coverage of the DFT-s-OFDM signal.

[0059] In an implementation manner of the second communication device according to the second aspect, further based on N sc or N data To determine N e .

[0060] The advantage of this implementation is that the PAPR of the DFT-s-OFDM signal is further reduced according to the bandwidth allocation. This is because the N e The values ​​increase roughly in proportion to the bandwidth allocation, but vary slightly.

[0061] In an implementation manner of the second communication device according to the second aspect, the N data The modulation symbol constellation of the data symbols is a π / 2-BPSK constellation or a QAM constellation.

[0062] The advantage of this implementation is that the constellation used in the 3GPP standard is that π / 2-BPSK is specifically designed to achieve very low PAPR transmission and can be achieved with a very different N than the QAM constellation. e value for further improvement.

[0063] In an implementation manner of the second communication device according to the second aspect,

[0064] N e is predetermined; or

[0065] The second communication device is used to send an indication N e The first control signal.

[0066] The advantage of this implementation is that it has a predetermined N e It is easy to minimize the signaling overhead; or if N is indicated by signaling e , then it is possible to better reduce the PAPR based on other transmission parameters or device-specific characteristics.

[0067] In an implementation manner of the second communication device according to the second aspect, the N e The Fourier coefficients are repeated:

[0068] included in resources allocated for transmission of the DFT-s-OFDM signal; or

[0069] added to the resources allocated for transmitting the DFT-s-OFDM signal.

[0070] The advantage of this implementation is that for adjacent multiple user bandwidths, if repeated Fourier coefficients are included in the allocated resources for each user, interference is avoided and repeated symbols can be used to improve demodulation performance; or if repeated Fourier coefficients are added to the allocated resources, the spectrum efficiency is improved.

[0071] In an implementation manner of the second communication device according to the second aspect, the second communication device is configured to:

[0072] receiving a second control signal, wherein the second control signal instructs the first communication device to repeat the N e Fourier coefficients.

[0073] The advantage of this implementation is that the N indicated by the first control signal can be determined based on the capability of the first communication device indicated by the second control signal. e This enables optimized PAPR reduction based on several configuration parameters and implementation aspects of the first communication device.

[0074] In an implementation manner of the second communication device according to the second aspect, when obtaining the N data Fourier coefficients before the second communication device is used to:

[0075] The L Fourier coefficients are used to represent the N sc Fourier coefficients are cyclically shifted to obtain N sc cyclically shifted Fourier coefficients, where L is a positive integer;

[0076] Based on the N sc The N cyclically shifted Fourier coefficients are obtained data Fourier coefficients.

[0077] The advantage of this implementation is that when the PAPR varies as a function of the cyclic shift coefficient L, the PAPR can be further reduced.

[0078] In an implementation manner of the second communication device according to the second aspect,

[0079] L is predetermined; or

[0080] The second communication device is used to send a second control signal indicating L.

[0081] The advantage of this implementation is that having a predetermined L can easily minimize the signaling overhead; or if L is indicated by signaling, it enables better PAPR reduction based on other transmission parameters or device-specific characteristics.

[0082] In an implementation manner of the second communication device according to the second aspect, L is determined based on any one of the following:

[0083] N e 、N data or N sc ;

[0084] The N data A modulation symbol constellation for data symbols;

[0085] formula or or or or or Where k is a positive integer, round(x) gives the integer closest to x. and They are the ceiling and floor operators for x respectively.

[0086] The advantage of this implementation is that the PAPR can be optimally reduced by selecting the optimal shift function.

[0087] In an implementation manner of the second communication device according to the second aspect, Less than the maximum allowed Fourier coefficient repetition capability of the first communication device (100) and is based on the FDSS window, L, and N sc and N data Any one of the items is determined.

[0088] The advantage of this implementation is that a smaller N e To improve spectral efficiency and further avoid choosing too large N which may increase PAPR e .

[0089] In an implementation manner of the second communication device according to the second aspect, based on the FDSS window, the N data The modulation symbol constellation of the data symbols, L, and N sc or N data Any of So as to minimize the PAPR of transmitting the DFT-s-OFDM signal.

[0090] The advantage of this implementation is that N can be selected to optimize PAPR performance based on transmission parameters or device-specific characteristics. e .

[0091] In an implementation manner of the second communication device according to the second aspect, the second communication device is configured to:

[0092] Receiving instructions The second control signal.

[0093] The advantage of this implementation is that the selected N is ensured according to the specific first communication device capabilities and implementation. e There will be no adverse impact on PAPR.

[0094] According to a third aspect of the present invention, the above and other objects are achieved by a method for a first communication device, the method comprising:

[0095] Based on N data Data symbols are obtained N data Fourier coefficients, where N data is a positive integer;

[0096] Repeat N data N of the Fourier coefficients e Fourier coefficients to obtain N sc Fourier coefficients, where N e and N sc is a positive integer, and N e is determined based on at least one of the following: data a modulation symbol constellation of data symbols, and a FDSS window of the first communication device;

[0097] The N sc Fourier coefficients and size N sc The FDSS window is multiplied to obtain N sc frequency-shaping Fourier coefficients;

[0098] The N sc The frequency-shaping Fourier coefficients are mapped to N sc subcarriers to obtain DFT-s-OFDM signal;

[0099] The DFT-s-OFDM signal is transmitted.

[0100] The method provided according to the third aspect can be extended to an implementation corresponding to the implementation of the first communication device provided according to the first aspect. Therefore, the implementation of the method includes one or more features of the corresponding implementation of the first communication device.

[0101] The advantages of the method provided according to the third aspect are the same as the advantages of the corresponding implementation of the first communication device provided according to the first aspect.

[0102] According to a fourth aspect of the present invention, the above and other objects are achieved by a method for a second communication device, the method comprising:

[0103] A DFT-s-OFDM signal is received from a first communication device, wherein the DFT-s-OFDM signal includes a sc N subcarriers sc Fourier coefficients, where N sc is a positive integer;

[0104] The N is obtained based on the DFT-s-OFDM signal sc Fourier coefficients, the Nsc Fourier coefficients include N data Fourier coefficients and the N data N of the Fourier coefficients e Repeated Fourier coefficients, where N data and N e is a positive integer; data The Fourier coefficients are based on N data data symbols are obtained, and N e is determined based on at least one of the following: data a modulation symbol constellation of data symbols, and a FDSS window of the first communication device;

[0105] Based on the N sc Fourier coefficients to obtain the N data Fourier coefficients;

[0106] For the N data Fourier coefficients are decoded to obtain the N data data symbol.

[0107] The method provided according to the fourth aspect can be extended to an implementation corresponding to the implementation of the second communication device provided according to the second aspect. Therefore, the implementation of the method includes one or more features of the corresponding implementation of the second communication device.

[0108] The advantages of the method provided according to the fourth aspect are the same as the advantages of the corresponding implementation of the second communication device provided according to the second aspect.

[0109] An embodiment of the present invention further relates to a computer program, characterized by a program code, when the program code is executed by at least one processor, the program code causes the at least one processor to perform any one of the methods provided according to the embodiments of the present invention. In addition, an embodiment of the present invention further relates to a computer program product comprising a computer-readable medium and the computer program, wherein the computer program is included in the computer-readable medium and may include one or more of the following groups: read-only memory (ROM), programmable ROM (PROM), erasable PROM (EPROM), flash memory, electrically erasable PROM (EEPROM), hard disk drive, etc.

[0110] Other applications and advantages of embodiments of the present invention will be apparent from the following detailed description. BRIEF DESCRIPTION OF THE DRAWINGS

[0111] The accompanying drawings are intended to illustrate and explain different embodiments of the present invention, in which:

[0112] Figure 1 A first communication device according to an embodiment of the present invention is shown;

[0113] Figure 2 A flowchart of a method for a first communication device according to an embodiment of the present invention is shown;

[0114] Figure 3 A second communication device according to an embodiment of the present invention is shown;

[0115] Figure 4 A flow chart of a method for a second communication device according to an embodiment of the present invention is shown;

[0116] Figure 5 A communication system according to an embodiment of the present invention is shown;

[0117] Figure 6 A first implementation example of a first communication device with FDSS and spectrum spreading including a shift parameter L is shown;

[0118] Figure 7 An example of implementation of a second communication device is shown;

[0119] Figure 8 shows the N values ​​for different constellations, FDSS windows and L values. e 99th percentile PAPR [dB] of the function;

[0120] Fig. 9 shows that at a fixed N data =96, as N e The 99th percentile PAPR [dB] of QPSK as a function of

[0121] Figure 10 shows the optimal number of REs for spectrum spreading The number of resource blocks N as the total allocated bandwidth RB =N sc / 12 function to minimize PAPR;

[0122] Fig.11 The optimal number of REs for spectrum spreading is shown in With N sc Compared with the fixed case, data Minimize the PAPR of QPSK when the parameters are fixed.

[0123] Fig.12 Some signaling aspects of embodiments of the present invention are shown;

[0124] Fig.13 shows the spectrum spread data sequence as a function of the shift parameter L, where N data = 10 symbols and N e =4;

[0125] Fig.14 A second implementation example of a first communication device with FDSS and spectrum spreading including a shift parameter L is shown;

[0126] Fig.15 A third implementation example of a first communication device with FDSS and periodic spectrum spreading including a shift parameter L is shown;

[0127] Fig.16 The DMRS subcarrier positions of DFT-s-OFDM are shown;

[0128] Fig.17 RE allocation with three different DMRS design options is shown;

[0129] Fig.18 The CCDF of PAPR with three different options of DMRS design is shown, where N data =72, N sc =96 and RRC window (ρ=0.5, β=-0.65);

[0130] Fig.19 The maximum PAPR for option C is shown as a function of SE size with three different FDSS windows. DETAILED DESCRIPTION

[0131] Currently, in 3GPP NR, FDSS is only implicitly supported for π / 2-BPSK uplink transmissions through RAN4 requirements to limit the distortion caused by the FDSS window. In addition, in current NR, the FDSS window is proprietary and may be different for each UE. Therefore, the SE cannot be predetermined for any user equipment (UE), otherwise the size of the SE for a specific UE may be too large and degrade the PAPR performance. At the same time, the resource allocation and signal configuration of each UE are usually scheduled by the base station (BS).

[0132] Therefore, the embodiment of the present invention relates to the first communication device 100 and the second communication device 300, and solves the problem that the conventional solution cannot provide the minimum PAPR because the size of the SE is constant or not optimized relative to the PAPR performance and the UE-specific transmission configuration and capability. In addition, the problem that the Fourier coefficient shift is not selected in order to minimize the PAPR in the conventional solution is also solved.

[0133] Figure 1 FIG. 1 shows a first communication device 100 according to an embodiment of the present invention. Figure 1 In the illustrated embodiment, the first communication device 100 includes a processor 102, a transceiver 104, and a memory 106. The processor 102 is coupled to the transceiver 104 and the memory 106 via a communication component 108 known in the art. The first communication device 100 can be used for wireless communication and / or wired communication in a communication system. Wireless communication capabilities can be provided by an antenna or antenna array 110 coupled to the transceiver 104, while wired communication capabilities can be provided by a wired communication interface 112 coupled to the transceiver 104, etc.

[0134] The processor 102 may be referred to as one or more general-purpose central processing units (CPUs), one or more digital signal processors (DSPs), one or more application-specific integrated circuits (ASICs), one or more field programmable gate arrays (FPGAs), one or more programmable logic devices, one or more discrete gates, one or more transistor logic devices, one or more discrete hardware components, or one or more chipsets. The memory 106 may be a read-only memory, a random access memory (RAM), or a non-volatile RAM (NVRAM). The transceiver 304 may be a transceiver circuit, a power controller, or an interface that provides the ability to communicate with other communication modules or communication devices (e.g., network nodes and network servers). The transceiver 104, the memory 106, and / or the processor 102 may be implemented in a separate chipset or in a common chipset. In the present disclosure, the first communication device 100 for performing certain actions may be understood to mean that the first communication device 100 includes suitable components for performing the actions, such as a processor 102 and a transceiver 104.

[0135] According to an embodiment of the present invention, the first communication device 100 is configured to: data Data symbols are obtained N data Fourier coefficients, where N data is a positive integer. The first communication device 100 is further configured to repeat N data N of the Fourier coefficients e Fourier coefficients to obtain N sc Fourier coefficients, where N e and Nsc is a positive integer, and N e is determined based on at least one of the following: data The first communication device 100 is further configured to: sc Fourier coefficients and size N sc The FDSS window is multiplied to obtain N sc The first communication device 100 is further configured to convert N sc The frequency-shaping Fourier coefficients are mapped to N sc subcarriers to obtain a discrete Fourier transform spread orthogonal frequency division multiplexing (DFT-s-OFDM) signal 510. The first communication device 100 is further configured to send the DFT-s-OFDM signal 510.

[0136] In addition, in the embodiment of the present invention, the first communication device 100 used in the communication system 500 includes: a processor for data Data symbols are obtained N data Fourier coefficients, where N data Positive integer; repeat N data N of the Fourier coefficients e Fourier coefficients to obtain N sc Fourier coefficients, where N e and N sc is a positive integer, and N e is determined based on at least one of the following: data The modulation symbol constellation of the data symbols and the FDSS window of the first communication device 100; sc Fourier coefficients and size N sc The FDSS window is multiplied to obtain N sc frequency shaping Fourier coefficients; N sc The frequency-shaping Fourier coefficients are mapped to N sc subcarriers to obtain a DFT-s-OFDM signal 510; a transceiver, configured to send the DFT-s-OFDM signal 510.

[0137] In addition, in another embodiment of the present invention, the first communication device 100 used in the communication system 500 includes a processor and a memory having computer-readable instructions stored thereon, and when the instructions are executed by the processor, the processor: based on N data Data symbols are obtained N data Fourier coefficients, where N data Positive integer; repeat N data N of the Fourier coefficients e Fourier coefficients to obtain N sc Fourier coefficients, where N e and N sc is a positive integer, and N e is determined based on at least one of the following: data The modulation symbol constellation of the data symbols and the FDSS window of the first communication device 100; sc Fourier coefficients and size N sc The FDSS window is multiplied to obtain N sc frequency shaping Fourier coefficients; N sc The frequency-shaping Fourier coefficients are mapped to N sc subcarriers to obtain a DFT-s-OFDM signal 510; and send the DFT-s-OFDM signal 510.

[0138] Figure 2 It is shown that the first communication device 100 (such as Figure 1 The method 200 includes: data Data symbol acquisition (202) N data Fourier coefficients, where N data is a positive integer. Method 200 also includes repeating (204) N data N of the Fourier coefficients e Fourier coefficients to obtain N sc Fourier coefficients, where N e and N sc is a positive integer, and N e is determined based on at least one of the following: data The method 200 further includes: sc Fourier coefficients and size N sc The FDSS window is multiplied by (206) to obtain N sc The method 200 also includes converting N sc The frequency-shaped Fourier coefficients are mapped (208) to N scsubcarriers to obtain a DFT-s-OFDM signal 510. The method 200 also includes transmitting (210) the DFT-s-OFDM signal 510.

[0139] Figure 3 FIG. 3 shows a second communication device 300 according to an embodiment of the present invention. Figure 3 In the illustrated embodiment, the second communication device 300 includes a processor 302, a transceiver 304, and a memory 306. The processor 302 is coupled to the transceiver 304 and the memory 306 via a communication member 308 known in the art. The second communication device 300 also includes an antenna or antenna array 310 coupled to the transceiver 304, which indicates that the second communication device 300 can be used for wireless communication in a communication system.

[0140] The processor 302 may be referred to as one or more general-purpose CPUs, one or more DSPs, one or more ASICs, one or more FPGAs, one or more programmable logic devices, one or more discrete gates, one or more transistor logic devices, one or more discrete hardware components, and one or more chipsets. The memory 306 may be a read-only memory, a RAM, or an NVRAM. The transceiver 104 may be a transceiver circuit, a power controller, or an interface that provides the ability to communicate with other communication modules or communication devices. The transceiver 304, the memory 306, and / or the processor 302 may be implemented in a separate chipset or in a common chipset. In the present disclosure, the second communication device 300 being used to perform certain actions may be understood to mean that the second communication device 300 includes a suitable component for performing the action, for example, a processor 302 and a transceiver 304, etc.

[0141] According to an embodiment of the present invention, the second communication device 300 is configured to receive a DFT-s-OFDM signal 510 from the first communication device 100, wherein the DFT-s-OFDM signal 510 includes a DFT-s-OFDM signal mapped to N sc N subcarriers sc Fourier coefficients, where N sc is a positive integer. The second communication device 300 is further configured to obtain N based on the DFT-s-OFDM signal 510 sc Fourier coefficients, N sc Fourier coefficients include N data Fourier coefficients and N data N of the Fourier coefficients e Repeated Fourier coefficients, where N data and N e is a positive integer; N data The Fourier coefficients are based on N data data symbols are obtained, and N eis determined based on at least one of the following: data The second communication device 300 is further configured to: sc Fourier coefficients to obtain N data Fourier coefficients. The second communication device 300 is further configured to data Fourier coefficients are decoded to obtain N data data symbol.

[0142] In addition, in an embodiment of the present invention, the second communication device 300 used in the communication system 500 includes: a transceiver, configured to: receive a DFT-s-OFDM signal 510 from the first communication device 100, wherein the DFT-s-OFDM signal 510 includes a DFT-s-OFDM signal mapped on N sc N subcarriers sc Fourier coefficients, where N sc is a positive integer; a processor, configured to: obtain N based on the DFT-s-OFDM signal 510 sc Fourier coefficients, N sc Fourier coefficients include N data Fourier coefficients and N data N of the Fourier coefficients e Repeated Fourier coefficients, where N data and N e is a positive integer; N data The Fourier coefficients are based on N data data symbols are obtained, and N e is determined based on at least one of the following: data The modulation symbol constellation of the data symbols and the FDSS window of the first communication device 100; based on N sc Fourier coefficients to obtain N data Fourier coefficients; for N data Fourier coefficients are decoded to obtain N data data symbol.

[0143] In addition, in another embodiment of the present invention, the second communication device 300 used in the communication system 500 includes a processor and a memory having computer-readable instructions stored thereon, and when the instructions are executed by the processor, the processor: receives a DFT-s-OFDM signal 510 from the first communication device 100, and the DFT-s-OFDM signal 510 includes a DFT-s-OFDM signal mapped on N sc N subcarriers sc Fourier coefficients, where N sc is a positive integer; based on the DFT-s-OFDM signal 510, N is obtained sc Fourier coefficients, Nsc Fourier coefficients include N data Fourier coefficients and N data Fourier coefficients N e Repeated Fourier coefficients, where N data and N e is a positive integer; N data The Fourier coefficients are based on N data data symbols are obtained, and N e is determined based on at least one of the following: data The modulation symbol constellation of the data symbols and the FDSS window of the first communication device 100; based on N sc Fourier coefficients to obtain N data Fourier coefficients; for N data Fourier coefficients are decoded to obtain N data data symbol.

[0144] Figure 4 It is shown that the second communication device 300 (such as Figure 3 4. The method 400 includes receiving (402) a DFT-s-OFDM signal 510 from the first communication device 100, the DFT-s-OFDM signal 510 including a signal mapped to N sc N subcarriers sc Fourier coefficients, where N sc The method 400 further includes obtaining (404) N based on the DFT-s-OFDM signal 510 sc Fourier coefficients, N sc Fourier coefficients include N data Fourier coefficients and N data N of the Fourier coefficients e Repeated Fourier coefficients, where N data and N e is a positive integer; N data The Fourier coefficients are based on N data data symbols are obtained, and N e is determined based on at least one of the following: data The method 400 also includes a modulation symbol constellation of N data symbols and a FDSS window of the first communication device 100. sc Fourier coefficients are obtained (406) N data Fourier coefficients. Method 400 also includes data Fourier coefficients are decoded (408) to obtain N data data symbol.

[0145] Figure 5 A communication system 500 provided according to an embodiment of the present invention is shown. The communication system 500 in the disclosed example includes a first communication device 100 and a second communication device 300 for communicating and operating in the communication system 500. For simplicity, the communication system 500 shown includes only one first communication device 100 and one second communication device 300. However, the communication system 500 may include any number of first communication devices 100 and any number of second communication devices 300, which does not deviate from the scope of the present invention.

[0146] The first communication device 100 is used to generate and send a DFT-s-OFDM signal 510 according to the disclosed scheme in the communication system 500. Therefore, the second communication device 300 is used to receive the DFT-s-OFDM signal 510 sent by the first communication device 100. The communication system 500 is any suitable communication system, such as 3GPP 5G NR or 6G. The first communication device 100 can also be represented as a transmitter device or simply referred to as a transmitter, and can be a client device, such as a UE. Correspondingly, the second communication device 300 can also be represented as a receiver device or simply referred to as a receiver, and can be a network access node, such as a BS. The network access node can be connected to a network (network, NW), such as a core network, via a communication interface. However, embodiments of the present invention are also applicable to situations where the transmitter is a base station and the receiver is a UE, that is, the opposite situation. In addition, embodiments of the present invention are also applicable to other network nodes, such as repeaters, repeaters, etc. In addition, embodiments of the present invention are also applicable to direct communication between UEs, such as through a sidelink (sidelink, SL) interface.

[0147] In order to better understand the disclosed solution, the following will provide a signal definition, where a signal carrying N data The DFT-s-OFDM signal s[n] with SE of constellation symbols uses N sc =N data +N e OFDM subcarriers, where N e is the number of subcarriers used for symbol repetition only for the purpose of reducing PAPR. For ease of explanation and considering the symmetry of the actual FDSS window, we assume that N e is an even integer. However, by assuming that the left SE is Right SE is Or vice versa, the disclosed scheme can be directly extended to N e For odd number cases.

[0148] For 0≤n≤N fft -1 samples, the low-pass equivalent time discrete DFT-s-OFDM signal representation is defined as follows:

[0149]

[0150] With frequency domain symbols

[0151]

[0152] in

[0153] ·N fft is the IFFT size for OFDM modulation.

[0154] ·N data is the number of modulation constellation symbols.

[0155] ·N sc =N e +N data is the total number of OFDM subcarriers.

[0156] ·N e is the number of subcarriers used for SE, also called the size of SE.

[0157] ·{W[0],…,W[N sc -1]} is the FDSS window coefficient.

[0158] L is the Fourier coefficient shift parameter.

[0159] (mod N) is the modulo N operator.

[0160] It can be noted that except and Other types of normalization factors other than can be used in equations (1) and (2). Typically, W[k] is real-valued, but embodiments of the present invention are not limited thereto, and complex-valued FDSS can be used. Embodiments of the present invention also include the case where W[k]=1.

[0161] Therefore, according to an embodiment of the present invention, a cyclic shift with a shift parameter L may also be introduced. This means that according to an embodiment, when N sc Before multiplying the L Fourier coefficients by the FDSS window, the first communication device 100 is used to: multiply the L Fourier coefficients by the N sc Fourier coefficients are cyclically shifted to obtain N sc Fourier coefficients of cyclic shift, where L is a positive integer; N sc The Fourier coefficients of the cyclic shift and the size of N sc The FDSS window is multiplied to obtain N sc The first communication device 100 finally converts N sc The frequency-shaped and cyclically shifted Fourier coefficients are mapped to Nsc subcarriers to obtain a DFT-s-OFDM signal 510.

[0162] Accordingly, according to the embodiment, when obtaining N data Before the L Fourier coefficients, the second communication device 300 is used to: use L Fourier coefficients to N sc The Fourier coefficients are cyclically shifted (in the direction opposite to that performed in the first communication device 100) to obtain N sc Fourier coefficients of cyclic shift, where L is a positive integer; based on N sc The Fourier coefficients of the cyclic shift are obtained by data Fourier coefficients.

[0163] Figure 6 The block diagram of FIG. 1 shows a first implementation example of the first communication device 100. Another implementation example of the first communication device 100 will be given below. Figure 6 , the first communication device 100 includes N data Point DFT block 120 is used to receive N data data symbols (x[m]) and based on N data Data symbol output N data The first communication device 100 further includes an SE block 122 for repeating N data N of the Fourier coefficients e Fourier coefficients to provide N sc Fourier coefficients. Therefore, N sc The L Fourier coefficients are cyclically shifted in cyclic shift block 124 to output N Fourier coefficients which are fed to FDSS block 126. sc Thereafter, in FDSS block 126 N sc The Fourier coefficients of the cyclic shift and the size of N sc The FDSS window is multiplied to output the N N that is fed to the mapper block 128 sc frequency shaped and cyclically shifted Fourier coefficients (X′[k]). In mapper block 128, N sc The frequency-shaped and cyclically shifted Fourier coefficients are mapped to N fft N of the subcarriers sc subcarriers to output N in the frequency domain fftThe DFT-s-OFDM signal with coefficients is converted into the time domain in the IFFT block 130. In addition, a cyclic prefix (CP) may be added in the CP block 132. Thereafter, the DFT-s-OFDM signal with the CP is transmitted in the communication system 500 as a signal s[n].

[0164] The corresponding implementation of the second communication device 300 is exemplified as follows: Figure 7 The received DFT-s-OFDM signal 510 is first demodulated by the CP removal step in the CP removal block 320 and then fed to the N fft Thereafter, the demapper block 324 allocates the N subcarrier coefficients among these N subcarrier coefficients according to the bandwidth of the first communication device 100. fft Select N out of the subcarriers sc Fourier coefficients. The selected N sc The Fourier coefficients are fed to a cyclic shift block 326, which cyclically shifts the L Fourier coefficients to provide N data Fourier coefficients. A common method to mitigate transmission channel fading and FDSS window attenuation is to use N data For further improvement, and depending on whether the repeated symbols are included in the resources allocated for transmission of the DFT-s-OFDM signal or added to the resources allocated for transmission of the DFT-s-OFDM signal, the equalization and combiner block 328 may also be input with N corresponding to the repeated symbols. e Fourier coefficients. For example, the Fourier coefficients received from in-band and their repeated versions received from spectrum extension can be in phase and summed before equalization, thereby providing combining gain and frequency diversity gain. Finally, N data The equalized Fourier coefficients are fed to N data Point inverse DFT (IDFT) block 330, IDFT block 330 is used to data The equalized Fourier coefficients are pre-coded to provide the information sent to the decoder block for decoding ( Figure 7 N data demodulated data symbols.

[0165] Given a FDSS window, parameter N e Both L and L will affect the PAPR performance. In this article, we will show how to carefully select these parameters to optimally reduce the PAPR. Specifically, we will show:

[0166] 1. When using SE (ie N e >0), PAPR will usually decrease, but only to an optimum Then it increases again. Therefore, choosing too large N e This is not only detrimental to spectral efficiency, but also to PAPR reduction. This non-trivial optimization depends mainly on the FDSS window, but also on the shift values ​​L, N data The modulation symbol constellation and bandwidth allocation for each data symbol.

[0167] 2. The L value that provides the lowest PAPR depends on N data The modulation symbol constellation of data symbols and the spreading size N e and the number of constellation symbols, i.e. N data =N sc -N e .

[0168] There exists a global optimum that minimizes PAPR

[0169] It is known that SE can further reduce PAPR based on FDSS. However, it is not known whether there is an optimal SE size. The PAPR can be minimized, so a too large SE may even increase the PAPR. Therefore, the SE size should be selected so that That is, less than Should Corresponding to the specific SE capabilities of the first communication device 100 as a transmitter. It can be seen that the precise Depends on FDSS window, L, and N sc and N data . Please note that selecting It may also be to mitigate the reduced spectral efficiency of SE at the expense of a slightly increased PAPR. The system configuration may also need to select For example, N e Restricted to factors of 12 subcarriers to have an integer number of RBs.

[0170] exist Figure 8(a) to Figure 8(c) In the example, the 99th percentile PAPR is plotted as a function of the number of SE subcarriers N. e function, and the total bandwidth is fixed to N sc =96. This is plotted for different filters, modulation symbol constellations and shift values ​​L. In all these numerical comparisons, two FDSS windows were considered, namely: i) Kaiser window with β = 3; ii) Truncated RRC window with ρ = 0.5, β = -0.65. The conventional shift rule L = 0 and The following will explain why the parameters It is the best shift method that is closest to π / 2-BPSK. For QAM constellations, it is also used It will be shown later why this is close to the optimal shift for QPSK.

[0171] Figure 8(a) to Figure 8(c) It shows that the PAPR generated as N e The function of is neither a linear function nor a continuously decreasing function. Therefore, for all cases, there is an optimal SE size that minimizes PAPR. From this Further increasing SE will become detrimental. Therefore, the SE size should preferably be chosen so that The bit rate can be predetermined based on the transmission configuration (e.g., FDSS window, shift parameter L, modulation symbol constellation, and bandwidth allocation). from Figure 8(a) to Figure 8(c) It can also be seen that even without FDSS, the PAPR can be greatly reduced through SE.

[0172] In the simulation, for a given total bandwidth N sc , increased by N e The value of N data The value of N e +N data =N sc Therefore, for a given N sc Alternatively, we can consider optimizing the optimal N e , in order to fix N data and increase the total bandwidth N sc . Fig. 9 Comparison of different N e PAPR, where N data = 96 is fixed, with QPSK and from Fig. 9 It can be observed that the shapes of these curves are similar to N sc The shapes when fixed are similar and clearly show a global optimum.

[0173] FDSS window and Follow N sc Convergence of growth

[0174] SE Size The global optimum of N depends mainly on the FDSS window design. In fact, for a given FDSS window, sc growth, It seems to converge to N sc This means that to achieve a similar PAPR reduction, N e N should be allocated with bandwidth sc or N data increases linearly. Therefore, Ne N is best sc or equivalently N data In other words, in the embodiment, N e is further based on the parameter N sc or N data Sure.

[0175] Figures 10(a) and 10(b) show the numerical results for different FDSS windows. (N sc percentage), the YesN RB =N sc / 12 function. sc The percentage indicates the numerical value Should YesN RB =4, 8, 12, 24, 48, 96 and 100. For a given modulation symbol constellation, it can be seen that, depending on the FDSS window, the optimal can vary significantly. In these examples, when N e The lowest PAPR is likely to be achieved when the value is between 7% and 35% of the total bandwidth allocation. In general, the more edge attenuation the FDSS window provides, the lower the PAPR, with the optimal value being More importantly, for each FDSS window, it seems that as N sc Growth, optimal SE size Almost N sc A constant fraction.

[0176] In addition, Fig.11 The N data The best way to get fixed As obtained The QPSK modulation used here is ) and the FDSS Kaiser window (where β = 2). This is similar to N sc Obtained when fixed (As N sc It can be observed that compared with N sc Optimization when fixed In comparison, N data Optimization when fixed Get very similar ratio. It should also be noted that in N data In the case of fixed It may be more natural to express the optimal SE size as a ratio of In general, the relationship between these two ratios is always as follows: Assume is equivalent to

[0177] Semi-analytical approximation of

[0178] The optimal SE size that minimizes the PAPR can also be verified and approximated by a semi-analytical numerical search

[0179] Therefore, first reformulate the signal as multiple pulses By inserting equation (2) into (1), we can obtain:

[0180]

[0181] The pulse shaping filter is:

[0182]

[0183] In the absence of FDSS windowing, W[0]=…=W[N sc -1]=1, which further simplifies to have N sc Conventional DFT-s-OFDM pulse in the form of Dirichlet kernel for modulated subcarriers:

[0184]

[0185] Using this pulse formula and adjusting the bounds, we get the following bounds for φ-rotated BPSK on our system:

[0186]

[0187] in is the squared norm of the FDSS window.

[0188] For π / 2-BPSK, φ = π / 2. In addition, L = -N e / 2 is the best shift method, so Therefore, the bound of equation (6) is approximately equal to

[0189]

[0190]

[0191] Please note that when allowing N e When it is an odd integer, the equation holds, that is, L = -(Ne -1) / 2.

[0192] Similar bounds can be derived for QPSK, and indeed any PSK modulation. The bounds become

[0193]

[0194] By using |cosθ|≤1, it can be directly verified that Furthermore, if φ = 0 (as in normal BPSK), if we can have L = -(N e -1) / 2, then the two boundaries are the same, that is:

[0195] To calculate these bounds, we need to fft finds a maximum over the time samples. However, the computational complexity of this step can be greatly reduced by noting that the function inside the maximum operator is periodic, so only samples, we can find the maximum value on n.

[0196] The bounds in equations (7) and (8) can be compared with the simulated 99th percentile PAPR. The bounds in equations (7) and (8) are not tight, but nonetheless they appear to roughly hold the location of the optimal SE size that minimizes the PAPR. Therefore, the value N that minimizes these bounds can be numerically searched for. e , to approximate the optimal SE size. This is shown in Figure 10(a) and Figure 10(b), where we can see that This semi-analytic approximation of is usually very close, staying within 5% in the worst case, except for the case of π / 2-BPSK and Kaiser window (β=3). For this special case, the semi-analytic search is far from the numerical search, and it seems to correspond to the position of the minimum PAPR when L=0, see Figure 8(a). In this case, the problem is that SE extension can only reduce the PAPR by 0.32 dB at most, while without SE, the PAPR is already very low at only 1.84 dB. Therefore, the resulting bound is very flat over a large range of values, without a very sharp minimum.

[0197] In addition, an equivalent pulse shaping formula is used to provide a complementary intuitive explanation for the existence of this optimal SE size. The upper bound in equation (8) can be restated as

[0198]

[0199] When N e When increasing, the coefficient systematically increased. However, is the square of the maximum coherent combination of pulses, which can be used as N e For the typical window function W[k] as described above, the pulse shaping filter |g i [n]|Basically maintains the sinc shape, but with more or less attenuated sidelobes. The number of pulses is N data , as given by equation (3), and their time interval is From equation (4), we can further observe that the pulse shape depends only on the FDSS window and the total number of allocated subcarriers N sc If N is fixed sc , there will be N samples separated data =(N sc -N e ) pulses, that is, the pulse shape will not be affected by the increase of N e The difference between the two pulses will be different, but the time interval between pulses will increase. The degree of pulse lobe overlap depends on the design of the FDSS, and in any case the increased time interval will work to reduce this overlap and help reduce the peak power of the signal.

[0200] If it is changed to a fixed value N data , N sc Will follow N e increases, and there will be N data The pulse is In contrast, in this case, without FDSS, the pulse shape of equation (5) becomes narrower because the width of the main lobe is N samples. Narrower pulses also help reduce the peak power of the signal. In any case, the FDSS window must be set for N sc The bandwidth of each subcarrier is designed.

[0201] Modulation symbol constellation

[0202] In the embodiment of the present invention, N data The modulation symbol constellation of the data symbols is a π / 2-BPSK constellation or a QAM constellation. In the case where the modulation symbol constellation is QPSK, 16-QAM and 64-QAM, when the same FDSS window as shown in Figure 10(b) is used, the optimal They seem to be very close to each other. With the Kaiser window (β=3), the maximum difference between 16-QAM and QPSK is 1.5%. In some cases, the difference is not noticeable. Therefore, a good rule of thumb is to choose the same N for 16-QAM and 64-QAM as for QPSK. e Table 1 below highlights the selection of the π / 2-BPSK constellation and the QAM constellation. The difference.

[0203] Table 1 According to Figure 10 Numerical selection

[0204]

[0205] From Table 1, we can see that for the same FDSS window, the N when the best PAPR is obtained when using π / 2-BPSK is 0. e For π / 2-BPSK, a good rule of thumb to significantly reduce PAPR is to choose in is the best value for QPSK because Therefore, in summary, we can A single value of N is derived for all constellations e choose.

[0206] Shift parameter L

[0207] In FIG8 , one can see the shifting method derived in the present disclosure Provides the best PAPR, while from conventional solutions The worst PAPR is obtained in . It can be found that the two shifting methods have good SE size. The best results can be achieved when the values ​​are similar. Best SE size comparable Slightly reduce N sc 1% to 2%.

[0208] Now let's explain why the best shift method for π / 2-BPSK is The best shift method for QPSK and other QAM constellations is Starting from the multi-carrier signal in equation (3), the phase difference between two adjacent pulses indexed m and (m+1) is

[0209]

[0210] Then, if the FDSS window is real and symmetric, one can use

[0211]

[0212] This can also be directly verified to hold for the special case of equation (2) without FDSS.

[0213] Therefore, the pulse phase difference can be obtained

[0214]

[0215] The purpose of designing the rotated constellation as π / 2-BPSK is to ensure that the BPSK symbols of adjacent pulses are transmitted with a phase difference of almost π / 2 (modπ) so as to minimize their maximum power combination.

[0216] For π / 2-BPSK, the optimal value of L is found so that equation (12) is closest to zero, that is, If N e is an odd integer, then Provides the best performance. If N e is an even integer, then as in [7] or Therefore, as shown in Figure 8(a), we choose The shifting method can provide better PAPR performance than L=0.

[0217] For QPSK and other QAMs, select is suboptimal, and better PAPR can almost always be found using other values ​​of L. It is worth noting that the lowest PAPR is obtained by creating a phase difference between adjacent pulses close to π / 4 (mod π / 2).

[0218] In other words, it should be concluded

[0219]

[0220] Therefore, the optimal PAPR can always be achieved by choosing L approximately equal to

[0221]

[0222] Where k is an integer, because the phase difference between data symbols caused by L shift is always close to π / 4.

[0223] That is why in Figure 8(b) and Figure 8(c), The PAPR is the worst when The PAPR is optimal when L=0. Another value L=0 in the conventional scheme is a trade-off between the first two L values, because when i)N e = 0, then or when ii) N e =N sc / 5 hours (equivalent to ),but It can be simplified to the other two methods. e ≈N sc In 20% of the cases, L = 0 can achieve near-optimal performance.

[0224] Compared with conventional regimens, the use of The gain can be as high as 0.6dB, especially when the size of SE N e Less than optimal , as shown in Figure 8(b) and Figure 8(c). Please note that when It may actually be desirable, for example, to mitigate the spectral efficiency degradation of SE or to have an integer number of RBs, to set N e Limited to factors of 12 subcarriers.

[0225] To summarize, in order to achieve the best PAPR reduction effect, as shown in the figure above, L of some modulation symbol constellations between π / 2-BPSK and QAM should be different. It should also be based on N e and depending on the modulation symbol constellation, it may also be based on N data or N sc To determine L. Specifically, for QAM, the optimal PAPR value is obtained by or or or or or where k is a positive integer and round(x) gives the integer closest to x. and They are the ceiling and floor operators for x respectively.

[0226] Signaling

[0227] Fig.12 The signaling aspects according to an embodiment of the present invention are shown, because based on the gains shown above, the disclosed scheme assumes that the number of subcarriers used for SE is N e It is based on the FDSS capability of the first communication device 100, and possible other transmission configurations, such as but not limited to:

[0228] The total number of allocated subcarriers N sc , or the number of subcarriers containing data symbols N data .

[0229] · Fourier coefficient shift parameters of the L subcarriers.

[0230] Modulation symbol constellation.

[0231] Both the transmitter (e.g., UE) and the receiver (e.g., base station) need to know the selected parameter N eand the value of L. Such information can be provided by control signaling between the first communication device 100 (i.e., the transmitter) and the second communication device 300 (i.e., the receiver). For example, the control signaling can be performed by a higher protocol layer, such as radio resource control (RRC) signaling or medium access control (MAC) signaling. This is advantageous if the parameters do not need to be changed in a dynamic manner. Another option is to signal through the physical layer, such as through a control channel. The advantage of this is that the parameters can be changed on the fly. A combination of high-level signaling and physical layer signaling can also be used, for example, N e Certain values ​​of and L are configured by higher layers, and physical layer signaling selects among these values.

[0232] refer to Fig.12 In step I, the second communication device 300 sends a first control signal 520 to the first communication device 100. The first control signal 520 indicates the parameter N e and / or L. Fig.12 In step II, the first communication device 100 is used to receive an instruction N from the second communication device 300 e and / or a first control signal 520 of L.

[0233] However, the parameter N e and / or L, or part of the parameters may also be provided implicitly. For example, parameter N e and L may be predetermined and defined by standards for different modulation symbol constellations, for different numbers of allocated subcarriers, for different parameters of frequency domain filters, etc. This avoids the use of control signaling, thereby reducing overhead in the communication system 500 .

[0234] In addition, when the first communication device 100 is a UE, the UE should signal to the BS whether it supports the use of SE. This can be done through UE capability signaling. In other words, the first communication device 100 can be used to send an instruction indicating that the first communication device 100 repeats N e The second control signal 530 capable of Fourier coefficients, such as Fig.12 As shown in step III in Fig.12 In step IV, the second communication device 300 receives the second control signal 530 and derives information about the repetition capability of the first communication device 100 therefrom.

[0235] In addition, if the communication standard does not provide an FDSS window, the first communication device 100 (eg, UE) may set the value The value is N eThe maximum value of N is greater than the value above which PAPR cannot be reduced. In this way, the BS can avoid e The value is greater than required.

[0236] It may be noted that the signaling of the first control signal 520 and the second control signal 530 may be performed in the reverse order, ie, the first control signal 520 is sent after the second control signal 530, without departing from the scope of the disclosed solution. The first control signal 520 and the second control signal 530 may also be sent simultaneously.

[0237] The communication resources used by the UE to transmit on the uplink may be pre-configured, for example, by semi-static scheduling or by configuring a grant. Alternatively, the BS may send an uplink grant containing information about the transmission, including the resource blocks allocated for the transmission, to the UE via a physical downlink control channel (PDCCH). In an embodiment of the present invention, N e subcarriers may be included in a resource block allocated for transmission, i.e., N e Fourier coefficient repetitions are included in the resources allocated for transmitting the DFT-s-OFDM signal 510. In other embodiments of the present invention, N e subcarriers may be added to a resource block allocated for transmission, i.e., N e Fourier coefficient repetitions are added to the resources allocated for transmitting the DFT-s-OFDM signal 510.

[0238] Implementation Example

[0239] The DFT-s-OFDM signal discussed and proposed above can generally be realized by the concatenation of DFT-precoding, periodic SE with shift, FDSS window and OFDM modulation as shown below.

[0240] DFT precoding: for k = 0, 1, ..., N data -1, constellation symbol {x[0],…,x[N data -1]}First, DFT precoding is performed to obtain the Fourier coefficients:

[0241]

[0242] The transmitter chain Figure 6 As shown in the figure, the constellation symbols {x[0],…,x[N data First, DFT precoding is performed in DFT block 120 to obtain Fourier coefficients {X[0],…,X[N data -1]}.

[0243] The periodic SE with shift in the SE block 122 and the cyclic shift block 124: The frequency domain sequence is allocated to N using periodic repetition sc =N data +N e The spread spectrum of subcarriers is as follows

[0244] X (se) [k] = X[k + L(mod N data )] (16)

[0245] For k = 0, 1, ..., N sc -1, where L is an integer shift. Note that this definition of SE is for N e Odd numbers are also directly valid.

[0246] FDSS: FDSS is applied to the Fourier coefficients in FDSS block 126 before transmission via OFDM modulation as follows

[0247] X′[k]=W[k]X (se) [k] (17)

[0248] where {W[0],…,W[N sc -1]} is the FDSS window.

[0249] OFDM modulation: Finally, for n = 0, 1, ..., N fft -1, in IFFT block 130 by N fft The point IFFT obtains the (normalized) time-discrete low-pass equivalent signal as shown below:

[0250]

[0251] SE with shift

[0252] In addition, the steps of SE with cyclic shift can be implemented by different equivalent embodiments. Different examples of transmission frequency domain sequences as a function of the shift parameter L are as follows: Fig.13 For any value of L, the original symbol sequence {X[0],…,X[N data -1]} is always included in the in-band spectrum, with a maximum cyclic shift of symbols; the left excess-band symbol is always a repetition of the symbol at the inner edge of the right excess-band; the same is true for the right excess-band. A number of embodiments are disclosed for implementing SE according to equation (12).

[0253] In such Figure 6 In the first exemplary implementation shown, the original N dataThe end of the longest sequence X[k] is appended with its first N e symbols, and then expand N sc The long sequence is cyclically shifted by L symbols to achieve SE.

[0254] In such Fig.14 In the second exemplary implementation shown in FIG. data The long sequence X[k] is cyclically shifted by L symbols, and then its first N bits are appended to the end of the shifted sequence. e symbols to implement the cycle SE. Therefore, Figure 6 and Fig.14 The difference between the exemplary implementations in is the order of the cyclic shift and SE operations. Fig.14 In , the rotation is performed before the SE operation.

[0255] In such Fig.15 In the third exemplary implementation shown in FIG. 1 , the original N data Cyclic shift of the long sequence X[k] symbols to implement SE, as shown below

[0256]

[0257] Then in the shift sequence X (L) [k] after and before the addition After symbols, namely

[0258]

[0259] therefore, Figure 6 and Fig.15 The difference between the exemplary implementations in is the order of the cyclic shift and SE operations. Fig.15 In , the circular shift is performed before the SE operation, such as Fig.13 However, in Fig.15 SE is also implemented in another way, such as Figure 6 and Fig.14 Therefore, The third exemplary implementation corresponds to the symmetric SE of the conventional scheme, see Fig.13 L = -2. Note that the choice of the shift parameter L depends only on N e , in fact, with N data It doesn't matter, because L is in N data In addition, it should be noted that the choice of L is only for convenience and does not actually bring any additional benefits.

[0260] DMRS transmission

[0261] In 5G NR, demodulation reference symbols (DMRS) are multiplexed with data symbols (called physical uplink shared channel (PUSCH) in NR), where usually only a few OFDM symbols carry DMRS, for example, 1 symbol in 14 symbols carries DMRS.

[0262] When using DFT-s-OFDM, DMRS is time-multiplexed with PUSCH. In the OFDM symbol carrying DMRS, N sc A DMRS sequence without DFT precoding is inserted into every other subcarrier (called resource element (RE) in NR), while other REs are blocked for data transmission. Two possible configuration examples are as follows: Fig.16 shown.

[0263] When data transmission adopts π / 2-BPSK modulation, one of the DMRS sequences is DFT precoded pseudo noise π / 2-BPSK. Therefore, the characteristics of SE discussed for data transmission are directly extended to such DMRS sequences.

[0264] For other constellations, another DMRS sequence is the Zadoff-Chu (ZC) sequence. ZC The DMRS sequence has the following elements obtained by cyclic extension:

[0265]

[0266] The above cyclic extension is a cyclic extension of the following ZC sequence

[0267]

[0268] And the maximum prime length satisfies N ZC ≤M ZC The possible root indices are u=1,…,N ZC -1.

[0269] Note that the ZC sequence is a constant amplitude zero autocorrelation sequence (CAZAC) sequence, and the IDFT of the ZC sequence is also a CAZAC sequence. Due to this property, it can be expected that DMRS with a ZC sequence is a DFT-s-OFDM transmission, so the properties of SE discussed for data transmission are also directly extended to this DMRS sequence.

[0270] Since the FDSS window used for data transmission is usually unknown at the receiver, the DMRS sequence needs to be shaped by the same FDSS window as the data symbols. Depending on the capabilities of the receiver, three options can be considered.

[0271] Option A: The second communication device 300 as a receiver can only decode the in-band spectrum without extending it, so the DMRS sequence needs to be designed based on the in-band, that is, M ZC =N data / 2.

[0272] Option B: The second communication device 300 as a receiver can utilize superband symbol reception, and the DMRS sequence needs to be based on the entire allocated bandwidth M ZC =N sc / 2 to design.

[0273] Option C is to accommodate two types of receivers with the same DMRS design. The sequence is first designed based on the in-band, and then the sequence is spectrally extended to the entire allocated band. The DMRS sequence design becomes

[0274]

[0275] Where N ZC is to satisfy N ZC ≤N data / 2 maximum prime length. Sequences without SE start in-band, for which symmetric expansion is used, equivalent to shifting

[0276] These three options for DMRS design are as follows: Fig.17 As shown. They are in the RRC window, N sc =96 and N data = 72 under different PAPR Fig.18 shown. Fig.18 Medium data The choice corresponds to having N sc 25% SE, which is the best choice for PAPR reduction for QPSK data transmission for this FDSS window. It can be observed that Option C has the largest PAPR, but this is balanced by the fact that it produces more sequences: 31 with Option B and 23 with Options A and C. Otherwise, the SE DMRS design Option C would reduce the PAPR of Option A.

[0277] Finally, in Fig.19 In the comparison option C in N sc =96, different SE size N e =(N sc -N data) and the maximum PAPR under three different FDSS windows. Interestingly, the maximum PAPR of DMRS when using SE behaves similarly to that when using data transmission. It can be observed that the PAPR first decreases to a minimum value and then increases. The SE size that provides the lowest PAPR using DMRS is close to the SE size that minimizes the PAPR using QPSK data transmission.

[0278] The network access node in this article can also be expressed as a wireless network access node, access network access node, access point (AP) or base station (BS), for example, a radio base station (RBS), which can be called a transmitter, "gNB", "gNodeB", "eNB", "eNodeB", "NodeB" or "B node" in some networks, depending on the standards, technologies and terminology used. Based on the transmission power and cell size, wireless network access nodes can have different categories or types, such as macro eNodeB, home eNodeB or mini base station. A wireless network access node can also be a site, that is, any device that includes a media access control (MAC) and physical layer (PHY) interface to a wireless medium (WM) that complies with IEEE 802.11. The wireless network access node can be used to communicate in the following: 3GPP-related long term evolution (LTE), advanced LTE, fifth generation (5G) wireless systems, such as new radio (NR) and its evolutions, and IEEE-related Wi-Fi, worldwide interoperability for microwave access (WiMAX) and its evolutions.

[0279] The client device in this article can be represented as a user device, user equipment (UE), mobile station, Internet of Things (IoT) device, sensor device, wireless terminal and / or mobile terminal, capable of wireless communication in a wireless communication system (sometimes also referred to as a cellular wireless system). UE can be referred to as a mobile phone, cellular phone, tablet or laptop with wireless capabilities. For example, the UE in this context can be a portable, pocket storage, handheld, computer-based or vehicle-mounted mobile device that can communicate voice and / or data with another communication entity (such as another receiver or server) through a radio access network (RAN). UE can also be a site, that is, any device including a MAC and PHY interface connected to WM that complies with IEEE 802.11. UE can be used to communicate in the following: 3GPP-related LTE, Advanced LTE, 5G wireless systems, such as NR and its evolution, and in IEEE-related Wi-Fi, WiMAX and its evolution.

[0280] In addition, any method according to an embodiment of the present invention can be implemented in a computer program with a code component, which causes the processing component to perform the steps of the method when the processing component is run. The computer program is included in the computer-readable medium of the computer program product. The computer-readable medium can basically include any memory, such as the above-mentioned ROM, PROM, EPROM, flash memory, EEPROM, hard disk drive, etc.

[0281] In addition, it should be appreciated that the first communication device 100 and the second communication device 300 include necessary communication capabilities in the form of, for example, functions, components, units, elements, etc., for performing or implementing embodiments of the present invention. Examples of other such components, units, elements and functions are: processors, memories, buffers, control logic, encoders, decoders, rate matchers, de-rate matchers, mapping units, multipliers, decision units, selection units, switches, interleavers, deinterleavers, modulators, demodulators, inputs, outputs, antennas, amplifiers, receiving units, transmitting units, DSPs, TCM encoders, TCM decoders, power supply units, power feeds, communication interfaces, communication protocols, etc., which are appropriately arranged together to perform the scheme.

[0282] Therefore, the one or more processors of the first communication device 100 and the second communication device 300 may include, for example, one or more instances of a CPU, a processing unit, a processing circuit, a processor, an ASIC, a microprocessor, or other processing logic that can interpret and execute instructions. Therefore, the expression "processor" may refer to a processing circuit including a plurality of processing circuits, such as any, some, or all of the above-mentioned processing circuits. The processing circuit may also perform data processing functions for inputting, outputting, and processing data, including data buffering and device control functions, such as call processing control, user interface control, etc.

[0283] Finally, it should be understood that the present invention is not limited to the embodiments described above, but also relates to and encompasses all embodiments within the scope of the attached independent claims.

Claims

1. A first communication device (100) for a communication system (500), the first communication device (100) being used for: Based on N data Data symbols are obtained N data Fourier coefficients, where N data is a positive integer; Repeat N data N of the Fourier coefficients e Fourier coefficients to obtain N sc Fourier coefficients, where N e and N sc is a positive integer, and N e is determined based on at least one of the following: data a modulation symbol constellation of data symbols, and a frequency domain spectral shaping (FDSS) window of the first communication device (100); The N sc Fourier coefficients and size N sc The FDSS window is multiplied to obtain N sc frequency-shaping Fourier coefficients; The N sc The frequency-shaping Fourier coefficients are mapped to N sc subcarriers to obtain a discrete Fourier transform spread orthogonal frequency division multiplexing (DFT-s-OFDM) signal (510); The DFT-s-OFDM signal is transmitted (510).

2. The first communication device (100) according to claim 1, wherein: N e Is also based on N sc or N data Sure.

3. The first communication device (100) according to claim 1 or 2, wherein: The N data The modulation symbol constellation of the data symbols is a π / 2-BPSK constellation or a QAM constellation.

4. The first communication device (100) according to any one of the preceding claims, wherein: N e is predetermined; or The first communication device (100) is used to receive an indication N e The first control signal (520) is provided.

5. The first communication device (100) according to any one of the preceding claims, wherein: The N e The Fourier coefficients are repeated: included in resources allocated for transmission of the DFT-s-OFDM signal (510); or added to the resources allocated for transmitting the DFT-s-OFDM signal (510).

6. The first communication device (100) according to any one of the preceding claims, wherein: The first communication device (100) is used for: Send a second control signal (530), wherein the second control signal (530) indicates that the first communication device (100) repeats the N e Fourier coefficients.

7. The first communication device (100) according to any one of the preceding claims, wherein: In the case of sc Before multiplying the Fourier coefficients by the FDSS window, the first communication device (100) is used to: The N sc Fourier coefficients are cyclically shifted to obtain N sc cyclically shifted Fourier coefficients, where L is a positive integer; The N sc The cyclically shifted Fourier coefficients are related to the size N sc The FDSS window is multiplied to obtain N sc Fourier coefficients of frequency shaping and cyclic shifting; The N sc The frequency-shaped and cyclically shifted Fourier coefficients are mapped to the N sc subcarriers to obtain the DFT-s-OFDM signal (510).

8. The first communication device (100) according to claim 7, wherein: L is predetermined; or The first communication device (100) is used to receive a first control signal (520) indicating L.

9. The first communication device (100) according to claim 7 or 8, wherein: L is determined based on either: N e 、N data or N sc ; The N data A modulation symbol constellation for data symbols; formula or or or or or Where k is a positive integer, round(x) gives the integer closest to x. and They are the ceiling and floor operators for x respectively.

10. The first communication device (100) according to any one of the preceding claims, wherein: Less than the maximum allowed Fourier coefficient repetition capability of the first communication device (100) and is based on the FDSS window, L, and N sc and N data Any one of the items is determined.

11. The first communication device (100) according to claim 10, wherein: The first communication device (100) is used for: Based on the FDSS window, the N data The modulation symbol constellation of the data symbols, L, and N sc or N data Any one of the To minimize the peak-to-average power ratio (PAPR) of transmitting the DFT-s-OFDM signal (510).

12. The first communication device (100) according to claim 10 or 11, wherein: The first communication device (100) is used for: Send Instructions The second control signal (530) is provided.

13. A second communication device (300) for a communication system (500), wherein: The second communication device (300) is used for: A DFT-s-OFDM signal (510) is received from a first communication device (100), wherein the DFT-s-OFDM signal (510) includes a plurality of pixels mapped to N sc N subcarriers sc Fourier coefficients, where N sc is a positive integer; The N is obtained based on the DFT-s-OFDM signal (510). sc Fourier coefficients, the N sc Fourier coefficients include N data Fourier coefficients and the N data N of the Fourier coefficients e Repeated Fourier coefficients, where N data and N e is a positive integer; data The Fourier coefficients are based on N data data symbols are obtained, and N e is determined based on at least one of the following: data a modulation symbol constellation of data symbols, and a FDSS window of the first communication device (100); Based on the N sc Fourier coefficients to obtain the N data Fourier coefficients; For the N data Fourier coefficients are decoded to obtain the N data data symbol.

14. The second communication device (300) according to claim 13, wherein: N e Is also based on N sc or N data Sure.

15. The second communication device (300) according to claim 13 or 14, wherein: The N data The modulation symbol constellation of the data symbols is a π / 2-BPSK constellation or a QAM constellation.

16. The second communication device (300) according to any one of claims 13 to 15, wherein: N e is predetermined; or The second communication device (100) is used to send an indication N e The first control signal (520) is provided.

17. The second communication device (300) according to any one of claims 13 to 16, wherein: The N e The Fourier coefficients are repeated: included in resources allocated for transmission of the DFT-s-OFDM signal (510); or added to the resources allocated for transmitting the DFT-s-OFDM signal (510).

18. The second communication device (300) according to any one of claims 13 to 17, wherein: The second communication device (300) is used for: Receive a second control signal (530), the second control signal (530) indicating that the first communication device (100) repeats the N e Fourier coefficients.

19. The second communication device (300) according to any one of claims 13 to 18, wherein: In obtaining the N data Fourier coefficients before, the second communication device (300) is used for: The N sc Fourier coefficients are cyclically shifted to obtain N sc cyclically shifted Fourier coefficients, where L is a positive integer; Based on the N sc The N cyclically shifted Fourier coefficients are obtained data Fourier coefficients.

20. The second communication device (300) according to claim 19, wherein: L is predetermined; or The second communication device (300) is used to send a second control signal (530) indicating L.

21. The second communication device (300) according to claim 19 or 20, wherein: L is determined based on either: N e 、N data or N sc ; The N data A modulation symbol constellation for data symbols; formula or or or or or Where k is a positive integer, round(x) gives the integer closest to x. and They are the ceiling and floor operators for x respectively.

22. The second communication device (300) according to any one of claims 13 to 21, wherein: Less than the maximum allowed Fourier coefficient repetition capability of the first communication device (100) and is based on the FDSS window, L, and N sc and N data Any one of the items is determined.

23. The second communication device (300) according to claim 22, wherein: Based on the FDSS window, the N data The modulation symbol constellation of the data symbols, L, and N sc or N data Any of In order to minimize the PAPR of transmitting the DFT-s-OFDM signal (510).

24. The second communication device (300) according to claim 22 or 23, wherein: The second communication device (300) is used for: Receiving instructions The second control signal (530) is provided.

25. A method (200) for a first communication device (100), wherein: The method (200) comprises: Based on N data Data symbol acquisition (202) N data Fourier coefficients, where N data is a positive integer; Repeat (204) N data N of the Fourier coefficients e Fourier coefficients to obtain N sc Fourier coefficients, where N e and N sc is a positive integer, and N e is determined based on at least one of the following: data a modulation symbol constellation of data symbols, and a FDSS window of the first communication device (100); The N sc Fourier coefficients and size N sc The FDSS window is multiplied by (206) to obtain N sc frequency-shaping Fourier coefficients; The N sc The frequency-shaped Fourier coefficients are mapped (208) to N sc subcarriers to obtain a DFT-s-OFDM signal (510); The DFT-s-OFDM signal (510) is transmitted (210).

26. A method (400) for a second communication device (300), wherein: The method (400) comprises: A DFT-s-OFDM signal (510) is received (402) from a first communication device (100), the DFT-s-OFDM signal (510) comprising a plurality of pixels mapped to N sc N subcarriers sc Fourier coefficients, where N sc is a positive integer; The N is obtained (404) based on the DFT-s-OFDM signal (510). sc Fourier coefficients, the N sc Fourier coefficients include N data Fourier coefficients and the N data N of the Fourier coefficients e Repeated Fourier coefficients, where N data and N e is a positive integer; data The Fourier coefficients are based on N data data symbols are obtained, and N e is determined based on at least one of the following: data a modulation symbol constellation of data symbols, and a FDSS window of the first communication device (100); Based on the N sc Fourier coefficients are obtained (406) data Fourier coefficients; For the N data Fourier coefficients are decoded (408) to obtain the N data data symbol.

27. A computer program comprising program code, wherein: When the computer program is run on a computer, the program code is used to execute the method according to claim 25 or 26.

Images