DFT-s-ofdm signal with spectrum spreading

Abstract

Embodiments of the present disclosure relate to a DFT-s-OFDM signal with spectral spreading. A first communication device (100) obtains N data Fourier coefficients based on N data data symbols, and repeats N data of the N e Fourier coefficients to obtain N sc Fourier coefficients, wherein N e is determined based on at least one of a modulation symbol constellation of the N data data symbols and an FDSS window of the first communication device (100); multiplies the N sc Fourier coefficients with an FDSS window of size N sc to obtain N sc frequency-shaped Fourier coefficients mapped onto N sc subcarriers to obtain a transmitted DFT-s-OFDM signal (510). Thus, the DFT-s-OFDM signal can be transmitted with a lower PAPR. Furthermore, embodiments of the present disclosure also relate to a corresponding second communication device (300), a corresponding method and a computer program.

Description

Technical Field

[0001] Embodiments of this disclosure relate to a first communication device and a second communication device for DFT-s-OFDM signals with spectral spread. Furthermore, embodiments of this disclosure also relate to a corresponding method and a computer program. Background Technology

[0002] Due to the limited transmission power of user equipment (UE), uplink transmission is considered a coverage bottleneck in modern wireless systems. Therefore, enhancing uplink coverage is a crucial issue in the current 3GPP New Radio (NR) standardization. Under limited coverage conditions, the envelope ripple of the transmitted signal (typically measured by its cubic metric (CM) or peak-to-average power ratio (PAPR)) should be minimized. This is because signals with high CM / PAPR require power amplifiers to operate with greater power back-off to avoid signal distortion caused by their non-linear amplification range, thus directly impacting the transmission link budget.

[0003] Orthogonal frequency division multiplexing (OFDM) is the dominant modulation waveform in wireless communication systems to date, and it is also the fundamental waveform in the uplink and downlink of the 3GPP standard. Discrete Fourier transform spread OFDM (DFT-s-OFDM) is a variant of OFDM, primarily used to achieve a lower PAPR (PAPR) compared to standard OFDM. In DFT-s-OFDM, data symbols are first pre-coded using the DFT before OFDM modulation, resulting in a single-carrier waveform, because the energy of the subsequently modulated symbols is distributed across all subcarriers in the allocated spectrum. Conversely, in the time domain, most of the symbol energy is concentrated in shorter intervals, resulting in time-domain multiplexing of the symbols.

[0004] While DFT-s-OFDM can improve PAPR, the results may still be unsatisfactory, for example, in scenarios with limited indoor depth coverage. Therefore, additional low PAPR techniques have been considered in 3GPP for uplink transmission. For example, in addition to traditional LTE constellations, 5G NR supports π / 2-BPSK in the uplink, aiming to further reduce PAPR and thus improve the power efficiency of radio frequency (RF) amplifiers at lower data rates. In current 5G NR, π / 2-BPSK can also be used in conjunction with frequency domain spectrum shaping (FDSS). FDSS can further reduce PAPR at the cost of manageable self-interference. In fact, the NR specification does not define a specific FDSS window; its use is indirectly permitted through the more lenient RAN4 spectrum flatness requirements for NR uplinks employing π / 2-BPSK.

[0005] Spectrum extension (SE) with FDSS is another technique for reducing PAPR. It is not currently specified in NR but is listed as a potential solution. The main goal is also to achieve lower PAPR for higher-order modulation (such as QPSK), thereby improving coverage at higher data rates. Summary of the Invention

[0006] The purpose of the embodiments disclosed herein is to provide a solution to reduce or solve the disadvantages and problems of conventional solutions.

[0007] Another object of embodiments of this disclosure is to provide a solution that can reduce PAPR compared to conventional solutions.

[0008] The foregoing and other objectives are achieved through the subject matter of the independent claims. Further embodiments of this disclosure can be found in the dependent claims.

[0009] According to a first aspect of this disclosure, the above and other objectives are achieved by a first communication device for a communication system, the first communication device being used for:

[0010] Based on N data Get N data symbols data Fourier coefficients, where N data It is a positive integer;

[0011] Repeat the N data N in the Fourier coefficients e Fourier coefficients are used to obtain N sc Fourier coefficients, where N e and N scN is a positive integer, and N is a positive integer. e It is determined based on at least one of the following: the N data The modulation symbol constellation of each data symbol, and the frequency domain spectrum shaping (FDSS) window of the first communication device;

[0012] The N sc The Fourier coefficients are of size N. sc Multiply by the FDSS window to obtain N sc Frequency-adjusted Fourier coefficients;

[0013] The N sc Each frequency-shaped Fourier coefficient is mapped to N. sc Discrete Fourier transform spread orthogonal frequency division multiplexing (DFT-s-OFDM) signals are obtained on each subcarrier;

[0014] The DFT-s-OFDM signal is sent.

[0015] According to the first aspect, the advantage of the first communication device is that it can reduce the PAPR of the DFT-s-OFDM signal. This is because the optimal amount of PAPR reduction and the N that achieves this goal... e The value depends largely on the modulation symbol constellation and FDSS window used for the data symbols. Therefore, the first communication device, as the transmitter, can use lower power back-off, thereby increasing the coverage of the DFT-s-OFDM signal.

[0016] In one implementation of the first communication device according to the first aspect, it is also based on N sc or N data Determine N e .

[0017] The advantage of this implementation is that it further reduces the PAPR of the DFT-s-OFDM signal based on bandwidth allocation. This is because N achieves the optimal reduction in PAPR. e The value increases roughly proportionally with bandwidth allocation, but varies slightly.

[0018] In one implementation of the first communication device according to the first aspect, the N data The modulation symbol constellation for each data symbol is either a π / 2-BPSK constellation or a QAM constellation.

[0019] The advantage of this implementation is that it allows the use of such a constellation in the 3GPP standard, where π / 2-BPSK is specifically designed for very low PAPR transmissions, and it can operate through N constellations that are very different from QAM constellations. e The value needs further improvement.

[0020] In one implementation of the first communication device according to the first aspect,

[0021] N e It is predetermined; or

[0022] The first communication device is used to receive indication N e The first control signal.

[0023] The advantage of this implementation method is that it has a predetermined N. e It is possible to easily minimize signaling overhead; or if N is indicated via signaling. e This allows for better reduction of PAPR based on other transmission parameters or device-specific characteristics.

[0024] In one implementation of the first communication device according to the first aspect, the N e The Fourier coefficients are repeatedly:

[0025] Included in the allocated resources for transmitting the DFT-s-OFDM signal; or

[0026] Add to the allocated resources for transmitting the DFT-s-OFDM signal.

[0027] The advantage of this implementation is that for multiple adjacent user bandwidths, if repeated Fourier coefficients are included in the allocated resources for each user, interference is avoided, and repeated symbols can be used to improve demodulation performance; or if repeated Fourier coefficients are added to the allocated resources, spectral efficiency is improved.

[0028] In one implementation of the first communication device according to the first aspect, the first communication device is used for:

[0029] Send a second control signal, wherein the second control signal instructs the first communication device to repeat the N. e The ability to generate Fourier coefficients.

[0030] The advantage of this implementation is that the N indicated by the first control signal can be determined based on the capability of the first communication device indicated by the second control signal. e This allows for optimization and reduction of PAPR based on several configuration parameters and implementation aspects of the first communication device.

[0031] In one implementation of the first communication device according to the first aspect, when the N sc Before multiplying the Fourier coefficients by the FDSS window, the first communication device is used to:

[0032] Using L Fourier coefficients to represent the N sc The Fourier coefficients are cyclically shifted to obtain N. sc Fourier coefficients of cyclic shift, where L is a positive integer;

[0033] The N sc The Fourier coefficients of each cyclic shift and the value of N sc Multiply by the FDSS window to obtain N sc Fourier coefficients for frequency shaping and cyclic shifting;

[0034] The N sc The Fourier coefficients of frequency shaping and cyclic shifting are mapped to the N sc The DFT-s-OFDM signal is obtained on each subcarrier.

[0035] The advantage of this implementation is that it can further reduce PAPR when PAPR varies as a function of the cyclic shift coefficient L.

[0036] In one implementation of the first communication device according to the first aspect,

[0037] L is predetermined; or

[0038] The first communication device is used to receive a first control signal indicating L.

[0039] The advantage of this implementation is that having a predetermined L makes it easy to minimize signaling overhead; or, if L is indicated by signaling, it makes it possible to better reduce PAPR based on other transmission parameters or device-specific characteristics.

[0040] In one implementation of the first communication device according to the first aspect, L is determined based on any of the following:

[0041] N e N data or N sc ;

[0042] The N data A modulation symbol constellation of data symbols;

[0043] formula or or or or or Where k is a positive integer, and round(x) gives the integer closest to x. and These are the floor operations for x: floor up and floor down.

[0044] The advantage of this implementation is that it can optimally reduce PAPR by selecting the best shift function.

[0045] In one implementation of the first communication device according to the first aspect, Less than the maximum permissible Fourier coefficient repeatability of the first communication device (100) and Based on the FDSS window, L, and N sc and N data Any one of them is determined.

[0046] The advantage of this implementation method is that it allows for the selection of a smaller N. e To improve spectral efficiency and further avoid selecting an excessively large N that might increase PAPR. e .

[0047] In one implementation of the first communication device according to the first aspect, the first communication device is used for:

[0048] Based on the FDSS window, the N data The modulation symbol constellation of each data symbol, L, and N. sc or N data Determine any one of the following: To minimize the peak-to-average-power ratio (PAPR) of the transmitted DFT-s-OFDM signal.

[0049] The advantage of this implementation is that it allows selection of N to optimize PAPR performance based on transmission parameters or specific device characteristics. e .

[0050] In one implementation of the first communication device according to the first aspect, the first communication device is used for:

[0051] Send Instruction The second control signal.

[0052] The advantage of this implementation is that it ensures the selection of N based on the specific capabilities and implementation of the first communication device. e It will not have an adverse effect on PAPR.

[0053] According to a second aspect of this disclosure, the above and other objectives are achieved by a second communication device for a communication system, the second communication device being used for:

[0054] Receive a DFT-s-OFDM signal from a first communication device, the DFT-s-OFDM signal including a mapping to N sc N on each subcarrier sc Fourier coefficients, where N sc It is a positive integer;

[0055] Based on the DFT-s-OFDM signal, obtain the N sc The N Fourier coefficients sc The Fourier coefficients include N data The Fourier coefficients and the N data N in the Fourier coefficients e N repeated Fourier coefficients, where N data and N e N is a positive integer; data The Fourier coefficients are based on N. data The data symbols were obtained, and N e It is determined based on at least one of the following: the N data The modulation symbol constellation of the data symbols, and the FDSS window of the first communication device;

[0056] Based on the N sc The N is obtained from the Fourier coefficients. data Fourier coefficients;

[0057] For the N data Decode the Fourier coefficients to obtain the N. data Data symbols.

[0058] According to the second aspect, the advantage of the second communication device is that the first communication device is able to reduce the PAPR of the DFT-s-OFDM signal. This is because the optimal amount of PAPR reduction and the N that achieves this goal... e The value depends largely on the modulation symbol constellation and FDSS window used for the data symbols. Therefore, the first communication device, as the transmitter, can use lower power back-off, thereby increasing the coverage of the DFT-s-OFDM signal.

[0059] In one implementation of the second communication device according to the second aspect, it is also based on N sc or N data Determine N e .

[0060] The advantage of this implementation is that it further reduces the PAPR of the DFT-s-OFDM signal based on bandwidth allocation. This is because N achieves the optimal reduction in PAPR. e The value increases roughly proportionally with bandwidth allocation, but varies slightly.

[0061] In one implementation of the second communication device according to the second aspect, the N data The modulation symbol constellation for each data symbol is either a π / 2-BPSK constellation or a QAM constellation.

[0062] The advantage of this implementation is that it allows the use of such a constellation in the 3GPP standard, where π / 2-BPSK is specifically designed for very low PAPR transmissions, and it can operate through N constellations that are very different from QAM constellations. e The value needs further improvement.

[0063] In one implementation of the second communication device according to the second aspect,

[0064] N e It is predetermined; or

[0065] The second communication device is used to send indication N e The first control signal.

[0066] The advantage of this implementation method is that it has a predetermined N. e It is possible to easily minimize signaling overhead; or if N is indicated via signaling. e This allows for better reduction of PAPR based on other transmission parameters or device-specific characteristics.

[0067] In one implementation of the second communication device according to the second aspect, the N e The Fourier coefficients are repeatedly:

[0068] Included in the allocated resources for transmitting the DFT-s-OFDM signal; or

[0069] Add to the allocated resources for transmitting the DFT-s-OFDM signal.

[0070] The advantage of this implementation is that for multiple adjacent user bandwidths, if repeated Fourier coefficients are included in the allocated resources for each user, interference is avoided, and repeated symbols can be used to improve demodulation performance; or if repeated Fourier coefficients are added to the allocated resources, spectral efficiency is improved.

[0071] In one implementation of the second communication device according to the second aspect, the second communication device is used for:

[0072] Receive a second control signal, wherein the second control signal instructs the first communication device to repeat the N. e The ability to generate Fourier coefficients.

[0073] The advantage of this implementation is that the N indicated by the first control signal can be determined based on the capability of the first communication device indicated by the second control signal. e This allows for optimization and reduction of PAPR based on several configuration parameters and implementation aspects of the first communication device.

[0074] In one implementation of the second communication device according to the second aspect, when acquiring the N data Before the Fourier coefficients, the second communication device is used for:

[0075] Using L Fourier coefficients to represent the N sc The Fourier coefficients are cyclically shifted to obtain N. sc Fourier coefficients of cyclic shift, where L is a positive integer;

[0076] Based on the N sc The Fourier coefficients of the cyclic shift are used to obtain the N. data Fourier coefficients.

[0077] The advantage of this implementation is that it can further reduce PAPR when PAPR varies as a function of the cyclic shift coefficient L.

[0078] In one implementation of the second communication device according to the second aspect,

[0079] L is predetermined; or

[0080] The second communication device is used to send a second control signal indicating L.

[0081] The advantage of this implementation is that having a predetermined L makes it easy to minimize signaling overhead; or, if L is indicated by signaling, it makes it possible to better reduce PAPR based on other transmission parameters or device-specific characteristics.

[0082] In one implementation of the second communication device according to the second aspect, L is determined based on any of the following:

[0083] N e N data or N sc ;

[0084] The N data A modulation symbol constellation of data symbols;

[0085] formula or or or or or Where k is a positive integer, and round(x) gives the integer closest to x. and These are the floor operations for x: floor up and floor down.

[0086] The advantage of this implementation is that it can optimally reduce PAPR by selecting the best shift function.

[0087] In one implementation of the second communication device according to the second aspect, Less than the maximum permissible Fourier coefficient repeatability of the first communication device (100) and Based on the FDSS window, L, and N sc and N data Any one of them is determined.

[0088] The advantage of this implementation method is that it allows for the selection of a smaller N. e To improve spectral efficiency and further avoid selecting an excessively large N that might increase PAPR. e .

[0089] In one implementation of the second communication device according to the second aspect, the first communication device is based on the FDSS window and the N data The modulation symbol constellation of each data symbol, L, and N. sc or N data Determine any one of the following: To minimize PAPR of the transmitted DFT-s-OFDM signal.

[0090] The advantage of this implementation is that it allows selection of N to optimize PAPR performance based on transmission parameters or specific device characteristics. e .

[0091] In one implementation of the second communication device according to the second aspect, the second communication device is used for:

[0092] Receive Instruction The second control signal.

[0093] The advantage of this implementation is that it ensures the selection of N based on the specific capabilities and implementation of the first communication device. e It will not reduce the adverse effects on PAPR.

[0094] According to a third aspect of this disclosure, the above and other objectives are achieved by a method for a first communication device, the method comprising:

[0095] Based on N data Get N data symbols data Fourier coefficients, where N data It is a positive integer;

[0096] Repeat the N data N in the Fourier coefficients e Fourier coefficients are used to obtain N sc Fourier coefficients, where N e and N sc N is a positive integer, and N is a positive integer. e It is determined based on at least one of the following: the N data The modulation symbol constellation of the data symbols, and the FDSS window of the first communication device;

[0097] The N sc The Fourier coefficients are of size N. sc Multiply by the FDSS window to obtain N sc Frequency-adjusted Fourier coefficients;

[0098] The N sc Each frequency-shaped Fourier coefficient is mapped to N. sc DFT-s-OFDM signals are acquired on each subcarrier;

[0099] The DFT-s-OFDM signal is sent.

[0100] The method provided according to the third aspect can be extended to an implementation corresponding to the implementation of the first communication device provided according to the first aspect. Therefore, the implementation of the method includes one or more features of the corresponding implementation of the first communication device.

[0101] The advantages of the method provided according to the third aspect are the same as the advantages of the corresponding implementation of the first communication device provided according to the first aspect.

[0102] According to a fourth aspect of this disclosure, the above and other objectives are achieved by a method for a second communication device, the method comprising:

[0103] Receive a DFT-s-OFDM signal from a first communication device, the DFT-s-OFDM signal including a mapping to N sc N on each subcarrier sc Fourier coefficients, where N sc It is a positive integer;

[0104] Based on the DFT-s-OFDM signal, obtain the N sc The N Fourier coefficientssc The Fourier coefficients include N data The Fourier coefficients and the N data N in the Fourier coefficients e N repeated Fourier coefficients, where N data and N e N is a positive integer; data The Fourier coefficients are based on N. data The data symbols were obtained, and N e It is determined based on at least one of the following: the N data The modulation symbol constellation of the data symbols, and the FDSS window of the first communication device;

[0105] Based on the N sc The N is obtained from the Fourier coefficients. data Fourier coefficients;

[0106] For the N data Decode the Fourier coefficients to obtain the N. data Data symbols.

[0107] The method provided according to the fourth aspect can be extended to an implementation corresponding to the implementation of the second communication device provided according to the second aspect. Therefore, the implementation of the method includes one or more features of the corresponding implementation of the second communication device.

[0108] The advantages of the method provided according to the fourth aspect are the same as the advantages of the corresponding implementation of the second communication device provided according to the second aspect.

[0109] Embodiments of this disclosure also relate to a computer program characterized by program code that, when executed by at least one processor, causes the at least one processor to perform any of the methods provided according to embodiments of this disclosure. Furthermore, embodiments of this disclosure also relate to a computer program product comprising a computer-readable medium and the computer program therein, wherein the computer program is contained in the computer-readable medium and may include one or more of the following: read-only memory (ROM), programmable ROM (PROM), erasable PROM (EPROM), flash memory, electrically erasable PROM (EEPROM), hard disk drive, etc.

[0110] Other applications and advantages of the embodiments of this disclosure will become apparent from the following detailed description. Attached Figure Description

[0111] The accompanying drawings are intended to illustrate and explain different embodiments of this disclosure, in which:

[0112] Figure 1 A first communication device according to an embodiment of the present disclosure is shown;

[0113] Figure 2 A flowchart of a method for a first communication device according to an embodiment of the present disclosure is shown;

[0114] Figure 3 A second communication device according to an embodiment of the present disclosure is shown;

[0115] Figure 4 A flowchart of a method for a second communication device according to an embodiment of the present disclosure is shown;

[0116] Figure 5 A communication system according to an embodiment of the present disclosure is shown;

[0117] Figure 6 An example of a first implementation of a first communication device having FDSS and a spectrum spread including a shift parameter L is shown;

[0118] Figure 7 An example of how the second communication device is implemented is shown;

[0119] Figure 8 shows the N value under different constellations, FDSS windows, and L values. e The 99th percentile PAPR[dB] of the function;

[0120] Figure 9 It shows that at a fixed N data When = 96, as N e The 99th percentile PAPR[dB] of the QPSK function;

[0121] Figure 10 shows the optimal number of REs for spectrum spread. The number N of resource blocks that constitute the total allocated bandwidth RB =N sc A function of / 12 that minimizes PAPR;

[0122] Figure 11 The optimal number of REs for spectrum spread is shown. With N sc Compared to the fixed case, in N data Minimize the PAPR of QPSK under fixed conditions;

[0123] Figure 12 Some signaling aspects of embodiments of this disclosure are illustrated;

[0124] Figure 13 The diagram shows a spectral spread data sequence, which is a function of the shift parameter L, where N data =10 symbols and N e =4;

[0125] Figure 14 An example of a second implementation of a first communication device with FDSS and a spectrum spread including a shift parameter L is shown;

[0126] Figure 15 A third implementation example of a first communication device with FDSS and periodic spectrum spread including shift parameter L is shown;

[0127] Figure 16 The DMRS subcarrier positions of DFT-s-OFDM are shown;

[0128] Figure 17 The RE allocation with three different DMRS design options is shown;

[0129] Figure 18 The CCDF of PAPR with three different options for DMRS design is shown, where N data =72, N sc =96 and RRC window (ρ=0.5, β=-0.65);

[0130] Figure 19 The maximum PAPR of option C is shown, which is a function of the SE size, with three different FDSS windows. Detailed Implementation

[0131] Currently, in 3GPP NR, RAN4 requires limiting distortion caused by the FDSS window, and FDSS is implicitly supported only for π / 2-BPSK uplink transmissions. Furthermore, in the current NR, the FDSS window is proprietary and may differ for each UE. Therefore, the SE cannot be pre-determined for any user equipment (UE), otherwise the SE size for a particular UE may be too large and degrade PAPR performance. Meanwhile, resource allocation and signal configuration for each UE are typically scheduled by the base station (BS).

[0132] Therefore, embodiments of this disclosure relate to a first communication device 100 and a second communication device 300, solving the problem that conventional solutions cannot provide minimum PAPR because the size of the SE is constant or unoptimized relative to PAPR performance and UE-specific transmission configuration and capabilities. Furthermore, it also solves the problem in conventional solutions where Fourier coefficient shifting is not chosen to minimize PAPR.

[0133] Figure 1 A first communication device 100 according to an embodiment of the present disclosure is shown. Figure 1 In the illustrated embodiment, the first communication device 100 includes a processor 102, a transceiver 104, and a memory 106. The processor 102 is coupled to the transceiver 104 and the memory 106 via a communication component 108 known in the art. The first communication device 100 can be used for wireless and / or wired communication in a communication system. Wireless communication capability can be provided by an antenna or antenna array 110 coupled to the transceiver 104, while wired communication capability can be provided by a wired communication interface 112, etc., coupled to the transceiver 104.

[0134] Processor 102 may be referred to as one or more general-purpose central processing units (CPUs), one or more digital signal processors (DSPs), one or more application-specific integrated circuits (ASICs), one or more field-programmable gate arrays (FPGAs), one or more programmable logic devices, one or more discrete gates, one or more transistor logic devices, one or more discrete hardware components, or one or more chipsets. Memory 106 may be read-only memory, random access memory (RAM), or non-volatile RAM (NVRAM). Transceiver 104 may be transceiver circuitry, a power controller, or an interface providing the ability to communicate with other communication modules or communication devices (e.g., network nodes and network servers). Transceiver 104, memory 106, and / or processor 102 may be implemented in separate chipsets or in a common chipset. In this disclosure, the first communication device 100 performing certain actions can be understood to mean that the first communication device 100 includes suitable components for performing said actions, such as processor 102 and transceiver 104.

[0135] According to embodiments of this disclosure, the first communication device 100 is used for N-based communication. data Get N data symbols data Fourier coefficients, where N data It is a positive integer. The first communication device 100 is also used to repeat N. data N in the Fourier coefficients e Fourier coefficients are used to obtain N sc Fourier coefficients, where N e and Nsc N is a positive integer, and N is a positive integer. e It is determined based on at least one of the following: N data The first communication device 100 is also used to configure the modulation symbol constellation of N data symbols and the frequency domain spectrum shaping (FDSS) window of the first communication device 100. sc The Fourier coefficients are of size N. sc Multiply the FDSS window to obtain L sc The frequency-shaped Fourier coefficients. The first communication device 100 is also used to convert N... sc Each frequency-shaped Fourier coefficient is mapped to N. sc The first communication device 100 is also used to transmit the DFT-s-OFDM signal 510 on each subcarrier.

[0136] Furthermore, in embodiments of this disclosure, the first communication device 100 for the communication system 500 includes: a processor for N-based communication... data Get N data symbols data There are N Fourier coefficients, where N is the number of coefficients. data Positive integers; repeat N data N in the Fourier coefficients e Fourier coefficients are used to obtain N sc Fourier coefficients, where N e and N sc N is a positive integer, and N is a positive integer. e It is determined based on at least one of the following: N data The modulation symbol constellation of N data symbols, and the FDSS window of the first communication device 100; sc The Fourier coefficients are of size N. sc Multiply by the FDSS window to obtain N sc Number of frequency-shaping Fourier coefficients; N sc Each frequency-shaped Fourier coefficient is mapped to N. sc 510 is used to acquire DFT-s-OFDM signals on each subcarrier; a transceiver is used to transmit DFT-s-OFDM signals 510.

[0137] Furthermore, in another embodiment of this disclosure, a first communication device 100 for a communication system 500 includes a processor and a memory thereon storing computer-readable instructions, which, when executed by the processor, cause the processor to: based on N data Get N data symbols data There are N Fourier coefficients, where N is the number of coefficients. data Positive integers; repeat N data N in the Fourier coefficients e Fourier coefficients are used to obtain N sc Fourier coefficients, where N e and N sc N is a positive integer, and N is a positive integer. e It is determined based on at least one of the following: N data The modulation symbol constellation of N data symbols, and the FDSS window of the first communication device 100; sc The Fourier coefficients are of size N. sc Multiply by the FDSS window to obtain N sc Number of frequency-shaping Fourier coefficients; N sc Each frequency-shaped Fourier coefficient is mapped to N. sc Acquire DFT-s-OFDM signal 510 on each subcarrier; transmit DFT-s-OFDM signal 510.

[0138] Figure 2 This illustrates that it can be implemented in the first communication device 100 (e.g., Figure 1 A flowchart illustrating the corresponding method 200 executed in the first communication device 100 shown. Method 200 includes N-based... data The data symbols are obtained as (202)N data Fourier coefficients, where N data It is a positive integer. Method 200 also includes repeating (204)N. data N in the Fourier coefficients e Fourier coefficients are used to obtain N sc Fourier coefficients, where N e and N sc N is a positive integer, and N is a positive integer. e It is determined based on at least one of the following: N data The method 200 also includes a modulation symbol constellation of N data symbols and the FDSS window of the first communication device 100. sc The Fourier coefficients are of size N. sc Multiply the FDSS windows by (206) to obtain N sc The method also includes shaping the N frequency Fourier coefficients. sc The frequency-shaped Fourier coefficients are mapped (208) to N. scThe method 200 also includes transmitting (210) the DFT-s-OFDM signal 510 on each subcarrier.

[0139] Figure 3 A second communication device 300 according to an embodiment of the present disclosure is shown. Figure 3 In the illustrated embodiment, the second communication device 300 includes a processor 302, a transceiver 304, and a memory 306. The processor 302 is coupled to the transceiver 304 and the memory 306 via a communication component 308 known in the art. The second communication device 300 also includes an antenna or antenna array 310 coupled to the transceiver 304, indicating that the second communication device 300 can be used for wireless communication in a communication system.

[0140] Processor 302 may be referred to as one or more general-purpose CPUs, one or more DSPs, one or more ASICs, one or more FPGAs, one or more programmable logic devices, one or more discrete gates, one or more transistor logic devices, one or more discrete hardware components, and one or more chipsets. Memory 306 may be read-only memory, RAM, or NVRAM. Transceiver 304 may be transceiver circuitry, a power controller, or an interface providing the ability to communicate with other communication modules or communication devices. Transceiver 304, memory 306, and / or processor 302 may be implemented in separate chipsets or in a common chipset. In this disclosure, the second communication device 300 for performing certain actions can be understood to mean that the second communication device 300 includes suitable components for performing said actions, such as processor 302 and transceiver 304.

[0141] According to embodiments of this disclosure, a second communication device 300 is configured to receive a DFT-s-OFDM signal 510 from a first communication device 100, the DFT-s-OFDM signal 510 including signals mapped to N... sc N on each subcarrier sc Fourier coefficients, where N sc It is a positive integer. The second communication device 300 is also used to acquire N based on the DFT-s-OFDM signal 510. sc Fourier coefficients, N sc The Fourier coefficients include N data Fourier coefficients and N data N in the Fourier coefficients e N repeated Fourier coefficients, where N data and N e N is a positive integer; data The Fourier coefficients are based on N. data The data symbols were obtained, and N eIt is determined based on at least one of the following: N data The second communication device 300 is also used for N-based modulation symbol constellations and the FDSS window of the first communication device 100. sc Obtain N Fourier coefficients data Fourier coefficients. The second communication device 300 is also used for N. data Decode the Fourier coefficients to obtain N. data Data symbols.

[0142] Furthermore, in embodiments of this disclosure, a second communication device 300 for the communication system 500 includes a transceiver for receiving a DFT-s-OFDM signal 510 from the first communication device 100, the DFT-s-OFDM signal 510 including signals mapped to N... sc N on each subcarrier sc Fourier coefficients, where N sc A positive integer; a processor used to: acquire N based on DFT-s-OFDM signals 510 sc Fourier coefficients, N sc The Fourier coefficients include N data Fourier coefficients and N data N in the Fourier coefficients e N repeated Fourier coefficients, where N data and N e N is a positive integer; data The Fourier coefficients are based on N. data The data symbols were obtained, and N e It is determined based on at least one of the following: N data A modulation symbol constellation of data symbols, and the FDSS window of the first communication device 100; based on N sc Obtain N Fourier coefficients data Fourier coefficients; for N data Decode the Fourier coefficients to obtain N. data Data symbols.

[0143] Furthermore, in another embodiment of this disclosure, a second communication device 300 for the communication system 500 includes a processor and a memory storing computer-readable instructions thereon, which, when executed by the processor, cause the processor to: receive a DFT-s-OFDM signal 510 from the first communication device 100, the DFT-s-OFDM signal 510 including a mapping on N sc N on each subcarrier sc Fourier coefficients, where N sc N is a positive integer; N is obtained based on the DFT-s-OFDM signal 510. sc Fourier coefficients, Nsc The Fourier coefficients include N data Fourier coefficients and N data N Fourier coefficients e N repeated Fourier coefficients, where N data and N e N is a positive integer; data The Fourier coefficients are based on N. data The data symbols were obtained, and N e It is determined based on at least one of the following: N data A modulation symbol constellation of data symbols, and the FDSS window of the first communication device 100; based on N sc Obtain N Fourier coefficients data Fourier coefficients; for N data Decode the Fourier coefficients to obtain N. data Data symbols.

[0144] Figure 4 This illustrates that it can be used in a second communication device 300 (such as...) Figure 3 The flowchart illustrates the corresponding method 400 performed in the second communication device 300 shown. Method 400 includes receiving (402) a DFT-s-OFDM signal 510 from the first communication device 100, the DFT-s-OFDM signal 510 including signals mapped to N... sc N on each subcarrier sc Fourier coefficients, where N sc It is a positive integer. Method 400 also includes obtaining (404)N based on the DFT-s-OFDM signal 510. sc Fourier coefficients, N sc The Fourier coefficients include N data Fourier coefficients and N data N in the Fourier coefficients e N repeated Fourier coefficients, where N data and N e N is a positive integer; data The Fourier coefficients are based on N. data The data symbols were obtained, and N e It is determined based on at least one of the following: N data A modulation symbol constellation of data symbols, and the FDSS window of the first communication device 100. Method 400 also includes N-based... sc The Fourier coefficients are obtained as (406)N data Fourier coefficients. Method 400 also includes N data Decode the Fourier coefficients (408) to obtain N. data Data symbols.

[0145] Figure 5 A communication system 500 according to an embodiment of the present disclosure is illustrated. The communication system 500 in the disclosed example includes a first communication device 100 and a second communication device 300 for communication and operation within the communication system 500. For simplicity, the illustrated communication system 500 includes only one first communication device 100 and one second communication device 300. However, the communication system 500 may include any number of first communication devices 100 and any number of second communication devices 300 without departing from the scope of the present disclosure.

[0146] The first communication device 100 is used to generate and transmit a DFT-s-OFDM signal 510 according to the disclosed scheme in the communication system 500. Therefore, the second communication device 300 is used to receive the DFT-s-OFDM signal 510 transmitted by the first communication device 100. The communication system 500 is any suitable communication system, such as 3GPP 5G NR or 6G. The first communication device 100 can also be represented as a transmitter device or simply a transmitter, and can be a client device, such as a UE. Correspondingly, the second communication device 300 can also be represented as a receiver device or simply a receiver, and can be a network access node, such as a BS. The network access node can be connected to a network (NW), such as a core network, via a communication interface. However, embodiments of this disclosure are also applicable to cases where the transmitter is a base station and the receiver is a UE, i.e., the reverse. Furthermore, embodiments of this disclosure are also applicable to other network nodes, such as repeaters, relays, etc. Additionally, embodiments of this disclosure are also applicable to direct communication between UEs, such as via a sidelink (SL) interface.

[0147] To gain a deeper understanding of the disclosed scheme, a signal definition will be given below, considering the signal carrying N. data The DFT-s-OFDM signal s[n] with SE of constellation symbols uses a total of N sc =N data +N e N OFDM subcarriers, of which N e This refers to the number of subcarriers used for symbol repetition solely to reduce PAPR. For ease of explanation and considering the symmetry of the actual FDSS window, we assume N... e It is an even integer. However, by assuming the left SE is... The right SE is Or vice versa, the disclosed solution can be directly extended to N. e The case where the number is odd.

[0148] For 0≤n≤N fft For a sample of -1, the low-pass equivalent time discrete DFT-s-OFDM signal representation is defined as follows:

[0149]

[0150] With frequency domain symbol

[0151]

[0152] in

[0153] ·N fft It is the size of the OFDM modulated IFFT.

[0154] ·N data It refers to the number of constellation symbols used for modulation.

[0155] ●N sc =N e +N data It is the total number of OFDM subcarriers.

[0156] ·N e It is the number of subcarriers used for the SE, also known as the size of the SE.

[0157] ·{W[0],…,W[N sc -1]} is the FDSS window coefficient.

[0158] ·L is the Fourier coefficient shift parameter.

[0159] ·(modN) is the modulo N operator.

[0160] It can be noted that, except and Other types of normalization factors besides those in equations (1) and (2) can be used. Typically, W[k] is real-valued, but embodiments of this disclosure are not limited thereto, and complex-valued FDSS can be used. Embodiments of this disclosure also include the case where W[k] = 1.

[0161] Therefore, according to embodiments of this disclosure, a cyclic shift with a shift parameter L can also be introduced. This means that, according to embodiments, when N... sc Before multiplying the L Fourier coefficients by the FDSS window, the first communication device 100 is used to: multiply the L Fourier coefficients by the N... sc The Fourier coefficients are cyclically shifted to obtain N. sc The Fourier coefficients are obtained by cyclic shifting, where L is a positive integer; N sc The Fourier coefficients of each cyclic shift are of size N. sc Multiply by the FDSS window to obtain N sc Fourier coefficients of frequency shaping and cyclic shifting. The first communication device 100 will ultimately convert N... sc The Fourier coefficients of frequency shaping and cyclic shifting are mapped to N.sc DFT-s-OFDM signals are acquired on each subcarrier 510.

[0162] Accordingly, according to the embodiments, in obtaining N data Before L Fourier coefficients, the second communication device 300 is used to: use L Fourier coefficients to compare N sc The Fourier coefficients are cyclically shifted (in the opposite direction to that performed in the first communication device 100) to obtain L. sc Fourier coefficients of cyclic shifts, where L is a positive integer; based on N sc The Fourier coefficients of the cyclic shift are obtained as N. data Fourier coefficients.

[0163] Figure 6 The block diagram illustrates, for example, a first implementation example of the first communication device 100. Another implementation example of the first communication device 100 will be given below. References Figure 6 The first communication device 100 includes N data Point DFT block 120, used to receive N data Data symbols (x[m]) and based on N data Output N data symbols data The first communication device 100 also includes an SE block 122, which is used to repeat N Fourier coefficients. data N in the Fourier coefficients e N Fourier coefficients are used to provide N to the cyclic shift block 124. sc There are N Fourier coefficients. Therefore, N sc The L Fourier coefficients are cyclically shifted in cyclic shift block 124 to feed the output to FDSS block 126. sc Fourier coefficients of cyclic shift. Subsequently, N in FDSS block 126 sc The Fourier coefficients of each cyclic shift are of size N. sc The FDSS windows are multiplied together to feed the output to N in mapper block 128. sc The Fourier coefficients (X′[k]) of frequency shaping and cyclic shifting. In mapper block 128, N sc The Fourier coefficients of frequency shaping and cyclic shifting are mapped to N. fft N in subcarriers sc On each subcarrier, with N in the output frequency domain fftThe DFT-s-OFDM signal with coefficients is converted to the time domain in IFFT block 130. Furthermore, a cyclic prefix (CP) can be added in CP block 132. Thereafter, the DFT-s-OFDM signal with CP is transmitted in communication system 500 as signal s[n].

[0164] An example of the implementation of the corresponding second communication device 300 is as follows: Figure 7 The block diagram is shown. The received DFT-s-OFDM signal 510 is first demodulated by the CP removal step in the CP removal block 320, and then fed to the N-providing... fft FFT block 322 for each subcarrier coefficient. Subsequently, demapper block 324 allocates these N subcarrier coefficients according to the bandwidth of the first communication device 100. fft N is selected from the subcarriers. sc Fourier coefficients. The selected N sc The Fourier coefficients are fed to the cyclic shift block 326, and cyclically shifted with L Fourier coefficients to provide N. data Fourier coefficients. A common method to mitigate transmission channel fading and FDSS window attenuation is to adjust N in the equalizer and combiner block 328. data Equalization is performed using fourfold coefficients. For further improvement, and depending on whether the repeating symbol is included in or added to the allocated resources for transmitting the DFT-s-OFDM signal, the equalizer and combiner block 328 can also be input with N values ​​corresponding to the repeating symbol. e Fourier coefficients. For example, the Fourier coefficients received from in-band and their repeating versions received from the spectral spread can be in phase and summed before equalization, thus providing combined gain and frequency diversity gain. Finally, N data The balanced Fourier coefficients are fed to N. data Point-wise inverse DFT (IDFT) block 330, IDFT block 330 is used for N data The equalized Fourier coefficients are pre-encoded to provide the data for the decoder block to be sent for decoding. Figure 7 N (not shown in the image) data One demodulated data symbol.

[0165] Given an FDSS window, parameter N e Both L and L affect PAPR performance. This article will reveal how to carefully select these parameters to optimally reduce PAPR. Specifically, it will show:

[0166] 1. When using SE (i.e., N) e When PAPR is >0, it usually decreases, but can only reach an optimal level. Then increase it again. Therefore, choose an excessively large N. e This not only negatively impacts spectral efficiency but also hinders PAPR reduction. This non-trivial optimization primarily depends on the FDSS window, but also on the shift values ​​L and N. data Modulation symbol constellation and bandwidth allocation for each data symbol.

[0167] 2. The L value that provides the lowest PAPR depends on N. data Modulation symbol constellation of data symbols, extension size N e The number of constellation symbols, i.e., N data =N sc -N e .

[0168] There exists a global optimum that minimizes PAPR.

[0169] It is known that SE can further reduce PAPR based on FDSS. However, it is still unknown whether there exists an optimal SE size. This can minimize PAPR; therefore, an excessively large SE might even increase PAPR. Therefore, the SE size should be chosen to ensure... That is, less than Should This corresponds to the specific SE capability of the first communication device 100 as a transmitter. It can be seen that the precise capabilities specific to each transmitter... Depends on the FDSS window, L, and N sc and N data Please note that you need to select... This could also be to mitigate the reduction in SE spectral efficiency at the cost of a slightly increased PAPR. System configuration may also need to be selected. For example, N e The factor is limited to 12 subcarriers to have an integer number of RBs.

[0170] exist Figures 8(a) to 8(c) In the diagram, the 99th percentile PAPR is plotted as the number N of SE subcarriers. e The function, while the total bandwidth is fixed at N. sc =96. This is plotted for different filters, modulation symbol constellations, and shift values ​​L. In all these numerical comparisons, two FDSS windows are considered: i) the Kaiser window, where β = 3; ii) a truncated RRC window, where ρ = 0.5 and β = -0.65. The shift rule L = 0 and... (The sentence is incomplete and requires further context to translate accurately.) The following text will explain why parameters This is the best shift method closest to π / 2-BPSK. It is also used for QAM constellations. I will explain later why it is close to the optimal shift for QPSK.

[0171] Figures 8(a) to 8(c) This indicates that the generated PAPR is N e The function is neither linear nor continuously decreasing. Therefore, for all cases, there exists an optimal SE size that minimizes PAPR. From that Increasing the SE further will become harmful. Therefore, the size of the SE should preferably be chosen such that... This can be predetermined based on transmission configurations (such as FDSS window, shift parameter L, modulation symbol constellation, and bandwidth allocation). from Figures 8(a) to 8(c) It can also be seen that even without FDSS, PAPR can be significantly reduced through SE.

[0172] In the simulation, for a given total bandwidth N sc N was added e The value of N was reduced, thus decreasing N. data The value of N is such that e +N data =N sc Therefore, given N sc Perform SE within the total bandwidth of each subcarrier. Alternatively, consider optimizing for the optimal N. e In order to fix N data and increase total bandwidth N sc . Figure 9 Comparison of different N e PAPR, where N data =96 is fixed, has QPSK and from Figure 9 It can be observed that the shape of these curves is similar to N. sc The shapes are similar when fixed, and the global optimum is clearly shown.

[0173] FDSS window and With N sc Convergence of growth

[0174] SE size The global optimum depends primarily on the FDSS window design. In fact, for a given FDSS window, as N... sc growth, It seems to converge to N sc The percentage. This means that to achieve a similar PAPR reduction, N e N should be allocated according to bandwidth. sc or N data It increases linearly. Therefore, Ne It's best to be N sc Or equivalent N data The function. In other words, in the embodiment, N e It is further based on parameter N sc or N data It's confirmed.

[0175] Figures 10(a) and 10(b) show the numerical results for different FDSS windows. (N sc (percentage), the It is N RB =N sc A function of / 12. With N sc The percentage is expressed numerically. Should It is N RB =4, 8, 12, 24, 48, 96, and 100. For a given modulation symbol constellation, it can be seen that, depending on the FDSS window, the optimal... There could be significant differences. In these examples, when N e The lowest PAPR is likely to be obtained when the value is between 7% and 35% of the total bandwidth allocation. Generally, the more edge attenuation the FDSS window provides, the lower the PAPR; the optimal value... The larger it becomes. More importantly, for each FDSS window, it seems that as N increases... sc The growth of the optimal SE size Almost N sc A constant fraction.

[0176] In addition, Figure 11 The diagram in N is drawn data Optimal under fixed conditions As obtained The percentage. QPSK modulation is used here (where...). ) and FDSS Kaiser window (where β = 2). This is related to N sc Obtained at fixed time (as N) sc The percentages were compared. It can be observed that, compared to N... sc Optimization when fixed In comparison, N data Optimization when fixed Get very similar Ratio. It's also important to note that in N... data In a fixed case, use Using a ratio to represent the optimal SE size might be more natural, and this ratio is clearly different from... Obviously. Generally speaking, the relationship between these two ratios is always as follows: assuming Then it is equivalent to

[0177] semi-analytical approximation

[0178] The optimal SE size that minimizes PAPR can also be verified and approximated through a semi-analytical numerical search.

[0179] Therefore, the signal is first represented as multiple pulses. Time duplex. By inserting equation (2) into (1), we can obtain:

[0180]

[0181] The pulse shaping filter is as follows:

[0182]

[0183] Without FDSS windowing, W[0] = ... = W[N] sc -1] = 1, which further simplifies to having N sc A conventional DFT-s-OFDM pulse in the form of a Dirichlet kernel for each modulated subcarrier:

[0184]

[0185] Using this pulse formula and adjusting the boundaries, we can obtain the following boundaries for the φ-rotation BPSK on our system:

[0186]

[0187] in It is the squared norm of the FDSS window.

[0188] For π / 2 - BPSK, specifically φ = π / 2. Furthermore, we choose L = -N. e / 2 is the best shift method, therefore Therefore, the boundary of equation (6) is approximately equal to

[0189]

[0190]

[0191] Please note that when N is allowed e When the integer is an odd number, the equation holds, i.e., L = -(N)e -1) / 2.

[0192] For QPSK, and indeed any PSK modulation, a similar boundary can be derived. The boundary becomes...

[0193]

[0194] This can be directly verified by using |cosθ|≤1. Furthermore, if φ = 0 (as in a normal BPSK), then if it is possible to have L = -(N) e If -1) / 2, then these two boundaries are the same, that is:

[0195] To calculate these boundaries, it is necessary to... fft Find a maximum value on the time sample. Nevertheless, by noting that the function inside the maximum operator is periodic, the computational complexity of this step can be significantly reduced, thus requiring only the computation of... By finding a sample, we can find the maximum value over n.

[0196] The boundaries in equations (7) and (8) can be compared using simulated 99th percentile PAPR. The boundaries in equations (7) and (8) are not tight, but nonetheless, they appear to roughly maintain the position of the optimal SE size that minimizes PAPR. Therefore, a numerical search can be conducted for the value N that minimizes these boundaries. e To approximate the optimal SE size. This is shown in Figures 10(a) and 10(b), where it can be seen that... This semi-analytical approximation is usually very close, remaining within 5% of the difference even in the worst case, except for the case of π / 2-BPSK with a Kaiser window (β=3). For this particular case, the semi-analytical search deviates significantly from the numerical search, seemingly corresponding to the location of the minimum PAPR at L=0, as shown in Figure 8(a). The problem in this case is that the SE extension can only reduce PAPR by a maximum of 0.32 dB, while without SE, PAPR is already very low, at only 1.84 dB. Therefore, the resulting boundary is very flat over a large numerical range, without any very sharp minimums.

[0197] Furthermore, an equivalent pulse shaping formula is used to provide a supplementary intuitive explanation for the existence of this optimal SE size. The upper bound in equation (8) can be restated as

[0198]

[0199] When N e When increasing, the coefficient Systematically increase. However, It is the square of the maximum coherent combination of pulses, which can be used as N. e The function is used to decrease or increase the value. For the typical window function W[k] as described above, the pulse shaping filter |g i [n]| Basically retains the sinc shape, but has more or less attenuated sidelobes. The number of pulses is N. data As given in equation (3), and their time interval is A sample. Through equation (4), it can be further observed that the pulse shape depends only on the FDSS window and the total number of subcarriers N assigned. sc If N is fixed sc The value will then be... N samples separated data =(N sc -N e ) pulses, meaning the pulse shape will not change with the increase of N e However, the time interval between pulses will increase. The degree of pulse lobe overlap depends on the design of the FDSS, and in any case, the increased time interval will work to reduce this overlap, helping to reduce the peak power of the signal.

[0200] If changed to a fixed value N data N sc Will follow N e Increase and increase, and there will be N data A pulse was The samples are separated. Conversely, in this case without FDSS, the pulse shape in equation (5) becomes narrower because the width of the main lobe is Narrower pulses also help reduce the peak power of the signal. In any case, the FDSS window must be targeted at N. sc The bandwidth of each subcarrier is designed.

[0201] Modulation Symbol Constellation

[0202] In embodiments of this disclosure, N data The modulation symbol constellation for each data symbol is either a π / 2-BPSK constellation or a QAM constellation. When the modulation symbol constellation is QPSK, 16-QAM, or 64-QAM, the optimal [condition / performance] is achieved when using the same FDSS window as shown in Figure 10(b). They appear to be very close. With a Kaiser window (β=3), the maximum difference between 16-QAM and QPSK is 1.5%. In some cases, the difference is not noticeable. Therefore, a good rule of thumb is to choose the same N for both 16-QAM and 64-QAM as for QPSK. e Table 1 below highlights the selection when using π / 2-BPSK and QAM constellations. The difference.

[0203] Table 1 is based on Figure 10. Numerical selection

[0204]

[0205] As can be seen from Table 1, for the same FDSS window, compared with QPSK, the N value for obtaining the optimal PAPR when using π / 2-BPSK is higher. e The value is much smaller. For π / 2-BPSK, a good rule of thumb for significantly reducing PAPR is to choose... in It is the optimal value for QPSK because Therefore, in summary, it can be seen from... The single value is used to derive N for all constellations. e choose.

[0206] Shift parameter L

[0207] Figure 8 shows the shifting method derived in this disclosure. It provides the best PAPR, while from the conventional scheme The worst PAPR was obtained in the middle. It can be found that these two shifting methods have the worst performance in terms of SE size. Optimal results can be achieved when values ​​are similar. If Best SE size comparable Slightly reduce N sc 1% to 2%.

[0208] Now let's explain why the optimal shift method for π / 2-BPSK is... The optimal shifting method for QPSK and other QAM constellations is... Starting with the multicarrier signal in equation (3), the phase difference between two adjacent pulses with indices m and (m+1) is:

[0209]

[0210] Then, if the FDSS window is real and symmetric, it can be used.

[0211]

[0212] This can also be directly verified as the special case of equation (2) without FDSS.

[0213] Therefore, the pulse phase difference can be obtained.

[0214]

[0215] The purpose of designing the rotating constellation as π / 2-BPSK is to ensure that the BPSK symbols of adjacent pulses are transmitted with a phase difference of almost π / 2 (mod π), thereby minimizing their maximum power combination.

[0216] For π / 2 - BPSK, find the optimal value of L that makes equation (12) closest to zero, i.e. If N e If it is an odd integer, then Provides optimal performance. If N e If it is an even integer, then as in [7] or To provide optimal performance. Therefore, as shown in Figure 8(a), select close to The shifting method can provide better PAPR performance than L=0.

[0217] For QPSK and other QAMs, choose This is suboptimal; using other values ​​of L almost always yields a better PAPR. It's worth noting that the lowest PAPR is achieved by generating a phase difference close to π / 4 (mod π / 2) between adjacent pulses.

[0218] In other words, it should be concluded that

[0219]

[0220] Therefore, the optimal PAPR can always be achieved by choosing L that is approximately equal to the following formula:

[0221]

[0222] Where k is an integer, because the L shift causes the phase difference between data symbols to always be close to π / 4.

[0223] This is why in Figures 8(b) and 8(c), PAPR was at its worst. PAPR is optimal at this time. The other value L = 0 in the conventional scheme is a trade-off between the first two L values, because when i)N e When = 0, then Or when ii)N e =N sc / 5 o'clock (equivalent to ),but It can be simplified to the other two methods. Therefore, only in N e ≈N sc In 20% of cases, L=0 is required to achieve near-optimal performance.

[0224] Compared with the conventional approach, using The gain can be as high as 0.6 dB, especially when the size of SE N e Smaller than optimal As shown in Figures 8(b) and 8(c). Please note that the selection... In practice, it might be desirable, for example, to mitigate the spectral efficiency reduction of the SE, or to have an integer number of RBs by setting N. e The factor is limited to 12 subcarriers.

[0225] In summary, to achieve optimal PAPR reduction, as shown in the figure above, the L values ​​for certain modulation symbol constellations between π / 2-BPSK and QAM should be different. This should also be based on N. e And depending on the modulation symbol constellation, it may also be based on N. data or N sc To determine L. More precisely, for QAM, the optimal PAPR value is determined by... or or or or or Implemented, where k is a positive integer, and round(x) gives the integer closest to x. and These are the floor operations for x: floor up and floor down.

[0226] Signaling aspect

[0227] Figure 12 The signaling aspects according to an embodiment of this disclosure are shown. Since the disclosed scheme assumes the number N subcarriers used for the SE based on the gain shown above,... e This is based on the FDSS capability of the first communication device 100, and other possible transmission configurations, such as, but not limited to:

[0228] The total number of subcarriers N allocated sc or the number N of subcarriers containing data symbols data .

[0229] ● Shift parameters of the Fourier coefficients of L subcarriers.

[0230] • Modulation symbol constellation.

[0231] Both the transmitter (e.g., UE) and the receiver (e.g., base station) need to know the selected parameter N. eThe value of L. Such information can be provided through control signaling between the first communication device 100 (i.e., the transmitter) and the second communication device 300 (i.e., the receiver). For example, control signaling can be executed through a higher protocol layer, such as radio resource control (RRC) signaling or medium access control (MAC) signaling. This is advantageous if the parameters do not need to be changed dynamically. Another option is to notify by signaling through the physical layer, such as through a control channel. The advantage of doing so is that the parameters can be changed on the fly. A combination of higher-layer signaling and physical-layer signaling can also be used, for example, N e Some values ​​of L are configured by higher layers, and physical layer signaling selects from these values.

[0232] refer to Figure 12 In step I, the second communication device 300 sends a first control signal 520 to the first communication device 100. The first control signal 520 indicates parameter N. e And / or L. In Figure 12 In step II, the first communication device 100 is used to receive an indication N from the second communication device 300. e The first control signal 520 for L and / or L.

[0233] However, parameter N e And / or L, or a portion thereof, may also be provided implicitly. For example, parameter N e L can be predetermined and defined by standards for different modulation symbol constellations, different numbers of allocated subcarriers, and different parameters for frequency domain filters. This avoids the use of control signaling, thereby reducing the overhead in the communication system 500.

[0234] Furthermore, when the first communication device 100 is a UE, the UE should signal to the BS whether it supports using SE. This can be accomplished through UE capability signaling. In other words, the first communication device 100 can be used to send an instruction to the first communication device 100 to repeat N. e The second control signal 530, which has the capability to generate a Fourier coefficient, is as follows: Figure 12 Step III is shown in the diagram. Figure 12 In step IV, the second communication device 300 receives the second control signal 530 and thereby derives information about the repeatability of the first communication device 100.

[0235] Furthermore, if the communication standard does not provide an FDSS window, the first communication device 100 (e.g., UE) can specify the value. The signal is sent to the second communication device 300 (e.g., BS), and the value is N. eThe maximum value of PAPR cannot be exceeded, and PAPR cannot be reduced beyond this value. In this way, the N determined by BS can be avoided. e The value is greater than the required value.

[0236] It can be noted that the signaling of the first control signal 520 and the second control signal 530 can be executed in reverse order, that is, the first control signal 520 can be sent after the second control signal 530, without deviating from the scope of the disclosed scheme. The first control signal 520 and the second control signal 530 can also be sent simultaneously.

[0237] The communication resources used by the UE for uplink transmissions can be pre-configured, for example, through semi-static scheduling or configuration granting. Alternatively, the BS can send uplink grants containing information about the transmission, including resource blocks allocated for the transmission, to the UE via the physical downlink control channel (PDCCH). In embodiments of this disclosure, N e Each subcarrier can be included in a resource block allocated for transmission, i.e., N e Each Fourier coefficient is repeatedly included in the resources allocated for transmitting the DFT-s-OFDM signal 510. In other embodiments of this disclosure, N e N subcarriers can be added to the resource block allocated for transmission, i.e., N e Each Fourier coefficient is repeatedly added to the resources allocated for transmitting the DFT-s-OFDM signal 510.

[0238] Implementation example

[0239] The DFT-s-OFDM signal discussed and proposed above can typically be implemented by cascading DFT-precoding, a periodic SE with shift, an FDSS window, and OFDM modulation, as shown below.

[0240] DFT precoding: for k = 0, 1, ..., N data -1, constellation symbol {x[0],…,x[N]} data -1]} First, it undergoes DFT precoding to obtain the Fourier coefficients:

[0241]

[0242] The transmitter chain is Figure 6 As shown in the figure, the constellation symbols {x[0],…,x[N] are... data -1]} First, DFT precoding is performed in DFT block 120 to obtain the Fourier coefficients {X[0],…,X[N]} data -1]}.

[0243] The SE in SE block 122 and cyclic shift block 124 has a shifted periodic SE: this frequency domain sequence is assigned to N using periodic repetition. sc =N data +N e The spread spectrum of each subcarrier is shown below.

[0244] X (se) [k] = X[k + L(mod N) data (16) Suitable for k = 0, 1, ..., N sc -1, where L is the integer shift. Note that this definition of SE applies to N. e Odd numbers are also directly valid.

[0245] FDSS: Before transmission via OFDM modulation, FDSS is applied to the Fourier coefficients in FDSS block 126, as shown below.

[0246] X′[k]=W[k]X (se) [k] (17)

[0247] Where {W[0],…,W[N] sc -1]} is the FDSS window.

[0248] OFDM modulation: Finally, for n = 0, 1, ..., N fft -1, in IFFT block 130 via N fft Pointed IFFT is used to obtain the (normalized) time-discrete low-pass equivalent signal, as shown below:

[0249]

[0250] SE with shift

[0251] Furthermore, the steps of the SE with cyclic shift can be implemented through different equivalent embodiments. Different examples of the transmission frequency domain sequence as a function of the shift parameter L are as follows: Figure 13 As shown. For any value of L, the original symbol sequence {X[0],…,X[N]} data -1]} is always included in the in-band spectrum, with a maximum cyclic shift. The symbols are: the left excess-band symbol is always a repetition of the symbol at the inner edge of the right band; the same is true for the right excess-band. Several embodiments are disclosed for implementing SE according to equation (12).

[0252] In such Figure 6 In the first exemplary implementation shown, it can be achieved by using the original N data Append the first N bits to the end of a long sequence X[k].e A symbol, then the extended N sc SE is achieved by cyclically shifting a long sequence by L symbols.

[0253] In such Figure 14 In the second exemplary implementation shown, N can be first set to N. data A long sequence X[k] is cyclically shifted by L symbols, and then appended with the first N symbols after the end of the shifted sequence. e Each symbol is used to implement a cyclic SE. Therefore, Figure 6 and Figure 14 The difference between the exemplary implementations in [the document] lies in the order of the cyclic shift and SE operations. Figure 14 In this context, the circular shift is performed before the SE operation.

[0254] In such Figure 15 In the third exemplary implementation shown, the original N can be first... data Circular shift of a long sequence X[k] The SE is implemented using symbols, as shown below.

[0255]

[0256] Then, respectively in the shift sequence X ( L ) [k] followed by and before the prefix One and after A symbol, namely

[0257]

[0258] therefore, Figure 6 and Figure 15 The difference between the exemplary implementations in [the document] lies in the order of the cyclic shift and SE operations. Figure 15 In this context, the circular shift is performed before the SE operation, such as... Figure 13 That's how it is in the middle. However, in Figure 15 SE is also executed in another way, such as Figure 6 and Figure 14 As shown. Therefore, having The third exemplary implementation corresponds to the symmetric SE of the conventional scheme, see [link to relevant documentation]. Figure 13 L = -2. Note that the choice of the shift parameter L depends only on N. e In fact, it is related to N data It's irrelevant because L is in N. data It is calculated in modular arithmetic. Furthermore, it should be noted that the choice of L is purely for convenience and does not actually provide any additional benefit.

[0259] DMRS transmission

[0260] In 5G NR, the demodulation reference symbol (DMRS) is multiplexed with the data symbols (called the physical uplink shared channel (PUSCH) in NR). Typically, only a few OFDM symbols carry the DMRS, for example, one out of 14 symbols carries the DMRS.

[0261] When using DFT-s-OFDM, DMRS and PUSCH are time-multiplexed. In the OFDM symbol carrying DMRS, N of the entire transmission bandwidth... sc An un-DFT-precoded DMRS sequence is inserted on every other subcarrier (called a resource element (RE) in NR), while other REs are blocked from data transmission. Two possible configuration examples are as follows: Figure 16 As shown.

[0262] When data transmission employs π / 2-BPSK modulation, one type of DMRS sequence is DFT-precoded pseudo-noise π / 2-BPSK. Therefore, the characteristics of the SE discussed for data transmission are directly extended to such DMRS sequences.

[0263] For other constellations, another DMRS sequence is the Zadoff-Chu (ZC) sequence, with a length of M. ZC The DMRS sequence has elements obtained by the following cyclic expansion.

[0264]

[0265] The above cyclic expansion is a cyclic expansion of the following ZC sequence.

[0266]

[0267] And the length of the largest prime number satisfies N ZC ≤M ZC The possible root indices are u = 1, ..., N. ZC -1.

[0268] Please note that the ZC sequence is a constant amplitude zero autocorrelation sequence (CAZAC), and the IDFT of the ZC sequence is also a CAZAC sequence. Due to this property, DMRS with a ZC sequence can be expected to be a DFT-s-OFDM transmission, and therefore the properties of the SE discussed for data transmission are directly extended to this DMRS sequence.

[0269] Since the FDSS window used for data transmission is typically unknown at the receiver, the DMRS sequence needs to be shaped using the same FDSS window as the data symbols. Depending on the receiver's capabilities, three options can be considered.

[0270] Option A: The second communication device 300, acting as the receiver, can only decode the in-band spectrum without extending it. Therefore, the DMRS sequence needs to be designed based on the in-band spectrum, i.e., M ZC =N data / 2.

[0271] Option B: The second communication device 300, acting as the receiver, is capable of receiving superband symbols, and the DMRS sequence needs to be based on the entire allocated bandwidth M. ZC =N sc / 2 is used for design.

[0272] Option C is to accommodate two types of receivers with the same DMRS design. The sequence is first designed based on in-band conditions, and then the sequence is spectrally extended to the entire allocated band. The DMRS sequence design becomes...

[0273]

[0274] Where N ZC It satisfies N ZC ≤N data The maximum prime length of / 2. Sequences without SE begin in the band; this is addressed using symmetric expansion, equivalent to a shift.

[0275] The three options for DMRS design are as follows: Figure 17 As shown. They are in the RRC window, N sc =96 and N data Different PAPR values ​​at 72 are as follows Figure 18 As shown. Figure 18 N data The choice corresponds to having N sc With a 25% SE, this is the optimal choice for reducing PAPR of QPSK data transmission for this FDSS window. It can be observed that option C has the highest PAPR, but this is balanced by the fact that it produces more sequences: 31 with option B, compared to 23 with options A and C. Otherwise, the SE DMRS design for option C would reduce the PAPR of option A.

[0276] Finally, Figure 19 In the comparison, option C is in N. sc =96, different SE sizes N e =(N sc -N dataThe maximum PAPR under three different FDSS windows was also analyzed. Interestingly, the maximum PAPR of DMRS using SE behaves similarly to that using data transfer. PAPR can be observed to first decrease to a minimum and then increase. The SE size that provides the lowest PAPR using DMRS is close to the SE size that minimizes PAPR using QPSK data transfer.

[0277] The network access node in this article can also be referred to as a wireless network access node, access network access node, access point (AP), or base station (BS), such as a radio base station (RBS). In some networks, it may be called a transmitter, "gNB," "gNodeB," "eNB," "eNodeB," "NodeB," or "B node," depending on the standards, technologies, and terminology used. Based on transmission power and cell size, wireless network access nodes can have different categories or types, such as macro eNodeB, home eNodeB, or mini base station. A wireless network access node can also be a site, i.e., any device that includes an IEEE 802.11 compliant media access control (MAC) and physical layer (PHY) interface to the wireless medium (WM). Wireless network access nodes can be used for communication in the following: 3GPP-related long term evolution (LTE), LTE-Advanced, fifth generation (5G) wireless systems, such as new radio (NR) and its evolution, and IEEE-related Wi-Fi, worldwide interoperability for microwave access (WiMAX) and its evolution.

[0278] The client device in this document can be represented as a user device (UE), user equipment (UE), mobile station, Internet of Things (IoT) device, sensor device, wireless terminal, and / or mobile terminal, capable of wireless communication in a wireless communication system (sometimes also called a cellular wireless system). A UE can be referred to as a wirelessly capable mobile phone, cellular phone, tablet, or laptop. For example, in this context, a UE can be a portable, pocket-sized, handheld, computer-based, or vehicle-mounted mobile device capable of communicating voice and / or data with another communication entity (such as another receiver or server) via a radio access network (RAN). A UE can also be a site, i.e., any device connected to WM with IEEE 802.11 compliant MAC and PHY interfaces. A UE can be used to communicate in: 3GPP-related LTE, Advanced LTE, 5G wireless systems such as NR and its evolution, and IEEE-related Wi-Fi, WiMAX, and their evolution.

[0279] Furthermore, any method according to embodiments of this disclosure can be implemented in a computer program having code components that, when run by a processing component, cause the processing component to perform the steps of the method. The computer program is included in a computer-readable medium of the computer program product. The computer-readable medium can substantially include any memory, such as the ROM, PROM, EPROM, flash memory, EEPROM, hard disk drive, etc., described above.

[0280] Furthermore, it should be recognized that the first communication device 100 and the second communication device 300 include the necessary communication capabilities, in the form of, for example, functions, components, units, elements, etc., for performing or implementing embodiments of this disclosure. Examples of other such components, units, elements, and functions include: processors, memories, buffers, control logic, encoders, decoders, rate matchers, de-rate matchers, mapping units, multipliers, decision units, selection units, switches, interleavers, deinterleavers, modulators, demodulators, inputs, outputs, antennas, amplifiers, receiving units, transmitting units, DSPs, TCM encoders, TCM decoders, power supply units, power feeders, communication interfaces, communication protocols, etc., which are suitably arranged together to execute the scheme.

[0281] Therefore, one or more processors in the first communication device 100 and the second communication device 300 may include, for example, a CPU, processing unit, processing circuitry, processor, ASIC, microprocessor, or one or more instances of other processing logic capable of interpreting and executing instructions. Thus, the term "processor" may refer to a processing circuitry comprising multiple processing circuits, such as any, some, or all of the aforementioned processing circuits. The processing circuitry may also perform data processing functions for inputting, outputting, and processing data, including data buffering and device control functions such as call processing control, user interface control, etc.

[0282] Finally, it should be understood that this disclosure is not limited to the embodiments described above, but also relates to and covers all embodiments within the scope of the appended independent claims.

Claims

1. A first communication device (100) for a communication system (500), characterized in that, The first communication device (100) is used for: Based on N data Get N data symbols data Fourier coefficients, where N data It is a positive integer; Repeat the N data N in the Fourier coefficients e Fourier coefficients are used to obtain N sc Fourier coefficients, where N e and N sc N is a positive integer, and N is a positive integer. e It is determined based on at least one of the following: the N data The modulation symbol constellation of data symbols, and the frequency domain spectrum shaping (FDSS) window of the first communication device (100); The N sc The Fourier coefficients are of size N. sc Multiply by the FDSS window to obtain N sc Frequency-adjusted Fourier coefficients; The N sc Each frequency-shaped Fourier coefficient is mapped to N. sc Discrete Fourier Transform Extended Orthogonal Frequency Division Multiplexing (DFT-s-OFDM) signals are obtained on each subcarrier (510); The DFT-s-OFDM signal (510) is transmitted.

2. The first communication device (100) according to claim 1, characterized in that, N e Is it still based on N? sc or N data It's confirmed.

3. The first communication device (100) according to claim 1 or 2, characterized in that, The N data The modulation symbol constellation for each data symbol is either a π / 2-BPSK constellation or a QAM constellation.

4. The first communication device (100) according to any one of claims 1 to 3, characterized in that, N e It is predetermined; or The first communication device (100) is used to receive instruction N e The first control signal (520).

5. The first communication device (100) according to any one of claims 1 to 4, characterized in that, The N e The Fourier coefficients are repeatedly: Included in the allocated resources for transmitting the DFT-s-OFDM signal (510); or Add to the allocated resources for transmitting the DFT-s-OFDM signal (510).

6. The first communication device (100) according to any one of claims 1 to 5, characterized in that, The first communication device (100) is used for: Send a second control signal (530), the second control signal (530) instructing: the first communication device (100) to repeat the N e The ability to generate Fourier coefficients.

7. The first communication device (100) according to any one of claims 1 to 6, characterized in that, In the N sc Before multiplying the Fourier coefficients by the FDSS window, the first communication device (100) is used to: Using L Fourier coefficients to represent the N sc The Fourier coefficients are cyclically shifted to obtain N. sc Fourier coefficients of cyclic shift, where L is a positive integer; The N sc The Fourier coefficients of each cyclic shift and the value of N sc Multiply by the FDSS window to obtain N sc Fourier coefficients for frequency shaping and cyclic shifting; The N sc The Fourier coefficients of frequency shaping and cyclic shifting are mapped to the N sc The DFT-s-OFDM signal (510) is acquired on each subcarrier.

8. The first communication device (100) according to claim 7, characterized in that, L is predetermined; or The first communication device (100) is used to receive a first control signal (520) indicating L.

9. The first communication device (100) according to claim 7 or 8, characterized in that, L is determined based on any of the following: N e N data or N sc ; The N data A modulation symbol constellation of data symbols; formula or or or or or Where k is a positive integer, and round(x) gives the integer closest to x. and These are the floor operations for x: floor up and floor down.

10. The first communication device (100) according to any one of claims 1 to 9, characterized in that, Less than the maximum permissible Fourier coefficient repeatability of the first communication device (100) and Based on the FDSS window, L, and N sc and N data Any one of them is determined.

11. The first communication device (100) according to claim 10, characterized in that, The first communication device (100) is used for: Based on the FDSS window, the N data The modulation symbol constellation of each data symbol, L, and N. sc or N data Determine any one of the following: To minimize the peak-to-average power ratio (PAPR) of the transmitted DFT-s-OFDM signal (510).

12. The first communication device (100) according to claim 10 or 11, characterized in that, The first communication device (100) is used for: Send Instruction The second control signal (530).

13. A second communication device (300) for a communication system (500), characterized in that, The second communication device (300) is used for: A DFT-s-OFDM signal (510) is received from a first communication device (100), the DFT-s-OFDM signal (510) comprising a mapping to N sc N on each subcarrier sc Fourier coefficients, where N sc It is a positive integer; The N is obtained based on the DFT-s-OFDM signal (510). sc The N Fourier coefficients sc The Fourier coefficients include N data The Fourier coefficients and the N data N in the Fourier coefficients e N repeated Fourier coefficients, where N data and N e N is a positive integer; data The Fourier coefficients are based on N. data The data symbols were obtained, and N e It is determined based on at least one of the following: the N data The modulation symbol constellation of the data symbols, and the FDSS window of the first communication device (100); Based on the N sc The N is obtained from the Fourier coefficients. data Fourier coefficients; For the N data Decode the Fourier coefficients to obtain the N. data Data symbols.

14. The second communication device (300) according to claim 13, characterized in that, N e Is it still based on N? sc or N data It's confirmed.

15. The second communication device (300) according to claim 13 or 14, characterized in that, The N data The modulation symbol constellation for each data symbol is either a π / 2-BPSK constellation or a QAM constellation.

16. The second communication device (300) according to any one of claims 13 to 15, characterized in that, N e It is predetermined; or The second communication device (100) is used to send an instruction N e The first control signal (520).

17. The second communication device (300) according to any one of claims 13 to 16, characterized in that, The N e The Fourier coefficients are repeatedly: Included in the allocated resources for transmitting the DFT-s-OFDM signal (510); or Add to the allocated resources for transmitting the DFT-s-OFDM signal (510).

18. The second communication device (300) according to any one of claims 13 to 17, characterized in that, The second communication device (300) is used for: Receive a second control signal (530), the second control signal (530) indicating that the first communication device (100) repeats the N e The ability to generate Fourier coefficients.

19. The second communication device (300) according to any one of claims 13 to 18, characterized in that, In obtaining the N data Before the Fourier coefficients, the second communication device (300) is used for: Using L Fourier coefficients to represent the N sc The Fourier coefficients are cyclically shifted to obtain N. sc Fourier coefficients of cyclic shift, where L is a positive integer; Based on the N sc The Fourier coefficients of the cyclic shift are used to obtain the N. data Fourier coefficients.

20. The second communication device (300) according to claim 19, characterized in that, L is predetermined; or The second communication device (300) is used to send a second control signal (530) indicating L.

21. The second communication device (300) according to claim 19 or 20, characterized in that, L is determined based on any of the following: N e N data or N sc ; The N data A modulation symbol constellation of data symbols; formula or or or or or Where k is a positive integer, and round(x) gives the integer closest to x. and These are the floor operations for x: floor up and floor down.

22. The second communication device (300) according to any one of claims 13 to 21, characterized in that, Less than the maximum permissible Fourier coefficient repeatability of the first communication device (100) and Based on the FDSS window, L, and N sc and N data Any one of them is determined.

23. The second communication device (300) according to claim 22, characterized in that, The first communication device (100) is based on the FDSS window and the N data The modulation symbol constellation of each data symbol, L, and N. sc or N data Determine any one of the following: To minimize the PAPR of transmitting the DFT-s-OFDM signal (510).

24. The second communication device (300) according to claim 22 or 23, characterized in that, The second communication device (300) is used for: Receive Instruction The second control signal (530).

25. A method (200) for a first communication device (100), characterized in that, The method (200) includes: Based on N data The data symbols are obtained as (202)N data Fourier coefficients, where N data It is a positive integer; Repeat N as described in (204) data N in the Fourier coefficients e Fourier coefficients are used to obtain N sc Fourier coefficients, where N e and N sc N is a positive integer, and N is a positive integer. e It is determined based on at least one of the following: the N data The modulation symbol constellation of the data symbols, and the FDSS window of the first communication device (100); The N sc The Fourier coefficients are of size N. sc Multiply the FDSS windows by (206) to obtain N sc Frequency-adjusted Fourier coefficients; The N sc The frequency-shaped Fourier coefficients are mapped (208) to N. sc DFT-s-OFDM signals are acquired on each subcarrier (510); Send (210) the DFT-s-OFDM signal (510).

26. A method (400) for a second communication device (300), characterized in that, The method (400) includes: Receive (402) DFT-s-OFDM signal (510) from the first communication device (100), the DFT-s-OFDM signal (510) comprising a mapping to N sc N on each subcarrier sc Fourier coefficients, where N sc It is a positive integer; Based on the DFT-s-OFDM signal (510), obtain (404) the N sc The N Fourier coefficients sc The Fourier coefficients include N data The Fourier coefficients and the N data N in the Fourier coefficients e N repeated Fourier coefficients, where N data and N e N is a positive integer; data The Fourier coefficients are based on N. data The data symbols were obtained, and N e It is determined based on at least one of the following: the N data The modulation symbol constellation of the data symbols, and the FDSS window of the first communication device (100); Based on the N sc The Fourier coefficients are obtained from the N (406). data Fourier coefficients; For the N data The Fourier coefficients are decoded (408) to obtain the N. data Data symbols.

27. A computer program product including program code, characterized in that, When the program code is run on a computer, the program code is used to perform the method according to claim 25 or 26.

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