Induction motor model predictive current control method based on generalized flux linkage error and closed-loop flux weakening strategy

Abstract

The invention discloses an induction motor model predictive current control method based on a generalized flux linkage error and a closed-loop flux weakening strategy, and belongs to the field of induction motor speed regulation control. According to the method, the problem that the existing control method cannot simultaneously meet the requirements of not adding extra flux weakening weight factors in a value function, not depending on motor parameters, carrying out closed-loop control and not causing current following error increase under single-vector model predictive current control is solved. The method specifically comprises the steps of 1, defining a generalized flux linkage vector reference value and a generalized flux linkage predicted value, and obtaining a relation between a basic generalized flux linkage vector and a basic voltage vector according to the generalized flux linkage predicted value; step 2, obtaining a value function of single vector model predictive current control of the induction motor according to a result of the step 1; and step 3, obtaining an optimal voltage vector according to the value function, and generating a flux weakening current component for compensation according to the basic generalized flux linkage vector corresponding to the optimal voltage vector. The method can be applied to model predictive current control of the induction motor.

Description

Technical Field

[0001] The invention belongs to the field of induction motor speed control, and in particular relates to an induction motor model prediction current control method based on generalized flux linkage error and closed-loop magnetic weakening strategy. Background Art

[0002] Single vector model predictive current control (SV-MPCC) has attracted widespread attention due to its high transient performance and small amount of calculation. Introducing the weakening magnetic control strategy into the MPC strategy is an important issue for high-speed operation of traction motors. Compared with vector control, the control object of SV-MPCC is no longer the fundamental current component, but directly generates the inverter switch state with reference to the current vector prediction value. Therefore, SV-MPCC can only obtain the actual output fundamental voltage amplitude according to the switch state to construct a voltage closed-loop weakening magnetic structure. However, due to the voltage limitation of the inverter, the actual fundamental voltage in the weakening magnetic region is often smaller than the expected fundamental voltage, which means that compared with the expected fundamental voltage, the actual fundamental voltage as feedback cannot reflect the actual voltage saturation situation. However, the expected fundamental voltage cannot be directly obtained in the SV-MPCC system, which makes the existing closed-loop weakening magnetic strategy have dynamic problems. For this reason, the existing SV-MPCC system mostly adopts the open-loop calculation of the excitation current based on the motor model to weaken the magnetic strategy, but it has the problems of poor robustness and low modulation ratio control accuracy.

[0003] In order to achieve weak magnetic operation in the MPC strategy, some scholars have proposed some solutions, but most of these methods are open-loop strategies, and the load capacity and the ability to further increase the motor speed are not strong. For example, the reference (Predictive Torque Control of Induction Motor Sensorless Drive Fedby a 3L-NPC Inverter) sets the reference torque and flux size to be proportional to the inverse of the rotor speed. The reference (IPMSM Model Predictive Control in Flux-Weakening Operation Using an Improved Algorithm) proposes a voltage-current linear weak magnetic method based on the MPCC algorithm, which has strong robustness to changes in motor parameters. The reference (Low Switching Frequency Model Predictive Control of Three-Level Inverter-Fed IM Drives With Speed-Sensorless and Field-Weakening Operations) adjusts the torque and stator flux reference values ​​online to improve the torque capacity during FW operation. However, some existing methods only consider voltage limitation and ignore the change of stator current; some methods generate unstable voltage when the magnetic field is weakened, which will reduce the control performance; or due to the limitation of switching loss and neutral point voltage error, more weight factors are introduced into the value function. Single vector model predictive control directly generates the switch state according to the following of the control variable, and cannot obtain voltage feedback information. The more mature weak magnetic control strategy based on voltage closed loop in vector control strategy cannot be directly applied. At present, there is no better method that can simultaneously meet the following requirements in SV-MPCC system: 1) No need to add additional weak magnetic weight factors to the value function; 2) Not dependent on motor parameters; 3) Closed-loop control can be performed; 4) Will not cause the increase of current following error under single vector model predictive current control. Therefore, it is urgent to develop a new control method to improve the versatility and practicality of the method while meeting the above requirements. Summary of the invention

[0004] The purpose of the present invention is to solve the problem that the existing control methods cannot simultaneously meet the requirements of not needing to add additional weakening magnetic weight factors in the value function, not relying on motor parameters, being able to perform closed-loop control, and not causing an increase in current following error under single-vector model predicted current control. An induction motor model predictive current control method based on generalized flux error and closed-loop weakening magnetic strategy is proposed.

[0005] The technical solution adopted by the present invention to solve the above technical problems is:

[0006] A method for predictive current control of an induction motor model based on generalized flux linkage error and closed-loop flux weakening strategy, the method specifically comprising the following steps:

[0007] Step 1: Define the generalized flux vector reference value According to the current in the current control of the single vector model of the induction motor, a generalized flux linkage prediction value is defined, and the relationship between the basic generalized flux linkage vector and the basic voltage vector is obtained according to the generalized flux linkage prediction value;

[0008] Step 2: Based on the generalized flux vector reference value The generalized flux prediction value and the relationship between the basic generalized flux vector and the basic voltage vector are used to obtain the value function of the induction motor single vector model predictive current control;

[0009] Step 3: Obtain the optimal voltage vector according to the value function in step 2 The field weakening controller uses the optimal voltage vector The corresponding basic generalized flux vector generates the weakening current component i sdFW , and then use the weak magnetic current component i sdFW Compensate the excitation current;

[0010] And use the optimal voltage vector Perform model predictive current control.

[0011] Furthermore, the generalized flux vector reference value for:

[0012]

[0013] in, represents the reference current, σ is the leakage inductance factor, L s is the stator inductance.

[0014] Furthermore, in the step 1, a generalized flux prediction value is defined according to the current in the current control of the induction motor single vector model prediction, and a relationship between a basic generalized flux vector and a basic voltage vector is obtained according to the generalized flux prediction value; the specific process is:

[0015] Step 1. The discrete current expression in the single vector model predictive current control of the induction motor is:

[0016]

[0017] in, represents the current in the induction motor single vector model predictive current control at time k+1, represents the current in the single vector model predictive current control of the induction motor at time k+2, T sis the sampling period, R s is the stator resistance, L m is the mutual inductance, L r is the rotor inductance, R r is the rotor resistance, L s is the stator inductance, T r is the rotor time constant, ω r is the mechanical speed, σ is the leakage inductance factor, j represents the imaginary unit, It represents the predicted value of rotor flux;

[0018] Define the generalized flux prediction value ψ s1 :

[0019]

[0020] And set the initial value of generalized magnetic flux ψ s0 for:

[0021]

[0022] Then the discrete current expression after deformation is:

[0023]

[0024] Then the basic generalized magnetic flux vector The relationship with the basic voltage vector is:

[0025]

[0026] in,

[0027] Furthermore, the rotor time constant is:

[0028] T r =L r / R r .

[0029] Furthermore, the leakage inductance factor σ is:

[0030]

[0031] Furthermore, the specific process of step 2 is as follows:

[0032] Define the value function J as:

[0033]

[0034] Among them, |·| means taking the absolute value;

[0035] According to ψ s1 and Transform the value function J into:

[0036]

[0037] in, represents the error value,

[0038] Furthermore, the specific process of step three is:

[0039] Step 3. Calculate the basic generalized flux vector corresponding to each voltage vector:

[0040] Δψ n =T s u n (4)

[0041] Among them, Δψ n represents the basic generalized flux vector corresponding to the nth voltage vector, u n represents the nth voltage vector, n=0,1,...,7;

[0042] Step 32: From the basic generalized flux vectors corresponding to each voltage vector, select the basic generalized flux vector that minimizes the value function value.

[0043] Step 3: According to Calculate the optimal voltage vector Using the optimal voltage vector Generate a switching pulse signal for the inverter;

[0044] According to the selected basic generalized magnetic flux vector Generates field weakening current component i sdFW , and then use the weak magnetic current component i sdFW Compensate the excitation current.

[0045] Furthermore, according to Calculate the optimal voltage vector Specifically:

[0046]

[0047] Furthermore, the field weakening controller is based on the optimal voltage vector The corresponding basic generalized flux vector generates the weakening current component i sdFW ; Specifically:

[0048] 2×3 0.5 U dc T s / 9 is used as the given value of the PI controller in the field weakening control, and the optimal voltage vector Corresponding As the feedback value of PI controller in weak magnetic control;

[0049] when When the field weakening control generates the field weakening current component i sdFW :

[0050]

[0051] Among them, K p represents the proportional parameter of the PI controller, K i represents the integral parameter of the PI controller, s represents the complex frequency domain variable, and U dc Indicates the DC bus voltage;

[0052] when When , the weakening current component generated by the weakening control is 0.

[0053] The beneficial effects of the present invention are:

[0054] 1. The magnetic field weakening controller designed by the present invention uses the generalized flux linkage error as the control variable to perform closed-loop magnetic field weakening control, thereby expanding the speed range of the induction motor system and improving the load carrying capacity of the system;

[0055] 2. Compared with the traditional weak magnetic field control method based on the motor model, the weak magnetic field control strategy of the present invention does not introduce motor parameters, so it does not need to rely on motor parameters and has higher robustness;

[0056] 3. The method of the present invention performs real-time compensation for the current through a closed-loop control strategy, that is, when the speed change causes the expected generalized flux error to change, the real-time compensation of the excitation current in the weak magnetic region is achieved by feeding back the generalized flux error. Therefore, the present invention will not cause an increase in the current following error under the SV-MPCC control strategy;

[0057] 4. The present invention can be implemented based on the traditional value function without adding additional weak magnetic weight factors to the value function. BRIEF DESCRIPTION OF THE DRAWINGS

[0058] Figure 1 It is a control block diagram of the flux weakening control strategy system based on generalized flux linkage error;

[0059] The given value of generalized flux is obtained from the given current, and according to the initial value of generalized flux ψ s0 The definition of can be obtained by the feedback current to obtain the initial value of the generalized flux linkage, and the subtraction of the two can be obtained to obtain the generalized flux linkage error expected in the current cycle. according to The basic generalized flux vector Δψ corresponding to different basic output voltage vectors is obtained s0～7 , and finally in Δψ s0～7 Choose one and The closest vector is output as the current cycle Depend on It can be seen that is the basic voltage vector output in this cycle, and the switching pulse signal of the inverter is generated according to the basic voltage vector. With the basic generalized magnetic linkage vector relationship, get The maximum value is 2×3 0.5 U dc T s / 9. 2×3 0.5 U dc T s / 9 is the given value of the weak magnetic controller, is the feedback value, and the weak magnetic current component i is obtained through the PI controller sdFW , realize real-time compensation of excitation current in the weak magnetic field area and complete closed-loop weak magnetic field control;

[0060] Figure 2 is the spatial flux vector plane and the basic flux vector;

[0061] Figure 3 It is a schematic diagram of the change of magnetic flux vector in the weak magnetic region;

[0062] Figure 4a It is a performance diagram without the field weakening method when the motor initial speed is given as the rated speed and a step speed of 1.6 times the rated speed (2400rpm) is added in 0.2 seconds;

[0063] In the figure, ω r represents the speed, i sd represents the d-axis component of the stator current, i sq represents the q-axis component of the stator current, and M represents the modulation ratio;

[0064] Figure 4b It is a performance diagram of the open-loop field weakening method based on the motor model calculation when the motor initial speed is given as the rated speed and a step speed of 1.6 times the rated speed (2400rpm) is added in 0.2 seconds;

[0065] Figure 4c It is a performance diagram of the closed-loop magnetic weakening method proposed in the present invention accelerating to the magnetic weakening region when the motor's initial speed is given as the rated speed and a step speed of 1.6 times the rated speed (2400 rpm) is added in 0.2 seconds;

[0066] Figure 5a It is a schematic diagram of the load capacity of the system in the weak magnetic field area before the method proposed by the present invention is added;

[0067] Figure 5bIt is a schematic diagram of the load carrying capacity of the system in the weak magnetic field area after the method proposed by the present invention is added;

[0068] Figure 6a It is a schematic diagram of the control performance of the closed-loop magnetic weakening method proposed in the present invention when the stator inductance parameter deviates from the actual value;

[0069] Figure 6b It is a schematic diagram of the control performance of the traditional model-based field weakening method when the stator inductance parameter deviates from the actual value;

[0070] Figure 7a This is a test result diagram of the step acceleration capability of the weak magnetic field method proposed by the present invention;

[0071] Figure 7b This is a test result diagram of the weak magnetic region reversal capability of the weak magnetic method proposed in the present invention. DETAILED DESCRIPTION

[0072] Specific implementation method 1: Combination Figure 1 The present embodiment describes an induction motor model predictive current control method based on generalized flux error and closed-loop flux weakening strategy, and the method specifically includes the following steps:

[0073] Step 1: Define the generalized flux vector reference value A generalized flux prediction value is defined according to the current in the single vector model predictive current control (SV-MPCC) of the induction motor, and the relationship between the basic generalized flux vector and the basic voltage vector is obtained according to the generalized flux prediction value;

[0074] Step 2: Based on the generalized flux vector reference value The generalized flux prediction value and the relationship between the basic generalized flux vector and the basic voltage vector are used to obtain the value function of the induction motor single vector model predictive current control;

[0075] Step 3: Obtain the optimal voltage vector according to the value function in step 2 The field weakening controller uses the optimal voltage vector The corresponding basic generalized flux vector generates the weakening current component i sdFW , and then use the weak magnetic current component i sdFW Compensate the excitation current;

[0076] And use the optimal voltage vector Perform model predictive current control (i.e. complete inverter control based on the optimal voltage vector).

[0077] Specific implementation method 2: This implementation method is different from the specific implementation method 1 in that the generalized flux vector reference value for:

[0078]

[0079] in, represents the reference current, σ is the leakage inductance factor, L s is the stator inductance.

[0080] The other steps and parameters are the same as those in the first embodiment.

[0081] Specific implementation method three: This implementation method is different from specific implementation method one or two in that, in step one, a generalized flux linkage prediction value is defined according to the current in the single vector model predictive current control (SV-MPCC) of the induction motor, and the relationship between the basic generalized flux linkage vector and the basic voltage vector is obtained according to the generalized flux linkage prediction value; the specific process is:

[0082] Step 1. The discrete current expression in the single vector model predictive current control of the induction motor is:

[0083]

[0084] in, represents the current in the induction motor single vector model predictive current control at time k+1, represents the current in the single vector model predictive current control of the induction motor at time k+2, T s is the sampling period, R s is the stator resistance, L m is the mutual inductance, L r is the rotor inductance, R r is the rotor resistance, L s is the stator inductance, T r is the rotor time constant, ω r is the mechanical speed, σ is the leakage inductance factor, j represents the imaginary unit, It represents the predicted value of rotor flux;

[0085] Define the generalized flux prediction value ψ s1 :

[0086]

[0087] And set the initial value of generalized magnetic flux ψ s0 for:

[0088]

[0089] Then the discrete current expression after deformation is:

[0090]

[0091] Then the basic generalized magnetic flux vector The relationship with the basic voltage vector is:

[0092]

[0093] in,

[0094] The other steps and parameters are the same as those in the first or second embodiment.

[0095] Specific implementation method 4: This implementation method is different from any one of the specific implementation methods 1 to 3 in that the rotor time constant is:

[0096] T r =L r / R r

[0097] The other steps and parameters are the same as those in Specific Embodiments 1 to 3.

[0098] Specific implementation mode 5: This implementation mode is different from any one of the specific implementation modes 1 to 4 in that the leakage inductance factor σ is:

[0099]

[0100] The other steps and parameters are the same as those in Specific Embodiments 1 to 4.

[0101] Specific implementation method 6: This implementation method is different from the specific implementation methods 1 to 5 in that the specific process of step 2 is as follows:

[0102] In the classic single vector model predictive current control of induction motor, the cost function J is defined as:

[0103]

[0104] Among them, |·| means taking the absolute value;

[0105] According to ψ s1 and Transform the value function J into:

[0106]

[0107] in, represents the error value,

[0108] The other steps and parameters are the same as those in Specific Implementations 1 to 5.

[0109] In this implementation, the defined generalized flux is introduced into the classical value function in the single vector model predictive current control of the induction motor, thereby obtaining a new value function without adding an additional magnetic weakening weight factor to the value function.

[0110] Specific implementation method 7: This implementation method is different from the specific implementation methods 1 to 6 in that the specific process of step 3 is as follows:

[0111] Step 3. Calculate the basic generalized flux vector corresponding to each voltage vector:

[0112] Δψ n =T s u n (4)

[0113] Among them, Δψ n represents the basic generalized flux vector corresponding to the nth voltage vector, u n represents the nth voltage vector, n=0,1,...,7;

[0114] Step 32: From the basic generalized flux vectors corresponding to each voltage vector, select the basic generalized flux vector that minimizes the value function value. (Right now From Δψ n , n=0,1,...,7);

[0115] Step 3: According to Calculate the optimal voltage vector Using the optimal voltage vector Generate switching pulse signals for the inverter to control the inverter;

[0116] According to the selected basic generalized magnetic flux vector (i.e. optimal voltage vector The corresponding basic generalized flux vector) generates the weak magnetic current component i sdFW , and then use the weak magnetic current component i sdFW Compensate the excitation current.

[0117] The other steps and parameters are the same as those in Specific Embodiments 1 to 6.

[0118] In this embodiment, by selecting the basic generalized flux vector that minimizes the value of the cost function, the tracking error of the stator current vector can be minimized.

[0119] Specific implementation eight: This implementation differs from specific implementations one to seven in that the Calculate the optimal voltage vector Specifically:

[0120]

[0121] The other steps and parameters are the same as those in Specific Embodiments 1 to 7.

[0122] Specific implementation method 9: This implementation method is different from any one of the specific implementation methods 1 to 8 in that the field weakening controller is based on the optimal voltage vector The corresponding basic generalized flux vector generates the weakening current component i sdFW ; Specifically:

[0123] 2×3 0.5 U dc T s / 9 is used as the given value of the PI controller in the field weakening control, and the optimal voltage vector Corresponding As the feedback value of PI controller in weak magnetic control;

[0124] when When the field weakening control generates the field weakening current component i sdFW :

[0125]

[0126] Among them, K p represents the proportional parameter of the PI controller, K i represents the integral parameter of the PI controller, s represents the complex frequency domain variable, and U dc Indicates the DC bus voltage;

[0127] when When , the weakening current component generated by the weakening control is 0.

[0128] The other steps and parameters are the same as those in Specific Embodiments 1 to 8.

[0129] like Figure 2 and Figure 3 As shown, Represents the length of line segment OA. s1 , Δψ s2 and Δψ s0 / Δψ s7 In the triangle formed by the generalized flux error, for any end point not at point A Generalized flux error The difference between the four basic generalized flux vectors will definitely be less than Therefore, the present invention uses the length of line segment OA as the given value of the PI controller in the field weakening control. The field weakening current component i is obtained by the PI controller. sdFW , limiting the output of the field weakening controller to [-2×3 0.5 U dc T s / 9,0] range; and use this as the compensation value to build a weak magnetic closed loop to achieve real-time compensation of the excitation current in the weak magnetic area.

[0130] The effectiveness of the method proposed in the present invention is verified by the experimental results of the accompanying drawings:

[0131] Figure 4a , Figure 4b and Figure 4c They are performance comparison diagrams of acceleration to the weak magnetic field area by the open-loop weak magnetic field method without weak magnetic field, the open-loop weak magnetic field method based on motor model calculation, and the closed-loop weak magnetic field method proposed in the present invention. The initial speed of the motor is given as the rated speed, and a step speed of 1.6 times the rated speed (2400rpm) is added in 0.2 seconds. It can be seen that the weak magnetic field strategies of the traditional method and the method proposed in the present invention can both improve the operating range of the motor, but the acceleration time of the method proposed in the present invention is faster and the voltage saturation duration is shorter.

[0132] Figure 5a and Figure 5b It is a comparison of the load capacity of the system in the weak magnetic zone before and after the method proposed in the present invention is added. The motor speed is 1650 rpm, and the load suddenly increases from 35% of the rated torque to 70% of the rated torque in 0.2 seconds, which means that the addition of the method proposed in the present invention improves the load capacity of the system in the weak magnetic zone.

[0133] Figure 6a and Figure 6b The control performance comparison of the closed-loop field weakening method proposed by the present invention and the traditional model-based field weakening method when the stator inductance parameter deviates from the actual value is shown in Figure 2. In the field weakening area (1800rpm), L s From 1.0s, it becomes 1.2 times the actual value (1.2L s ), and at 3.0s it starts to become 0.8 times the actual value (0.8L s ), and returns to the initial value (L s ) experimental results; it can be seen that when the parameters change, the speed, current and voltage of the closed-loop weak magnetic method proposed in the present invention do not fluctuate significantly. Therefore, the weak magnetic method proposed in the present invention is much less dependent on parameters than the traditional method and has better parameter robustness.

[0134] Figure 7a and Figure 7b This is a test of the step acceleration and reversal capability of the weak magnetic field method proposed in the present invention. During acceleration, the speed increases from the base speed, and is gradually increased by 1.2 times, 1.4 times, and 1.6 times the rated speed. The initial speed during reversal is 1.2 times the rated speed (1800rpm), and the speed reverses to -1.2 times the rated speed at 0.2s. It can be seen that the weak magnetic field method based on error voltage proposed in the present invention can meet the requirements of acceleration and reversal in the weak magnetic field area at the same time.

[0135] The above calculation examples of the present invention are only used to explain the calculation model and calculation process of the present invention in detail, and are not intended to limit the implementation methods of the present invention. For ordinary technicians in the relevant field, other different forms of changes or modifications can be made based on the above description. It is impossible to list all the implementation methods here. All obvious changes or modifications derived from the technical solution of the present invention are still within the scope of protection of the present invention.

Claims

1. An induction motor model predictive current control method based on generalized flux error and closed-loop field weakening strategy, characterized in that: The method specifically comprises the following steps: Step 1: Define the generalized flux vector reference value According to the current in the current control of the single vector model of the induction motor, a generalized flux linkage prediction value is defined, and the relationship between the basic generalized flux linkage vector and the basic voltage vector is obtained according to the generalized flux linkage prediction value; Step 2: Based on the generalized flux vector reference value The generalized flux prediction value and the relationship between the basic generalized flux vector and the basic voltage vector are used to obtain the value function of the induction motor single vector model predictive current control; Step 3: Obtain the optimal voltage vector according to the value function in step 2 The field weakening controller uses the optimal voltage vector The corresponding basic generalized flux vector generates the weakening current component i sdFW , and then use the weak magnetic current component i sdFW Compensate the excitation current; And use the optimal voltage vector Perform model predictive current control.

2. The induction motor model predictive current control method based on generalized flux error and closed-loop flux weakening strategy according to claim 1, characterized in that: The generalized flux vector reference value for: in, represents the reference current, σ is the leakage inductance factor, L s is the stator inductance.

3. The induction motor model predictive current control method based on generalized flux error and closed-loop flux weakening strategy according to claim 2, characterized in that: In the step 1, a generalized flux prediction value is defined according to the current in the current control of the induction motor single vector model prediction, and the relationship between the basic generalized flux vector and the basic voltage vector is obtained according to the generalized flux prediction value; the specific process is: Step 1. The discrete current expression in the single vector model predictive current control of the induction motor is: in, represents the current in the induction motor single vector model predictive current control at time k+1, represents the current in the single vector model predictive current control of the induction motor at time k+2, T s is the sampling period, R s is the stator resistance, L m is the mutual inductance, L r is the rotor inductance, R r is the rotor resistance, L s is the stator inductance, T r is the rotor time constant, ω r is the mechanical speed, σ is the leakage inductance factor, j represents the imaginary unit, It represents the predicted value of rotor flux; Define the generalized flux prediction value ψ s1 : And set the initial value of generalized magnetic flux ψ s0 for: Then the discrete current expression after deformation is: Then the basic generalized magnetic flux vector The relationship with the basic voltage vector is: in, 4. The induction motor model predictive current control method based on generalized flux error and closed-loop flux weakening strategy according to claim 3 is characterized in that: The rotor time constant is: T r =L r / R r 。 5. The induction motor model predictive current control method based on generalized flux error and closed-loop flux weakening strategy according to claim 4, characterized in that: The leakage inductance factor σ is:

6. The induction motor model predictive current control method based on generalized flux error and closed-loop flux weakening strategy according to claim 5, characterized in that: The specific process of step 2 is as follows: Define the value function J as: Among them, |·| means taking the absolute value; According to ψ s1 and Transform the value function J into: in, represents the error value, 7. The induction motor model predictive current control method based on generalized flux error and closed-loop flux weakening strategy according to claim 6, characterized in that: The specific process of step three is: Step 31. Calculate the basic generalized flux vector corresponding to each voltage vector: Dp n =T s you n (4) Among them, Δψ n represents the basic generalized flux vector corresponding to the nth voltage vector, u n represents the nth voltage vector, n=0,1,...,7; Step 32: From the basic generalized flux vectors corresponding to each voltage vector, select the basic generalized flux vector that minimizes the value function value. Step 3: According to Calculate the optimal voltage vector Using the optimal voltage vector Generate a switching pulse signal for the inverter; According to the selected basic generalized magnetic flux vector Generates field weakening current component i sdFW , and then use the weak magnetic current component i sdFW Compensate the excitation current.

8. The induction motor model predictive current control method based on generalized flux error and closed-loop flux weakening strategy according to claim 7, characterized in that: The basis Calculate the optimal voltage vector Specifically:

9. The induction motor model predictive current control method based on generalized flux error and closed-loop flux weakening strategy according to claim 8, characterized in that: The field weakening controller is based on the optimal voltage vector The corresponding basic generalized flux vector generates the weakening current component i sdFW ; Specifically: 2×3 0.5 U dc T s / 9 is used as the given value of the PI controller in the field weakening control, and the optimal voltage vector Corresponding As the feedback value of PI controller in weak magnetic control; when When the field weakening control generates the field weakening current component i sdFW : Among them, K p represents the proportional parameter of the PI controller, K i represents the integral parameter of the PI controller, s represents the complex frequency domain variable, and U dc Indicates the DC bus voltage; when When , the weakening current component generated by the weakening control is 0.

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