Fast phase synchronization method based on model prediction flux linkage control under low carrier wave ratio

Abstract

The invention belongs to the technical field of motor control, and provides a rapid phase synchronization method based on model prediction flux linkage control under a low carrier wave ratio in order to solve the problem that a phase error exists between a voltage instruction phase of a closed-loop controller and a preset sampling point of synchronous optimization pulse width modulation. Comprising the following steps: S1, calculating a voltage instruction without sampling period compensation at the current moment; s2, determining a preset sampling position closest to the phase of the voltage instruction at the current moment; s3, calculating a sampling period compensation value; and S4, updating the voltage instruction after sampling period compensation. According to the rapid phase synchronization method designed by the invention, the accurate correspondence between the voltage instruction phase and the preset sampling point position is ensured, and the low harmonic characteristic of synchronous optimization pulse width modulation can be reproduced when the control system operates in a quasi-steady state. The method is established on derivation and analysis of an equivalent relation between a motor equation and an electromagnetic torque, so that the method is suitable for all AC motor closed-loop control based on synchronous optimization pulse width modulation.

Description

Technical Field

[0001] The present invention belongs to the technical field of motor control, and in particular relates to a fast phase synchronization method based on model prediction flux control under low carrier ratio. Background Art

[0002] For high-power or high-speed induction motor drive systems, the switching frequency is usually low due to switching losses and heat dissipation conditions. At this time, the carrier ratio (the ratio of switching frequency to fundamental frequency) is low. Under low carrier ratio, the low-order harmonic content in the traditional asynchronous modulation output voltage waveform is high. Therefore, in order to improve the control performance of the drive system, reduce current harmonics, and meet the constraints of the highest switching frequency, multi-mode pulse width modulation is usually used, that is, asynchronous modulation is used in the low-speed area, and synchronous optimized pulse width modulation strategy is used in the medium and high-speed area.

[0003] Under low carrier ratio, the proportional-integral regulator in traditional vector control does not consider the cross-coupling of the motor model itself and the one-shot delay problem in the digital control system, and the extension of the sampling period will aggravate the coupling effect, resulting in the deterioration of the dynamic response capability of the system. Even when the system is in quasi-steady-state operation, its dynamic response capability is still poor, which seriously affects the control performance of the motor drive system. Therefore, model predictive control (MPC) has attracted widespread attention in academia and industry in recent years due to its simple principle, multivariable control, and easy processing of nonlinear constraints such as low carrier ratio and common mode voltage.

[0004] However, the essence of model predictive control is to select the optimal voltage vector through online rolling optimization based on the principle of minimizing the objective function, so as to achieve optimal control of the system. When the control variables have different unit dimensions in the objective function, it is necessary to introduce weight coefficients to achieve simultaneous control of multiple variables. Since the design of weight coefficients lacks theoretical support, it is usually necessary to go through a large number of simulation experiments to select the weight coefficients of the optimized objective function, and the process is complicated and cumbersome. Model predictive flux control (MPFC) is improved on the basis of MPC, taking the equivalent stator flux vector as the control target, avoiding the complex design of weight coefficients and further improving practicality.

[0005] The main synchronous optimization pulse width modulation strategies include specific subharmonic elimination pulse width modulation (SHEPWM), current harmonic minimum pulse width modulation (CHMPWM) and synchronous space vector pulse width modulation (SSVM). The three synchronous modulation strategies must meet the symmetry requirements of synchronous modulation, namely synchronization, three-phase symmetry, and half-wave symmetry to eliminate the even harmonics of the output voltage waveform. Among them, the optimization switch angle position calculation of SHEPWM and CHMPWM is very complicated. Usually, the output voltage waveform is optimized offline within the fundamental period under the assumption of steady-state operating conditions, and it is difficult to match high-performance closed-loop controllers. SSVM calculates the switch angle in real time based on the volt-second balance principle. There is no harmonic component at the sampling position, and there is no need to separate the fundamental and harmonic components. Compared with the other two synchronous modulation strategies, it is easier to achieve closed-loop control and is often widely used as a quasi-optimal modulation strategy.

[0006] However, in digital control systems, the voltage command phase may change dramatically at any time during the dynamic process. There is a phase error between the voltage command phase and the preset sampling position, which makes it difficult to meet the symmetry requirements of synchronous modulation.

[0007] In order to reproduce the low harmonic characteristics of SSVM, the existing method calculates the sampling period compensation value based on the phase error and applies it to the voltage command calculation of the next beat. Although the traditional phase correction method has a good phase correction effect in steady state, there is still a large phase error in the dynamic process, especially when the torque increases suddenly, and it takes a long time to achieve phase synchronization.

[0008] Therefore, it is necessary to develop a phase synchronization method with faster dynamic response speed and better compensation effect. Summary of the invention

[0009] In view of the above-mentioned technical problems, the present invention proposes a phase synchronization method with faster dynamic response speed and better compensation effect. The present invention effectively eliminates the phase synchronization error between the voltage command phase and the preset sampling position by dynamic time compensation of the sampling period. Compared with the traditional method that can only remedy the phase error after it occurs, the present invention achieves faster phase correction by reconstructing the voltage command, ensuring that the control system can reproduce the low harmonic characteristics of SSVM during quasi-steady-state operation.

[0010] In order to achieve the above technical effects, the present invention proposes a fast phase synchronization method based on model prediction flux control under low carrier ratio, comprising:

[0011] S1: Calculate the voltage command without sampling period compensation at the current moment;

[0012] S2: Based on S1, confirm the preset sampling position that is closest to the voltage command phase at the current moment;

[0013] S3: Based on S2, calculate the sampling period compensation value;

[0014] S4: Based on S3, update the voltage command after sampling period compensation.

[0015] Preferably, S1 includes:

[0016] S1.1: Calculate the phase of the stator flux vector reference value without sampling period compensation, specifically:

[0017]

[0018] In the formula, is the modulus of the stator flux reference value, is the stator flux vector reference phase without sampling period compensation, is the rotor flux phase at time k+1, N p is the number of pole pairs of the motor, is the modulus of the rotor flux vector at time k+1, is the electromagnetic torque, θ T is the load angle of electromagnetic torque, λ=1 / L s L r -L m 2 , L S , L r , L M They are the motor stator inductance, rotor inductance and stator-rotor mutual inductance;

[0019] S1.2: Stator flux vector reference value without sampling period compensation for:

[0020]

[0021] Where, e is the exponential function of the natural logarithm, and j is the imaginary unit;

[0022] S1.3: Voltage command without sampling period compensation for:

[0023]

[0024] In the formula, is the sampling period at time k+2, is the stator flux vector at time k+1, is the stator current vector at time k+1, R s is the stator resistance;

[0025] Preferably, in S2:

[0026] The preset sampling position θ closest to the S1 voltage command rfor:

[0027]

[0028] Where m is the sector where the voltage command is closest to the preset sampling position, and round is a function used to round the number to the nearest integer. is the voltage command phase, θ0 is the first preset sampling position corresponding to the voltage command in the first sector, and N is the number of sampling positions in a sector.

[0029] Preferably, S3 is specifically:

[0030] S3.1: Assume that the sampling period is k+1 After compensation, the voltage command phase is eliminated With the preset sampling position θ r The phase deviation between them achieves precise synchronization. The sampling period at time k+1 after compensation It is expressed as:

[0031]

[0032] In the formula, f e is the fundamental frequency, is the time compensation value of the sampling period at time k+1;

[0033] S3.2: Time compensation value It can be expressed as:

[0034]

[0035] In the formula, ω e Expressed as the rotational angular velocity of the rotor flux vector, and They represent the real and imaginary parts of the stator flux vector at time k+1 respectively.

[0036] Preferably, the voltage command after S4 compensation for:

[0037]

[0038] In the formula, is the phase increment of the stator flux reference value after sampling period compensation at time k+1.

[0039] Compared with the prior art, the present invention has the following beneficial effects:

[0040] 1. The present invention effectively eliminates the phase synchronization error between the voltage command phase and the preset sampling position by dynamically compensating the sampling period. Compared with the traditional method that can only remedy the phase error after it occurs, the present invention achieves faster phase correction by reconstructing the voltage command, ensuring that the control system can reproduce the low harmonic characteristics of SSVM during quasi-steady-state operation.

[0041] 2. When solving the compensation value of the sampling period, the present invention does not require parameter setting and has good parameter robustness.

[0042] 3. The fast phase synchronization method designed in the present invention is based on the derivation and analysis of the equivalent relationship between the motor equation and the electromagnetic torque, and is therefore applicable to all AC motor closed-loop controls based on synchronous optimized pulse width modulation. BRIEF DESCRIPTION OF THE DRAWINGS

[0043] The accompanying drawings are used to provide further understanding of the present invention and constitute a part of the specification. They are used to explain the present invention together with the embodiments of the present invention and do not constitute a limitation of the present invention.

[0044] In the attached picture:

[0045] Figure 1 This is a block diagram of model-predictive flux control based on fast phase synchronization;

[0046] Figure 2 is a flow chart of the proposed fast phase synchronization method;

[0047] exist Figure 3 In the figure, (a) is the simulation waveform of SSVM when the torque suddenly increases from 0 to 500Nm under BBCS_11 when using MPFC based on the traditional phase correction method; (b) is the simulation waveform of SSVM when the torque suddenly increases from 0 to 500Nm under BBCS_11 when using MPFC based on the fast phase synchronization method.

[0048] exist Figure 4 In the figure, (a) is the dynamic process of SSVM switching from BBCS_7 to BBCS_5 when using MPFC based on the traditional phase correction method; (b) is the dynamic process of SSVM switching from BBCS_7 to BBCS_5 when using MPFC based on the proposed fast phase synchronization method.

[0049] exist Figure 5In the figure, (a) is the experimental waveform of SSVM when the torque suddenly increases to 14Nm under BBCS_11 when using MPFC based on traditional phase correction method; (b) is the experimental waveform of SSVM when the torque suddenly increases to 14Nm under BBCS_11 when using MPFC based on the proposed fast phase synchronization method; (c) is the experimental waveform of SSVM when the torque suddenly increases to 14Nm under BBCS_7 when using MPFC based on traditional phase correction method; (d) is the experimental waveform of SSVM when the torque suddenly increases to 14Nm under BBCS_7 when using MPFC based on the proposed fast phase synchronization method.

[0050] exist Figure 6 In the figure, (a) is the experimental waveform when SSVM switches from BBCS_11 to BBCS_7 when using MPFC based on the traditional phase correction method; (b) is the experimental waveform when SSVM switches from BBCS_11 to BBCS_7 when using MPFC based on the proposed fast phase synchronization method; (c) is the experimental waveform when SSVM switches from BBCS_7 to BBCS_5 when using MPFC based on the traditional phase correction method; (d) is the experimental waveform when SSVM switches from BBCS_7 to BBCS_5 when using MPFC based on the proposed fast phase synchronization method.

[0051] Figure 7 The present invention is a flow chart of the method. DETAILED DESCRIPTION

[0052] The following is combined with Figure 1 -Attached Figure 7 The preferred embodiments of the present invention are described. It should be understood that the preferred embodiments described herein are only used to illustrate and explain the present invention, and are not used to limit the present invention.

[0053] Example:

[0054] A fast phase synchronization method based on model prediction flux control under low carrier ratio, comprising:

[0055] S1: Calculate the voltage command without sampling period compensation at the current moment;

[0056] In the two-phase stationary reference coordinate system, the stator flux ψ s and the stator current i s As a state variable, the mathematical model of the induction motor in the two-phase stationary coordinate system is as follows:

[0057] px=Ax+Bu s ,

[0058] In the formula,

[0059]

[0060] x=[ψ s i s ] T ;

[0061] p represents a differential operator; R s , R r are the stator and rotor resistance of the motor respectively; L s , L r , L m They are the motor stator inductance, rotor inductance and stator-rotor mutual inductance; ω r is the motor speed; u s is the voltage command;

[0062] λ=1 / L s L r -L m 2 .

[0063] According to the above formula, the continuous domain is constructed with the stator flux ψ s and the stator current i s is the full-order observer model for the state variables:

[0064]

[0065] In the formula, is represented by the observed value of ; G is the feedback matrix, preferably, G can be expressed as:

[0066]

[0067] The full-order observer model is discretized using the trapezoidal integration method and expressed as:

[0068]

[0069] In the formula, is the predicted correction variable at time k+1; x k+1 is the predicted state variable at time k+1; x k is the state variable at time k; is the sampling period at time k.

[0070] According to the discretized formula, the stator current at time k+1 is estimated and stator flux Solve for the rotor flux at time k+1 Specifically:

[0071]

[0072] S1.1: Calculate the phase of the stator flux reference value based on the equivalent relationship between the stator flux, rotor flux and torque, specifically:

[0073]

[0074] In the formula, is the modulus of the stator flux reference value; is the phase of the stator flux vector reference value without sampling period compensation; is the rotor flux phase at time k+1; N p is the number of pole pairs of the motor, is the modulus of the rotor flux vector at time k+1; is the electromagnetic torque; θ T is the load angle of the electromagnetic torque.

[0075] S1.2: Stator flux vector reference value without sampling period compensation for:

[0076]

[0077] Where e is the exponential function of the natural logarithm; j is the imaginary unit.

[0078] S1.6: Current voltage command without sampling period compensation for:

[0079]

[0080] In the formula, is the sampling period at time k+2, is the stator flux vector at time k+1, is the stator current vector at time k+1, R s is the stator resistance.

[0081] S2: Based on S1, confirm the preset sampling position that is closest to the voltage command phase at the current moment; S2 is specifically:

[0082] The preset sampling position θ that is closest to the current voltage command phase of S1 r for:

[0083]

[0084] Where m is the sector where the voltage command is closest to the preset sampling position, and round is a function used to round the number to the nearest integer. is the voltage command phase at the current moment, θ0 is the first preset sampling position corresponding to the voltage command in the first sector, and N is the number of sampling positions in a sector.

[0085] S3: Based on S2, calculate the sampling period compensation value. Specifically:

[0086] S3.1: Assume that the sampling period is k+1 After compensation, the voltage command phase is eliminated With the preset sampling position θ r The phase deviation between them achieves precise synchronization. The sampling period at time k+1 after compensation It is expressed as:

[0087]

[0088] In the formula, f e is the fundamental frequency, is the time compensation value of the sampling period at time k+1;

[0089] S3.2: Time compensation value It can be expressed as:

[0090]

[0091] In the formula, ω e Expressed as the rotational angular velocity of the rotor flux vector, and They represent the real and imaginary parts of the stator flux vector at time k+1 respectively.

[0092] S4: Based on S3, update the voltage command after sampling period compensation. Specifically:

[0093] Based on the time compensation value obtained by S3.2, the compensated voltage command for:

[0094]

[0095] In the formula, is the phase increment of the stator flux reference value after sampling period compensation at time k+1.

[0096] Verification experiment:

[0097] exist Figure 3 In the figure, (a) is the simulation waveform when the torque of SSVM suddenly increases from 0 to 500Nm under BBCS_11 when the MPFC based on the traditional phase correction method is used; (b) is the simulation waveform when the torque of SSVM suddenly increases from 0 to 500Nm under BBCS_11 when the MPFC based on the proposed fast phase synchronization method is used; from top to bottom in the figure are line voltage pulse, voltage command phase, phase error and sampling period. Figure 3As shown in the figure, both methods can achieve fast torque tracking, and the stabilization time is about 2ms. When the traditional phase correction method is used, the voltage command θ u The phase of the sample is obviously different from the preset sampling position, and the phase error Δθ u It takes about 13ms to recover to 0. In contrast, the fast phase synchronization method can adjust the sampling period in time to prevent a large phase error from occurring.

[0098] exist Figure 4 In the figure, (a) is the dynamic process of SSVM switching from BBCS_7 to BBCS_5 when using MPFC based on the traditional phase correction method; (b) is the dynamic process of SSVM switching from BBCS_7 to BBCS_5 when using MPFC based on the proposed fast phase synchronization method. From top to bottom in the figure are line voltage pulse, voltage command phase, phase error and sampling period. Figure 4 As shown in the figure, the results show that during the PWM mode switching process, the traditional phase correction method has obvious phase errors and takes a long time to correct, while the proposed fast phase synchronization method is almost unaffected during the entire switching process. This test further confirms the good transient performance of the proposed fast phase synchronization method.

[0099] exist Figure 5 In the figure, (a) is the experimental waveform when the torque of SSVM suddenly increases to 14Nm under BBCS_11 when using MPFC based on the traditional phase correction method; (b) is the experimental waveform when the torque of SSVM suddenly increases to 14Nm under BBCS_11 when using MPFC based on the proposed fast phase synchronization method; (c) is the experimental waveform when the torque of SSVM suddenly increases to 14Nm under BBCS_7 when using MPFC based on the traditional phase correction method; (d) is the experimental waveform when the torque of SSVM suddenly increases to 14Nm under BBCS_7 when using MPFC based on the proposed fast phase synchronization method. From top to bottom in the figure are the voltage command phase, phase error, torque calculation value and reference value, line voltage pulse and three-phase current. Figure 5 As shown in the figure, in all tests, the torque can quickly track its reference value, while the MPFC based on the traditional phase correction method has a large phase error and a slow error decay. In contrast, the MPFC based on the proposed phase synchronization method is almost not affected by torque mutations, can respond quickly in the dynamic process, and achieve accurate phase synchronization.

[0100] exist Figure 6In the figure, (a) is the experimental waveform when SSVM switches from BBCS_11 to BBCS_7 when using MPFC based on the traditional phase correction method; (b) is the experimental waveform when SSVM switches from BBCS_11 to BBCS_7 when using MPFC based on the proposed fast phase synchronization method; (c) is the experimental waveform when SSVM switches from BBCS_7 to BBCS_5 when using MPFC based on the traditional phase correction method; (d) is the experimental waveform when SSVM switches from BBCS_7 to BBCS_5 when using MPFC based on the proposed fast phase synchronization method. From top to bottom in the figure are voltage command phase, phase error, modulation strategy identifier, line voltage pulse and three-phase current. Figure 6 As shown in the figure, in all tests, smooth switching between different PWM strategies can be achieved. In steady state, the voltage command phase is consistent with the preset sampling point, and the phase error Δθ u Almost 0.

[0101] The above shows and describes the basic principles, main features and advantages of the present invention. It should be understood by those skilled in the art that the present invention is not limited to the above embodiments, and the above embodiments and descriptions are only for explaining the principles of the present invention. Without departing from the spirit and scope of the present invention, the present invention may have various changes and improvements, which fall within the scope of the present invention to be protected. The scope of protection of the present invention is defined by the attached claims and their equivalents.

Claims

1. A fast phase synchronization method based on model prediction flux control under low carrier ratio, characterized by: include: S1: Calculate the voltage command without sampling period compensation at the current moment; S2: Based on S1, confirm the preset sampling position that is closest to the voltage command phase at the current moment; S3: Based on S2, calculate the sampling period compensation value; S4: Based on S3, update the voltage command after sampling period compensation.

2. According to claim 1, a fast phase synchronization method based on model prediction flux control under low carrier ratio, characterized in that: S1 includes: S1.1: Calculate the phase of the stator flux vector reference value without sampling period compensation, specifically: In the formula, is the modulus of the stator flux reference value, is the stator flux vector reference phase without sampling period compensation, is the rotor flux phase at time k+1, N p is the number of pole pairs of the motor, is the modulus of the rotor flux vector at time k+1, is the electromagnetic torque reference value, θ T is the load angle of electromagnetic torque, λ=1 / L s L r -L m 2 , L S , L r , L M They are the motor stator inductance, rotor inductance and stator-rotor mutual inductance; S1.2: Stator flux vector reference value without sampling period compensation for: Where, e is the exponential function of the natural logarithm, and j is the imaginary unit; S1.3: Current voltage command without sampling period compensation for: In the formula, is the sampling period at time k+2, is the stator flux vector at time k+1, is the stator current vector at time k+1, R s is the stator resistance.

3. According to claim 2, a fast phase synchronization method based on model prediction flux control under low carrier ratio, characterized in that: S2 is specifically: The preset sampling position θ that is closest to the current voltage command phase of S1 r for: Where m is the sector where the voltage command is closest to the preset sampling position, and round is a function used to round the number to the nearest integer. is the voltage command phase at the current moment, θ0 is the first preset sampling position corresponding to the voltage command in the first sector, and N is the number of sampling positions in a sector.

4. The fast phase synchronization method based on model prediction flux control under low carrier ratio according to claim 3 is characterized in that: S3 is specifically: S3.1: Assume that the sampling period is k+1 After compensation, the voltage command phase is eliminated With the preset sampling position θ r The phase deviation between them achieves precise synchronization. The sampling period at time k+1 after compensation It is expressed as: In the formula, f e is the fundamental frequency, is the time compensation value of the sampling period at time k+1; S3.2: Time compensation value It can be expressed as: In the formula, ω e Expressed as the rotational angular velocity of the rotor flux vector, and They represent the real and imaginary parts of the stator flux vector at time k+1 respectively.

5. The fast phase synchronization method based on model prediction flux control under low carrier ratio according to claim 4 is characterized in that: S4 compensated voltage command for: In the formula, is the phase increment of the stator flux reference value after sampling period compensation at time k+1.

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