Permanent magnet synchronous motor current control method of adjustable switching frequency FCS-MPC

Abstract

The invention relates to a permanent magnet synchronous motor current control method of an adjustable switching frequency FCS-MPC. Compared with the prior art, the defect that the switching frequency of an inverter is not fixed in current control of an existing FCS-MPC permanent magnet synchronous motor system is overcome. The method comprises the following steps: acquiring data of a driving system; constructing a discrete current prediction control model; determining a cost function and a delay compensation strategy; calculating a current error control band degree; selecting an optimal voltage vector; and outputting the voltage vector. According to the invention, the unique advantages of the FCS-MPC are reserved, and the defect that the inherent switching frequency is not fixed is overcome, so that the FCS-MPC can operate under the specified switching frequency in a wide speed regulation range; by introducing the PI type switching frequency controller, the zero-static-error tracking of the switching frequency reference value can be realized, and the control method realizes the effective regulation and control of the switching frequency on the basis of ensuring the current control performance of the permanent magnet motor.

Description

Technical Field

[0001] The invention relates to the technical field of permanent magnet synchronous motor current control, in particular to a permanent magnet synchronous motor current control method with adjustable switching frequency FCS-MPC. Background Art

[0002] Permanent magnet synchronous motors have the advantages of high efficiency and high power density, and have been widely used in new energy vehicles, servo drives and other fields. In order to achieve high-performance control of permanent magnet synchronous motors, many scholars and engineering researchers have proposed a variety of advanced control strategies. Among them, the Finite Control Set Model Predictive Control (FCS-MPC) algorithm, as a potential advanced control algorithm, has been applied in the field of motor drives. FCS-MPC can make full use of the inherent discrete characteristics of the inverter, directly control the switching state of the inverter, and obtain superior current dynamic control performance. In addition, the cost function design of FCS-MPC has a high degree of design flexibility, and the cost function design can achieve the optimal control of multiple control objectives and nonlinear terms of the system under the constraints.

[0003] However, the permanent magnet synchronous motor system of FCS-MPC does not have an inverter modulation link, and whether the inverter switching state is switched depends on the cost function value, which causes its inverter switching frequency to often change with the operating conditions. For example, the DC bus voltage, the motor's operating speed, the load, and the sampling time will all affect the switching frequency, resulting in the system being unable to achieve fixed switching frequency operation within a wide speed regulation range and the inverter switching loss and thermal stress being difficult to quantify. This poses technical challenges to the current control of the FCS-MPC permanent magnet synchronous motor system.

[0004] There are currently two main technical solutions to solve this problem. The first technical solution is to reintroduce the modulation link and use multiple basic voltage vectors in one control cycle to solve the problem of non-fixed switching frequency in the permanent magnet synchronous motor system of FCS-MPC. The use of multiple basic voltage vectors also effectively improves the current control quality of the permanent magnet synchronous motor system of FCS-MPC, but at the same time weakens the technical advantages of FCS-MPC's high dynamic response and high flexibility. The second technical solution is to achieve fixed switching frequency control by adding additional switching frequency control targets to the cost function. This solution better retains the inherent advantages of FCS-MPC, but it needs to introduce weight factors to achieve simultaneous control of current and inverter switching frequency. Fixed weight factors are difficult to obtain relatively balanced control performance in a wide speed regulation range. Therefore, the reasonable determination of weight factors has become a new technical problem.

[0005] In summary, how to overcome the problem of unstable switching frequency in the permanent magnet synchronous motor system of FCS-MPC, while cleverly avoiding the technical difficulty of difficult adjustment of weight factors, and ensuring the current control performance of the permanent magnet synchronous motor, has become a key technical problem that needs to be urgently solved in the current control of the FCS-MPC permanent magnet synchronous motor system. Summary of the invention

[0006] The purpose of the present invention is to solve the problem of non-fixed inverter switching frequency in the current control of the existing FCS-MPC permanent magnet synchronous motor system, and to provide a permanent magnet synchronous motor current control method with adjustable switching frequency FCS-MPC, which not only cleverly avoids the technical difficulty of difficult adjustment of weight factors, but also can achieve precise regulation of the switching frequency while ensuring the current control performance.

[0007] In order to achieve the above object, the technical solution of the present invention is as follows:

[0008] A permanent magnet synchronous motor current control method with adjustable switching frequency FCS-MPC comprises the following steps:

[0009] 11) Collection of drive system data: Collection of DC bus voltage, three-phase current, and rotor position data of permanent magnet synchronous motor drive system;

[0010] 12) Construction of discrete current prediction control model: coordinate transformation is performed according to the rotor position, and the dq axis current in the synchronous speed rotating dq coordinate system is obtained based on the synchronous speed rotating dq coordinate system oriented by the rotor magnetic field and the collected three-phase current. A discrete current prediction control model is established, and the eight basic voltage vectors of the two-level inverter are substituted into the discrete current prediction control model to obtain the dq axis predicted current of each voltage vector at time k+2;

[0011] 13) Determine the cost function and delay compensation strategy: Design the cost function, determine the performance evaluation criteria of the finite control set model predictive control, substitute the dq axis current prediction value obtained in step 12) into the designed cost function, and obtain the cost function size corresponding to each voltage vector;

[0012] 14) Calculate the current error control band: Calculate the switching frequency of the finite control set model current prediction controller based on the sliding window method, establish a switching frequency controller, and determine the current error control band;

[0013] 15) Selecting the optimal voltage vector: According to the cost function result of step 13), selecting the switch state with the least number of switch switching times and the cost function being within the current error control width determined in step 14) as the optimal voltage vector;

[0014] 16) Voltage vector output: The inverter outputs the optimal voltage vector, generates an inverter drive signal according to the inverter switch state corresponding to the optimal voltage vector, completes the voltage vector update of the control cycle, and realizes the real-time tracking of the dq axis current of the permanent magnet synchronous motor.

[0015] The acquisition of the drive system data comprises the following steps:

[0016] 21) Collect the DC bus voltage, three-phase current signal and rotor position signal of the permanent magnet synchronous motor drive system at time k;

[0017] 22) The three-phase current signal is transformed into a synchronous speed rotating dq coordinate system oriented by the rotor magnetic field, expressed as:

[0018]

[0019] Among them, i d 、i q is the dq axis current, i a 、i b 、i c is the three-phase current in the three-phase stationary coordinate system, θ is the electrical angle of the rotor position, sinθ and cosθ represent the sine and cosine values ​​of the rotor position;

[0020] 23) Obtain eight switching states of the three-phase two-level inverter. Different combinations of the upper arm switching states of the three-phase two-level inverter [Sa Sb Sc] are expressed as {000, 100, 110, 010, 011, 001, 101, 111}. In the synchronous speed rotating dq axis coordinate system, the dq axis voltage corresponding to each switching state is expressed as:

[0021]

[0022] Among them, u d ,u q Indicates the dq axis voltage output by the inverter. Sa, Sb, and Sc indicate the switch status of the upper bridge arm of the inverter. 1 indicates on and 0 indicates off. V dc is the DC bus voltage.

[0023] The construction of the discrete current predictive control model includes the following steps:

[0024] 31) Assuming the synchronous speed rotating dq coordinate system with rotor magnetic field orientation, the voltage equation of the permanent magnet synchronous motor is:

[0025]

[0026] In the formula, i d 、i q 、u d 、uq is the dq axis current and voltage, L s is the stator inductance, R s is the stator resistance, ψ f is the permanent magnet flux, ω r is the rotor electrical angular velocity, represents the d-axis current differential, represents the q-axis current differential;

[0027] 32) Based on the Euler discretization method, the discrete current prediction model of the permanent magnet synchronous motor is obtained as follows:

[0028]

[0029] Where k is the kth sampling period, T s is the sampling period, i d (k+1), i q (k+1) represents the predicted value of the dq axis current at time k+1, ω r (k) represents the rotor electrical angular velocity sampling value at time k, u d (k),u q (k) represents the dq axis voltage output value of the inverter at time k, L s is the stator inductance, R s is the stator resistance, ψ f is the permanent magnet flux;

[0030] 33) Substitute the eight basic voltage vectors of the three-phase two-level inverter into formula (4) respectively to obtain the dq-axis predicted current value corresponding to each basic voltage vector.

[0031] Determining the cost function and the delay compensation strategy comprises the following steps:

[0032] 41) In order to compensate for the timing mismatch caused by the time-consuming calculation, the voltage vector updated at time k and the sampling signal at time k are used to estimate the current at time k+1, and the current estimation value at time k+1 is used as the starting point for the current prediction at time k+2, and the current at time k+2 is predicted, thereby compensating for the "one-beat lag" of digital control by "one-beat ahead" in control;

[0033] 42) The design cost function is the sum of squares of the dq current tracking errors g at time k+2, which is:

[0034]

[0035] in, is the dq axis current reference value, i d (k+2), i q (k+2) represents the predicted value of dq axis current at time k+2, C limUsed to prevent the motor from overcurrent, expressed as:

[0036]

[0037] Where k is the kth sampling period, is the dq axis current reference value, i d (k+2), i q (k+2) represents the predicted value of the dq axis current at time k+2, I sm Indicates the maximum allowable current of the permanent magnet synchronous motor;

[0038] 43) Substitute the predicted dq-axis current value corresponding to each basic voltage vector in step 33) into formula (5) to obtain the cost function size corresponding to each basic voltage vector.

[0039] The calculation of the current error control band comprises the following steps:

[0040] 51) Based on the idea of ​​sliding window, the ratio of the total number of switching times of the inverter upper bridge arm in a certain time window to the duration of the time window is expressed as the average switching frequency of the inverter.

[0041] The average switching frequency of the inverter at time k is expressed as:

[0042]

[0043] Where, k is the kth sampling period, f sw (k) is the average switching frequency of the inverter calculated at time k, T w is the duration of the sliding window, n is the number of time periods of the sliding window, S = [Sa, Sb, Sc] represents the switch state of the upper bridge arm of the inverter, 1 represents on, 0 represents off, ‖‖1 represents the total number of switches of the upper bridge arm of the three-phase two-level inverter, S(ki) is the switch state of the upper bridge arm of the inverter at time ki, and S(ki-1) is the switch state of the upper bridge arm of the inverter at time ki-1;

[0044] In order to eliminate the quantization noise in the switching frequency calculation, the real-time calculated value of the switching frequency is further processed by a low-pass filter. The final real-time feedback of the switching frequency is:

[0045]

[0046] Where k is the kth sampling period, T s represents the control period, α is the filter coefficient, ω c is the cutoff frequency of the first-order low-pass filter, f sw (k) is the average switching frequency of the inverter calculated at time k, f swL(k) is the average switching frequency of the inverter after filtering at time k, f swL (k-1) is the average switching frequency of the inverter calculated at time k-1;

[0047] 52) Based on the feedback calculation of the real-time switching frequency using formulas (7)-(8), a switching frequency control loop is established.

[0048] The switching frequency controller uses a PI regulator. The output of the PI regulator is the current control error band width. The switching frequency controller is expressed as:

[0049]

[0050] Where k is the kth sampling period, T s represents the control period, δ(k) is the current control error band width at time k, is the reference value of the average switching frequency of the inverter at time k, f swL (k) is the average switching frequency of the inverter after filtering at time k, k p With k i is the proportional gain and integral gain of the switching frequency controller, e f (k) is the average switching frequency error of the inverter at time k, e f (k-1) is the average switching frequency error of the inverter at time k-1;

[0051] The range of the switching frequency controller output variable δ is limited to:

[0052] δ min <δ(k)≤δ max (10)

[0053] Where k is the kth sampling period, δ(k) is the current control error band width at time k, δ min Determined by the maximum value of the average switching frequency of the inverter allowed by the operating point, δ max The maximum value of current pulsation allowed by the operating point;

[0054] 53) The real-time average switching frequency of the inverter is achieved by adjusting the bandwidth of the current control error band;

[0055] When the average switching frequency reference value of the inverter is lower than the actual switching frequency, the positive error is input into the PI regulator, and the PI gradually increases the width of the current control error band to achieve accurate tracking of the switching frequency reference value;

[0056] When the switching frequency reference value is higher than the actual switching frequency, the negative error is input into the PI regulator, and the PI regulator gradually reduces the width of the current control error band;

[0057] Under the regulation of the integrator, the actual switching frequency of the inverter can track the switching frequency reference value without static error.

[0058] The selecting of the optimal voltage vector comprises the following steps:

[0059] 61) Based on the voltage vector applied at time k+1, the candidate voltage vectors at time k+2 are divided into three categories according to the number of switch switching times:

[0060]

[0061] Where k is the kth sampling period, S(k+1) and S(k+2) represent the inverter switch states at time k+1 and k+2, S=[Sa,Sb,Sc] represents the switch state of the upper bridge arm of the inverter, 1 represents on, 0 represents off, ‖‖1 represents the total number of switching times of the upper bridge arm switch of the three-phase two-level inverter, V I Represents the switch state set without bridge arm switch state switching, V II Represents a switch state set of a bridge arm switch state switching, V III A switch state set representing the switching of the switch states of the two bridge arms;

[0062] If the voltage vector at the previous moment is still used at time (k+2), the switch state does not change. At this time, the number of switch switching times is zero, and the voltage vector at the previous moment is confirmed to be V I Group;

[0063] If a new voltage vector is applied at time (k+2) and only one bridge arm’s switch state changes, then the vector is confirmed to be V II If a new voltage vector is applied at time (k+2) and the switch states of two bridge arms change, the vector is confirmed to be V III Group;

[0064] 62) Based on the classification result of the candidate voltage vector defined by formula (11), the selection method of the candidate voltage vector at time k+2 is expressed as:

[0065]

[0066] Where k is the kth sampling period, V(k+2) represents the optimal voltage vector to be output at time k+2, and V optI Indicates V I The voltage vector with the smallest cost function, V optII Indicates {V I ∪V II The voltage vector with the minimum cost function, V optIII Indicates {V I ∪V II ∪V III}The voltage vector with the minimum cost function, g[] represents the cost function obtained in step 43), and δ is the current error control band width determined in step 52);

[0067] That is, V is evaluated first. I If the cost function of the voltage vector of the VI group is less than δ 2 , it means that applying this voltage vector will not cause the actual current to exceed the current control error band, and the voltage vector at the previous moment will continue to be used. At this time, the switching state of the inverter does not need to change;

[0068] If V I The cost function of the group voltage vector is greater than δ 2 , then for {V I ∪V II}Evaluate the voltage vectors in the set and select the voltage vector with the smallest cost function. If the cost function of the voltage vector is less than δ 2 , then the voltage vector is used as the final output;

[0069] If the cost function of the voltage vector is still greater than δ 2 , then for {V I ∪V II ∪V III The voltage vectors of the set are evaluated, and the voltage vector with the smallest cost function is selected as the final output.

[0070] Beneficial Effects

[0071] The present invention discloses a permanent magnet synchronous motor current control method with adjustable switching frequency FCS-MPC, which retains the unique advantages of FCS-MPC compared with the prior art, overcomes its inherent defect of non-fixed switching frequency, and enables FCS-MPC to operate at a specified switching frequency within a wide speed regulation range; by introducing a PI type switching frequency controller, it is possible to achieve zero-static error tracking of the switching frequency reference value, and the control method achieves effective regulation of the switching frequency on the basis of ensuring the current control performance of the permanent magnet motor. BRIEF DESCRIPTION OF THE DRAWINGS

[0072] Figure 1 is a method sequence diagram of the present invention;

[0073] Figure 2 It is the overall control structure diagram of the present invention;

[0074] Figure 3 This is an experimental effect diagram of the present invention when the switching frequency is 0.8kHz and the power is stepped from 0A to 10A at 100rpm;

[0075] Figure 4This is an experimental effect diagram of the present invention when the switching frequency is 1kHz and the power is stepped from 0A to 10A at 500rpm;

[0076] Figure 5 This is an experimental effect diagram of the present invention when the switching frequency is stepped from 1kHz to 3kHz at 300rpm-10A;

[0077] Figure 6 This is an experimental effect diagram of the present invention when the switching frequency is stepped from 1kHz to 2kHz at 500rpm-20A;

[0078] Figure 7 This is a flowchart of the algorithm implementation involved in the present invention. DETAILED DESCRIPTION

[0079] In order to have a further understanding and recognition of the structural features and the effects achieved by the present invention, a preferred embodiment and accompanying drawings are used for detailed description as follows:

[0080] like Figure 1 As shown, the current control method of a permanent magnet synchronous motor with an adjustable switching frequency FCS-MPC according to the present invention comprises the following steps:

[0081] The first step is to collect drive system data: collect DC bus voltage, three-phase current, and rotor position data of the permanent magnet synchronous motor drive system.

[0082] (1) Collect the DC bus voltage, three-phase current signal, and rotor position signal of the permanent magnet synchronous motor drive system at time k.

[0083] (2) The three-phase current signal is transformed into a synchronous speed rotating dq coordinate system oriented by the rotor magnetic field, expressed as:

[0084]

[0085] Among them, i d 、i q is the dq axis current, i a 、i b 、i c is the three-phase current in the three-phase stationary coordinate system, θ is the electrical angle of the rotor position, sinθ and cosθ represent the sine and cosine values ​​of the rotor position.

[0086] (3) The eight switching states of the three-phase two-level inverter are obtained. The different combinations of the upper arm switching states of the three-phase two-level inverter [Sa Sb Sc] are expressed as {000, 100, 110, 010, 011, 001, 101, 111}. In the synchronous speed rotating dq axis coordinate system, the dq axis voltage corresponding to each switching state is expressed as:

[0087]

[0088] Among them, u d ,u q Indicates the dq axis voltage output of the inverter. Sa, Sb, and Sc indicate the switch status of the upper bridge arm of the inverter. 1 indicates on and 0 indicates off. V dc is the DC bus voltage.

[0089] The second step is to construct a discrete current prediction control model: according to the rotor position, coordinate transformation is performed, and the dq-axis current in the synchronous speed rotating dq coordinate system is obtained based on the rotor magnetic field oriented synchronous speed rotating dq coordinate system and the collected three-phase current. A discrete current prediction control model is established, and the eight basic voltage vectors of the two-level inverter are substituted into the discrete current prediction control model to obtain the dq-axis predicted current of each voltage vector at time k+2. This can effectively utilize the discrete switching characteristics of the inverter and effectively improve the current dynamic response.

[0090] (1) Assuming the synchronous speed rotating dq coordinate system oriented by the rotor magnetic field, the voltage equation of the permanent magnet synchronous motor is:

[0091]

[0092] In the formula, i d 、i q 、u d 、u q is the dq axis current and voltage, L s is the stator inductance, R s is the stator resistance, ψ f is the permanent magnet flux, ω r is the rotor electrical angular velocity, represents the d-axis current differential, Represents the q-axis current differential.

[0093] (2) Based on the Euler discretization method, the discrete current prediction model of the permanent magnet synchronous motor is obtained as follows:

[0094]

[0095] Where k is the kth sampling period, T s is the sampling period, i d (k+1), i q (k+1) represents the predicted value of the dq axis current at time k+1, ω r (k) represents the rotor electrical angular velocity sampling value at time k, u d (k),u q (k) represents the dq axis voltage output of the inverter at time k, L s is the stator inductance, R sis the stator resistance, ψ f is the permanent magnet flux.

[0096] (3) Substitute the eight basic voltage vectors of the three-phase two-level inverter into formula (4) respectively to obtain the dq-axis predicted current value corresponding to each voltage vector.

[0097] The third step is to determine the cost function and delay compensation strategy: design the cost function, determine the performance evaluation criteria of the finite control set model predictive control, substitute the dq-axis current prediction value obtained in the second step into the designed cost function, and obtain the cost function size corresponding to each voltage vector.

[0098] The cost function can include multiple control objectives and has the advantages of being simple and intuitive. Through the cost function enumeration evaluation, the FCS-MPC controller can directly select the optimal voltage vector without going through the voltage modulation link. The delay compensation strategy can accurately compensate for the timing mismatch caused by the calculation delay and effectively improve the current control performance.

[0099] (1) In order to compensate for the timing mismatch caused by the time-consuming calculation, the voltage vector updated at time k and the sampling signal at time k are used to estimate the current at time k+1. The current estimation value at time k+1 is used as the starting point for the current prediction at time k+2, and the current at time k+2 is predicted, thereby achieving accurate compensation for the "one-beat lag" of digital control by "one-beat ahead" in control.

[0100] (2) The design cost function is the sum of squares of the dq current tracking errors g at time (k+2), which is:

[0101]

[0102] in, is the dq axis current reference value, i d (k+2), i q (k+2) represents the predicted value C of the dq axis current at time k+2 lim , used to prevent the motor from overcurrent, expressed as:

[0103]

[0104] Where k is the kth sampling period, is the dq axis current reference value, i d (k+2), i q (k+2) represents the predicted value of the dq axis current at time k+2, I sm Indicates the maximum allowable current of the permanent magnet synchronous motor;

[0105] (3) Substitute the predicted dq-axis current value corresponding to each basic voltage vector in the third step into formula (5) to obtain the cost function size corresponding to each basic voltage vector. The cost function effectively compensates for the delay effect caused by the system calculation time by advancing the current at time (k+2), thus solving the timing mismatch problem.

[0106] The fourth step is to calculate the current error control band: calculate the switching frequency of the finite control set model current prediction controller based on the sliding window method, establish the switching frequency controller, and determine the current error control band.

[0107] This step innovatively establishes a real-time control loop for the average switching frequency, and decomposes the coupled control of the current control and switching frequency into two parts: the switching frequency outer loop and the current control inner loop through the current error band. The introduced switching frequency controller can achieve error-free tracking of the switching frequency command, overcoming the difficulty in eliminating the static error of the traditional switching frequency control scheme. This step fully utilizes the advantages of the feedback control system, has a small computational load, and is easy to implement.

[0108] (1) Based on the idea of ​​sliding window, the ratio of the total number of switching times of the inverter upper bridge arm in a certain time window to the duration of the time window is expressed as the average switching frequency of the inverter.

[0109] The average switching frequency of the inverter at time k is expressed as:

[0110]

[0111] Where, k is the kth sampling period, f sw (k) is the average switching frequency of the inverter calculated at time k, T w is the duration of the sliding window, n is the number of time periods of the sliding window, S = [Sa, Sb, Sc] represents the switch state of the upper bridge arm of the inverter, 1 represents on, 0 represents off, ‖‖1 represents the total number of switches of the upper bridge arm of the three-phase two-level inverter, S(ki) is the switch state of the upper bridge arm of the inverter at time ki, and S(ki-1) is the switch state of the upper bridge arm of the inverter at time ki-1;

[0112] In order to eliminate the quantization noise in the switching frequency calculation, the real-time calculated value of the switching frequency is further processed by a low-pass filter. The final real-time feedback of the switching frequency is:

[0113]

[0114] Where k is the kth sampling period, T s represents the control period, α is the filter coefficient, ω c is the cutoff frequency of the first-order low-pass filter, f sw(k) is the average switching frequency of the inverter calculated at time k, f swL (k) is the average switching frequency of the inverter after filtering at time k, f swL (k-1) is the average switching frequency of the inverter calculated at time k-1.

[0115] (2) Based on the feedback calculation of the real-time switching frequency using formulas (7)-(8), a switching frequency control loop is established.

[0116] The switching frequency controller uses a PI regulator. The output of the PI regulator is the current control error band width. The switching frequency controller is expressed as:

[0117]

[0118] Where k is the kth sampling period, T s represents the control period, δ(k) is the current control error band width at time k, is the reference value of the average switching frequency of the inverter at time k, f swL (k) is the average switching frequency of the inverter after filtering at time k, k p With k i is the proportional gain and integral gain of the switching frequency controller, e f (k) is the average switching frequency error of the inverter at time k, e f (k-1) is the average switching frequency error of the inverter at time k-1;

[0119] The range of the switching frequency controller output variable δ is limited to:

[0120] δ min <δ(k)≤δ max (10)

[0121] Where k is the kth sampling period, δ(k) is the current control error band width at time k, δ min Determined by the maximum value of the average switching frequency of the inverter allowed by the operating point, δ max The maximum value of current pulsation allowed by the operating point.

[0122] (3) The real-time average switching frequency of the inverter is achieved by adjusting the bandwidth of the current control error band;

[0123] When the average switching frequency reference value of the inverter is lower than the actual switching frequency, the positive error is input into the PI regulator, and the PI gradually increases the width of the current control error band to achieve accurate tracking of the switching frequency reference value;

[0124] When the switching frequency reference value is higher than the actual switching frequency, the negative error is input into the PI regulator, and the PI regulator gradually reduces the width of the current control error band;

[0125] Under the regulation of the integrator, the actual switching frequency of the inverter can track the switching frequency reference value without static error.

[0126] Step 5: Select the optimal voltage vector: Based on the cost function result of step 3, select the switch state with the cost function within the current error control width determined in step 4 and the least number of switch switching as the optimal voltage vector. This step uses the current error band as a hard constraint to achieve quantitative control of current performance. The candidate voltage vectors are evaluated sequentially according to different priorities, and the voltage vector that satisfies both current control performance and the least number of switch switching is evaluated with the minimum computational load.

[0127] (1) Based on the voltage vector applied at time k+1, the candidate voltage vectors at time k+2 are divided into three categories according to the number of switch switching times:

[0128]

[0129] Where k is the kth sampling period, S(k+1) and S(k+2) represent the inverter switch states at time k+1 and k+2, S=[Sa,Sb,Sc] represents the switch state of the upper bridge arm of the inverter, 1 represents on, 0 represents off, ‖‖1 represents the total number of switching times of the upper bridge arm switch of the three-phase two-level inverter, V I Represents the switch state set without bridge arm switch state switching, V II Represents a switch state set of a bridge arm switch state switching, V III A switch state set representing the switching of the switch states of the two bridge arms;

[0130] If the voltage vector at the previous moment is still used at time (k+2), the switch state does not change. At this time, the number of switch switching times is zero, and the voltage vector at the previous moment is confirmed to be V I Group;

[0131] If a new voltage vector is applied at time (k+2) and only one bridge arm’s switch state changes, then the vector is confirmed to be V II If a new voltage vector is applied at time (k+2) and the switch states of two bridge arms change, the vector is confirmed to be V III Group.

[0132] For example, if the voltage vector applied at time (k+1) is V5(101), then the candidate voltage vector V5(101) at time (k+2) belongs to V I Group, V1(001), V4(100), V7(111) belong to V II Group, V3(011), V6(110) belongs to V III Group.

[0133] (2) Based on the classification result of the candidate voltage vector defined by formula (11), the selection method of the candidate voltage vector at time k+2 is expressed as:

[0134]

[0135] Where k is the kth sampling period, V(k+2) represents the optimal voltage vector to be output at time k+2, and V optI Indicates V I The voltage vector with the smallest cost function, V optII Indicates {V I ∪V II The voltage vector with the minimum cost function, V optIII Indicates {V I ∪V II ∪V III}The voltage vector with the minimum cost function, g[] represents the cost function, and δ is the current error control bandwidth;

[0136] That is, V is evaluated first I If the cost function of the voltage vector of the VI group is less than δ 2 , it means that applying this voltage vector will not cause the actual current to exceed the current control error band, and the voltage vector at the previous moment will continue to be used. At this time, the switching state of the inverter does not need to change;

[0137] If V I The cost function of the group voltage vector is greater than δ 2 , then for {V I ∪V II}Evaluate the voltage vectors in the set and select the voltage vector with the smallest cost function. If the cost function of the voltage vector is less than δ 2 , then the voltage vector is used as the final output;

[0138] If the cost function of the voltage vector is still greater than δ 2 , then for {V I ∪V II ∪V III} The voltage vectors of the set are evaluated, and the voltage vector with the smallest cost function is selected as the final output. Current is the target of permanent magnet motor control, voltage is the control input, and cost function is the performance function. The smaller the value, the better the current control. The idea of ​​predictive current control is to select a voltage vector that minimizes the cost function at the future moment as the final output. The process of voltage vector selection is the process of current control. Current control is realized here, and the final output is performed in the sixth step.

[0139] Step 6: Voltage vector output: The inverter outputs the optimal voltage vector, generates an inverter drive signal according to the inverter switch state corresponding to the optimal voltage vector, completes the voltage vector update of this control cycle, and realizes real-time tracking of the dq axis current by the permanent magnet synchronous motor.

[0140] The specific implementation flow chart of the control method of the present invention is as follows Figure 7 As shown in the figure, the control structure diagram of the overall solution is as follows Figure 2 shown.

[0141] Table 1 Nominal parameters of permanent magnet synchronous motor

[0142]

[0143] The nominal parameters of the controlled motor are shown in Table 1. The inverter is powered by a programmable DC power supply, and the output voltage is set to 48V. The controlled motor works in torque control mode and is powered by a three-phase two-level inverter. The controlled motor side uses dSPACE / DS1007 as the motor controller under test, and a pair of poles of the rotary transformer is used to detect the rotor position of the motor under test. The motor phase current sensor is LEM LA25-P. The current sensor samples the three-phase current value in real time, and the sampling time is set to 50μs. The inverter dead time is set to 2μs. The dynamometer is a 2.2kW three-phase asynchronous motor, which works in speed control mode and is driven by an ACS 800 inverter.

[0144] Figure 3 The actual waveform of the permanent magnet motor running at 100 rpm and the switching frequency reference value set to 0.8 kHz, when the q-axis current command changes from 0 A to 10 A.

[0145] Figure 4 The actual waveform of the permanent magnet motor running at 500 rpm and the switching frequency reference value set to 1.0 kHz, when the q-axis current command changes from 0 A to 10 A.

[0146] Figure 5 The actual waveform is that the permanent magnet motor runs at a speed of 300 rpm, the q-axis current command is maintained at 10 A, and the switching frequency reference value changes from 1.0 kHz to 3.0 kHz in a step manner.

[0147] Figure 6 The actual waveform is that the permanent magnet motor runs at a speed of 500 rpm, the q-axis current command is maintained at 20 A, and the switching frequency reference value changes from 1.0 kHz to 2.0 kHz.

[0148] from Figure 3 and Figure 4It can be seen that when the dq axis current command changes, the FCS-MPC current controller can accurately track the current command, and the switching frequency will be stably maintained at the reference value and will not change with the change of the operating point.

[0149] from Figure 5 and Figure 6 It can be seen that when the dq-axis current command remains constant, the actual switching frequency can accurately track the step change of the reference value without steady-state error, indicating that this strategy can effectively control the switching frequency while ensuring the current control performance.

[0150] Based on the measured waveform, the present invention can simultaneously accurately track the current command and the switching frequency command, operate at a fixed switching frequency within a wide speed range, and overcome the inherent problem of the finite control set model predicting the switching frequency to change with the operating point. In addition, the present invention does not require a weight factor, eliminating the complex weight factor setting process.

[0151] The above shows and describes the basic principles, main features and advantages of the present invention. Those skilled in the art should understand that the present invention is not limited to the above embodiments. The above embodiments and descriptions only describe the principles of the present invention. The present invention may be subject to various changes and improvements without departing from the spirit and scope of the present invention. These changes and improvements fall within the scope of the present invention. The scope of protection claimed by the present invention is defined by the attached claims and their equivalents.

Claims

1. A permanent magnet synchronous motor current control method with adjustable switching frequency FCS-MPC, characterized in that: The following steps are involved: 11) Collection of drive system data: Collection of DC bus voltage, three-phase current, and rotor position data of permanent magnet synchronous motor drive system; 12) Construction of discrete current prediction control model: coordinate transformation is performed according to the rotor position, and the dq axis current in the synchronous speed rotating dq coordinate system is obtained based on the synchronous speed rotating dq coordinate system oriented by the rotor magnetic field and the collected three-phase current. A discrete current prediction control model is established, and the eight basic voltage vectors of the two-level inverter are substituted into the discrete current prediction control model to obtain the dq axis predicted current of each voltage vector at time k+2; 13) Determine the cost function and delay compensation strategy: Design the cost function, determine the performance evaluation criteria of the finite control set model predictive control, substitute the dq axis current prediction value obtained in step 12) into the designed cost function, and obtain the cost function size corresponding to each voltage vector; 14) Calculate the current error control band: Calculate the switching frequency of the finite control set model current prediction controller based on the sliding window method, establish a switching frequency controller, and determine the current error control band; 15) Selecting the optimal voltage vector: According to the cost function result of step 13), selecting the switch state with the least number of switch switching times and the cost function being within the current error control width determined in step 14) as the optimal voltage vector; 16) Voltage vector output: The inverter outputs the optimal voltage vector, generates an inverter drive signal according to the inverter switch state corresponding to the optimal voltage vector, completes the voltage vector update of the control cycle, and realizes the real-time tracking of the dq axis current of the permanent magnet synchronous motor.

2. The method for controlling the current of a permanent magnet synchronous motor with an adjustable switching frequency FCS-MPC according to claim 1 is characterized in that: The acquisition of the drive system data comprises the following steps: 21) Collect the DC bus voltage, three-phase current signal and rotor position signal of the permanent magnet synchronous motor drive system at time k; 22) The three-phase current signal is transformed into a synchronous speed rotating dq coordinate system oriented by the rotor magnetic field, expressed as: Among them, i d 、i q is the dq axis current, i a 、i b 、i c is the three-phase current in the three-phase stationary coordinate system, θ is the electrical angle of the rotor position, sinθ and cosθ represent the sine and cosine values ​​of the rotor position; 23) Obtain eight switching states of the three-phase two-level inverter. Different combinations of the upper arm switching states of the three-phase two-level inverter [Sa Sb Sc] are expressed as {000, 100, 110, 010, 011, 001, 101, 111}. In the synchronous speed rotating dq axis coordinate system, the dq axis voltage corresponding to each switching state is expressed as: Among them, u d ,u q Indicates the dq axis voltage output by the inverter. Sa, Sb, and Sc indicate the switch status of the upper bridge arm of the inverter. 1 indicates on and 0 indicates off. V dc is the DC bus voltage.

3. The method for controlling the current of a permanent magnet synchronous motor with an adjustable switching frequency FCS-MPC according to claim 1, characterized in that: The construction of the discrete current predictive control model includes the following steps: 31) Assuming the synchronous speed rotating dq coordinate system with rotor magnetic field orientation, the voltage equation of the permanent magnet synchronous motor is: In the formula, i d 、i q 、u d 、u q is the dq axis current and voltage, L s is the stator inductance, R s is the stator resistance, ψ f is the permanent magnet flux, ω r is the rotor electrical angular velocity, represents the d-axis current differential, represents the q-axis current differential; 32) Based on the Euler discretization method, the discrete current prediction model of the permanent magnet synchronous motor is obtained as follows: Where k is the kth sampling period, T s is the sampling period, i d (k+1), i q (k+1) represents the predicted value of the dq axis current at time k+1, ω r (k) represents the rotor electrical angular velocity sampling value at time k, u d (k),u q (k) represents the dq axis voltage output value of the inverter at time k, L s is the stator inductance, R s is the stator resistance, ψ f is the permanent magnet flux; 33) Substitute the eight basic voltage vectors of the three-phase two-level inverter into formula (4) respectively to obtain the dq-axis predicted current value corresponding to each basic voltage vector.

4. The method for controlling the current of a permanent magnet synchronous motor with an adjustable switching frequency FCS-MPC according to claim 1 is characterized in that: Determining the cost function and the delay compensation strategy comprises the following steps: 41) In order to compensate for the timing mismatch caused by the time-consuming calculation, the voltage vector updated at time k and the sampling signal at time k are used to estimate the current at time k+1, and the current estimation value at time k+1 is used as the starting point for the current prediction at time k+2, and the current at time k+2 is predicted, thereby compensating for the "one-beat lag" of digital control by "one-beat ahead" in control; 42) The design cost function is the sum of squares of the dq current tracking errors g at time k+2, which is: in, is the dq axis current reference value, i d (k+2), i q (k+2) represents the predicted value of dq axis current at time k+2, C lim Used to prevent the motor from overcurrent, expressed as: Where k is the kth sampling period, is the dq axis current reference value, i d (k+2), i q (k+2) represents the predicted value of the dq axis current at time k+2, I sm Indicates the maximum allowable current of the permanent magnet synchronous motor; 43) Substitute the predicted dq-axis current value corresponding to each basic voltage vector in step 33) into formula (5) to obtain the cost function size corresponding to each basic voltage vector.

5. The method for controlling the current of a permanent magnet synchronous motor with an adjustable switching frequency FCS-MPC according to claim 1, characterized in that: The calculation of the current error control band comprises the following steps: 51) Based on the idea of ​​sliding window, the ratio of the total number of switching times of the inverter upper bridge arm in a certain time window to the duration of the time window is expressed as the average switching frequency of the inverter. The average switching frequency of the inverter at time k is expressed as: Where, k is the kth sampling period, f sw (k) is the average switching frequency of the inverter calculated at time k, T w is the duration of the sliding window, n is the number of time periods of the sliding window, S = [Sa, Sb, Sc] represents the switch state of the upper bridge arm of the inverter, 1 represents on, 0 represents off, ‖‖1 represents the total number of switches of the upper bridge arm of the three-phase two-level inverter, S(ki) is the switch state of the upper bridge arm of the inverter at time ki, and S(ki-1) is the switch state of the upper bridge arm of the inverter at time ki-1; In order to eliminate the quantization noise in the switching frequency calculation, the real-time calculated value of the switching frequency is further processed by a low-pass filter. The final real-time feedback of the switching frequency is: Where k is the kth sampling period, T s represents the control period, α is the filter coefficient, ω c is the cutoff frequency of the first-order low-pass filter, f sw (k) is the average switching frequency of the inverter calculated at time k, f swL (k) is the average switching frequency of the inverter after filtering at time k, f swL (k-1) is the average switching frequency of the inverter calculated at time k-1; 52) Based on the feedback calculation of the real-time switching frequency using formulas (7)-(8), a switching frequency control loop is established. The switching frequency controller uses a PI regulator. The output of the PI regulator is the current control error band width. The switching frequency controller is expressed as: Where k is the kth sampling period, T s represents the control period, δ(k) is the current control error band width at time k, is the reference value of the average switching frequency of the inverter at time k, f swL (k) is the average switching frequency of the inverter after filtering at time k, k p With k i is the proportional gain and integral gain of the switching frequency controller, e f (k) is the average switching frequency error of the inverter at time k, e f (k-1) is the average switching frequency error of the inverter at time k-1; The range of the switching frequency controller output variable δ is limited to: d min <δ(k)≤δ max (10) Where k is the kth sampling period, δ(k) is the current control error band width at time k, δ min Determined by the maximum value of the average switching frequency of the inverter allowed by the operating point, δ max The maximum value of current pulsation allowed by the operating point; 53) The real-time average switching frequency of the inverter is achieved by adjusting the bandwidth of the current control error band; When the average switching frequency reference value of the inverter is lower than the actual switching frequency, the positive error is input into the PI regulator, and the PI gradually increases the width of the current control error band to achieve accurate tracking of the switching frequency reference value; When the switching frequency reference value is higher than the actual switching frequency, the negative error is input into the PI regulator, and the PI regulator gradually reduces the width of the current control error band; Under the regulation of the integrator, the actual switching frequency of the inverter can track the switching frequency reference value without static error.

6. The method for controlling the current of a permanent magnet synchronous motor with an adjustable switching frequency FCS-MPC according to claim 1, characterized in that: The selecting of the optimal voltage vector comprises the following steps: 61) Based on the voltage vector applied at time k+1, the candidate voltage vectors at time k+2 are divided into three categories according to the number of switch switching times: Where k is the kth sampling period, S(k+1) and S(k+2) represent the inverter switch states at time k+1 and k+2, S=[Sa,Sb,Sc] represents the switch state of the upper bridge arm of the inverter, 1 represents on, 0 represents off, ‖‖1 represents the total number of switching times of the upper bridge arm switch of the three-phase two-level inverter, V I Represents the switch state set without bridge arm switch state switching, V II Represents a switch state set of a bridge arm switch state switching, V III A switch state set representing the switching of the switch states of the two bridge arms; If the voltage vector at the previous moment is still used at time (k+2), the switch state does not change. At this time, the number of switch switching times is zero, and the voltage vector at the previous moment is confirmed to be V I Group; If a new voltage vector is applied at time (k+2) and only one bridge arm’s switch state changes, then the vector is confirmed to be V II If a new voltage vector is applied at time (k+2) and the switch states of two bridge arms change, the vector is confirmed to be V III Group; 62) Based on the classification result of the candidate voltage vector defined by formula (11), the selection method of the candidate voltage vector at time k+2 is expressed as: Where k is the kth sampling period, V(k+2) represents the optimal voltage vector to be output at time k+2, and V optI Indicates V I The voltage vector with the smallest cost function, V optII Indicates {V I ∪V II The voltage vector with the minimum cost function, V optIII Indicates {V I ∪V II ∪V III }The voltage vector with the minimum cost function, g[] represents the cost function obtained in step 43), and δ is the current error control band width determined in step 52); That is, V is evaluated first I If the cost function of the voltage vector of the VI group is less than δ 2 , it means that applying this voltage vector will not cause the actual current to exceed the current control error band, and the voltage vector at the previous moment will continue to be used. At this time, the switching state of the inverter does not need to change; If V I The cost function of the group voltage vector is greater than δ 2 , then for {V I ∪V II }Evaluate the voltage vectors in the set and select the voltage vector with the smallest cost function. If the cost function of the voltage vector is less than δ 2 , then the voltage vector is used as the final output; If the cost function of the voltage vector is still greater than δ 2 , then for {V I ∪V II ∪V III The voltage vectors of the set are evaluated, and the voltage vector with the smallest cost function is selected as the final output.

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