Low-temperature-drift segmented temperature compensation bandgap reference circuit, chip and control device

Abstract

The present invention discloses a low-temperature-drift segmented temperature compensation bandgap reference circuit in the field of circuit design technology, which includes a first-order temperature compensation module, a segmented temperature compensation module, and a high-temperature trimming module. The first-order temperature compensation module is used to generate a reference voltage according to the temperature characteristics of a triode. The output end of the segmented temperature compensation module is connected to the first-order temperature compensation module, and the segmented temperature compensation module is used to superimpose the generated compensation current on the first-order temperature compensation module to reduce the temperature drift of the reference voltage. The output end of the high-temperature trimming module is connected to the input end of the segmented temperature compensation module. Based on the segmented temperature compensation module of the first-order compensation bandgap reference circuit, the present invention corrects the temperature of the compensation current in some high-temperature segments of the segmented temperature compensation module, which can effectively solve the problem of excessive compensation in the high-temperature segment, broaden the high-precision temperature range of the bandgap reference, and achieve a lower temperature drift coefficient.

Description

Technical Field

[0001] The present invention relates to the technical field of circuit design, and specifically to a low-temperature-drift segmented temperature compensation bandgap reference circuit, a chip and a control device. Background Art

[0002] As an important part of an integrated circuit system, a bandgap reference circuit can provide a reference voltage, and its accuracy and temperature characteristics will directly affect the performance of the entire circuit system. Among them, the compensation principle of a common bandgap reference circuit is to eliminate through first-order compensation, that is, the difference between the bandgap voltages of two triodes is added to the bandgap voltage of the triode in a specific ratio to eliminate the first-order temperature term and approximately obtain a bandgap reference voltage independent of temperature.

[0003] Although the first-order compensated bandgap reference circuit can provide a reference voltage that is basically unchanged with temperature, for some high-precision circuit systems, for example, for high-precision ADCs, due to the failure to compensate the middle-temperature term, its temperature characteristics are far from meeting the requirements; on the other hand, in the existing segmented temperature compensation methods, most use currents with an exponential relationship with temperature in the high-temperature section for compensation. Therefore, when performing high-order compensation in the high-temperature section, overcompensation will occur. If the compensation current is simply reduced in proportion, the compensation effect in the middle and high-temperature sections will be reduced. Summary of the Invention

[0004] The purpose of the present invention is to provide a low-temperature-drift segmented temperature compensation bandgap reference circuit, a chip and a control device. By performing temperature correction on part of the high-temperature section compensation current of the segmented temperature compensation module based on the segmented temperature compensation module of the first-order compensated bandgap reference circuit, the problem of overcompensation in the high-temperature section can be effectively solved, the high-precision temperature range of the bandgap reference is broadened, and a lower temperature drift coefficient is achieved.

[0005] To achieve the above purpose, the present invention provides the following technical solutions:

[0006] A low-temperature-drift segmented temperature compensation bandgap reference circuit includes a first-order temperature compensation module, a segmented temperature compensation module and a high-temperature trimming module. The first-order temperature compensation module is used to generate a reference voltage according to the temperature characteristics of a triode;

[0007] The output end of the segmented temperature compensation module is connected to the first-order temperature compensation module. The segmented temperature compensation module is used to superimpose the generated compensation current on the first-order temperature compensation module to reduce the temperature drift of the reference voltage;

[0008] The output end of the high-temperature trimming module is connected to the input end of the segmented temperature compensation module. The high-temperature trimming module is used to generate a reverse compensation current containing quadratic term, logarithmic term and linear term in the high-temperature section to reduce the slope of the compensation current in the high-temperature section of the segmented temperature compensation module. The high-temperature trimming module includes a first current generation circuit, a second current generation circuit, a third current generation circuit and a high-temperature section current output circuit.

[0009] As a further aspect of the present invention: The first-order temperature compensation module includes a first resistor R1 and a second resistor R2. The first ends of the first resistor R1 and the second resistor R2 are both connected to a high level. The first end of the first resistor R1 is connected to the first end of the second resistor R2. The second end of the first resistor R1 is connected to the positive input end of the first differential amplifier AMP1. The positive input end of the first differential amplifier AMP1 is electrically connected to the drain of the first transistor M1. The source of the first transistor M1 is connected to the collector of the first triode Q1. The emitter of the first triode Q1 is connected to the first end of the third resistor R3. The second end of the third resistor R3 is connected to the emitter of the second triode Q2. The emitter of the second triode Q2 is connected to the first end of the fourth resistor R4. The second end of the fourth resistor R4 is connected to the first end of the fifth resistor R5. The second end of the fifth resistor R5 is grounded. The gate of the second triode Q2 is connected to the first end of the sixth resistor R6. The first end of the sixth resistor R6 is connected to the first end of the seventh resistor R7. The second end of the seventh resistor R7 is connected to the first end of the eighth resistor R8. The second end of the eighth resistor R8 is connected to the first end of the ninth resistor R9. The second end of the ninth resistor R9 is connected to the second end of the sixth resistor R6. The collector of the second triode Q2 is connected to the source of the second transistor M2. The drain of the second transistor M2 is connected to the second end of the second resistor R2. The second end of the sixth resistor R6 is grounded. The second end of the second resistor R2 is connected to the negative input end of the first differential amplifier AMP1. The output end of the first differential amplifier AMP1 is connected to the gate of the third transistor M3. The source of the third transistor M3 is connected to the first end of the second resistor R2. The drain of the third transistor M3 is connected to the source of the fourth transistor M4. The drain of the fourth transistor M4 is connected to the gate of the second triode Q2. The gate of the fourth transistor M4 is connected to the bias voltage Vbias;

[0010] The gates of the first transistor M1 and the second transistor M2 are both connected to the bias voltage Vbias to improve the power supply rejection ratio of the bandgap reference circuit.

[0011] As a further aspect of the present invention: The resistance values of the first resistor R1 and the second resistor R2 are equal. The sizes of the first transistor M1 and the second transistor M2 are the same. The area ratio of the emitter of the first triode Q1 to the emitter of the second triode Q2 is 8:1.

[0012] As a further solution of the present invention: the bandgap reference voltage output of the first-order temperature compensation module is:

[0013]

[0014] In the formula, △V BE is the voltage difference between the base and emitter of the first transistor Q1 and the second transistor Q2, R3 is the resistance value of the third resistor R3, R4 is the resistance value of the fourth resistor R4, R5 is the resistance value of the fifth resistor R5, and V BE is the voltage between the base and emitter of the second transistor Q2.

[0015] As a further solution of the present invention: the seventh resistor R7, the eighth resistor R8, and the ninth resistor R9 form a voltage-dividing resistor unit, and the voltage-dividing resistor unit is used to perform resistor voltage division on the bandgap reference voltage and output the voltage-divided bandgap reference voltage to the segmented temperature compensation module.

[0016] As a further solution of the present invention: the third transistor M3 and the fourth transistor M4 form a set of cascode structures, and the gates of the third transistor M3 and the fourth transistor M4 are connected with several sets of cascode structures, and the cascode structures are used to generate a reference current.

[0017] As a further solution of the present invention: the segmented temperature compensation module includes a current source, a current replication unit, and a reference voltage compensation unit. The output end of the current source is connected to the input end of the current replication unit. The current replication unit is used to replicate the output current of the current source, and the reference voltage compensation unit is used to compensate a part of the reference voltage of the first-order temperature compensation module.

[0018] As a further solution of the present invention: the current source includes a second differential amplifier AMP2, a fifth transistor M5, a sixth transistor M6, and a tenth resistor R10. The positive input end of the second differential amplifier AMP2 is connected to the bandgap reference voltage V ref , the negative input end of the second differential amplifier AMP2 is connected to the first end of the tenth resistor R10, the output end of the second differential amplifier AMP2 is connected to the gate of the fifth transistor M5, the drain of the fifth transistor M5 is connected to the source of the sixth transistor M6, and the drain of the sixth transistor M6 is connected to the negative input end of the second differential amplifier AMP2.

[0019] As a further solution of the present invention: the current replication unit includes a seventh transistor M7, an eighth transistor M8, a ninth transistor M9, and a tenth transistor M10. The source of the seventh transistor M7 is connected to the source of the fifth transistor M5. The drain of the seventh transistor M7 is connected to the source of the eighth transistor M8. The gate of the eighth transistor M8 is connected to the gate of the sixth transistor M6. The source of the ninth transistor M9 is connected to the source of the seventh transistor M7. The drain of the ninth transistor M9 is connected to the source of the tenth transistor M10. The gate of the tenth transistor M10 is connected to the gate of the eighth transistor M8.

[0020] As a further solution of the present invention: the reference voltage compensation unit includes an eleventh transistor M11, a twelfth transistor M12, a thirteenth transistor M13, and a fourteenth transistor M14. The source of the eleventh transistor M11 is connected to the source of the twelfth transistor M12. The source of the twelfth transistor M12 is connected to the drain of the eighth transistor M8. The drain of the eleventh transistor M11 is connected to the second end of the tenth resistor R10. The drain of the twelfth transistor M12 is connected to the drain of the fourteenth transistor M14. The source of the fourteenth transistor M14 is connected to the source of the thirteenth transistor M13. The drain of the thirteenth transistor M13 is connected to the drain of the eleventh transistor M11. The gates of the fourteenth transistor M14 and the eleventh transistor M11 are both connected to the positive temperature coefficient voltage V ptat , the gate of the twelfth transistor M12 is connected to the bandgap reference voltage V refx , the gate of the thirteenth transistor M13 is connected to the bandgap reference voltage V refy , the eleventh transistor M11 and the twelfth transistor M12 are the first pair of differential pair transistors, the thirteenth transistor M13 and the fourteenth transistor M14 are the second pair of differential pair transistors, and the eleventh transistor M11, the twelfth transistor M12, the thirteenth transistor M13, and the fourteenth transistor M14 all operate in the subthreshold region.

[0021] As a further solution of the present invention: the high-temperature trimming module includes a first current generation circuit, a second current generation circuit, a third current generation circuit, and a high-temperature section current output circuit;

[0022] The first current generation circuit is used to generate a current including a quadratic relationship. The current including a quadratic relationship changes with temperature, and the current including a quadratic relationship and the changing temperature present a quadratic relationship; the second current generation circuit is used to generate a current including a linear relationship. The current including a linear relationship changes with temperature, and the current including a linear relationship and the changing temperature present a linear relationship; the third current generation circuit is used to generate a current including a linear relationship and a logarithmic relationship. The current including a linear relationship and a logarithmic relationship changes with temperature, and the current including a linear relationship and a logarithmic relationship and the changing temperature present a linear relationship and a logarithmic relationship;

[0023] The current including a quadratic relationship, the current including a linear relationship, and the current including a logarithmic relationship are proportionally added to obtain a summed current. At an overcompensation temperature, the high-temperature section current output circuit subtracts the current including the linear relationship from the summed current to obtain a high-temperature corrected current.

[0024] As a further solution of the present invention: The first current generation circuit includes a fifteenth transistor M15. The drain of the fifteenth transistor M15 is connected to the drain of a sixteenth transistor M16. The source of the fifteenth transistor M15 is connected to the source of a seventeenth transistor M17. The gate of the fifteenth transistor M15 is connected to the gate of the seventeenth transistor M17. The drain of the seventeenth transistor M17 is connected to the drain of an eighteenth transistor M18. The source of the eighteenth transistor M18 is connected to the gate of the sixteenth transistor M16. The source of the sixteenth transistor M16 is connected to the source of a nineteenth transistor M19. The source of the nineteenth transistor M19 is grounded. The gate of the nineteenth transistor M19 is connected to the gate of the sixteenth transistor M16. The drain of the nineteenth transistor M19 is connected to the source of the eighteenth transistor M18. The gate of the eighteenth transistor M18 is connected to the gate of a twenty-first transistor M21. The drain of the twenty-first transistor M21 is connected to the drain of a twentieth transistor M20. The drain of the twentieth transistor M20 is connected to the gate of the twenty-first transistor M21. The source of the twenty-first transistor M21 is grounded. The gate of the twentieth transistor M20 is connected to the gate of the seventeenth transistor M17. The source of the twentieth transistor M20 is connected to a high level. The source of the twentieth transistor M20 is connected to the source of a twenty-second transistor M22. The gate of the twenty-second transistor M22 is connected to the gate of the twentieth transistor M20. The drain of the twenty-second transistor M22 is connected to the drain of a twenty-third transistor M23. The drain of the twenty-third transistor M23 is connected to the gate of the twenty-third transistor M23. The gate of the twenty-third transistor M23 is connected to the drain of the twenty-second transistor M22. The gate of the twenty-third transistor M23 is connected to the gate of a twenty-fourth transistor M24. The source of the twenty-fourth transistor M24 is connected to the source of the twenty-third transistor M23. The source of the twenty-third transistor M23 is grounded. The drain of the twenty-fourth transistor M24 is connected to the output terminal of a bias current source I bias of, and the output terminal of the bias current source I bias is connected to the drain of a twenty-fifth transistor M25. The drain of the twenty-fifth transistor M25 is connected to the gate of the twenty-fifth transistor M25. The gate of the twenty-fifth transistor M25 is connected to the gate of a twenty-sixth transistor M26. The source of the twenty-sixth transistor M26 is connected to the source of the twenty-fifth transistor M25. The source of the twenty-fifth transistor M25 is grounded;

[0025] The fifteenth transistor M15, the seventeenth transistor M17, and the twentieth transistor M20 have the same size, and the eighteenth transistor M18 and the twenty-first transistor M21 operate in the subthreshold region.

[0026] As a further aspect of the present invention: The second current generating circuit includes a twenty-seventh transistor M27. The source of the twenty-seventh transistor M27 is connected to the source of the twenty-second transistor M22. The drain of the twenty-seventh transistor M27 is connected to the drain of the twenty-eighth transistor M28. The drain of the twenty-eighth transistor M28 is connected to the gate of the twenty-seventh transistor M27. The gate of the twenty-eighth transistor M28 is connected to a bias voltage V bias , the source of the twenty-eighth transistor M28 is connected to the collector of a third triode Q3. The base of the third triode Q3 is connected to a reference voltage V bg , the emitter of the third triode Q3 is connected to the first end of an eleventh resistor R11. The second end of the eleventh resistor R11 is connected to the source of the twenty-sixth transistor M26.

[0027] As a further aspect of the present invention: The third current generating circuit includes a twenty-ninth transistor M29. The gate of the twenty-ninth transistor M29 is connected to the gate of the twenty-seventh transistor M27. The source of the twenty-ninth transistor M29 is connected to the source of the twenty-seventh transistor M27. The drain of the twenty-ninth transistor M29 is connected to the collector of a fourth triode Q4. The emitter of the fourth triode Q4 is connected to the second end of the eleventh resistor R11. The base of the fourth triode Q4 is connected to the source of a thirty-first transistor M31. The gate of the thirty-first transistor M31 is connected to the drain of the twenty-ninth transistor M29. The drain of the thirty-first transistor M31 is connected to the drain of a thirtieth transistor M30. The drain of the thirtieth transistor M30 is connected to the gate of the thirtieth transistor M30. The gate of the thirtieth transistor M30 is connected to the gate of a thirty-second transistor M32. The source of the thirty-second transistor M32 is the same as the source of the thirtieth transistor M30. The source of the thirty-first transistor M31 is connected to the first end of a twelfth resistor R12. The second end of the twelfth resistor R12 is connected to the emitter of the fourth triode Q4. The second end of the twelfth resistor R12 is connected to the source of a thirty-third transistor M33. The drain of the thirty-third transistor M33 is connected to the drain of the thirty-second transistor M32. The gate of the thirty-third transistor M33 is connected to the drain of the thirty-third transistor M33. The drain current of the thirty-third transistor M33 is current I3. The gate of the thirty-third transistor M33 is connected to the gate of a thirty-fourth transistor M34. The drain of the thirty-fourth transistor M34 is connected to the drain of the twenty-sixth transistor M26. The source of the thirty-fourth transistor M34 is connected to the source of the thirty-third transistor M33.

[0028] As a further solution of the present invention: The high-temperature section current output circuit includes a thirty-fifth transistor M35. The source of the thirty-fifth transistor M35 is connected to the source of the thirty-second transistor M32. The gate of the thirty-fifth transistor M35 is connected to the gate of the twenty-ninth transistor M29. The drain of the thirty-fifth transistor M35 is connected to the drain of the thirty-sixth transistor M36. The drain of the thirty-sixth transistor M36 is connected to its gate. The source of the thirty-sixth transistor M36 is connected to the source of the thirty-seventh transistor M37. The source of the thirty-seventh transistor M37 is connected to the source of the thirty-fourth transistor M34. The drain of the thirty-seventh transistor M37 is connected to the trimmed current. The gate of the thirty-seventh transistor M37 is connected to the gate of the thirty-sixth transistor M36.

[0029] In a second aspect, a chip is provided, which includes the low-temperature drift segmented temperature compensation bandgap reference circuit as described in the above solution.

[0030] In a third aspect, a control device is provided, which includes the low-temperature drift segmented temperature compensation bandgap reference circuit as described in the above solution.

[0031] Compared with the prior art, the beneficial effects of the present invention are as follows:

[0032] 1. In the present invention, based on the segmented temperature compensation module of the first-order compensation bandgap reference circuit, the temperature of the compensation current in the high-temperature section of the segmented temperature compensation module is corrected, which can effectively solve the problem of over-compensation in the high-temperature section, broaden the high-precision temperature range of the bandgap reference, and achieve a lower temperature drift coefficient.

[0033] 2. In the present invention, the high-temperature trimming module generates a reverse compensation current containing quadratic terms, logarithmic terms, and linear terms in the high-temperature section, which can reduce the slope of the compensation current in the high-temperature section of the segmented temperature compensation module. On the other hand, by adjusting the current slope proportionally, the adjustment accuracy of the current slope can be higher, which is beneficial to further improving the temperature compensation effect, avoiding reducing the compensation current in the medium-high temperature section during the process of reducing the compensation current in the high-temperature section, and preventing weakening the compensation effect in the medium-high temperature section.

[0034] 3. In the present invention, by modifying the parameters of the high-temperature trimming module, the effect of the segmented temperature compensation module can be adjusted, realizing the adaptability to different process platforms and designs. At the same time, based on the improvement of the traditional segmented temperature bandgap reference, the structure is simple and easy to implement in integrated circuits. BRIEF DESCRIPTION OF THE DRAWINGS

[0035] Figure 1 It is the structure diagram of the low-temperature drift segmented temperature compensation bandgap reference of the present invention;

[0036] Figure 2 It is the circuit diagram of the segmented temperature compensation module of the present invention;

[0037] Figure 3 is the circuit diagram of the high-temperature trimming module of the present invention;

[0038] Figure 4 is the schematic diagram of the output voltage of the low-temperature-drift segmented temperature compensation bandgap reference circuit of the present invention.

[0039] In the figure: 1. First-order temperature compensation module; 2. Segmented temperature compensation module; 3. High-temperature trimming module; 301. First current generation circuit; 302. Second current generation circuit; 303. Third current generation circuit; 304. High-temperature section current output circuit. Specific embodiments

[0040] Next, the technical solutions in the embodiments of the present invention will be clearly and completely described in conjunction with the accompanying drawings in the embodiments of the present invention. Obviously, the described embodiments are only a part of the embodiments of the present invention, rather than all the embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those of ordinary skill in the art without creative efforts shall fall within the protection scope of the present invention.

[0041] Embodiment:

[0042] Please refer to Figures 1-4 , in the embodiment of the present invention, a low-temperature-drift segmented temperature compensation bandgap reference circuit includes a first-order temperature compensation module 1, a segmented temperature compensation module 2, and a high-temperature trimming module 3. The first-order temperature compensation module 1 is used to generate a reference voltage according to the temperature characteristics of the triode. The temperature characteristics of the triode are the negative temperature characteristics of the triode reference voltage in the first-order temperature compensation module 1 and the positive temperature characteristics of the difference between the two triode reference voltages. The output end of the segmented temperature compensation module 2 is connected to the first-order temperature compensation module 1. The segmented temperature compensation module 2 is used to superimpose the generated compensation current on the first-order temperature compensation module 1 to reduce the temperature drift of the reference voltage. The compensation current decreases with the increase of temperature in the low-temperature section and increases with the increase of temperature in the high-temperature section. The output end of the high-temperature trimming module 3 is connected to the input end of the segmented temperature compensation module 2. The high-temperature trimming module 3 is used to generate a reverse compensation current containing quadratic terms, logarithmic terms, and linear terms in the high-temperature section to reduce the slope of the compensation current in the high-temperature section of the segmented temperature compensation module 2. The high-temperature trimming module 3 includes a first current generation circuit 301, a second current generation circuit 302, a third current generation circuit 303, and a high-temperature section current output circuit 304. By adjusting the current slope in proportion, the adjustment accuracy of the current slope can be higher, which is beneficial to further improving the temperature compensation effect.

[0043] Preferably, the first-order temperature compensation module 1 includes a first resistor R1 and a second resistor R2. The first ends of both the first resistor R1 and the second resistor R2 are connected to a high level. The first end of the first resistor R1 is connected to the first end of the second resistor R2. The second end of the first resistor R1 is connected to the positive input terminal of the first differential amplifier AMP1. The positive input terminal of the first differential amplifier AMP1 is electrically connected to the drain of the first transistor M1. The source of the first transistor M1 is connected to the collector of the first triode Q1. The emitter of the first triode Q1 is connected to the first end of the third resistor R3. The second end of the third resistor R3 is connected to the emitter of the second triode Q2. The emitter of the second triode Q2 is connected to the first end of the fourth resistor R4. The second end of the fourth resistor R4 is connected to the first end of the fifth resistor R5. The second end of the fifth resistor R5 is grounded. The gate of the second triode Q2 is connected to the first end of the sixth resistor R6. The first end of the sixth resistor R6 is connected to the first end of the seventh resistor R7. The second end of the seventh resistor R7 is connected to the first end of the eighth resistor R8. The second end of the eighth resistor R8 is connected to the first end of the ninth resistor R9. The second end of the ninth resistor R9 is connected to the second end of the sixth resistor R6. The collector of the second triode Q2 is connected to the source of the second transistor M2. The drain of the second transistor M2 is connected to the second end of the second resistor R2. The second end of the sixth resistor R6 is grounded. The second end of the second resistor R2 is connected to the negative input terminal of the first differential amplifier AMP1. The output terminal of the first differential amplifier AMP1 is connected to the gate of the third transistor M3. The source of the third transistor M3 is connected to the first end of the second resistor R2. The drain of the third transistor M3 is connected to the source of the fourth transistor M4. The drain of the fourth transistor M4 is connected to the gate of the second triode Q2. The gate of the fourth transistor M4 is connected to the bias voltage Vbias;

[0044] The gates of the first transistor M1 and the second transistor M2 are both connected to the bias voltage Vbias, which is used to improve the power supply rejection ratio of the bandgap reference circuit.

[0045] The reference voltage is the voltage V between the base and the emitter of the triode BE：

[0046] In the formula, V bg0 is the bandgap voltage, V bg0 = 1.2V, V BE (T 0 ) is the V 0 value at temperature T BE , V BE (T 0 ) is a constant, η is a constant related to the process, and m depends on the collector current of the triode;

[0047] The difference △V BE of V BE between the two triodes is:

[0048] Wherein, V T is the thermodynamic temperature, k is the Boltzmann constant, q is the electronic charge quantity, and the ratio of the emitter areas of the first triode Q1 and the second triode Q2 is N.

[0049] Preferably, the resistance value of the first resistor R1 is equal to that of the second resistor R2, the sizes of the first transistor M1 and the second transistor M2 are the same, and the ratio of the area of the emitter of the first triode Q1 to that of the emitter of the second triode Q2 is 8:1.

[0050] Preferably, the bandgap reference voltage output of the first-order temperature compensation module 1 is:

[0051]

[0052] Wherein, △V BE is the voltage difference between the base and the emitter of both the first triode Q1 and the second triode Q2, R3 is the resistance value of the third resistor R3, R4 is the resistance value of the fourth resistor R4, R5 is the resistance value of the fifth resistor R5, and V BE is the voltage between the base and the emitter of the second triode Q2. By adjusting the resistance values of the third resistor R3, the fourth resistor R4, and the fifth resistor R5 to appropriate values, a bandgap reference voltage that hardly changes with temperature can be obtained. The appropriate values are the resistance values of the third resistor R3, the fourth resistor R4, and the fifth resistor R5 when it is measured that the bandgap reference voltage does not change with temperature.

[0053] When both the first triode Q1 and the second triode Q2 are NPN-type triodes, the voltage difference △V BE between the base and the emitter of both the first triode Q1 and the second triode Q2 is a positive temperature coefficient voltage. When both the first triode Q1 and the second triode Q2 are PNP-type triodes, the voltage difference △V BE between the base and the emitter of both the first triode Q1 and the second triode Q2 is a positive temperature coefficient voltage.

[0054] Preferably, the seventh resistor R7, the eighth resistor R8, and the ninth resistor R9 form a voltage-dividing resistor unit. The voltage-dividing resistor unit is used to perform resistor voltage division on the bandgap reference voltage and output the voltage-divided bandgap reference voltage to the segmented temperature compensation module 2.

[0055] Preferably, the third transistor M3 and the fourth transistor M4 form a set of cascode structures. The gates of the third transistor M3 and the fourth transistor M4 are connected with several sets of cascode structures. The cascode structures are used to generate the reference current I ref :

[0056] The base currents of the first triode Q1 and the second triode Q2 are both set to 0, which is beneficial to improving the negative feedback coefficient and the stability of the system. By adjusting the number of cascode structures in the first-order temperature compensation module 1, the effect of outputting the reference current in the required proportion can be achieved.

[0057] Preferably, the segmented temperature compensation module 2 includes a current source 201, a current replication unit 202, and a reference voltage compensation unit 203. The output end of the current source 201 is connected to the input end of the current replication unit 202. The current replication unit 202 is used to replicate the output current of the current source 201, and the reference voltage compensation unit 203 is used to compensate the reference voltage of the first-order temperature compensation module 1.

[0058] Preferably, the current source 201 includes a second differential amplifier AMP2, a fifth transistor M5, a sixth transistor M6, and a tenth resistor R10. The positive input end of the second differential amplifier AMP2 is connected to the bandgap reference voltage V ref , the negative input end of the second differential amplifier AMP2 is connected to the first end of the tenth resistor R10, the output end of the second differential amplifier AMP2 is connected to the gate of the fifth transistor M5, the drain of the fifth transistor M5 is connected to the source of the sixth transistor M6, and the drain of the sixth transistor M6 is connected to the negative input end of the second differential amplifier AMP2.

[0059] Preferably, the current replication unit 202 includes a seventh transistor M7, an eighth transistor M8, a ninth transistor M9, and a tenth transistor M10. The source of the seventh transistor M7 is connected to the source of the fifth transistor M5, the drain of the seventh transistor M7 is connected to the source of the eighth transistor M8, the gate of the eighth transistor M8 is connected to the gate of the sixth transistor M6, the source of the ninth transistor M9 is connected to the source of the seventh transistor M7, the drain of the ninth transistor M9 is connected to the source of the tenth transistor M10, and the gate of the tenth transistor M10 is connected to the gate of the eighth transistor M8.

[0060] Preferably, the reference voltage compensation unit 203 includes an eleventh transistor M11, a twelfth transistor M12, a thirteenth transistor M13, and a fourteenth transistor M14. The source of the eleventh transistor M11 is connected to the source of the twelfth transistor M12, the source of the twelfth transistor M12 is connected to the drain of the eighth transistor M8, the drain of the eleventh transistor M11 is connected to the second end of the tenth resistor R10, the drain of the twelfth transistor M12 is connected to the drain of the fourteenth transistor M14, the source of the fourteenth transistor M14 is connected to the source of the thirteenth transistor M13, the drain of the thirteenth transistor M13 is connected to the drain of the eleventh transistor M11, and the gates of the fourteenth transistor M14 and the eleventh transistor M11 are both connected to the positive temperature coefficient voltage V ptat , the gate of the twelfth transistor M12 is connected to the bandgap reference voltage Vrefx , the gate of the thirteenth transistor M13 is connected to the bandgap reference voltage V refy , the eleventh transistor M11 and the twelfth transistor M12 are the first pair of differential pair transistors, the thirteenth transistor M13 and the fourteenth transistor M14 are the second pair of differential pair transistors, and the eleventh transistor M11, the twelfth transistor M12, the thirteenth transistor M13, and the fourteenth transistor M14 all operate in the subthreshold region. At room temperature, the magnitude relationship of V refx , V refy and V ptat is:

[0061] Vrefx > V ptat > Vrefy;

[0062] The drain of the twelfth transistor M12 generates a compensation current I comp1 for the high-temperature section, and the drain of the fourteenth transistor M14 generates a compensation current I comp2 for the low-temperature section. The total compensation current I comp is the sum of the compensation current I comp1 and the compensation current I comp2 for the low-temperature section, that is:

[0063] I comp = I comp1 + I comp2 .

[0064] For the first pair of differential pair transistors or the second pair of differential pair transistors, if Vrefx > V ptat , then V GS_M11 > V GS_M12 , and all the current provided by the tail current source flows into the branch where the eleventh transistor M11 is located, that is:

[0065] The current formula for a transistor operating in the subthreshold region can be expressed as:

[0066] In the formula, η is the subthreshold slope factor, I 0 is the subthreshold current pre-factor, so:

[0067]

[0068] It is not easy to see the trend of the above formula I comp1 changing with temperature, and it can be transformed. Let V ptat = aT + b, where a and b are both constants, so:

[0069]

[0070] Since V refx > Vptat , so b - V refx <0, so I comp1 increases with the increase of temperature.

[0071] It should be noted that the formula is only valid when V ptat is close to V refx i.e., it is valid in the high - temperature range. So in the full - temperature range:

[0072]

[0073] So the overall trend of I comp1 changing with temperature can be obtained: in the normal - temperature and low - temperature ranges, i.e., T < T 2 , I comp1 is approximately equal to 0. In the high - temperature range, i.e., T > T 2 , I comp1 increases with the increase of temperature. Similarly, it can be known that:

[0074]

[0075] For the design of V refy , in the low - temperature range, V refy > V ptat , in the normal - temperature and high - temperature ranges, V refy < V ptat , so in the low - temperature range, V refy - b > 0, so I comp2 decreases with the increase of temperature. So the overall trend of I comp2 changing with temperature can be obtained: in the low - temperature range, i.e., T < T 1 , I comp2 decreases with the increase of temperature. In the normal - temperature and high - temperature ranges, i.e., T > T 1 , I comp2 is approximately equal to 0. That is:

[0076]

[0077] The overall trend of I comp changing with temperature: it decreases with the increase of temperature in the low - temperature range, remains unchanged at normal temperature, and increases with the increase of temperature in the high - temperature range, so as to compensate for the reference voltage V bg whose output image of the first - order bandgap reference is a parabola opening downward, greatly reducing the temperature drift of the reference voltage output by the bandgap reference.

[0078] It should be noted that: for the traditional segmented - temperature - compensated bandgap reference, because the compensation current has an exponential relationship with temperature, there will be an over - compensation situation in the high - temperature range. Assume that at temperature T > T 3 (T 3 > T 2) When overcompensation occurs, if the compensation current is simply reduced proportionally, that is, the Ip in the above formula is reduced proportionally to reduce the compensation current, although the compensation current in the high-temperature section T>T 3 can be reduced, but at the same time, the compensation current in the medium-high temperature section T 2 <T<T 3 is also reduced, weakening the compensation effect of the medium-high temperature section without overcompensation. This problem can be solved by the high-temperature trimming module 3 in the present invention.

[0079] Preferably, the high-temperature trimming module 3 includes a first current generation circuit 301, a second current generation circuit 302, a third current generation circuit 303, and a high-temperature section current output circuit 304;

[0080] The first current generation circuit 301 is used to generate a current including a quadratic relationship. The current including a quadratic relationship changes with temperature, and the quadratic relationship current and the changing temperature present a quadratic relationship; the second current generation circuit 302 is used to generate a current including a linear relationship. The current including a linear relationship changes with temperature, and the linear relationship current and the changing temperature present a linear relationship; the third current generation circuit 303 is used to generate a current including a linear relationship and a logarithmic relationship. The current including a linear relationship and a logarithmic relationship changes with temperature, and the linear relationship and logarithmic relationship currents and the changing temperature present a linear relationship and a logarithmic relationship;

[0081] The current including a quadratic relationship, the current including a linear relationship, and the current including a logarithmic relationship are added proportionally to obtain an added current. At the temperature of overcompensation, the high-temperature section current output circuit 304 subtracts the added current from the current including a linear relationship to obtain a high-temperature correction current.

[0082] Preferably, the first current generating circuit 301 includes a fifteenth transistor M15. The drain of the fifteenth transistor M15 is connected to the drain of a sixteenth transistor M16. The source of the fifteenth transistor M15 is connected to the source of a seventeenth transistor M17. The gate of the fifteenth transistor M15 is connected to the gate of the seventeenth transistor M17. The drain of the seventeenth transistor M17 is connected to the drain of an eighteenth transistor M18. The source of the eighteenth transistor M18 is connected to the gate of the sixteenth transistor M16. The source of the sixteenth transistor M16 is connected to the source of a nineteenth transistor M19. The source of the nineteenth transistor M19 is grounded. The gate of the nineteenth transistor M19 is connected to the gate of the sixteenth transistor M16. The drain of the nineteenth transistor M19 is connected to the source of the eighteenth transistor M18. The gate of the eighteenth transistor M18 is connected to the gate of a twenty-first transistor M21. The drain of the twenty-first transistor M21 is connected to the drain of a twentieth transistor M20. The drain of the twentieth transistor M20 is connected to the gate of the twenty-first transistor M21. The source of the twenty-first transistor M21 is grounded. The gate of the twentieth transistor M20 is connected to the gate of the seventeenth transistor M17. The source of the twentieth transistor M20 is connected to a high level. The source of the twentieth transistor M20 is connected to the source of a twenty-second transistor M22. The gate of the twenty-second transistor M22 is connected to the gate of the twentieth transistor M20. The drain of the twenty-second transistor M22 is connected to the drain of a twenty-third transistor M23. The drain of the twenty-third transistor M23 is connected to the gate of the twenty-third transistor M23. The gate of the twenty-third transistor M23 is connected to the drain of the twenty-second transistor M22. The gate of the twenty-third transistor M23 is connected to the gate of a twenty-fourth transistor M24. The source of the twenty-fourth transistor M24 is connected to the source of the twenty-third transistor M23. The source of the twenty-third transistor M23 is grounded. The drain of the twenty-fourth transistor M24 is connected to the output terminal of a bias current source Ibi as The output terminal of bias is connected to the drain of a twenty-fifth transistor M25. The drain of the twenty-fifth transistor M25 is connected to the gate of the twenty-fifth transistor M25. The gate of the twenty-fifth transistor M25 is connected to the gate of a twenty-sixth transistor M26. The source of the twenty-sixth transistor M26 is connected to the source of the twenty-fifth transistor M25. The source of the twenty-fifth transistor M25 is grounded;

[0083] The fifteenth transistor M15, the seventeenth transistor M17, and the twentieth transistor M20 have the same size. The eighteenth transistor M18 and the twenty-first transistor M21 operate in the subthreshold region. The drain current of the eighteenth transistor M18 is current I1. The current between the gate and the drain of the twenty-fifth transistor M25 is current I2. The voltage difference △V GS between the source and the drain of the eighteenth transistor M18 and the twenty-first transistor M21 BEIf the voltage characteristics are similar, then;

[0084]

[0085] where α M18 and α M21 respectively represent the products of the transconductance factor K and the width-to-length ratio W / L of the eighteenth transistor M18 and the twenty-first transistor M21. Since the nineteenth transistor M19 and the sixteenth transistor M16 are both operating in the saturation region, the on-resistance of the nineteenth transistor M19 is:

[0086]

[0087] Substituting back into the formula for I 1 yields:

[0088]

[0089] It can be seen that I 1 is a current that varies quadratically with temperature. In the first current generation circuit 301, the ratio of the sizes of the twenty-second transistor M22 and the seventeenth transistor M17 is m; the sizes of the twenty-third transistor M23 and the twenty-fourth transistor M24 are the same; the sizes of the twenty-fifth transistor M25 and the twenty-sixth transistor M26 are the same; I bias is the reference current, and in this example, the reference current output by the bandgap reference is taken out according to the designed ratio. Therefore:

[0090] I 2 = I bias - mk 1 T 2 ,

[0091] That is, I 2 is the CTAT 2 current.

[0092] Preferably, the second current generation circuit 302 includes a twenty-seventh transistor M27. The source of the twenty-seventh transistor M27 is connected to the source of the twenty-second transistor M22. The drain of the twenty-seventh transistor M27 is connected to the drain of the twenty-eighth transistor M28. The drain of the twenty-eighth transistor M28 is connected to the gate of the twenty-seventh transistor M27. The gate of the twenty-eighth transistor M28 is connected to the bias voltage V bias . The source of the twenty-eighth transistor M28 is connected to the collector of the third triode Q3. The base of the third triode Q3 is connected to the reference voltage V bg . The emitter of the third triode Q3 is connected to the first end of the eleventh resistor R11. The second end of the eleventh resistor R11 is connected to the source of the twenty-sixth transistor M26.

[0093] Preferably, the third current generation circuit 303 includes a twenty-ninth transistor M29. The gate of the twenty-ninth transistor M29 is connected to the gate of the twenty-seventh transistor M27. The source of the twenty-ninth transistor M29 is connected to the source of the twenty-seventh transistor M27. The drain of the twenty-ninth transistor M29 is connected to the collector of a fourth triode Q4. The emitter of the fourth triode Q4 is connected to the second end of an eleventh resistor R11. The base of the fourth triode Q4 is connected to the source of a thirty-first transistor M31. The gate of the thirty-first transistor M31 is connected to the drain of the twenty-ninth transistor M29. The drain of the thirty-first transistor M31 is connected to the drain of a thirtieth transistor M30. The drain of the thirtieth transistor M30 is connected to the gate of the thirtieth transistor M30. The gate of the thirtieth transistor M30 is connected to the gate of a thirty-second transistor M32. The source of the thirty-second transistor M32 is the same as the source of the thirtieth transistor M30. The source of the thirty-first transistor M31 is connected to the first end of a twelfth resistor R12. The second end of the twelfth resistor R12 is connected to the emitter of the fourth triode Q4. The second end of the twelfth resistor R12 is connected to the source of a thirty-third transistor M33. The drain of the thirty-third transistor M33 is connected to the drain of the thirty-second transistor M32. The gate of the thirty-third transistor M33 is connected to the drain of the thirty-third transistor M33. The drain current of the thirty-third transistor M33 is current I3. The gate of the thirty-third transistor M33 is connected to the gate of a thirty-fourth transistor M34. The drain of the thirty-fourth transistor M34 is connected to the drain of the twenty-sixth transistor M26. The source of the thirty-fourth transistor M34 is connected to the source of the thirty-third transistor M33.

[0094] The second current generation circuit 302 is a positive temperature coefficient current generation circuit. The base of the third triode Q3 is connected to the reference voltage V of the bandgap reference output terminal. bg , and the twenty-eighth transistor M28 is connected to a bias voltage. Therefore, the branch current of the second current generation circuit 302 is:

[0095]

[0096] In the third current generation circuit 303 that generates a current including a linear relationship and a logarithmic relationship, the twenty-ninth transistor M29 and the twenty-seventh transistor M27 have the same size. The thirtieth transistor M30 and the thirty-fourth transistor M34 have the same size. The thirty-third transistor M33 and the thirty-fourth transistor M34 have the same size. Therefore:

[0097]

[0098] That is, I 3 is a current that decreases with increasing temperature and includes a linear term and a logarithmic term.

[0099] Preferably, the high-temperature section current output circuit 304 includes a thirty-fifth transistor M35. The source of the thirty-fifth transistor M35 is connected to the source of the thirty-second transistor M32. The gate of the thirty-fifth transistor M35 is connected to the gate of the twenty-ninth transistor M29. The drain of the thirty-fifth transistor M35 is connected to the drain of the thirty-sixth transistor M36. The drain of the thirty-sixth transistor M36 is connected to its gate. The source of the thirty-sixth transistor M36 is connected to the source of the thirty-seventh transistor M37. The source of the thirty-seventh transistor M37 is connected to the source of the thirty-fourth transistor M34. The drain of the thirty-seventh transistor M37 is connected to the trimming current. The gate of the thirty-seventh transistor M37 is connected to the gate of the thirty-sixth transistor M36.

[0100] In the high-temperature section current output circuit 304, the size ratio of the thirty-fifth transistor M35 to the twenty-seventh transistor M27 is n. Therefore, the drain current of the thirty-fifth transistor M35:

[0101] I D_M35 = nI PTAT ,

[0102] The thirty-sixth transistor M36 and the thirty-seventh transistor M37 have the same size. Assume that the temperature when over-compensation occurs for high-temperature is T3. It is set that when the temperature is less than T3, I 2 + I 3 > I D_M35 . At this time, if the twenty-sixth transistor M26 and the thirty-fourth transistor M34 are operating in the saturation region, then: I D_M26 + I D_M34 = I 2 + I 3 > I D_M35 , but according to Kirchhoff's current law, this is obviously incorrect. Therefore, the twenty-sixth transistor M26 and the thirty-fourth transistor M34 must be operating in the linear region. At this time, the potential at point c is pulled down, and the thirty-sixth transistor M36 and the thirty-seventh transistor M37 are turned off. It is set that when the temperature is greater than T3, I 2 + I 3 < I D_M35 . The twenty-sixth transistor M26 and the thirty-fourth transistor M34 are operating in the saturation region, and the thirty-sixth transistor M36 and the thirty-seventh transistor M37 are turned on:

[0103]

[0104] Rearranging the above formula gives:

[0105] I trim = AT 2 + BTlnT + CT - D,

[0106] Where A, B, C, and D are the simplified coefficient terms and constant terms in the formula, and all are greater than 0. Among them, A, B, and C can reach the designed values by adjusting the ratio of the sizes of the nineteenth transistor M19 and the sixteenth transistor M16, the resistance value of the twelfth resistor R12, and the resistance value of the eleventh resistor R11 respectively. According to the properties of the mathematical function itself, the correction effect T in the high-temperature section 2 >TlnT>T, so appropriate coefficients can be selected according to the actual over-compensation situation in high-temperature compensation to achieve more accurate correction. For example, when the over-compensation situation in high-temperature compensation is relatively serious, adjust the circuit parameters to make the coefficients A, B, and C larger and the coefficient A the largest; when the over-compensation situation in high-temperature compensation is relatively mild, adjust the circuit parameters to make the coefficients A, B, and C smaller and the coefficient A the smallest. According to the simulation results, it is determined that the temperature at which over-compensation occurs in the high-temperature section of the segmented temperature compensation module is T3. At the same time, by adjusting the value of n, it can be realized that when T>T2, the compensation current is:

[0107]

[0108] It can be realized that in the medium-high temperature section T2<T<T3, the compensation effect is not weakened, and in the high-temperature section, the over-compensation current is reduced, so as to obtain a bandgap reference voltage with lower temperature drift.

[0109] Please refer to Figure 4 , which shows the schematic diagram of the output voltage of the low-temperature-drift segmented temperature compensation bandgap reference. The high-temperature compensation current correction module of the present invention makes the output voltage of the bandgap reference no longer show an exponential trend of increase in the high-temperature section T>130°C. The design of a bandgap reference with a low temperature drift of 1 ppm / °C in a wide temperature range of -60°C to 150°C is realized.

[0110] The implementation scenario of the present invention is that when designing a segmented temperature compensation bandgap reference, there is an over-compensation situation in the high-temperature section. If the designer simply reduces the high-temperature compensation current proportionally, although the over-compensation situation in the high-temperature section is improved, the compensation effect in the medium-high temperature section will be weakened at the same time. In order to meet the design requirements of a high-precision bandgap reference with low temperature drift, the present invention introduces currents that change in a quadratic relationship, a logarithmic relationship, and a linear relationship with temperature in the high-temperature section, only weakening the over-compensation current part in the high-temperature section while keeping the compensation effect in the medium-high temperature section unchanged.

[0111] It can be obtained from Figure 1In summary, the first-order temperature compensation module 1 generates a bandgap reference voltage that is almost independent of temperature. At this time, the bandgap reference voltage cannot meet the requirements of low temperature drift. Therefore, the segmented temperature compensation module 2 is introduced. By introducing a compensation current that decreases with increasing temperature in the low-temperature section and a compensation current that increases with increasing temperature in the high-temperature section, a reference voltage with lower temperature drift is achieved at the output of the bandgap reference circuit. However, during segmented temperature compensation, due to the exponential relationship, overcompensation will occur in the high-temperature section. If the high-temperature compensation current is simply reduced proportionally, the compensation effect in the middle and high-temperature sections will be weakened simultaneously. Therefore, the high-temperature trimming module 3 is introduced. By introducing a current that shows a higher-order term change with temperature in the high-temperature section, only the overcompensation current part in the high-temperature section is weakened, while the compensation effect in the middle and high-temperature sections remains unchanged, enabling the segmented temperature compensation bandgap reference to obtain a lower temperature coefficient and a wider temperature range, thus meeting the design requirements.

[0112] In a second aspect, the present invention also provides a chip, including the low-temperature-drift segmented temperature compensation bandgap reference circuit as described in the above solution.

[0113] In a third aspect, the present invention also provides a control device, including the low-temperature-drift segmented temperature compensation bandgap reference circuit as described in the above solution.

[0114] The above is only a preferred specific embodiment of the present invention, but the protection scope of the present invention is not limited thereto. Any person skilled in the art within the technical scope disclosed by the present invention, according to the technical solution and inventive concept of the present invention, making equivalent substitutions or changes, should be covered within the protection scope of the present invention.

Claims

1. A low temperature drift segmented temperature compensation bandgap reference circuit, characterized in that: include: A first-order temperature compensation module, the first-order temperature compensation module is used to generate a reference voltage according to the temperature characteristics of the transistor; A segmented temperature compensation module, wherein the output end of the segmented temperature compensation module is connected to the first-order temperature compensation module, and the segmented temperature compensation module is used to superimpose the generated compensation current on the first-order temperature compensation module to reduce the temperature drift of the reference voltage; A high temperature adjustment module, wherein the output end of the high temperature adjustment module is connected to the input end of the segmented temperature compensation module, and the high temperature adjustment module is used to generate a reverse compensation current containing a quadratic term, a logarithmic term and a linear term in the high temperature segment to reduce the slope of the compensation current in the high temperature segment of the segmented temperature compensation module, and the high temperature adjustment module includes a first current generating circuit, a second current generating circuit, a third current generating circuit and a high temperature segment current output circuit; The first current generating circuit is used to generate a current having a quadratic relationship, the current having a quadratic relationship varies with temperature, and the current having a quadratic relationship and the varying temperature present a quadratic relationship; The second current generating circuit is used to generate a current including a linear relationship, the current including the linear relationship changes with temperature, and the current including the linear relationship presents a linear relationship with the changing temperature; the third current generating circuit is used to generate a current including a linear relationship and a logarithmic relationship, the current including the linear relationship and the logarithmic relationship changes with temperature, and the current including the linear relationship and the logarithmic relationship presents a linear relationship and a logarithmic relationship with the changing temperature; The current containing a quadratic relationship, the current containing a linear relationship and the current containing a logarithmic relationship are proportionally added to obtain an added current. At the over-compensation temperature, the high-temperature section current output circuit subtracts the added current from the current containing a linear relationship to obtain a high-temperature corrected current.

2. The low temperature drift segmented temperature compensation bandgap reference circuit according to claim 1, characterized in that: The first-order temperature compensation module includes a first resistor R1 and a second resistor R2, wherein the first end of the first resistor R1 and the first end of the second resistor R2 are both connected to a high level, the first end of the first resistor R1 is connected to the first end of the second resistor R2, the second end of the first resistor R1 is connected to the positive input end of the first differential amplifier AMP1, the positive input end of the first differential amplifier AMP1 is electrically connected to the drain of the first transistor M1, the source of the first transistor M1 is connected to the collector of the first triode Q1, the emitter of the first triode Q1 is connected to the first end of the third resistor R3, the second end of the third resistor R3 is connected to the emitter of the second triode Q2, the emitter of the second triode Q2 is connected to the first end of the fourth resistor R4, the second end of the fourth resistor R4 is connected to the first end of the fifth resistor R5, the second end of the fifth resistor R5 is grounded, the base of the second triode Q2 is connected to the first end of the sixth resistor R6, the base of the first triode Q1 is connected to The first end of the resistor R6, the first end of the sixth resistor R6 is connected to the first end of the seventh resistor R7, the second end of the seventh resistor R7 is connected to the first end of the eighth resistor R8, the second end of the eighth resistor R8 is connected to the first end of the ninth resistor R9, the second end of the ninth resistor R9 is connected to the second end of the sixth resistor R6, the collector of the second transistor Q2 is connected to the source of the second transistor M2, the drain of the second transistor M2 is connected to the second end of the second resistor R2, the second end of the sixth resistor R6 is grounded, the second end of the second resistor R2 is connected to the negative input end of the first differential amplifier AMP1, the output end of the first differential amplifier AMP1 is connected to the gate of the third transistor M3, the source of the third transistor M3 is connected to the first end of the second resistor R2, the drain of the third transistor M3 is connected to the source of the fourth transistor M4, the drain of the fourth transistor M4 is connected to the base of the second transistor Q2, and the gate of the fourth transistor M4 is connected to the bias voltage V bias ; The gate of the first transistor M1 and the gate of the second transistor M2 are both connected to the bias voltage V bias , used to improve the power supply rejection ratio of the bandgap reference circuit.

3. The low temperature drift segmented temperature compensation bandgap reference circuit according to claim 2, characterized in that: The resistance value of the first resistor R1 is equal to the resistance value of the second resistor R2, the sizes of the first transistor M1 and the second transistor M2 are the same, and the area ratio of the emitter of the first transistor Q1 to the emitter of the second transistor Q2 is 8:

1.

4. The low temperature drift segmented temperature compensation bandgap reference circuit according to claim 3, characterized in that: The reference voltage output of the first-order temperature compensation module is: Where, △V BE is the voltage difference between the base and emitter voltages of the two transistors, R3 is the resistance value of the third resistor R3, R4 is the resistance value of the fourth resistor R4, R5 is the resistance value of the fifth resistor R5, and VBE is the voltage between the base and emitter of the second transistor Q2.

5. The low temperature drift segmented temperature compensation bandgap reference circuit according to claim 4, characterized in that: The seventh resistor R7, the eighth resistor R8 and the ninth resistor R9 constitute a voltage-dividing resistor unit, which is used to divide the reference voltage by resistors and output the divided reference voltage to the segmented temperature compensation module.

6. The low temperature drift segmented temperature compensation bandgap reference circuit according to claim 5, characterized in that: The third transistor M3 and the fourth transistor M4 form a group of cascode structures. The gates of the third transistor M3 and the fourth transistor M4 are connected to a plurality of groups of cascode structures. The cascode structures are used to generate a reference current.

7. The low temperature drift segmented temperature compensation bandgap reference circuit according to claim 1, characterized in that: The segmented temperature compensation module includes a current source, a current replication unit and a reference voltage compensation unit. The output end of the current source is connected to the input end of the current replication unit. The current replication unit is used to replicate the output current of the current source. The reference voltage compensation unit is used to compensate for the reference voltage of the first-order temperature compensation module.

8. The low temperature drift segmented temperature compensation bandgap reference circuit according to claim 7, characterized in that: The current source includes a second differential amplifier AMP2, a fifth transistor M5, a sixth transistor M6 and a tenth resistor R10, the positive input terminal of the second differential amplifier AMP2 is connected to the reference voltage, the negative input terminal of the second differential amplifier AMP2 is connected to the first end of the tenth resistor R10, the second end of the tenth resistor R10 is connected to the first end of the reference voltage compensation unit, the output terminal of the second differential amplifier AMP2 is connected to the gate of the fifth transistor M5, the drain of the fifth transistor M5 is connected to the source of the sixth transistor M6, the drain of the sixth transistor M6 is connected to the negative input terminal of the second differential amplifier AMP2, and the source of the fifth transistor M5 is connected to the first end of the current replication unit.

9. The low temperature drift segmented temperature compensation bandgap reference circuit according to claim 8, characterized in that: The current copy unit includes a seventh transistor M7, an eighth transistor M8, a ninth transistor M9 and a tenth transistor M10, a source of the seventh transistor M7 is connected to the source of the fifth transistor M5, the source of the seventh transistor M7 is the first end of the current copy unit, the drain of the seventh transistor M7 is connected to the source of the eighth transistor M8, the gate of the eighth transistor M8 is connected to the gate of the sixth transistor M6, the source of the ninth transistor M9 is connected to the source of the seventh transistor M7, the drain of the ninth transistor M9 is connected to the source of the tenth transistor M10, the gate of the tenth transistor M10 is connected to the gate of the eighth transistor M8, and the gate of the seventh transistor M7 is connected to the gate of the ninth transistor M9 and the gate of the fifth transistor M5.

10. The low temperature drift segmented temperature compensation bandgap reference circuit according to claim 9, characterized in that: The reference voltage compensation unit includes an eleventh transistor M11, a twelfth transistor M12, a thirteenth transistor M13 and a fourteenth transistor M14, the source of the eleventh transistor M11 is connected to the source of the twelfth transistor M12, the source of the twelfth transistor M12 is connected to the drain of the eighth transistor M8, the drain of the eleventh transistor M11 is connected to the second end of the tenth resistor R10, the drain of the twelfth transistor M12 is connected to the drain of the fourteenth transistor M14, the source of the fourteenth transistor M14 is connected to the source of the thirteenth transistor M13, the source of the thirteenth transistor M13 is connected to the drain of the tenth transistor M10, the drain of the thirteenth transistor M13 is connected to the drain of the eleventh transistor M11, and the gate of the fourteenth transistor M14 and the gate of the eleventh transistor M11 are both connected to the positive temperature coefficient voltage V ptat The gate of the twelfth transistor M12 is connected to the reference voltage V refx The gate of the thirteenth transistor M13 is connected to the reference voltage V refy The eleventh transistor M11 and the twelfth transistor M12 are a first pair of differential pair transistors, the thirteenth transistor M13 and the fourteenth transistor M14 are a second pair of differential pair transistors, the eleventh transistor M11, the twelfth transistor M12, the thirteenth transistor M13 and the fourteenth transistor M14 all operate in a subthreshold region, and the first end of the current replication unit is the drain of the eleventh transistor M11.

11. The low temperature drift segmented temperature compensation bandgap reference circuit according to claim 10, characterized in that: The first current generating circuit includes a fifteenth transistor M15, a drain of the fifteenth transistor M15 is connected to the drain of the sixteenth transistor M16, a source of the fifteenth transistor M15 is connected to the source of the seventeenth transistor M17, a gate of the fifteenth transistor M15 is connected to the gate of the seventeenth transistor M17, a drain of the seventeenth transistor M17 is connected to the drain of the eighteenth transistor M18, a source of the eighteenth transistor M18 is connected to the gate of the sixteenth transistor M16, a source of the sixteenth transistor M16 is connected to the source of the nineteenth transistor M19, a source of the nineteenth transistor M19 is grounded, a gate of the nineteenth transistor M19 is connected to the gate of the sixteenth transistor M16, a drain of the nineteenth transistor M19 is connected to the source of the eighteenth transistor M18, a gate of the eighteenth transistor M18 is connected to the gate of the twenty-first transistor M21, a drain of the twenty-first transistor M21 is connected to the drain of the twentieth transistor M20, and the twentieth transistor M20 is connected to the drain of the twentieth transistor M20. The drain of the 21st transistor M21 is connected to the gate of the 21st transistor M21, the source of the 21st transistor M21 is grounded, the gate of the 20th transistor M20 is connected to the gate of the 17th transistor M17, the source of the 20th transistor M20 is connected to a high level, the source of the 20th transistor M20 is connected to the source of the 22nd transistor M22, the gate of the 22nd transistor M22 is connected to the gate of the 20th transistor M20, the drain of the 22nd transistor M22 is connected to the drain of the 23rd transistor M23, the drain of the 23rd transistor M23 is connected to the gate of the 23rd transistor M23, the gate of the 23rd transistor M23 is connected to the drain of the 22nd transistor M22, the gate of the 23rd transistor M23 is connected to the gate of the 24th transistor M24, the source of the 24th transistor M24 is connected to the source of the 23rd transistor M23, the source of the 23rd transistor M23 is grounded, and the drain of the 24th transistor M24 is connected to the bias current source I bias The output terminal of the bias current source I bias The output end is connected to the drain of the twenty-fifth transistor M25, the drain of the twenty-fifth transistor M25 is connected to the gate of the twenty-fifth transistor M25, the gate of the twenty-fifth transistor M25 is connected to the gate of the twenty-sixth transistor M26, the source of the twenty-sixth transistor M26 is connected to the source of the twenty-fifth transistor M25, the source of the twenty-fifth transistor M25 is grounded, and the sources of the fifteenth transistor M15 and the seventeenth transistor M17 are connected to a high level; The fifteenth transistor M15, the seventeenth transistor M17 and the twentieth transistor M20 have the same size, and the eighteenth transistor M18 and the twenty-first transistor M21 operate in the subthreshold region.

12. The low temperature drift segmented temperature compensation bandgap reference circuit according to claim 11, characterized in that: The second current generating circuit includes a twenty-seventh transistor M27, the source of the twenty-seventh transistor M27 is connected to the source of the twenty-second transistor M22, the drain of the twenty-seventh transistor M27 is connected to the drain of the twenty-eighth transistor M28, the drain of the twenty-eighth transistor M28 is connected to the gate of the twenty-seventh transistor M27, and the gate of the twenty-eighth transistor M28 is connected to the bias voltage V bias The source of the twenty-eighth transistor M28 is connected to the collector of the third transistor Q3, the base of the third transistor Q3 is connected to the reference voltage, the emitter of the third transistor Q3 is connected to the first end of the eleventh resistor R11, and the second end of the eleventh resistor R11 is connected to the source of the twenty-sixth transistor M26.

13. The low temperature drift segmented temperature compensation bandgap reference circuit according to claim 12, characterized in that: The third current generating circuit includes a twenty-ninth transistor M29, the gate of the twenty-ninth transistor M29 is connected to the gate of the twenty-seventh transistor M27, the source of the twenty-ninth transistor M29 is connected to the source of the twenty-seventh transistor M27, the drain of the twenty-ninth transistor M29 is connected to the collector of the fourth transistor Q4, the emitter of the fourth transistor Q4 is connected to the second end of the eleventh resistor R11, the base of the fourth transistor Q4 is connected to the source of the thirty-first transistor M31, the gate of the thirty-first transistor M31 is connected to the drain of the twenty-ninth transistor M29, the drain of the thirty-first transistor M31 is connected to the drain of the thirtieth transistor M30, the drain of the thirtieth transistor M30 is connected to the gate of the thirtieth transistor M30, the gate of the thirtieth transistor M30 is connected to the gate of the thirty-second transistor M32, and the source of the thirty-second transistor M32 is connected to the thirty-first transistor M31. The source of transistor M30 and the source of the thirty-first transistor M31 are connected to the first end of the twelfth resistor R12, the second end of the twelfth resistor R12 is connected to the emitter of the fourth transistor Q4, the second end of the twelfth resistor R12 is connected to the source of the thirty-third transistor M33, the drain of the thirty-third transistor M33 is connected to the drain of the thirty-second transistor M32, the gate of the thirty-third transistor M33 is connected to the drain of the thirty-third transistor M33, the drain current of the thirty-third transistor M33 is current I3, the gate of the thirty-third transistor M33 is connected to the gate of the thirty-fourth transistor M34, the drain of the thirty-fourth transistor M34 is connected to the drain of the twenty-sixth transistor M26, the source of the thirty-fourth transistor M34 is connected to the source of the thirty-third transistor M33, and the sources of the thirtieth transistor M30 and the thirty-second transistor M32 are connected to a high level.

14. The low temperature drift segmented temperature compensation bandgap reference circuit according to claim 13, characterized in that: The high temperature section current output circuit includes a thirty-fifth transistor M35, the source of the thirty-fifth transistor M35 is connected to the source of the thirty-second transistor M32, the gate of the thirty-fifth transistor M35 is connected to the gate of the twenty-ninth transistor M29, the drain of the thirty-fifth transistor M35 is connected to the drain of the thirty-sixth transistor M36, the drain of the thirty-sixth transistor M36 is connected to the gate of the thirty-sixth transistor M36, the source of the thirty-sixth transistor M36 is connected to the source of the thirty-seventh transistor M37, the source of the thirty-seventh transistor M37 is connected to the source of the thirty-fourth transistor M34, the drain of the thirty-seventh transistor M37 is connected to the adjustment current, and the gate of the thirty-seventh transistor M37 is connected to the gate of the thirty-sixth transistor M36.

15. A chip, characterized in that: It comprises the low temperature drift segmented temperature compensation bandgap reference circuit as described in any one of claims 1 to 14.

16. A control device, characterized in that: It comprises the low temperature drift segmented temperature compensation bandgap reference circuit as described in any one of claims 1 to 14.

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