Resonant llc switching power converter and method of controlling the same

Abstract

The application discloses a resonant LLC switching power supply converter and a control method thereof. The resonant LLC switching power supply converter comprises a first switching module, a second switching module, a third switching module, a fourth switching module, a first capacitor module, a second capacitor module, a third capacitor module, an energy storage and isolation module and a control module. The control module controls the first switching module and the second switching module through a first control signal and a second control signal respectively, acquires an output current, compares the output current with a preset current, controls the on-off state of the third switching module according to the comparison result and the state of the first control signal, and controls the on-off state of the fourth switching module according to the comparison result and the state of the second control signal. Through real-time monitoring of the output current, dynamic adjustment of the resonant LLC switching power supply converter is realized, zero-voltage turn-off or turn-on is realized under a large-current load, the capacitor module is turned off under a small-current load, and power loss is reduced.

Description

Technical Field

[0001] This invention relates to the field of switching power supply technology, and in particular to a resonant LLC switching power converter and its control method. Background Technology

[0002] In existing technologies, PC power supplies, server power supplies, LED power supplies, adapters, DIN rail power supplies, and high-end industrial power supplies are all high-frequency switching power supplies with high conversion efficiency requirements. The main power topology of the DC-to-DC / DC / DC isolation converters in these industry power supplies widely uses high-efficiency, low-cost resonant half-bridge LLC switching power supply converters. These converters mainly consist of input filter capacitors, high-voltage side switching transistors of the switching bridge arms, low-voltage side switching transistors of the switching bridge arms, resonant inductors, resonant capacitors, isolation transformers, secondary output rectifier diodes (or synchronous rectifier MOSFETs), output filter capacitors, and control circuitry.

[0003] like Figure 1 As shown, in existing resonant half-bridge LLC switching power converters, the first and second switching modules typically use N-channel field-effect transistors. During the complementary turn-on or turn-off process of the first and second switching modules, the high-current hard-turn-off state of one transistor is a necessary condition for achieving zero-voltage turn-on of the other transistor. Therefore, this serious problem is an inherent defect of existing resonant half-bridge LLC switching power converter topologies and cannot be solved by adjusting component parameters.

[0004] To address the high-current hard-turn-off loss issue of the first and second switching modules of the bridge arm in a resonant LLC switching power converter, the following method is generally adopted:

[0005] 1. Adjust the frequency of the converter when it is operating at full load to near the second resonant frequency of the resonant cavity, so as to minimize the current value when the first or second switching module is turned off and minimize its hard turn-off loss.

[0006] The above technical solution has the following technical drawbacks: Due to the rapid development of the industry economy, the demand for instantaneous peak output power of switching power supplies has increased in industries such as personal computers, automation, and robotic arms. Generally, the peak output power of the power supply is required to reach approximately 2-3 times the average output power. Under this requirement, as mentioned above, in order to ensure that the LLC switching power supply converter can operate normally under any load, its operating frequency f at peak load must be designed. w It reliably exceeds its first resonant frequency f1 of the resonant cavity, thereby causing its operating frequency f under average load to be higher than that of the resonant cavity. wAlso will be much higher than its resonant cavity second resonant frequency f2, at this time, the first switch module or the second switch module is hard off at a large current, so that the switching power supply generates an average shutdown loss increase, electromagnetic interference EMI radiation enhancement, efficiency decline and other serious adverse problems.

[0007] 2. The capacitor is connected in parallel between the drain D and the source S of the first switch module and the second switch module of the resonant half-bridge LLC switching power supply converter, so as to reduce the loss of the switch tube when it is hard off at a large current.

[0008] The above technical solution has the following technical defects: the capacitor is directly connected in parallel between the drain D and the source S of the first switch module and the second switch module, and when the driving pulse is a discontinuous pulse group output, the first switch module and the second switch module are in a hard-on state at the first pulse after the driving pulse is interrupted, at this time, the above-mentioned auxiliary capacitor of the first switch module and the second switch module will increase the switching loss. Because in general, when the resonant LLC switching power supply converter is in an idle standby state and a small load state, the control module adjusts the working frequency to increase to the set maximum safe frequency value in order to stabilize the output voltage, and still cannot realize the stable voltage requirement. At this time, the control module will enter the driving pulse group output state of not stopping interruption in order to stabilize the output voltage, therefore, the above-mentioned prior art method will increase the loss and reduce the efficiency of the LLC switching power supply converter in the standby and small load state. SUMMARY

[0009] The embodiment of the present application provides a resonant LLC switching power supply converter and a control method thereof, so as to solve the above technical problems.

[0010] The first aspect of the embodiment of the present application provides a resonant LLC switching power supply converter, which comprises: the resonant LLC switching power supply converter comprises a first switch module, a second switch module, a third switch module, a fourth switch module, a first capacitor module, a second capacitor module, a third capacitor module, an energy storage and isolation module and a control module.

[0011] The first end of the first switch module is connected to the first end of the second capacitor module, the second end of the first switch module is connected to the first end of the second switch module, the second end of the third switch module, the first end of the third capacitor module and the first end of the energy storage and isolation module respectively, the second end of the second capacitor module is connected to the first end of the third switch module, the second end of the second switch module is connected to the second end of the fourth switch module and the first end of the first capacitor module respectively, the second end of the first capacitor module is connected to the second end of the energy storage and isolation module, the second end of the third capacitor module is connected to the first end of the fourth switch module, and the control module is connected to the control end and the second end of the first switch module, the control end and the second end of the second switch module, the control end and the second end of the third switch module and the control end and the second end of the fourth switch module respectively.

[0012] The control module is configured to output a first control signal to the first switch module to control the on-off state of the first switch module, and output a second control signal to the second switch module to control the on-off state of the second switch module.

[0013] The control module is further configured to obtain an output current of the resonant LLC switching power supply converter, compare the output current with a preset current, and control the on-off state of the third switch module according to the comparison result and the state of the first control signal, and control the on-off state of the fourth switch module according to the comparison result and the state of the second control signal.

[0014] Optionally, the control module comprises:

[0015] A first control module connected to the control end of the first switch module and the control end of the second switch module respectively, configured to output a first control signal to the first switch module to control the on-off state of the first switch module, and output a second control signal to the second switch module to control the on-off state of the second switch module.

[0016] A second control module connected to the control end of the first switch module, the control end of the second switch module, the control end of the third switch module and the control end of the fourth switch module respectively, configured to obtain an output current of the resonant LLC switching power supply converter, compare the output current with a preset current, and control the on-off state of the third switch module according to the comparison result and the state of the first control signal, and control the on-off state of the fourth switch module according to the comparison result and the state of the second control signal.

[0017] Optionally, the second control module comprises:

[0018] a reference voltage module, configured to output a reference voltage;

[0019] a current detection module, a first input terminal of which is connected to a second terminal of the first capacitor module, and a second input terminal of which is connected to an output terminal of the reference voltage module, configured to collect an output current of the resonant LLC switching power supply converter, convert the output current into a collection voltage, and compare the collection voltage with the reference voltage to output a first comparison signal;

[0020] a first control submodule, a first input terminal of which receives the first control signal, a second input terminal of which is connected to the output terminal of the reference voltage module, a third input terminal of which is connected to the output terminal of the current detection module, and an output terminal of which is connected to a control terminal of the third switching module, configured to output a first on-off control signal to the third switching module according to the first control signal, the reference voltage and the first comparison signal;

[0021] a second control submodule, a first input terminal of which receives the second control signal, a second input terminal of which is connected to the output terminal of the reference voltage module, a third input terminal of which is connected to the output terminal of the current detection module, and an output terminal of which is connected to a control terminal of the fourth switching module, configured to output a second on-off control signal to the fourth switching module according to the second control signal, the reference voltage and the first comparison signal.

[0022] Optionally, the first control submodule comprises:

[0023] a first delay trigger module, a first input terminal of which receives the first control signal, a second input terminal of which receives the reference voltage, and a third input terminal of which receives the first comparison signal, configured to output a first weak current on-off control signal according to the first control signal, the reference voltage and the first comparison signal;

[0024] a first driving module, an input terminal of which is connected to an output terminal of the first delay trigger module, configured to output a first on-off control signal after amplifying and isolating the first weak current on-off control signal.

[0025] Optionally, the first delay trigger module comprises:

[0026] a first comparison module, a first input terminal of which receives the first control signal, and a second input terminal of which receives the reference voltage, configured to convert the first control signal into a first collection voltage after time delay, and output a first trigger signal when the first collection voltage is greater than the reference voltage;

[0027] a first trigger module, a first input terminal of which receives the first comparison signal, and a second input terminal of which is connected to an output terminal of the first comparison module, configured to output the first weak current on-off control signal according to the first trigger signal.

[0028] Optionally, the second control submodule comprises:

[0029] a second delay trigger module, a first input end of which receives the second control signal, a second input end of which receives the reference voltage, and a third input end of which receives the first comparison signal, for outputting a second weak current on-off control signal according to the second control signal, the reference voltage and the first comparison signal;

[0030] a second drive module, an input end of which is connected to an output end of the second delay trigger module, for outputting a second on-off control signal after amplification and isolation of the output second weak current on-off control signal.

[0031] Optionally, the second delay trigger module comprises:

[0032] a second comparison module, a first input end of which receives the second control signal, and a second input end of which receives the reference voltage, for converting the second control signal into a second acquisition voltage through time delay, and outputting a second trigger signal when the second acquisition voltage is greater than the reference voltage;

[0033] a second trigger module, a first input end of which receives the first comparison signal, and a second input end of which is connected to an output end of the second comparison module, for outputting the second weak current on-off control signal according to the second trigger signal.

[0034] Optionally, the first switch module is a first MOS tube, the second switch module is a second MOS tube, the first control signal comprises a first voltage control signal and a second voltage control signal, and the second control signal comprises a third voltage control signal and a fourth voltage control signal;

[0035] The resonant LLC switching power supply converter further comprises:

[0036] a first voltage dividing unit, a first end of which receives the first voltage control signal, a second end of which is connected to a gate of the first MOS tube, and a third end of which receives the second voltage control signal after being connected to a source of the first MOS tube, for controlling on-off of the first MOS tube according to the first voltage control signal and the second voltage control signal;

[0037] a second voltage dividing unit, a first end of which receives the third voltage control signal, a second end of which is connected to a gate of the second MOS tube, and a third end of which receives the fourth voltage control signal after being connected to a source of the second MOS tube, for controlling on-off of the second MOS tube according to the third voltage control signal and the fourth voltage control signal;

[0038] The second control module is configured to control the third switch module to start conducting when it is detected that the output current is greater than the preset current and a high-level signal is formed between the first voltage control signal and the second voltage control signal or between the second voltage control signal and the ground terminal of the second control module.

[0039] The second control module is further configured to control the fourth switch module to start conducting when the output current is greater than the preset current and a high-level signal is formed between the third voltage control signal and the fourth voltage control signal.

[0040] The second aspect of the embodiment of the present application provides a control method of the resonant LLC switching power supply converter according to claim 1, which is applied to the control module and includes:

[0041] outputting a first control signal to the first switch module to control the on-off state of the first switch module and outputting a second control signal to the second switch module to control the on-off state of the second switch module;

[0042] obtaining the output current of the resonant LLC switching power supply converter, comparing the output current with the preset current, and controlling the on-off state of the third switch module according to the comparison result and the state of the first control signal and controlling the on-off state of the fourth switch module according to the comparison result and the state of the second control signal.

[0043] Optionally, the obtaining of the output current of the resonant LLC switching power supply converter includes:

[0044] the output current of the resonant LLC switching power supply converter is obtained by detecting the effective value of the alternating component of the voltage across the second capacitor module.

[0045] Optionally, the controlling of the on-off state of the third switch module according to the current comparison result and the state of the first control signal includes:

[0046] when the output current is greater than the preset current and the first control signal is switched from a first level signal to a second level signal, the third switch module is controlled to be switched from a first switch state to a second switch state with a delay and maintained in the second switch state;

[0047] when the output current is not greater than the preset current and the first control signal is switched from a first level signal to a second level signal, the third switch module is controlled to be switched from a second switch state to a first switch state with a delay and maintained in the first switch state.

[0048] Optionally, the controlling the on-off state of the fourth switch module according to the current comparison result and the state of the second control signal comprises:

[0049] When the output current is greater than the preset current, and the second control signal is switched from the first level signal to the second level signal, the fourth switch module is controlled to switch from the first switch state to the second switch state with a delay, and the second switch state is maintained;

[0050] When the output current is not greater than the preset current, and the second control signal is switched from the first level signal to the second level signal, the fourth switch module is controlled to switch from the second switch state to the first switch state with a delay, and the first switch state is maintained.

[0051] The technical effect of the embodiment of the present application is that by monitoring the output current in real time and comparing it with the preset current, dynamic adjustment of the resonant LLC switching power supply converter is realized. When the load state changes, the control module can intelligently adjust the on-off state of the third switch module and the fourth switch module, ensuring zero-voltage turn-off or turn-on under large current load, and turning off the capacitor module under small current load to reduce power loss. This design effectively avoids the problems of efficiency reduction due to increased hard turn-off loss of the first switch module and the second switch module under large load current in the prior art, and the problem of increased hard turn-on loss due to the output driving waveform of the control circuit being an interval discontinuous pulse group and a skip cycle state under standby and small load conditions. The circuit of the present application greatly reduces the hard turn-on and hard turn-off loss of the first switch module and the second switch module, can maintain low standby power consumption and high efficiency operation, and significantly improves the overall efficiency and adaptability of the power supply converter. BRIEF DESCRIPTION OF DRAWINGS

[0052] In order to more clearly illustrate the technical solutions of the embodiments of the present application, the following will briefly introduce the drawings needed to be used in the description of the embodiments of the present application. Obviously, the drawings in the following description are only some embodiments of the present application, and other drawings can also be obtained by those skilled in the art without creative labor.

[0053] Figure 1 is a structure schematic diagram of a resonant LLC switching power supply converter provided by the prior art;

[0054] Figure 2 is a first structure schematic diagram of a resonant LLC switching power supply converter provided by the embodiment one of the present application;

[0055] Figure 3 is a second structure schematic diagram of a resonant LLC switching power supply converter provided by the embodiment one of the present application;

[0056] Figure 4 is a first structure diagram of a second control module in a resonant LLC switching power supply converter provided by embodiment one of the present application;

[0057] Figure 5 is a structure diagram of a first control submodule in the second control module in the resonant LLC switching power supply converter provided by embodiment one of the present application;

[0058] Figure 6 is a structure diagram of a first delay trigger module in the first control submodule in the second control module in the resonant LLC switching power supply converter provided by embodiment one of the present application;

[0059] Figure 7 is a structure diagram of a second control submodule in the second control module in the resonant LLC switching power supply converter provided by embodiment one of the present application;

[0060] Figure 8 is a structure diagram of a second delay trigger module in the second control submodule in the second control module in the resonant LLC switching power supply converter provided by embodiment one of the present application;

[0061] Figure 9 is a circuit diagram of the resonant LLC switching power supply converter provided by embodiment one of the present application;

[0062] Figure 10 is a transistor drive voltage waveform timing diagram output by the first control module and the second control module of the resonant LLC switching power supply converter provided by embodiment one of the present application;

[0063] Figure 11 is a second structure diagram of the second control module in the resonant LLC switching power supply converter provided by embodiment one of the present application;

[0064] Figure 12 is a circuit diagram of the second control module in the resonant LLC switching power supply converter provided by embodiment one of the present application;

[0065] Figure 13 is a same time axis comparison diagram of a drive voltage waveform and a main element current waveform of the main switching transistors Q1 and Q2 in the resonant LLC switching power supply converter provided by embodiment one of the present application;

[0066] Figure 14 is a same time axis comparison diagram of a drive voltage waveform and a main element voltage waveform of the main switching transistors Q1 and Q2 in the resonant LLC switching power supply converter provided by embodiment one of the present application;

[0067] Figure 15 is Figure 13 , Figure 14 a detailed expansion diagram of the same time axis comparison chart of the voltage current waveforms of the four working modes identified in

[0068] Figure 16 is a flow chart of a control method of a resonant LLC switching power supply converter provided by the second embodiment of the present application;

[0069] Figure 17 is a specific flow chart of step S20 in the control method of a resonant LLC switching power supply converter provided by the second embodiment of the present application;

[0070] Figure 18 is another specific flow chart of step S20 in the control method of a resonant LLC switching power supply converter provided by the second embodiment of the present application;

[0071] In the figure: 101, first switch module; 102, second switch module; 103, third switch module; 104, fourth switch module; 105, first capacitor module; 106, second capacitor module; 107, third capacitor module; 108, energy storage and isolation module; 110, control module; 111, first control module; 112, second control module; 201, current detection module; 202, reference voltage module; 203, first delay trigger module; 204, first drive module; 205, second delay trigger module; 206, second drive module; 211, first control submodule; 212, second control submodule; 241, first comparison module; 242, first trigger module; 243, second comparison module; 244, second trigger module. DETAILED DESCRIPTION

[0072] The technical solutions in the embodiments of the present application will be described clearly and completely below with reference to the drawings in the embodiments of the present application. Obviously, the described embodiments are only part of the embodiments of the present application, rather than all the embodiments of the present application. Based on the embodiments in the present application, all other embodiments obtained by those of ordinary skill in the art without creative work fall within the protection scope of the present application.

[0073] It should be understood that the present application can be implemented in different forms and should not be interpreted as being limited to the embodiments presented herein. On the contrary, these embodiments are provided to make the disclosure complete and full, and to fully convey the scope of the present application to those skilled in the art. In the drawings, the sizes and relative sizes of layers and regions may be exaggerated for clarity throughout the same reference numerals represent the same elements.

[0074] It will be understood that when an element or layer is referred to as being "on" or "connected to" or "coupled to" another element or layer, it can be directly on or connected or coupled to the other element or layer or intervening elements or layers can be present. In contrast, when an element is referred to as being "directly on" or "directly connected to" or "directly coupled to" another element or layer, there are no intervening elements or layers present. It will be understood that, although the terms first, second, third, etc. can be used herein to describe various elements, components, regions, layers and / or sections, these elements, components, regions, layers and / or sections should not be limited by these terms. These terms are only used to distinguish one element, component, region, layer or section from another element, component, region, layer or section. Thus, a first element, component, region, layer or section discussed below could be termed a second element, component, region, layer or section without departing from the teachings of the present application.

[0075] For a thorough understanding of the present application, reference is made to the following detailed description taken in conjunction with the accompanying drawings, in which:

[0076] Embodiment One

[0077] Embodiment One provides a resonant LLC switching power converter, as shown in FIG. 1, comprising: Figure 2

[0078] The resonant LLC switching power converter comprises a first switching module 101, a second switching module 102, a third switching module 103, a fourth switching module 104, a first capacitor module 105, a second capacitor module 106, a third capacitor module 107, an energy storage and isolation module 108, and a control module 110.

[0079] ​The first end of the first switch module 101 is connected to the first end of the second capacitor module 106, the second end of the first switch module 101 is respectively connected to the first end of the second switch module 102, the second end of the third switch module 103, the first end of the third capacitor module 107 and the first end of the energy storage and isolation module 108, the second end of the second capacitor module 106 is connected to the first end of the third switch module 103, the second end of the second switch module 102 is respectively connected to the second end of the fourth switch module 104 and the first end of the first capacitor module 105, the second end of the first capacitor module 105 is connected to the second end of the energy storage and isolation module 108, the second end of the third capacitor module 107 is connected to the first end of the fourth switch module 104, and the control module 110 is respectively connected to the control end and the second end of the first switch module 101, the control end and the second end of the second switch module 102, the control end and the second end of the third switch module 103 and the control end and the second end of the fourth switch module 104.

[0080] The control module 110 is configured to output a first control signal to the first switch module 101 to control the on-off state of the first switch module 101, and output a second control signal to the second switch module 102 to control the on-off state of the second switch module 102.

[0081] The control module 110 is further configured to obtain an output current of the resonant LLC switching power supply converter, compare the output current with a preset current, and control the on-off state of the third switch module 103 according to a comparison result and a state of the first control signal, and control the on-off state of the fourth switch module 104 according to the comparison result and a state of the second control signal.

[0082] The first switch module 101 receives the first control signal output by the control module 110 for control. When receiving the start control signal, the first switch module 101 is turned on to make the current flow to the energy storage and isolation module 108. When receiving the off control signal, the first switch module 101 is turned off, and the current path is interrupted. The second switch module 102 works with the first switch module 101 to provide another phase of switching signal to form a complete bridge topology. The second switch module 102 receives the second control signal output by the control module 110 for control. When the first switch module 101 is off, the second switch module 102 is on to ensure the continuity of the current. The third switch module 103 and the fourth switch module 104 are dynamically adjusted according to the output current state to realize zero-voltage soft turn-on and reduce loss and interference. The turn-on and turn-off of the third switch module 103 and the fourth switch module 104 are strictly controlled by the load current and the main switch state to ensure no interference to the main circuit and maintain high efficiency operation. The first capacitor module 105 forms an oscillation circuit with the energy storage and isolation module 108 when the circuit is turned on. The second capacitor module 106 is part of the resonant circuit and provides the resonant capacitance required by the circuit during the switching transition of the switch module. Together with the energy storage and isolation module 108 and the switch module, it forms a resonant circuit to generate a resonant signal, store energy and release energy during the resonant process, and reduce the turn-off loss of the first switch module 101. The third capacitor module 107 is part of the resonant circuit and provides the resonant capacitance required by the circuit during the switching transition of the first switch module 101. Together with the energy storage and isolation module 108 and the first switch module 101, it forms a resonant circuit to generate a resonant signal, store energy and release energy during the resonant process, and reduce the turn-off loss of the second switch module 102. The energy storage and isolation module 108 includes a transformer or an inductor element for energy storage, energy transfer and input / output isolation. When the switch module is turned on, the energy storage module stores energy, realizes electrical isolation between input and output through the transformer, and releases the stored energy to the output at the resonant frequency. The control module 110 is used to generate switching control signals and adjust the switching state according to the feedback signal of the output current. The control module 110 acquires the output current of the resonant LLC power supply converter, compares the output current with the preset current, and adjusts the state of the third switch module 103 and the fourth switch module 104 according to the result. High-frequency control signals are output to the first switch module 101 and the second switch module 102 to ensure that the entire circuit works in a reliable frequency range. The control module 110 monitors the output current of the resonant LLC switching power supply converter in real time. The output current reflects the load state and working condition of the power supply and is a key parameter for adjusting power conversion efficiency and control strategy. The control module 110 compares the real-time collected output current with the preset reference current. The preset current is the ideal current value set according to the design requirements or load characteristics, which is used to judge whether the system is in a large current load state or a small current load state.According to the comparison result of the output current and the preset current, the control module 110 determines whether the state of the third switch module 103 needs to be adjusted. If the output current exceeds the preset value, it is determined that the large current load state is in, and the control module 110 adjusts the on-off of the third switch module 103 based on the state (such as level change) of the first control signal to ensure that the third switch module 103 performs zero voltage turn-off or turn-on. If the output current does not exceed the preset value, it is determined that the small current load state is in, and the control module 110 adjusts the on-off of the third switch module 103 based on the state (such as level change) of the first control signal to disconnect the second capacitor module 106 from the circuit. Similarly, the control module 110 also controls the on-off state of the fourth switch module 104 based on the same output current and preset current comparison result combined with the state of the second control signal. If the output current exceeds the preset value, it is determined that the large current load state is in, and the control module 110 adjusts the on-off of the fourth switch module 104 based on the state (such as level change) of the second control signal to ensure that the fourth switch module 104 performs zero voltage turn-off or turn-on. If the output current does not exceed the preset value, it is determined that the small current load state is in, and the control module 110 adjusts the on-off of the fourth switch module 104 based on the state (such as level change) of the second control signal to disconnect the third capacitor module 107 from the circuit.

[0083] The technical effect of the technical solution provided by the embodiment one is that by monitoring the output current in real time and comparing it with the preset current, dynamic adjustment of the resonant LLC switching power supply converter is realized. When the load state changes, the control module can intelligently adjust the on-off state of the third switch module and the fourth switch module, ensuring that zero voltage turn-off or turn-on is realized in large current load, and unnecessary capacitor modules are disconnected in small current load, reducing power loss. This design effectively avoids the problem of efficiency reduction caused by the increase of hard turn-off loss of the first switch module and the second switch module in large load current in the prior art, and the problem of hard turn-on loss increase caused by the control circuit output driving waveform being an interval discontinuous pulse group and a skip period state in standby and small load conditions. The circuit of the present application greatly reduces the hard turn-on and hard turn-off loss of the first switch module and the second switch module, can maintain low standby power consumption and high efficiency operation, and significantly improves the overall efficiency and adaptability of the power supply converter.

[0084] As an implementation manner, as shown in Figure 3 The control module 110 includes:

[0085] The first control module 111 is connected to the control end of the first switch module 101 and the control end of the second switch module 102 respectively, and is used to output a first control signal to the first switch module 101 to control the on-off state of the first switch module 101, and output a second control signal to the second switch module 102 to control the on-off state of the second switch module 102.

[0086] The second control module 112 is connected to the control end of the first switch module 101, the control end of the second switch module 102, the control end of the third switch module 103 and the control end of the fourth switch module 104 respectively, and is used to obtain the output current of the resonant LLC switching power supply converter, compare the output current with a preset current, and control the on-off state of the third switch module 103 according to the comparison result and the state of the first control signal, and control the on-off state of the fourth switch module 104 according to the comparison result and the state of the second control signal.

[0087] The first control module 111 controls the on-off state of the main switch module (the first switch module 101 and the second switch module 102). The first control signal and the second control signal are generated to drive the first switch module 101 and the second switch module 102 to work alternately. The first control module 111 generates the first control signal and the second control signal according to the system running state, outputs the first control signal to the first switch module 101 to control the on-off state thereof, and outputs the second control signal to the second switch module 102 to control the on-off state thereof. The first control signal and the second control signal are alternately turned on to ensure that the first switch module 101 and the second switch module 102 work alternately. The second control module 112 realizes dynamic control of the auxiliary switch module (the third switch module 103 and the fourth switch module 104). Based on the real-time collected output current signal, in combination with the state of the first control signal and the second control signal, the on-off state of the third switch module 103 and the fourth switch module 104 is adjusted to realize zero-voltage turn-on of the first switch module 101 and the second switch module 102.

[0088] The second control module 112 acquires the output current of the resonant LLC converter in real time, compares it with the preset current, and generates a comparison result. According to the comparison result of the output current and the preset current, in combination with the state of the first control signal, the on-off state of the third switch module 103 is controlled, and according to the comparison result of the output current and the preset current, in combination with the state of the second control signal, the on-off state of the fourth switch module 104 is controlled. Specifically, when the output current is greater than the preset current, and the first control signal is switched from the first level signal to the second level signal, the third switch module 103 is controlled to switch from the first switch state to the second switch state with a delay and maintain the second switch state. When the output current is greater than the preset current, and the second control signal is switched from the first level signal to the second level signal, the fourth switch module 104 is controlled to switch from the first switch state to the second switch state with a delay and maintain the second switch state. The second switch state is the on state, realizing zero-voltage turn-on or turn-off of the first switch module 101 and the second switch module 102 in the case of large current. When the output current is not greater than the preset current, and the first control signal is switched from the first level signal to the second level signal, the third switch module 103 is controlled to switch from the second switch state to the first switch state with a delay and maintain the first switch state. When the output current is not greater than the preset current, and the second control signal is switched from the first level signal to the second level signal, the fourth switch module 104 is controlled to switch from the second switch state to the first switch state with a delay and maintain the first switch state. The first switch state is the off state, realizing the disconnection of the third switch module 103 and the fourth switch module 104 in the case of small current, so that the second capacitor module 106 and the third capacitor module 107 are in the off state, thereby avoiding the increase of the hard turn-on power loss of the LLC switch power converter during the interval intermittent pulse group and the skip cycle state. w The problem of increasing the hard turn-on loss power of the LLC switch power converter during the interval intermittent pulse group and the skip cycle state.

[0089] The technical effect of the embodiment is that through the cooperative control of the first control module and the second control module, the resonant LLC switch power converter realizes zero-voltage turn-on and turn-off under large current load and high-efficiency operation under small current load. The first control module ensures the alternating turn-on of the main switch module, realizing stable resonant operation; the second control module dynamically adjusts the state of the auxiliary switch module through real-time current feedback, optimizing energy conversion under large current and reducing switching loss, avoiding the additional increase of hard turn-on power loss caused by frequency discontinuity under small current, and effectively improving the efficiency, stability and adaptability of the converter as a whole.

[0090] As an embodiment, as shown in Figure 4 The second control module 112 includes:

[0091] The reference voltage module 202 is configured to output a reference voltage.

[0092] a current detection module 201, a first input end of which is connected to a second end of the first capacitor module 105, a second input end of which is connected to an output end of the reference voltage module 202, for collecting an output current of the resonant LLC switching power supply converter, converting the output current into a collection voltage, and comparing the collection voltage with the reference voltage to output a first comparison signal;

[0093] a first control submodule 211, a first input end of which receives the first control signal, a second input end of which is connected to the output end of the reference voltage module 202, a third input end of which is connected to the output end of the current detection module 201, and an output end of which is connected to the control end of the third switch module 103, for outputting a first on-off control signal to the third switch module 103 according to the first control signal, the reference voltage and the first comparison signal;

[0094] a second control submodule 212, a first input end of which receives the second control signal, a second input end of which is connected to the output end of the reference voltage module 202, a third input end of which is connected to the output end of the current detection module 201, and an output end of which is connected to the control end of the fourth switch module 104, for outputting a second on-off control signal to the fourth switch module 104 according to the second control signal, the reference voltage and the first comparison signal.

[0095] The reference voltage module 202 outputs a stable reference voltage as a reference value for the current detection module 201, the first control submodule 211 and the second control submodule 212, which is used to determine the judgment standard of current control and comparison in the resonant LLC switching power supply converter. The reference voltage module 202 generates and outputs a fixed reference voltage, which is provided to the current detection module 201 for comparison with the collected voltage, and is also provided to the first control submodule 211 and the second control submodule 212 to participate in the generation of the on-off control signal. The current detection module 201 collects the output current signal of the resonant LLC switching power supply converter, converts the collected output current into a corresponding collected voltage, compares the collected voltage with the reference voltage, and generates a first comparison signal as the basis for subsequent control. The current detection module 201 receives the output current signal from the second end of the first capacitor module 105, converts the output current signal into a collected voltage through an internal conversion circuit, and compares the collected voltage with the reference voltage in real time. When the collected voltage exceeds the reference voltage, a first comparison signal is output to indicate whether the current state meets the trigger condition. The first control submodule 211 generates a first on-off control signal according to the first control signal, the reference voltage and the first comparison signal, controls the on-off operation of the third switch module 103, and participates in the dynamic adjustment of the resonant circuit at the switching transition moment of the first switch module 101. The first control submodule 211 receives the first control signal (used to indicate the starting point of the on-off logic), the reference voltage (as a reference value) and the first comparison signal (indicating the current state), judges whether the first control signal meets the trigger condition, and performs logical processing on the reference voltage and the first comparison signal. According to the comprehensive judgment, a first on-off control signal is generated. The signal is output to the control end of the third switch module 103 to trigger the on-off action of the third switch module 103, so that the switching loss of the main switch module 101 in different working states is greatly reduced. The second control submodule 212 generates a second on-off control signal according to the second control signal, the reference voltage and the first comparison signal. The second control submodule 212 controls the on-off operation of the fourth switch module 104 and participates in the dynamic adjustment of the resonant circuit at the switching transition moment of the second switch module 102. The second control submodule 212 receives the second control signal, the reference voltage and the first comparison signal, judges whether the second control signal meets the trigger condition, and performs logical processing on the reference voltage and the first comparison signal. According to the comprehensive judgment, a second on-off control signal is generated. The signal is output to the control end of the fourth switch module 104 to trigger the on-off action of the fourth switch module 104, so that the switching loss of the main switch module 101 in different working states is greatly reduced.

[0096] The technical effect of the embodiment is that through the cooperation of the reference voltage module, the current detection module, the first control submodule and the second control submodule, the switching loss dynamic control of the main switch module of the LLC switching power supply converter is realized and greatly reduced. The reference voltage module provides a stable reference voltage, the current detection module acquires current in real time and generates a comparison signal to provide a basis for subsequent control. The first control submodule and the second control submodule generate accurate on-off control signals according to the input control signal, the reference voltage and the current comparison signal, and drive the actions of the third switch module and the fourth switch module respectively, so as to dynamically adjust and reduce the switching loss of the main switch module. The design effectively ensures the high-efficiency operation of the system under different load conditions, and effectively reduces the circuit interference.

[0097] As an embodiment, as shown in Figure 5 The first control submodule 211 includes:

[0098] The first delay trigger module 203 receives the first control signal at the first input end, the reference voltage at the second input end and the first comparison signal at the third input end, and outputs the first weak current on-off control signal according to the first control signal, the reference voltage and the first comparison signal.

[0099] The first drive module 204 is connected to the output end of the first delay trigger module 203, amplifies and isolates the first weak current on-off control signal, and then outputs the first on-off control signal.

[0100] The first delay trigger module 203 delays the first control signal, combines the reference voltage and the first comparison signal, and outputs a first weak current on-off control signal for realizing accurate on-off control of the output weak current. The first delay trigger module 203 receives three input signals, the first control signal as the core control basis for indicating the starting point of the trigger logic, the reference voltage for comparing with the internally generated signal to determine the trigger condition. The first comparison signal is used to assist in determining whether the trigger condition is met, and the first control signal is delayed to generate a first acquisition voltage, which is used to monitor the size relationship between the first acquisition voltage and the reference voltage in real time. When the first acquisition voltage exceeds the reference voltage and the first comparison signal meets the trigger condition, the first weak current on-off control signal is outputted for controlling the operation of the subsequent circuit module. If the condition is not met, no signal is outputted to ensure the safe operation of the circuit. The first drive module 204 receives the first weak current on-off control signal outputted by the first delay trigger module 203, amplifies and electrically isolates the first weak current on-off control signal, and then outputs a first on-off control signal for driving the subsequent circuit. The electrical isolation circuit is used to realize the electrical isolation between the control signal and the power circuit, avoid the interference of high voltage or large current on the control loop, and output the first on-off control signal to directly drive the on-off operation of the power device and complete the control target.

[0101] The technical effect of the embodiment is that the accurate on-off control and safe driving of the third switch module are realized through the cooperative work of the first delay trigger module and the first drive module. The first delay trigger module is based on the delay processing and the multi-signal judgment mechanism, and only outputs a reliable weak current on-off control signal when the trigger condition is met, thereby ensuring the accuracy and safety of the control. The first drive module amplifies and electrically isolates the weak current signal, converts it into a high current or high voltage signal that can drive the power device, and avoids the interference of high voltage or large current on the control circuit. The design effectively improves the driving ability, anti-interference ability and stability of the system operation of the control signal, and realizes efficient and safe circuit operation.

[0102] As an embodiment, as shown in Figure 6 The first delay trigger module 203 includes:

[0103] The first comparison module 241 receives the first control signal at the first input end and the reference voltage at the second input end, and is used to convert the first control signal into a first acquisition voltage through delay, and output a first trigger signal when the first acquisition voltage is greater than the reference voltage;

[0104] The first trigger module 242 receives the first comparison signal at the first input end and is connected to the output end of the first comparison module 241 at the second input end, and is used to output a first weak current on-off control signal according to the first trigger signal.

[0105] The first comparison module 241 delays the first control signal to generate a first acquisition voltage, and monitors the size relationship between the first acquisition voltage and the reference voltage in real time. When the first acquisition voltage exceeds the reference voltage, the first comparison module 241 outputs a first trigger signal. The first trigger module 242 detects the state of the first trigger signal, and activates the trigger logic when receiving a valid first trigger signal. According to the trigger logic, a first weak current on-off control signal is outputted to instruct the subsequent circuit module to perform weak current on-off control. When the trigger signal is invalid or does not meet the condition, no control signal is outputted, ensuring that the circuit is in a safe state.

[0106] The technical effect of the embodiment is that the first comparison module and the first trigger module work together to realize the delay processing and accurate trigger control of the first control signal. The first comparison module converts the first control signal into a first acquisition voltage and monitors its size relationship with the reference voltage in real time, and outputs the first trigger signal only when the trigger condition is met. The first trigger module generates a first weak current on-off control signal according to the trigger signal state, ensuring that the subsequent circuit module performs weak current on-off operation at the right time. This design has the technical effects of accurate delay, rapid response and high safety, effectively avoids false triggering, and ensures the stability and reliability of the circuit.

[0107] As an embodiment, as shown in Figure 7 The second control submodule 212 includes:

[0108] The second delay trigger module 205 receives the second control signal at the first input end, receives the reference voltage at the second input end, and receives the first comparison signal at the third input end, and is configured to output a second weak current on-off control signal according to the second control signal, the reference voltage and the first comparison signal;

[0109] The second drive module 206 is connected to the output end of the second delay trigger module, and is configured to amplify and isolate the output second weak current on-off control signal to output a second on-off control signal.

[0110] The main function of the second delay trigger module 205 is to output a second weak current on-off control signal at the appropriate time according to the input second control signal, reference voltage and first comparison signal. The second control signal is used to indicate the trigger logic, usually triggered when the level state changes. The reference voltage is used as a reference voltage for comparison with the internally generated voltage to ensure that the circuit works within a predetermined voltage range. The first comparison signal is used to assist in determining whether the trigger condition is met, and the second control signal is processed by delay, to generate a second acquisition voltage, which monitors the size relationship between the second acquisition voltage and the reference voltage in real time. When the second acquisition voltage exceeds the reference voltage and the first comparison signal meets the trigger condition, the second weak current on-off control signal is output to control the operation of the subsequent circuit module. The main function of the second drive module 206 is to receive the second weak current on-off control signal output by the second delay trigger module 205, and amplify and electrically isolate the signal to output a second on-off control signal for driving the subsequent circuit. The second drive module 206 outputs the processed second on-off control signal to control the on-off operation of the fourth switch module 104.

[0111] The technical effect of the embodiment is that through the cooperative work of the second delay trigger module and the second drive module, accurate on-off control and safe driving of the fourth switch module are realized. The second delay trigger module outputs a reliable weak current on-off control signal only when the trigger condition is met based on the delay processing and multi-signal judgment mechanism, ensuring the accuracy and safety of the control. The second drive module amplifies and electrically isolates the weak current signal, and converts it into a high current or high voltage signal that can drive the power device, while avoiding the interference of high voltage or large current on the control circuit. This design effectively improves the driving ability, anti-interference and stability of the system operation of the control signal, and realizes efficient and safe circuit operation.

[0112] As an embodiment, as shown in Figure 8 The second delay trigger module 205 includes:

[0113] The second comparison module 243 receives the second control signal at the first input end and the reference voltage at the second input end, and is used to convert the second control signal into a second acquisition voltage by delay, and output a second trigger signal when the second acquisition voltage is greater than the reference voltage;

[0114] The second trigger module 244 receives the first comparison signal at the first input end and is connected to the output end of the second comparison module at the second input end, and is used to output a second weak current on-off control signal according to the second trigger signal.

[0115] The main function of the second comparison module 243 is to compare the input second control signal with the reference voltage and generate the second collection voltage according to the comparison result, which is used to trigger the subsequent control signal output. The second control signal is the input signal of the second switch module 102, which is used to indicate whether the delay trigger is needed, usually triggered when the level state changes. The reference voltage provides a reference voltage for comparison with the voltage converted from the second control signal. The second comparison module 243 converts the received second control signal into the second collection voltage through delay processing. The delay processing is used to ensure the timing accuracy of the system and avoid unstable control caused by too fast signal change. The converted second collection voltage is compared with the reference voltage. If the second collection voltage is greater than the reference voltage, it is considered that the system meets the trigger condition, and the module will output the second trigger signal. If the condition is not met, the subsequent operation will not be triggered. The main function of the second trigger module 244 is to further output the second weak current on-off control signal for controlling the subsequent circuit according to the second trigger signal and the first comparison signal output by the second comparison module 243, which is used to control the on-off state of the fourth switch module 104 in the subsequent circuit.

[0116] The technical effect of the embodiment is that through the cooperative work of the second comparison module and the second trigger module, the delay processing and accurate trigger control of the second control signal are realized. The second comparison module converts the second control signal into the second collection voltage and monitors its size relationship with the reference voltage in real time, and outputs the second trigger signal only when the trigger condition is met; the second trigger module generates the second weak current on-off control signal according to the trigger signal state, ensuring that the subsequent circuit module performs the weak current on-off operation at the right time. The design has the technical effects of delay accuracy, rapid response and high safety, effectively avoids false triggering, and ensures the stability and reliability of the circuit.

[0117] As an embodiment, the first switch module 101 is a first MOS tube, the second switch module 102 is a second MOS tube, the first control signal includes a first voltage control signal and a second voltage control signal, and the second control signal includes a third voltage control signal and a fourth voltage control signal.

[0118] The resonant LLC switching power supply converter further comprises:

[0119] The first voltage control signal is received at the first end of the first voltage dividing unit, the second voltage control signal is received at the third end of the first voltage dividing unit, and the gate of the first MOS tube is connected to the second end of the first voltage dividing unit. The source of the first MOS tube is connected to the third end of the first voltage dividing unit, and the on-off of the first MOS tube is controlled according to the first voltage control signal and the second voltage control signal.

[0120] The second voltage dividing unit has a first end receiving a third voltage control signal, a second end connected to the gate of the second MOS tube, and a third end receiving a fourth voltage control signal after being connected to the source of the second MOS tube, and controls the on-off of the second MOS tube according to the third voltage control signal and the fourth voltage control signal.

[0121] The second control module 112 is configured to control the third switch module to start conducting when it is detected that the output current is greater than the preset current, and a high-level signal is formed between the first voltage control signal and the second voltage control signal or between the second voltage control signal and the ground end of the second control module.

[0122] The second control module 112 is further configured to control the fourth switch module to start conducting when the output current is greater than the preset current, and a high-level signal is formed between the third voltage control signal and the fourth voltage control signal.

[0123] The first voltage control signal and the second voltage control signal jointly act on the first MOS tube through the voltage division principle and the current discharge of the voltage division resistor to adjust the gate voltage of the first MOS tube. When a high-level signal is formed between the first voltage control signal and the second voltage control signal, the control signal triggers the first MOS tube to turn on; when the control signal becomes low, the first MOS tube is turned off; when the control circuit is not powered on, the voltage division resistor reliably discharges the gate capacitance voltage of the MOS tube to 0V. The second voltage control signal and the fourth voltage control signal jointly act on the second MOS tube through the voltage division principle and the current discharge of the voltage division resistor to adjust the gate voltage of the second MOS tube. When a high-level signal is formed between the third voltage control signal and the fourth voltage control signal, the control signal triggers the second MOS tube to turn on; when the control signal becomes low, the second MOS tube is turned off; when the control circuit is not powered on, the voltage division resistor reliably discharges the gate capacitance voltage of the MOS tube to 0V. According to the above signals, the second voltage division unit accurately controls the switching state (turning on or turning off) of the second MOS tube, and ensures that the MOS tube is in a reliable off state when not powered on. The second control module 112 controls the conduction state of the third switch module and the fourth switch module according to the level state of each control signal when the output current is greater than the preset current. The second control module 112 first monitors the output current and compares it with the preset reference current. When the output current is greater than the preset current, it is determined that the load is in a large current state, and subsequent dynamic control is prepared. When a high-level signal is formed between the first voltage control signal and the second voltage control signal, or a high-level signal is formed between the second voltage control signal and the ground, the second control module 112 starts the conduction of the third switch module. This control ensures that the third switch module 103 can work effectively under the condition of large load current, supporting the zero-voltage conduction or turn-off of the first switch module 101. When a high-level signal is formed between the third voltage control signal and the fourth voltage control signal, the second control module 112 starts the conduction of the fourth switch module 104. This operation ensures that the fourth switch module 104 can enter the conduction state under the preset condition, supporting the zero-voltage conduction or turn-off of the second switch module 102.

[0124] The technical effect of the embodiment is that by precisely controlling the gate voltages of the first MOS tube and the second MOS tube, efficient regulation of the power converter is realized, and stability and reliability are improved. The first voltage dividing unit and the second voltage dividing unit control the switching states of the first MOS tube and the second MOS tube respectively, ensuring accurate control of current flow. When the load current is large, the second control module can adjust the conduction states of the third switch module and the fourth switch module in time according to the feedback signal of the current, realizing zero-voltage conduction and disconnection of the first switch module and the second switch module, reducing the loss in the switching process, and optimizing the power conversion efficiency.

[0125] The embodiment of the application will be described in detail below with reference to a specific circuit structure:

[0126] The first switch module 101 is a main switch transistor Q1, the second switch module 102 is a main switch transistor Q2, the third switch module 103 is an auxiliary switch transistor Q3, the fourth switch module 104 is an auxiliary switch transistor Q4, the first capacitor module 105 is a resonance capacitor C2, the second capacitor module 106 is an auxiliary capacitor C4, the third capacitor module 107 is an auxiliary capacitor C5, the energy storage isolation module includes an inductor L1 and a transformer T1, and the resonant LLC switching power converter further includes resistors R1, R2, R3, R4, R5, R6, R7, R8, capacitors C1, C3, diodes D1 and D2.

[0127] One end of the auxiliary capacitor C4 is connected with the drain D of the auxiliary switch transistor Q3, the source S of the auxiliary switch transistor Q3, the source S of the main switch transistor Q1, the drain D of the main switch transistor Q2, the positive pole of the auxiliary capacitor C5, one end of the resistor R2, one end of the resistor R6, one end of the inductor L1, the second output end (output driving signal VS-Q1) of the first control module 111, the second input end (driving signal VS-Q1 voltage sampling end) of the second control module 112, the negative pole of the second auxiliary DC power supply VCC2-GND, the second output end (output driving signal VS-Q3) of the second control module 112, the seventh input end (second path power supply negative pole end) of the second control module 112, and twelve points are commonly connected. The negative pole of the auxiliary capacitor C5 is connected with the drain D of the auxiliary switch transistor Q4, the other end of the resistor R2 is connected with the gate G of the switch transistor Q1 and one end of the resistor R1, the other end of the resistor R6 is connected with the gate G of the auxiliary switch transistor Q3 and one end of the resistor R5, the other end of the resistor R1 is connected with the first output end (output driving signal VG-Q1) of the first control module 111, the other end of the resistor R5 is connected with the first output end (output driving signal VG-Q3) of the second control module 112; the other end of the inductor L1 is connected with one end of the primary coil of the transformer T1, the other end of the primary coil of the transformer T1, one end of the resonance capacitor C2, the second input end (I-S current sampling end) of the first control module 111, the first input end (I-S current sampling end) of the second control module 112, and four points are commonly connected; the other end of the resonance capacitor C2, the source S of the main switch transistor Q2, the source S of the auxiliary switch transistor Q4, one end of the resistor R4, one end of the resistor R8, the negative pole of the input capacitor C1, the negative pole of the input DC voltage VI-GND, the negative pole of the first auxiliary DC power supply VCC1-GND, the fourth output end (output driving signal VS-Q2) of the first control module 111, the fourth output end (output driving signal VS-Q4) of the second control module 112, the fifth input end (first path power supply negative pole end) of the first control module 111, the fourth input end (first path power supply negative pole end) of the first control module 111, the fifth input end (first path power supply negative pole end) of the second control module 112, and thirteen points are commonly connected; the other end of the resistor R4 is connected with the gate G of the main switch transistor Q2 and one end of the resistor R3, the other end of the resistor R8 is connected with the gate G of the auxiliary switch transistor Q4 and one end of the resistor R7; the other end of the resistor R3 is connected with the third output end VG-Q2 of the first control module 111 and the third input end (VG-Q2 voltage sampling end) of the second control module 112, and the other end of the resistor R7 is connected with the third output end (output driving signal VG-Q4) of the second control module 112. The positive pole of the first auxiliary DC power supply VCC1, the third input end (the positive pole of the first auxiliary DC power supply VCC1) of the first control module 111, and the fourth input end (the positive pole of the first path power supply) of the second control module 112 are commonly connected.The second auxiliary DC power supply positive pole VCC2 is commonly connected with the sixth input end (second path power supply positive pole end) of the second control module 112. The first end of the transformer T1 secondary coil is connected with the positive pole of the diode D1, the negative pole of the diode D1, the negative pole of the diode D2, the positive pole of the output capacitor C3, the positive pole of the switch power supply converter output end VO, the fifth input end (output voltage sampling end positive pole) of the first control module 111, and the five points are commonly connected; the second end of the transformer T1 secondary coil is connected with the positive pole of the diode D2, the negative pole of the output capacitor C3, the third end (coil center tap) of the transformer T1 secondary coil, the negative pole of the switch power supply converter output end VO-GND, the sixth input end (output voltage sampling end negative pole) of the first control module 111, and the four points are commonly connected.

[0128] wherein, Figure 9 and Figure 10The signal meanings in the formula are as follows: U(GS-Q*) is the voltage difference waveform between the gate G and the source S of the switch transistor Q*; * represents a digital label; U(VS-Q1) is the voltage difference waveform between the node A (the signal received by the node is the driving signal VS-Q1) of the source S of the switch transistor Q1 and VCC1-GND (0 point reference point); VG-Q1 is the first positive driving signal output by the first control module 111, which is connected to the gate G of the main switch transistor Q1 through a resistor R1; VS-Q1 is the first negative driving signal output by the first control module 111, which is connected to the source S of the main switch transistor Q1 and has the same potential as the negative electrode VCC2-GND of the second auxiliary power supply; VG-Q2 is the second positive driving signal output by the first control module 111, which is connected to the gate G of the main switch transistor Q2 through a resistor R3; VS-Q2 is the second negative driving signal output by the first control module 111, which is connected to the source S of the main switch transistor Q2 and has the same potential as the negative electrode VCC1-GND of the first auxiliary power supply; VG-Q3 is the third positive driving signal output by the second control module 112, which is connected to the gate G of the auxiliary switch transistor Q3 through a resistor R5; VS-Q3 is the third negative driving signal output by the second control module 112, which is connected to the source S of the auxiliary switch transistor Q3 and has the same potential as the first negative driving signal VS-Q1 and the negative electrode VCC2-GND of the second auxiliary power supply; VG-Q4 is the fourth positive driving signal output by the second control module 112, which is connected to the gate G of the auxiliary switch transistor Q4 through a resistor R7; VS-Q4 is the fourth negative driving signal output by the second control module 112, which is connected to the source S of the auxiliary switch transistor Q4 and has the same potential as the negative electrode VCC1-GND of the first auxiliary power supply; I(VO) is the current value flowing out of the output terminal VO of the switching power supply converter; V-P is the output voltage signal waveform of the current detection module; I-H represents a larger current value, at which time the driving pulse signal output by the switching power supply converter is a continuous pulse; I-S represents a set current value. When the output current value I(VO) of the innovative resonant LLC switching power supply converter is less than the set value I-S, the switching power supply converter works in a small current load state 1, at which time the driving pulse signal output by the first control module 111 is a continuous pulse or a discontinuous pulse group with a higher frequency, and the voltage difference values U(GS-Q3) and U(GS-Q3) of the G electrode and the S electrode of the auxiliary switch transistors Q3 and Q4 are both low (close to 0V); when the output current value I(VO) is greater than the set value I-S, the switching power supply converter works in a large current load state 2, and the driving pulse signal output by the first control module 111 is a continuous pulse with a lower frequency, at which time the voltage difference values U(GS-Q3) and U(GS-Q3) of the G electrode and the S electrode of the auxiliary switch transistors Q3 and Q4 are both high (close to the auxiliary power supply voltage value).IL represents a smaller current value, at which point the drive pulse signal output by the first control module 111 is a discontinuous pulse group or a continuous pulse with a higher frequency. VH represents a high signal level. Ton is the high-level period of the drive signal voltage within one switching cycle T, i.e., the turn-on period of the switching transistor within one cycle T. Toff is the low-level period of the drive signal voltage within one switching cycle T, i.e., the turn-off period of the switching transistor within one cycle T. 180° is the 180° phase difference between the waveform of the first drive signal voltage difference U(GS-Q1) and the waveform of the second drive signal voltage difference U(GS-Q2) output by the first control module 111.

[0129] like Figure 11 As shown, the second control module 112 includes a current detection module 201, a reference voltage module 202, a first delay trigger module 203, a second delay trigger module 205, a first drive module 204, and a second drive module 206. The input and output signals of each module are as follows:

[0130] The input signals of the current detection module 201 include: the voltage waveform IS across the resonant capacitor C2 (the amplitude of the AC component in this voltage waveform is proportional to the maximum current flowing through the resonant capacitor C2) and the reference voltage V-REF. Both signals use the negative terminal VCC1-GND of the first auxiliary DC power supply as the reference zero point. Its output signal is the switching power converter output power status signal VP, also using the negative terminal VCC1-GND of the first auxiliary DC power supply as the reference zero point.

[0131] The output signal of the reference voltage module 202 includes: a reference voltage source V-REF generated by the positive terminal VCC1 of the first auxiliary DC power supply and regulated, with the negative terminal VCC1-GND of the first auxiliary DC power supply as the reference 0 point.

[0132] The input signals of the first delay trigger module 203 include: reference voltage V-REF, voltage difference between the source S circuit node A signal VS-Q1 of the main switching transistor Q1 and the negative terminal VCC1-GND of the first auxiliary DC power supply, and power status signal VP output by the current detection module 201. Both signal VS-Q1 and signal VP take the negative terminal VCC1-GND of the auxiliary DC power supply as the reference 0 point. Its output signal is the small current drive level signal V-Q3 of the auxiliary switching transistor Q3, which takes the negative terminal VCC1-GND of the first auxiliary DC power supply as the reference 0 point. Its working power supply positive terminal is VCC1 and its working power supply negative terminal is VCC1-GND.

[0133] The input signals of the second delay trigger module 205 include a reference voltage V-REF, a driving voltage signal VG-Q2 of the main switch transistor Q2, and an output power state signal V-P of the current detection module 201, both of which are referenced to the first auxiliary DC power negative pole VCC1-GND as the 0 point. The output signal is a small current driving level signal V-Q4 of the auxiliary switch transistor Q4, which is referenced to the first auxiliary DC power negative pole VCC1-GND as the 0 point. The working power positive pole is VCC1, and the working power negative pole is VCC1-GND.

[0134] The input signal of the first driving module 204 is a small current driving level signal V-Q3 of the auxiliary switch transistor Q3, and the output signal is a large current driving voltage signal VG-Q3 of the auxiliary switch transistor Q3, which is referenced to the second auxiliary DC power negative pole VCC2-GND (or circuit node VS-Q3, VS-Q1) as the 0 point.

[0135] The input signal of the second driving module 206 is a small current driving signal level signal of the auxiliary switch transistor Q4, and the output signal is a large current driving voltage signal VG-Q4 of the auxiliary switch transistor Q4, which is referenced to the first auxiliary DC power negative pole VCC1-GND as the 0 point.

[0136] Among them, VCC1 is the positive pole of the first auxiliary DC power supply; VCC1-GND is the negative pole of the first auxiliary DC power supply; VCC2 is the positive pole of the second auxiliary DC power supply; VCC2-GND is the negative pole of the second auxiliary DC power supply.

[0137] In Figure 9 The circuit and Figure 11 In the circuit block diagram shown in the figure, the positive pole of the resonance capacitor C2 is connected to the first input end of the current detection module 201, and the current signal I-S is input. The output end of the current detection module 201 is connected to the first input end of the first delay trigger module 203 and the first input end of the second delay trigger module 205, and outputs a voltage signal waveform V-P.

[0138] The second input end of the first delay trigger module 203 is connected with the source S circuit node of the main switch transistor Q1, and receives the first negative pole driving signal VS-Q1; the third input end of the first delay trigger module 203, the third input end of the second delay trigger module 205 and the output end of the reference voltage module 202 are commonly connected, and receive the reference voltage signal V-REF; the output end of the first delay trigger module 203 is connected with the first input end of the first driving module 204, and outputs the small current driving level signal V-Q3; the first output end of the first driving module 204 is connected with one end of the resistor R5, the other end of the resistor R5 is connected with the gate G of the auxiliary switch transistor Q3, and the third positive pole driving signal VG-Q3 is output; the second output end of the first driving module 204 is connected with the source S of the auxiliary switch transistor Q3, and the third negative pole driving signal VS-Q3 is output; the second auxiliary power supply positive pole is connected with the second input end of the first driving module 204, and the power supply signal VCC2 is output; the second auxiliary power supply negative pole VCC2-GND is connected with the third input end of the first driving module 204.

[0139] The second input end of the second delay trigger module 205 is connected with the main switch transistor Q2 driving voltage output end of the first control module 111, and receives the second positive pole driving signal VG-Q2; the output end of the second delay trigger module 205 is connected with the first input end of the second driving module 206, and outputs the small current driving level signal V-Q4; the first output end of the second driving module 206 is connected with one end of the resistor R7, and outputs the fourth positive pole driving signal VG-Q4; the other end of the resistor R7 is connected with the gate G of the auxiliary switch transistor Q4, and the second output end of the second driving module 206 is commonly connected with the input direct current voltage negative pole VI-GND, the auxiliary power supply negative pole VCC1-GND and the source S of the auxiliary switch transistor Q4, and receives the fourth negative pole driving signal VS-Q4.

[0140] The first auxiliary power supply positive pole is connected with the second input end of the current detection module 201, the first input end of the reference voltage module 202, the fourth input end of the first delay trigger module 203, the fourth input end of the second delay trigger module 205, the fourth input end of the first driving module 204 and the second input end of the second driving module 206, and outputs the power supply signal.

[0141] The first auxiliary power supply negative pole VCC-GND is connected with the third input end of the current detection module 201, the second input end of the reference voltage module 202, the fifth input end of the first delay trigger module 203, the fifth input end of the second delay trigger module 205, the fifth input end of the first driving module 204 and the third input end of the second driving module 206.

[0142] The elements in the current detection module 201 include: comparator IC201, resistor R201, resistor R202, resistor R203, resistor R204, resistor R205, resistor R206, capacitor C201, capacitor C202, capacitor C203. The connection mode is as follows: one end of capacitor C201 is connected to current signal I-S, the other end of capacitor C201 is connected to the common connection of one end of resistor R201, one end of resistor R202, one end of resistor R203, and one end of capacitor C202, the other end of resistor R201 is connected to the common connection of one end of resistor R204 and the non-inverting input terminal of comparator IC201, the other end of resistor R202 is connected to the negative electrode of the first auxiliary power supply VCC1-GND (reference 0 potential point), the other end of resistor R203 is connected to the common connection of one end of resistor R205 and reference voltage V-REF, the other end of resistor R205 is connected to the common connection of one end of resistor R206 and the inverting input terminal of comparator IC201, the other end of resistor R204 is connected to the cathode of diode D201, the anode of diode D201 is connected to the output terminal of comparator IC201, and the output signal V-P is output. The positive electrode of the power supply of comparator IC201 is connected to one end of capacitor C203, and receives the power supply signal VCC1, and the negative electrode of the power supply of comparator IC201 is connected to the other end of capacitor C203 and the negative electrode of the first auxiliary power supply VCC1-GND.

[0143] The elements in the reference voltage module 202 include: resistor R207, three-terminal voltage regulator IC202, capacitor C204. The connection mode is as follows: one end of resistor R207 is connected to the positive electrode of the first auxiliary power supply and receives the power supply signal VCC1, the other end of resistor R207, the cathode of three-terminal voltage regulator IC202, and one end of capacitor C204 are commonly connected and output the reference voltage signal V-REF, and the other end of capacitor C204 is commonly connected to the anode of IC202 and the negative electrode of the first auxiliary power supply VCC1-GND.

[0144] The elements in the first delay trigger module 203 include: a comparator IC 204, a D flip-flop IC 203, a resistor R208, a resistor R209, a resistor R210, a capacitor C205, a capacitor C206, a capacitor C207. The connection mode is as follows: one end of the resistor R208 receives the first negative driving signal VS-Q1, the other end of the resistor R208 is commonly connected with the resistor R209, the resistor R210, one end of the capacitor C205, the non-inverting input end of the comparator IC 204, and the five points of the comparator IC 204 are commonly connected with the first auxiliary power negative electrode VCC1-GND, the other end of the resistor R209 and the capacitor C205, the set 0 end R and the set 1 end S of the flip-flop IC 203, the power negative electrode of the IC 203, one end of the capacitor C206, one end of the capacitor C207, and the power negative electrode eight points of the comparator IC 204 are commonly connected, the other end of the resistor R210 is connected with the cathode of the diode D202, the anode of the diode D202, the output end of the comparator IC 204, and the clock input end C three points of the D flip-flop IC 203 are commonly connected and receive the signal V-T1, the data input end D of the D flip-flop IC 203 is connected with the output end of the comparator IC 201 in the current detection module 201 and receives the signal V-P, the other end of the capacitor C206, the other end of the C207, and the power positive electrode four points of the comparator IC 204 and the D flip-flop IC 203 are commonly connected with the first auxiliary power positive electrode and receive the power signal VCC1, and the data output end Q of the D flip-flop IC 203 commonly outputs the small current driving level signal V-Q3.

[0145] The elements in the first driving module 204 include: resistors R211, R212, R213, R214, R215, R216, R217, NPN type switching transistors Q203, Q205, PNP type switching transistors Q204, Q206, and a capacitor C208. The connection mode is as follows: one end of the resistor R211 receives a small current driving level signal V-Q3, the other end of the resistor R211 is connected with one end of the resistor R212 and the base B of the NPN type switching transistor Q205, the emitter E of the NPN type switching transistor Q205 is connected with the other end of the resistor R211 to the first auxiliary power negative pole VCC1-GND, the collector C of the NPN type switching transistor Q205 is connected with one end of the resistor R213, the other end of the resistor R213 is connected with the base B of the PNP type switching transistor Q206 and one end of the resistor R214, the other end of the resistor R214, the emitter E of the PNP type switching transistor Q206, the collector C of the NPN type switching transistor Q203 and one end of the capacitor C208 are connected with four points to the second auxiliary power positive pole to receive a power signal VCC2, the collector C of the PNP type switching transistor Q206 is connected with one end of the resistor R215, the other end of the resistor R215, the base B of the PNP type switching transistor Q204, the base B of the NPN type switching transistor Q203 and one end of the resistor R216 are connected with four points, the emitter E of the PNP type switching transistor Q204 and the emitter E of the NPN type switching transistor Q203 are connected with one end of the resistor R217 with three points, the other end of the resistor R217 is an output end, outputs the third positive pole driving signal VG-Q3, the collector C of the PNP type switching transistor Q204 and the other end of the resistor R216 are connected with the second auxiliary power negative pole VCC2-GND, and outputs the third negative pole driving signal VS-Q3.

[0146] The elements in the second delay trigger module 205 include: a comparator IC 205, a D flip-flop IC 206, a resistor R218, a resistor R219, a resistor R220, a capacitor C209, a capacitor C210, and a capacitor C211. The connection mode is as follows: one end of the resistor R218 receives the second positive driving signal VG-Q2, the other end of the resistor R218 is commonly connected with the five points of the one end of the resistor R219, the one end of the resistor R220, the one end of the capacitor C209, the non-inverting input terminal of the comparator IC 205, and the power negative electrode of the comparator IC 205, the other end of the resistor R219, the other end of the capacitor C209, the one end of the capacitor C210, the one end of the capacitor C211, the power negative electrode of the comparator IC 205, the set 0 terminal R and the set 1 terminal S of the D flip-flop IC 206, and the power negative electrode of the IC 206 are commonly connected with the first auxiliary power negative electrode VCC1-GND, the other end of the resistor R220 is connected with the cathode of the diode D203, the anode of the diode D203, the output terminal of the comparator IC 205, and the clock input terminal C of the D flip-flop IC 206 are commonly connected with the output signal V-T2, the data input terminal D of the D flip-flop IC 206 is connected with the output terminal of the comparator IC 201 in the current detection module 201 to receive the signal V-P, the other end of the capacitor C210 and the other end of the capacitor C211, and the four points of the power terminals of the comparator IC 205 and the D flip-flop IC 206 are commonly connected with the first auxiliary power positive electrode to receive the power signal VCC1, and the data output terminal Q of the D flip-flop IC 206 outputs the small current driving level signal V-Q4.

[0147] The elements in the second driving module 206 include: a resistor R221, a resistor R222, a resistor R223, an NPN type switching transistor Q201, a PNP type switching transistor Q202, and a capacitor C212. The connection mode is as follows: one end of the resistor R221 receives the small current driving level signal V-Q4, the other end of the resistor R221, the base B of the PNP type switching transistor Q202, the base B of the NPN type switching transistor Q201, and the one end of the resistor R222 are commonly connected with the four points, the emitter E of the NPN type switching transistor Q201 and the emitter E of the PNP type switching transistor Q202 are commonly connected with the three points of the one end of the resistor R223, the other end of the resistor R223 is connected with the output fourth positive driving signal VG-Q4, the collector C of the NPN type switching transistor Q201 is connected with the first auxiliary power positive electrode to receive the power signal VCC1, the collector C of the PNP type switching transistor Q202 and the other end of the resistor R216 are connected with the first auxiliary power negative electrode VCC1-GND to output the fourth negative driving signal VS-Q4.

[0148] The first control module 111 outputs the driving signals of the two main switch transistors Q1 and Q2 in the resonant LLC switching power converter of the application, and the reference 0 potentials of the two main driving signals are different. The first positive driving signal is VG-Q1, and the first negative driving signal is VS-Q1, which is the same as the negative potential of the second auxiliary DC power supply VCC2-GND; the second positive driving signal is VG-Q2, and the second negative driving signal is VS-Q2, which is the same as the negative potential of the first auxiliary DC power supply VCC1-GND.

[0149] The second control module 112 outputs the driving signals of the two auxiliary switch transistors Q3 and Q4 in the application, and the reference 0 potentials of the two auxiliary driving signals are different. The third positive driving signal is VG-Q3, and the third negative driving signal is VS-Q3, which is the same as the first negative driving signal VS-Q1 and the negative potential of the second auxiliary DC power supply VCC2-GND; the fourth positive driving signal is VG-Q4, and the fourth negative driving signal is VS-Q4, which is the same as the negative potential of the second auxiliary DC power supply VCC1-GND.

[0150] As shown in Figure 10 , the first driving signal waveform U(GS-Q1) output by the first control module 111 (i.e. the voltage difference between the first positive driving signal VG-Q1 and the first negative driving signal VS-Q1) and the second driving signal waveform U(GS-Q2) (i.e. the voltage difference between the second positive driving signal VG-Q2 and the second negative driving signal VS-Q2) have a phase difference of 180° in a switching cycle.

[0151] As shown in Figure 9 , Figure 10 , the third driving signal waveform U(GS-Q3) output by the second control module 112 is converted from low level to high level, and two necessary conditions are required for the auxiliary switch transistor Q3 to be triggered on.

[0152] The first condition: the LLC power converter output current value I(VO) is a large current value I-H, as shown in Figure 10 , the current value I-H needs to meet the condition: I-H is greater than the set current value I-S, i.e. I-H>I-S. At this time, the driving signal waveforms U(GS-Q1) and U(GS-Q2) output by the first control module 111 are continuous square waves.

[0153] The second condition is that the first drive signal waveform U(GS-Q1) is high or the source drive signal VS-Q1 of the main switch transistor Q1 is high voltage (reference 0 potential is VCC1-GND), which is only used as a trigger condition for the starting time when U(GS-Q3) is converted from low to high. When U(GS-Q1) is maintained high, the second condition is no longer used as a basis for judgment.

[0154] As shown in the above, Figure 10 The fourth drive signal waveform U(GS-Q4) output by the second control module 112 is converted from low to high, and the auxiliary switch Q4 is triggered to be turned on, which requires two necessary conditions.

[0155] The first condition is that the output current value I(VO) of the converter is a large current value I-H, as shown in the above, Figure 4 The current value I-H needs to meet the condition that I-H is greater than the set current value I-S, that is, I-H>I-S. At this time, the drive signal waveforms U(GS-Q1) and U(GS-Q2) output by the first control module 111 are continuous square waveforms.

[0156] The second condition is that the second drive signal waveform U(GS-Q2) is high, which is only used as a trigger condition for the starting time when U(GS-Q4) is converted from low to high. When U(GS-Q4) is maintained high, the second condition is no longer used as a basis for judgment.

[0157] As described above, when the power converter enters a large current load state 2 from a small current load state 1, the auxiliary switch transistors Q3 and Q4 cannot be turned on immediately. At this time, the second control module 112 needs to determine the state of the main switch transistors Q1 and Q2, and wait until the main switch transistors Q1 and Q2 enter a stable on state before triggering the auxiliary switch transistors Q3 and Q4 to be turned on. After the auxiliary switch transistors Q3 and Q4 are triggered to be turned on, they will be locked in the on state. At this time, only when the following conditions for triggering the auxiliary switch transistors Q3 and Q4 to be turned off are met, the auxiliary switch transistors Q3 and Q4 can be turned off. Therefore, the auxiliary switch transistors Q3 and Q4 both realize the high-efficiency zero-voltage soft turn-on function, and the turn-on of the auxiliary switch transistors Q3 and Q4 does not interfere with the main circuit.

[0158] As shown in the above, Figure 10 The third drive signal waveform U(GS-Q3) output by the second control module 112 is converted from high to low, and the auxiliary switch Q3 is triggered to be turned off, which requires two necessary conditions.

[0159] The first condition is that the output current value I(VO) of the power converter is a small current value I-L, as shown in the above, Figure 10As shown, the current value I-L needs to meet the condition: I-L is less than the set current value I-S, that is, I-L < I-S. At this time, the driving signal waveforms U(GS-Q1) and U(GS-Q2) output by the first control module 111 are discontinuous square wave groups (that is, the low-level time intervals repeatedly appear in the square wave and exceed one square wave period) or continuous square waves with a relatively high frequency.

[0160] Second condition: The first driving signal waveform U(GS-Q1) is at a high level or the source potential VS-Q1 of the switching transistor Q1 is at a high voltage (referring to the reference 0 potential as VCC1-GND). This condition only serves as the triggering condition for the starting moment when U(GS-Q3) changes from a high level to a low level. When U(GS-Q1) maintains a low level, this second condition no longer serves as the judgment basis.

[0161] Similarly, as Figure 10 shown, for the fourth driving signal waveform U(GS-Q4) output by the second control module 112 to change from a high level to a low level, and at the same time for the auxiliary switching transistor Q4 to be triggered to turn off, two necessary conditions are required:

[0162] First condition: The output current value I(VO) of the power converter is a small current value I-L. As Figure 10 shown, the current value I-L needs to meet the condition: I-L is less than the set current value I-S, that is, I-L < I-S. At this time, the driving signal waveforms U(GS-Q1) and U(GS-Q2) output by the first control module 111 are discontinuous square wave groups (that is, the low-level time intervals repeatedly appear in the square wave and exceed one square wave period) or continuous square waves with a relatively high frequency.

[0163] Second condition: The second driving signal waveform U(GS-Q2) is at a high level. This condition only serves as the triggering condition for the starting moment when U(GS-Q4) changes from a high level to a low level. When U(GS-Q4) maintains a low level, this second condition no longer serves as the judgment basis.

[0164] As above, when the power converter enters the "small current load state 1" from the "large current load state 2", it cannot immediately turn off the auxiliary switching transistors Q3 and Q4. At this time, the second control module 112 still needs to judge the states of the main switching transistors Q1 and Q2, and wait until the main switching transistors Q1 and Q2 enter a stable on state before immediately triggering to turn off the auxiliary switching transistors Q3 and Q4. After the auxiliary switching transistors Q3 and Q4 are triggered to turn off, they will be locked in the off state. At this time, only when the above conditions for triggering the auxiliary switching transistors Q3 and Qo to turn on are met can they be switched to the on state. Therefore, both the auxiliary switching transistors Q3 and Q4 have achieved the high-efficiency zero-voltage soft turn-off function, and the turn-off of the auxiliary switching transistors Q3 and Q4 does not cause any interference to the main circuit.

[0165] The meaning of the component symbols in the above schematic diagram includes the series and parallel combination of the individual components and the components with equivalent properties.

[0166] Some components in the schematic diagram can be selected with limit values, for example:

[0167] a. The resistance values of the resistors R2, R4, R6, R8, R204, R206, R209, R210, R212, R214, R216, R219, R220, and R222 can be infinite, equivalent to open circuit, which can be removed, and the resistor symbols can not be drawn in the schematic diagram.

[0168] b. The resistance values of the resistors R1, R3, R5, R7, R201, R205, R208, R215, R217, R218, and R223 can be 0Ω, at which time the resistor symbols can be replaced by a short-circuit connection line.

[0169] c. The diodes D201, D202, and D203 can be removed, equivalent to short circuit, and replaced by a short-circuit connection line.

[0170] d. The capacitance values of the capacitors C203, C204, C205, C206, C207, C208, C209, C210, C211, and C212 can be 0F, at which time the capacitors are equivalent to open circuit, and the capacitor symbols can not be drawn in the schematic diagram.

[0171] The working principle of the implementation circuit of the resonant LLC switching power supply converter of the present application is as shown in Figure 9 、 Figure 11 、 Figure 12 The working principle of the implementation circuit of the resonant LLC switching power supply converter of the present application is as shown in

[0172] As shown in Figure 9 , the core control integrated circuit IC of the first control module 111 in the resonant LLC switching power supply converter of the present application can apply an existing LLC switching power supply converter dedicated control IC, or can apply an existing general-purpose digital signal processor DSP to write software programs to achieve.

[0173] As shown in Figure 12 , the principle analysis of a detailed implementation circuit of the second control module 112 according to the block diagram shown in Figure 11 is as follows:

[0174] (1) The working principle of the current detection module 201 is as follows:

[0175] The effective value of the AC component in the voltage waveform across the resonant capacitor C2 of the resonant LLC switching power converter is proportional to the effective value of the current flowing through the capacitor C2. Therefore, the current value of the resonant cavity of the LLC switching power converter can be represented by detecting the effective value of the AC component of the voltage across the capacitor C2, as shown in Equation 1:

[0176] When U(GS-Q1) is high level and U(GS-Q2) is low level:

[0177] I Q1-ON = I L1 = I T1-1 = I C2 = C2*dV C2 / dt Equation 1

[0178] When U(GS-Q1) is low level and U(GS-Q2) is high level:

[0179] I Q2-ON = I L1 = I T1-1 = I C2 = C2*dV C2 / dt Equation 1

[0180] In Equation 1, the symbols have the following meanings: C* is the capacitance value of the capacitor C*; * is an arbitrary positive integer or a letter label; I Q1-ON is the instantaneous current value flowing through the drain D and the source S of the main switching transistor Q1 when the voltage U(GS-Q1) is high level and the main switching transistor Q1 is turned on, with the current flowing into the drain D as the positive direction; I Q2-ON is the instantaneous current value flowing through the drain D and the source S of the main switching transistor Q2 when the voltage U(GS-Q2) is high level and the main switching transistor Q2 is turned on, with the current flowing into the drain D as the positive direction; I L1 is the instantaneous current value flowing through the inductor L1, with the current flowing out from one end of the inductor L1 and flowing into one end of the primary winding of the transformer T1 as the positive direction; I T1-1 is the instantaneous current value flowing through the primary winding of the transformer T1, with the current flowing out from one end of the inductor L1 and flowing into one end of the primary winding of the transformer T1 as the positive direction; I C2 is the instantaneous current value flowing through the capacitor C2, with the current flowing out from one end of the primary winding of the transformer T1 and flowing into the positive voltage end of the capacitor C2 as the positive direction; dV C2 / dt is the rate of change of the instantaneous voltage value across the capacitor C2, with an increase in voltage value as positive rate of change and a decrease in voltage value as negative rate of change.

[0181] In the above Equation 1, applying Fourier analysis, the resonant cavity current I C2 can be approximately expressed as a sinusoidal wave I C2-MAXSubstituting *sin(ωt) into Formula 1, we can obtain an approximate expression for the AC component of the voltage waveform across capacitor C2, as shown in Formula 2 below:

[0182]

[0183] From Equation 2, we can see that the maximum voltage amplitude V of the AC component in the voltage waveform across capacitor C2 is... C2-AC-MAX The maximum current amplitude I of the current waveform of the resonant cavity circuit C2-MAX It is a fixed proportional relationship.

[0184] like Figure 12 As shown, capacitors C201, C202, and resistor R202 in the current detection circuit 201 form an AC voltage divider circuit, where the capacitive reactances of capacitors C201 and C202 are X respectively. C201 =1 / (2πf) W *C201),X C202 =1 / (2πf) W *C202), this voltage divider circuit increases the AC voltage amplitude V across capacitor C202. C202-AC-MAX The amplitude V of the AC component in the voltage waveform across capacitor C2 C2-AC-MAX It is a fixed proportional relationship.

[0185] like Figure 12 As shown, resistors R215 and R202 form a DC reference voltage divider circuit. Since resistor R202 is connected in parallel with capacitor C202, this DC voltage is the DC component of the voltage waveform across capacitor C202.

[0186] Therefore, as shown above, the voltage across capacitor C202 (resistor R202) is composed of the DC and AC components, and the resulting voltage is calculated as shown in Formula 3 below:

[0187]

[0188] The symbols in Formula 2-3 have the following meanings: X C201 For capacitor C201 at the operating switching frequency f W Capacitive reactance value; X C202 For capacitor C202 at the operating switching frequency f W Capacitive reactance value; f W ΔV represents the operating switching frequency of the resonant LLC switching power converter. C2 ΔV is the instantaneous value of the AC component of the voltage waveform across capacitor C2. C2-MAX V represents the maximum amplitude of the AC component of the voltage waveform across capacitor C2. C202-MAX V represents the maximum amplitude of the voltage waveform across capacitor C202. C202-DCV is the DC component voltage value of the voltage waveform across capacitor C202; V C202-AC-MAX V is the maximum voltage amplitude of the AC component of the voltage waveform across capacitor C202; V C202-MAX V is the maximum amplitude of the voltage waveform across capacitor C202; I C2-MAX I is the maximum amplitude of the current flowing through capacitor C2, i.e., the maximum amplitude of the resonant cavity current waveform of the resonant LLC switching power supply converter; U V-REF V is the voltage difference between the positive terminal of the reference voltage V-REF and the reference 0 potential VCC1-GND.

[0189] The current detection method described above does not have any current detection element connected in series in the main resonant loop, thus does not increase significant power consumption, and has a smaller volume and easier printed circuit board (PCB) layout, so that the current detection method is widely applied. Other current detection methods of various prior arts, such as the series resistance direct current detection method and the current transformer isolation detection method, can also be applied to the second control module 112 of the resonant LLC switching power supply converter of the present application, and will not be described here.

[0190] The working principle of the reference voltage module 202 is described as follows: the positive terminal VCC of the auxiliary power supply supplies power to the three-terminal voltage regulator IC202 through the resistor R207, the reference terminal R of the three-terminal voltage regulator IC202 is connected to the cathode K, and the anode A of the three-terminal voltage regulator IC202 is connected to the negative terminal VCC1-GND (reference 0 potential) of the auxiliary power supply, so that the cathode K of the three-terminal voltage regulator IC202 generates a reference voltage V-REF relative to the reference 0 potential VCC1-GND after voltage stabilization and adjustment.

[0191] The working principle of the first delay trigger module 203 is described as follows: as shown in Figure 3 , Figure 6 , the high-voltage input signal U VS-Q1 between the circuit node VS-Q1 of the resonant LLC switching power supply converter of the present application and the 0 potential reference point VCC1-GND is divided by the resistor R208 and the resistor R209 to form a low-voltage waveform in proportion, and the calculation is as formula 4:

[0192] U R209 = U VS-Q1 *R 209 (R 208 +R 209 ) Formula 4

[0193] As shown in Figure 10 and Figure 12 , when the first driving signal waveform U(GS-Q1) output by the first control module 111 is converted from low level to high level, the main switching transistor Q1 is converted from off state to on state, at this time U VS-Q1 ≈UVI By selecting the proper resistance value of resistance R208 and resistance R209, the voltage value across resistance R209 at this time satisfies the condition: U VCC > U R209 > U V-REF At this time, the output voltage V-T1 of comparator IC204 is converted from low level to high level, and the flip-flop IC203 starts triggering at the rising edge of the V-T1 voltage waveform, and transmits the signal V-P inputted at the input end to the output end and outputs the small current driving level signal V-Q3.

[0194] When the signal V-P is low level, the small current driving level signal V-Q3 is also converted to low level at this time, indicating that the power converter is in small current load state 1 at this time. When the signal V-P is high level, the small current driving level signal V-Q3 is also converted to high level at this time, indicating that the converter is in large current load state 2 at this time. And before the next rising edge of V-T1 signal appears, the current output state of small current driving level signal V-Q3 is locked.

[0195] When the first driving signal waveform U(GS-Q1) outputted by the first control module 111 is converted from high level to low level, the main switch transistor Q1 is converted from on state to off state, at this time U VS-Q1 ≈ 0V, U R209 ≈ 0V. At this time, the signal V-T1 voltage outputted by the output end of comparator IC204 is converted from high level to low level, and at this time the flip-flop IC203 still locks the current output state of small current driving level signal V-Q3.

[0196] The resistance R210 and diode D202 constitute the positive feedback hysteresis circuit of comparator IC204. When the output signal V-T1 of IC204 output end is high level, the current flows through the anode and cathode of diode D202 and the two ends of resistance R210 to the two ends of resistance R209, so that the voltage value U R209 of the two ends of resistance R209 rises, so that the comparator IC204 can also stabilize the high level state of its output signal V-T1 when disturbed.

[0197] The working principle of the first driving module 204 is explained as follows: as Figure 9 , Figure 12As shown, when the small current drive level signal V-Q3 is converted from low level to high level, the current flows out from the output pin of the flip-flop IC203 in the first delay trigger module 203, and is divided into two paths after passing through the resistor R211, one of which flows into the base B of the NPN switch transistor Q205, and then is merged into the first auxiliary power negative pole VCC1-GND through the collector C; the other path is merged into the first auxiliary power negative pole VCC1-GND through the resistor R212. At this time, the NPN switch transistor Q205 is converted from the off state to the on state.

[0198] As shown above, due to the current amplification function of the NPN switch transistor Q205, more current than that flowing into its base B flows out from the second auxiliary power positive pole, and is divided into two paths, one of which flows into the emitter E of the PNP switch transistor Q206, and then is merged into the resistor R213 through its base B; the other path is merged into the resistor R213 through the resistor R214. This current flows into the collector C of the NPN switch transistor Q205 through the resistor R213, and then is merged into the first auxiliary power negative pole VCC1-GND through its emitter E. At this time, the PNP switch transistor Q206 is converted from the off state to the on state.

[0199] As shown above, due to the current amplification function of the PNP switch transistor Q206, more current than that flowing into its base B flows out from the second auxiliary power positive pole VCC2, and flows into the base B of the NPN switch transistor Q203 in turn through the emitter E of the PNP switch transistor Q206, the collector C of the PNP switch transistor Q206, and the resistor R215, and is divided into two paths, one of which is merged into the resistor R217 through the emitter E, and the other path flows into the second auxiliary power negative pole VCC2-GND through the resistor R216. At this time, the NPN switch transistor Q203 is converted from the off state to the on state. Due to the current amplification function of the NPN switch transistor Q203, more current than that flowing into its base B flows out from the second auxiliary power positive pole VCC2, and is merged into the resistor R217 through the collector C of the NPN switch transistor Q203 and the emitter E of the NPN switch transistor Q203, and the third positive pole drive signal VG-Q3 is outputted through the resistor R217.

[0200] As described above, the third positive drive signal VG-Q3 flows through resistor R5 into the gate G of the NMOSFET auxiliary switching transistor Q3 and one end of resistor R6, then through the source S of the auxiliary switching transistor Q3 and the other end of resistor R6 to the common connection point between the output terminal VS-Q3 and the negative terminal VCC2-GND of the second auxiliary power supply. When the charging voltage of the parasitic capacitance between the gate G and the source S of Q3 exceeds the turn-on threshold voltage Vth of Q3, the auxiliary switching transistor Q3 switches from the off state to the on state. When the state transition is stable, since its input voltage V-Q3 remains at a high level, the voltage difference U(GS-Q3) between its output positive drive signal VG-Q3 and negative drive signal VS-Q3 will also remain at a high level, and the auxiliary switching transistor Q3 will remain in the on state.

[0201] like Figure 9 , Figure 12 As shown, when the low-current drive level signal V-Q3 transitions from high to low, the current flowing into the base B of the NPN switching transistor Q205 drops to 0A, at which point the NPN switching transistor Q205 transitions from the on state to the off state. Therefore, the current flowing through the base B of the PNP switching transistor Q206 drops to approximately 0A, and the PNP switching transistor Q206 also transitions from the on state to the off state. Consequently, the current flowing into the base B of the NPN switching transistor Q203 drops to approximately 0A, and the NPN switching transistor Q203 also transitions from the on state to the off state.

[0202] As above, since the NPN switch transistor Q203 is in the off state, the high-level driving voltage provided by the resistor R217 and the resistor R5 between the gate G and the source S of the auxiliary switch transistor Q3 disappears. At this time, the charge stored in the parasitic capacitor between the gate G and the source S of the auxiliary switch transistor Q3 flows out from the gate G and is divided into two paths, one of which flows back to the source S (i.e. the circuit node VS-Q3) of the auxiliary switch transistor Q3 through the resistor R6, and the other flows into the emitter E of the PNP switch transistor Q204 in turn through the resistor R5 and the resistor R217, flows out from the base B, and then flows back to the source S (i.e. the circuit node VS-Q3) of the auxiliary switch transistor Q3 through the resistor R216. At this time, due to the current amplification function of the PNP switch transistor Q204, a larger current than the base B will flow out from the collector C and flow back to the source S (i.e. the circuit node VS-Q3) of the auxiliary switch transistor Q3. When the discharge voltage value of the parasitic capacitor between the gate G and the source S of the auxiliary switch transistor Q3 is lower than the on threshold voltage Vth value of the auxiliary switch transistor Q3, the auxiliary switch transistor Q3 is switched from the on state to the off state. When the state transition is stable, since the input voltage V-Q3 is maintained at a low level, the voltage difference U(GS-Q3) between the positive electrode driving signal VG-Q3 and the negative electrode driving signal VS-Q3 at the output end will also be maintained at a low level, and the auxiliary switch transistor Q3 will be maintained in the off state.

[0203] The working principle of the second delay trigger module 203 is explained as follows: The circuit composition and internal working principle of the second delay trigger module 203 are exactly the same as those of the first driving module 204. The input reference voltage signal V-REF and the current state signal V-P of the second delay trigger module 203 are the same as the corresponding input signals of the first driving module 204. The working power positive electrode VCC1 and the working power negative electrode VCC1-GND of the second delay trigger module 203 are exactly the same as the working power of the first driving module 204.

[0204] The input voltage signal VG-Q2 of the second delay trigger module 203 is the driving voltage signal positive electrode of the main switch transistor Q2, which is different from the circuit node of the corresponding input voltage signal VS-Q3 of the first driving module 204. The signal output end connection point VS-Q4 of the second delay trigger module 203 is different from the signal output end connection point V-Q3 of the first driving module 204.

[0205] As described above, the working principle analysis process of the second delay trigger module 203 is similar to that of the first driving module 204, and will not be described here.

[0206] The working principle of the second driving module 206 is explained as follows: Figure 9 , Figure 12As shown, when the small current drive level signal V-Q4 is converted from low level to high level, the current flows out from the output terminal of the flip-flop IC 206 in the second delay trigger module 205, and the current is divided into two paths after passing through the resistor R221, one of which flows into the base B of the NPN switching transistor Q201 and then flows into the resistor R223 through the emitter E, and the other flows into the first auxiliary power supply negative pole VCC1-GND through the resistor R222. At this time, the NPN switching transistor Q201 is converted from the off state to the on state, and due to the current amplification function of the NPN switching transistor Q201, a larger current than that flowing into the base B flows out from the first auxiliary power supply positive pole VCC1, flows into the resistor R223 through the collector C of the NPN switching transistor Q201 and the emitter E of the NPN switching transistor Q201, flows to the output terminal through the resistor R223, and then flows into the gate G of the auxiliary switching transistor Q4 and one end of the resistor R8 through the resistor R7, flows into the common connection point of the output terminal and the first auxiliary power supply negative pole VCC1-GND through the other end of the resistor R8 and the source S of the auxiliary switching transistor Q4. When the charging voltage value of the parasitic capacitance between the gate G and the source S of the auxiliary switching transistor Q4 exceeds the opening threshold voltage Vth value of the auxiliary switching transistor Q4, the auxiliary switching transistor Q4 is converted from the off state to the on state. When the state conversion is stable, since the small current drive level signal V-Q4 is maintained at high level, the voltage difference U(GS-Q4) between the positive pole drive signal VG-Q4 and the negative pole drive signal VS-Q4 of the output terminal will be maintained at high level, and the auxiliary switching transistor Q4 will be maintained in the on state.

[0207] As shown in Figure 9 , Figure 12 When the small current drive level signal V-Q4 is converted from high level to low level, the current flowing into the base B of the NPN switching transistor Q201 decreases to 0A. At this time, the NPN switching transistor Q201 is converted from the on state to the off state.

[0208] As above, since the NPN switch transistor Q201 is in the off state, the high-level driving voltage provided by the resistor R223 and the resistor R7 between the gate G and the source S of the auxiliary switch transistor Q4 disappears. At this time, the charge stored in the parasitic capacitor between the gate G and the source S of the auxiliary switch transistor Q4 flows out from the gate G and is divided into two paths, one of which flows back to the source S (i.e. the circuit node VS-Q4) through the resistor R8, and the other flows into the emitter E of the PNP switch transistor Q202 in turn through the resistors R7 and R223, flows out from the base B, and then flows back to the source S (i.e. the circuit node B) of the auxiliary switch transistor Q4 through the resistor R222. At this time, due to the current amplification function of the PNP switch transistor Q202, a larger current than the base B will flow out from the collector C and flow back to the source S (i.e. the circuit node B) of the auxiliary switch transistor Q4. When the discharge voltage value of the parasitic capacitor between the gate G and the source S of the auxiliary switch transistor Q4 is lower than the on threshold voltage Vth value of the auxiliary switch transistor Q4, the auxiliary switch transistor Q4 is switched from the on state to the off state. When the state transition is stable, since the input voltage V-Q4 is maintained at a low level, the voltage difference U(GS-Q4) between the positive electrode driving signal VG-Q4 and the negative electrode driving signal VS-Q4 at the output end will be maintained at a low level, and the auxiliary switch transistor Q4 will be maintained in the off state.

[0209] As shown in Figure 10 , Figure 11 , Figure 12 The innovative resonant LLC switching power supply converter of the present application distinguishes between small current load state 1 and large current load state 2 by detecting the size of the resonant cavity current. In small current load state 1, the current detection module 201 outputs a low-level signal V-P, and in large current load state 2, the current detection module 201 outputs a high-level signal V-P, where each working state has 4 working modes respectively.

[0210] The working process and principle analysis of different load states and different working modes of the innovative resonant LLC switching power supply converter of the present application are as follows:

[0211] (1) Small current load state 1:

[0212] As shown in Figure 10 When the innovative resonant LLC switching power supply converter of the present application is converted from large current load state 2 to small current load state 1, the output signal V-P of the current detection module 201 will be converted from high level to low level.

[0213] At this time, according to the analysis of the first drive module 204, the voltage difference U(GS-Q3) between the drive signal VG-Q3 output by the positive output end of the first drive module 204 and the drive signal VS-Q3 output by the negative output end will be converted from high level to low level, and the auxiliary switch transistor Q3 will be converted from the on state to the off state. When the state conversion is stable, the voltage difference U(GS-Q3) will be maintained at low level, and the auxiliary switch transistor Q3 will be maintained at off state.

[0214] At this time, according to the analysis of the second drive module 206, the voltage difference U(GS-Q4) between the drive signal VG-Q4 output by the positive output end of the second drive module 206 and the drive signal VS-Q4 output by the negative output end will be converted from high level to low level, and the auxiliary switch transistor Q4 will be converted from the on state to the off state. When the state conversion is stable, the voltage difference U(GS-Q4) will be maintained at low level, and the auxiliary switch transistor Q4 will be maintained at off state.

[0215] As described above, in the small current load state 1, the auxiliary switch transistors Q3 and Q4 in the resonant LLC switching power converter of the present application are maintained in off state, and at this time, the working mode is the same as that of the prior art LLC switching power converter, which will not be described here.

[0216] (2) Large current load state 2:

[0217] As shown in Figure 10 When the power converter is converted from the small current load state 1 to the large current load state 2, the output signal V-P of the current detection module 201 will be converted from low level to high level.

[0218] At this time, according to the analysis of the first drive module 204, the voltage difference U(GS-Q3) between the drive signal VG-Q3 output by the positive output end of the first drive module 204 and the drive signal VS-Q3 output by the negative output end will be converted from high level to low level, and the auxiliary switch transistor Q3 will be converted from the on state to the off state. When the state conversion is stable, the voltage difference U(GS-Q3) will be maintained at low level, and the auxiliary switch transistor Q3 will be maintained at off state.

[0219] At this time, according to the analysis of the second drive module 206, the voltage difference U(GS-Q4) between the drive signal VG-Q4 output by the positive output end of the second drive module 206 and the drive signal VS-Q4 output by the negative output end will be converted from high level to low level, and the auxiliary switch transistor Q4 will be converted from the on state to the off state. When the state conversion is stable, the voltage difference U(GS-Q4) will be maintained at low level, and the auxiliary switch transistor Q4 will be maintained at off state.

[0220] As above, in the large current load state 2, both auxiliary switch transistors Q3 and Q4 in the resonant LLC switching power supply converter of the present application are maintained in the on state. At this time, the power supply converter of the present application has the following four working modes, the principles and process analysis of which are as follows:

[0221] Working mode 1:

[0222] As shown in FIG. 4, in the mode 1 period in the large current load state 2, the first output drive voltage U(GS-Q1) of the control circuit module 10 is converted from low level to high level, the second drive voltage U(GS-Q2) is maintained at low level, the main switch transistor Q1 is converted from off state to on state, and the main switch transistor Q2 is off. The output drive voltages U(GS-Q3) and U(GS-Q4) of the second control module 112 are maintained at high level, the auxiliary switch transistor Q3 is maintained on, and the auxiliary switch transistor Q4 is maintained on. As described in working mode 4, the on process of the main switch transistor Q1 is zero voltage on, so the on loss power is close to 0W. Figure 10 As shown in FIG. 4, due to the extremely low on resistance value Rds(on) of the auxiliary switch transistors Q3 and Q4, one end of the auxiliary capacitor C4 is equivalent to being connected to the source S of the main switch transistor Q1, and the other end of the auxiliary capacitor C4 is directly connected to the drain D of the main switch transistor Q1. Therefore, the auxiliary capacitor C4 at this time is equivalent to being connected in parallel between the drain D and the source S of the main switch transistor Q1.

[0223] Figure 10 Similarly, one end of the auxiliary capacitor C5 is equivalent to being connected to the source S of the main switch transistor Q2, and the other end of the auxiliary capacitor C5 is directly connected to the drain D of the main switch transistor Q2. Therefore, the auxiliary capacitor C5 at this time is equivalent to being connected in parallel between the drain D and the source S of the main switch transistor Q1.

[0224] At this time, the current flows out from the input DC voltage positive electrode VI, through the N-channel insulated gate field effect transistor Q1 drain D, the main switch transistor Q1 source S, the inductor L1, the transformer T1 primary coil, the capacitor C2, and flows into the input DC voltage negative electrode VI-GND. At this time, the capacitor C2 is charged, and the inductor L1 stores energy in the positive direction (here, the current flowing out from the inductor L1 into the transformer T1 is defined as positive current). At this time, the voltage U C4 ≈0V across the auxiliary capacitor C4, and the voltage U Q1-DS-ON ≈VI across the auxiliary capacitor C5. C5 Q2-DS-OFF

[0225] At this time, the current flows out from the input DC voltage positive electrode VI, through the N-channel insulated gate field effect transistor Q1 drain D, the main switch transistor Q1 source S, the inductor L1, the transformer T1 primary coil, the capacitor C2, and flows into the input DC voltage negative electrode VI-GND. At this time, the capacitor C2 is charged, and the inductor L1 stores energy in the positive direction (here, the current flowing out from the inductor L1 into the transformer T1 is defined as positive current). At this time, the voltage U

[0226] ​​​At this time, the output current from the positive terminal of the induced voltage of the secondary coil of transformer T1, after passing through the positive and negative terminals of diode D1, is divided into two paths. One path flows through the positive and negative terminals of capacitor C3 back to the negative terminal of the secondary coil voltage of T1. The other path flows through the positive terminal VO of the output DC voltage, the positive and negative terminals of the load, and the negative terminal VO-GND of the output DC voltage back to the negative terminal of the secondary coil voltage of T1. At this time, capacitor C3 is charging; diode D2 is turned off in reverse.

[0227] Working Mode 2:

[0228] like Figure 10 As shown, during mode 2 of the high-current load state 2, the first output drive voltage U(GS-Q1) of the control circuit module 10 changes from a high level to a low level in operating mode 1, while its second output drive voltage U(GS-Q2) remains at a low level. The main switch transistor Q1 changes from the on state to the off state, and the main switch transistor Q2 remains off. The output drive voltages U(GS-Q3) and U(GS-Q4) of the second control module 112 remain at a high level, while the auxiliary switch transistor Q3 remains on and the auxiliary switch transistor Q4 remains on.

[0229] At this point, after the current flows out from the positive terminal VI of the input DC voltage, it splits into two paths. One path passes through the drain D of the main switching transistor Q1, the semiconductor channel resistance between the drain D and the source S of the main switching transistor Q1, and the source S of the main switching transistor Q1, before converging into the circuit node VS-Q1. The other path passes through the positive terminal of the auxiliary capacitor C4, the negative terminal of C4, the drain D of the auxiliary switching transistor Q3, and the source S of the auxiliary switching transistor Q3, before also converging into the circuit node VS-Q1. Then, it passes through the two ends of the inductor L1, the two ends of the primary coil of the transformer T1, and the positive terminal of the capacitor C2, before converging into the common connection point between the negative terminal of the capacitor C2 and the source S of the main switching transistor Q2, and finally flows back to the negative terminal VI-GND of the DC voltage.

[0230] As described above, when the current flows through the common connection node between the drain D of the main switching transistor Q1 and the positive terminal of the auxiliary capacitor C4, it splits into two branches. One branch flows through the semiconductor channel resistance between the drain D and source S of the main switching transistor Q1, and merges into the circuit node VS-Q1. This resistance value changes rapidly from near zero ohms when the main switching transistor Q1 is turned on (the transition period is generally several hundred nanoseconds) to several megaohms when the main switching transistor Q1 is turned off. The other branch of the current flows into the positive terminal of the auxiliary capacitor C4, through the negative terminal of the auxiliary capacitor C4, the drain D of the auxiliary switching transistor Q3, and the source S of the auxiliary switching transistor Q3, and also merges into the circuit node VS-Q1. At this time, the auxiliary capacitor C4 is charged, and its voltage across it increases from near 0V in mode 1 to U. C4 ≈U Q1-DS-OFF ≈VI.

[0231] When the main switching transistor Q1 completes the turn-off process and the auxiliary capacitor C4 completes the charging process, since the current in the inductor L1 cannot change abruptly, the freewheeling current flows out from the positive terminal of the self-induced voltage of the inductor L1 (the positive direction current defined above), through the two ends of the primary coil of the transformer T1, the two ends of the capacitor C2, the source S of the main switching transistor Q2, the positive and negative terminals of the parasitic reverse body diode between the source S and the drain D of the main switching transistor Q2, and the drain D of the main switching transistor Q2, and flows back to the negative terminal of the self-induced voltage of the inductor L1.

[0232] At this time, the voltage between the source (S) and drain (D) of the main switching transistor Q2 is approximately -1V, and the main switching transistor Q2 meets the zero-voltage turn-on condition.

[0233] By selecting appropriate specifications for inductor L1 and transformer T1, the energy stored in inductor L1 is rapidly released, and the current I in the resonant cavity increases. L1 =I T1-1 =I C2 The current drops rapidly, and capacitor C2 continues to charge. When the resonant cavity current I... L1 The excitation current I of transformer T1 is less than or equal to T1-M At this time, diode D1 quickly switches from the conducting state to the reverse turning-off state. At this time, diode D2 is still in the reverse turning-off state, and the secondary coil of transformer T1 also quickly stops charging capacitor C3, and capacitor C3 quickly switches from the charging state to the discharging state.

[0234] At this time, the current flows out from the positive terminal of capacitor C3, through the positive terminal VO of the output DC voltage, the positive and negative terminals of the load, the negative terminal VO-GND of the output DC voltage, and back to the negative terminal of capacitor C3.

[0235] As described above, the current flowing from VI, after passing through the common node of the drain D of the main switching transistor Q1 and the positive terminal of capacitor C4, splits into two branches. By selecting an appropriate capacitance for capacitor C4 and increasing the turn-off speed of the main switching transistor Q1, during the turn-off process of the main switching transistor Q1 in operating mode 2, one branch has a very small current value and a short duration through the semiconductor channel between the drain D and source S of the main switching transistor Q1, while the other branch has a larger current value and a longer duration through the auxiliary capacitor C4. This allows for near-zero current soft turn-off of the main switching transistor Q1, thereby significantly reducing the turn-off loss of the main switching transistor Q1 and further improving the efficiency of the switching power supply converter to meet the growing demands for energy conservation and carbon reduction in today's socio-economic environment.

[0236] Working Mode 3:

[0237] like Figure 10As shown, during mode 3 of the high-current load state 2, the first output drive voltage U(GS-Q1) of the control circuit module 10 remains at a low level, while the second output drive voltage U(GS-Q2) changes from a low level to a high level during mode 2. The main switch transistor Q1 remains off, and the main switch transistor Q2 changes from off to on. The output drive voltages U(GS-Q3) and U(GS-Q4) of the second control module 112 remain at a high level, and the auxiliary switch transistors Q3 and Q4 remain on. As described in mode 2, the turn-on process of the main switch transistor Q2 is zero-voltage turn-on, therefore the turn-on power loss is close to 0W.

[0238] By appropriately selecting the inductance value of inductor L1 and the magnetizing inductance value of the primary coil of transformer T1, the forward energy stored in inductor L1 can be rapidly released during the initial brief period of operating modes 2 and 3. At this time, current flows out from the positive terminal of capacitor C2, through the two ends of the primary coil of transformer T1, the two ends of inductor L1, the drain D of main switching transistor Q2, the source S of main switching transistor Q1, and back to the negative terminal of capacitor C2. At this time, capacitor C2 discharges; the energy stored in inductor L1 increases in the reverse direction (defined here as current flowing from transformer T1 into inductor L1 being reverse current).

[0239] At this time, the output current from the positive terminal of the induced voltage of the secondary coil of transformer T1, after passing through the positive and negative terminals of diode D2, is split into two paths. One path flows through the positive and negative terminals of capacitor C3 back to the negative terminal of the secondary coil voltage of T1. The other path flows through the positive terminal VO of the output DC voltage, the positive and negative terminals of the load, and the negative terminal VO-GND of the output DC voltage back to the negative terminal of the secondary coil voltage of T1. At this time, capacitor C3 is charging; diode D1 is reverse-biased and turned off.

[0240] Working Mode 4:

[0241] like Figure 10 As shown, during mode 4 of the high-current load state 2, the first output drive voltage U(GS-Q1) of the control circuit module 10 remains at a low level, and the second output drive voltage U(GS-Q2) changes from a high level to a low level during the operating mode 3. The main switch transistor Q1 remains off, and the main switch transistor Q2 changes from the on state to the off state. The output drive voltages U(GS-Q3) and U(GS-Q4) of the second control module 112 remain at a high level, and the auxiliary switch transistor Q3 remains on, and the auxiliary switch transistor Q4 remains on.

[0242] At this time, the current flows out from the positive terminal of the capacitor C2, through the two terminals of the transformer T1 primary coil, the two terminals of the inductor L1, through the common connection node of the drain D of the main switch transistor Q2 and the positive terminal of the auxiliary capacitor C5, and then is divided into two branches, one of which flows through the semiconductor channel resistance between the drain D and the source S of the main switch transistor Q2, the drain D and the source S of the main switch transistor Q2, and the DC voltage negative terminal VI-GND, and the other branch flows through the positive terminal of the auxiliary capacitor C5, the negative terminal of the auxiliary capacitor C5, the drain D of the auxiliary switch transistor Q3, the source S of the auxiliary switch transistor Q3, and also flows into the DC voltage negative terminal VI-GND.

[0243] As described above, when the current flows through the common connection node of the drain D of the main switch transistor Q2 and the positive terminal of the auxiliary capacitor C5, it is divided into two branches, one of which flows through the semiconductor channel resistance between the drain D and the source S of the main switch transistor Q2, and flows into the DC voltage negative terminal VI-GND, and the other branch flows into the positive terminal of the auxiliary capacitor C5, through the negative terminal of the auxiliary capacitor C5, the drain D of the auxiliary switch transistor Q4, the source S of the auxiliary switch transistor Q4, and the input DC voltage negative terminal VI-GND. At this time, the auxiliary capacitor C5 is charged, and the voltage across the two terminals of the auxiliary capacitor C5 is charged from the maximum value of approximately 0V in mode 3 to U C5 ≈U Q2-DS-OFF ≈VI. At the same time, the auxiliary capacitor C4 is discharged, and the voltage across the two terminals of the auxiliary capacitor C4 is discharged from the minimum value of U C4 ≈U Q1-DS-OFF ≈VI to approximately 0V.

[0244] As described above, during the charging of the auxiliary capacitor C5, when the voltage U C5 of the auxiliary capacitor C5 is greater than the voltage U C2 across the two terminals of the capacitor C2, the voltage across the two terminals of the inductor L1 will be reversed, and the energy stored in the inductor L1 will be gradually released, and the self-induced electromotive force of the inductor L1 will continue to maintain the direction of the above-mentioned current (the opposite direction current defined above), but the current value will continue to decrease. The current flows out from the positive terminal of the self-induced voltage of the inductor L1, through the common connection node of the source S of the main switch transistor Q1 and the negative terminal of the auxiliary capacitor C4, the positive and negative electrodes of the parasitic reverse body diode between the source S and the drain D of the main switch transistor Q1, the common connection node of the drain D of the main switch transistor Q1 and the positive terminal of the auxiliary capacitor C4, the positive terminal of the capacitor C1, the negative terminal of the capacitor C1, the negative terminal of the capacitor C2, the positive terminal of the capacitor C2, the two terminals of the transformer T1 primary coil, and flows into the negative terminal of the self-induced voltage of the inductor L1.

[0245] At this time, the voltage between the source S and the drain D of the main switch transistor Q1 is about -1V, the main switch transistor Q1 meets the zero-voltage turn-on condition, and the main switch transistor Q1 will realize zero-voltage soft turn-on in the working mode 1 of the next cycle.

[0246] At this time, the energy stored in the inductor L1 is quickly released, the capacitor C2 continues to discharge, and the diode D2 quickly converts from the conducting state to the reverse off state. At this time, the diode D1 is still in the reverse off state, and the transformer T1 secondary coil also quickly stops charging the capacitor C3, and the capacitor C3 quickly converts from the charging state to the discharging state.

[0247] At this time, the current flows out from the positive electrode of the capacitor C3, passes through the output DC voltage positive electrode end VO, the positive and negative ends of the load, and the output DC voltage negative electrode end VO-GND, and flows back to the negative electrode of the capacitor C3.

[0248] As described above, in the resonant LLC switching power supply converter of the application, as can be known from the above working mode analysis, in the mode 2 and the mode 4 in the large current load state 2, the main switch transistors Q1 and Q2 realize minimum current soft turn-off in the turn-off process due to the current bypass effect of the auxiliary capacitors C4 and C5. Therefore, the application further reduces the power loss of the LLC switching power supply converter and further improves the conversion efficiency by using an innovative method.

[0249] In summary, the resonant LLC switching power supply converter of the application meets the economic and social development direction of energy saving, consumption reduction and low carbon, and has a broad application prospect.

[0250] An implementation circuit of the resonant LLC switching power supply converter of the application is as follows: Figure 9 As shown in the example, the parameters of the circuit elements in the example are as follows:

[0251] Resistance: R1=R3=R5=R7=10Ω, R2=R4=R6=R8=10KΩ;

[0252] Main switch transistors Q1 and Q2: N-channel field effect tube SPP20N60C3 (20A, 600V);

[0253] Auxiliary switch transistors Q1 and Q2: N-channel field effect tube SPP02N60C3 (2A, 600V);

[0254] Inductor L1=235uH; Capacitor C2=15nH; Transformer T1 primary coil magnetizing inductance LP1=1mH;

[0255] DC input voltage: U(VI)=380V;

[0256] Steady state DC output voltage: U(VO) = 24V; output power: 230W;

[0257] Capacitance: C1 = 220uF, C3 = 4400uF;

[0258] Auxiliary power supply voltage: U(VCC1, VCC1-GNG) = 12V;

[0259] Switching frequency of the switching converter when output is full load: 83.3kHz.

[0260] Dead time of the first control module 111 output first drive signal waveform U(GS-Q1) and the second drive signal waveform U(GS-Q2): 900nS.

[0261] When the above switching power converter applying the innovative technology of the present application works in full load (belongs to the above-mentioned large current load state 2), the measured as shown in the circuit, the driving voltage waveform of the main switch transistor Q1 and Q2 and the same time axis comparison diagram of the main element current waveform, as shown in the circuit, the driving voltage waveform of the main switch transistor Q1 and Q2 and the same time axis comparison diagram of the main element voltage waveform, as shown in the circuit, the voltage and current waveform of the four working modes identified in the above Figure 9 、 Figure 13 、 Figure 9 、 Figure 14 、 Figure 13 、 Figure 14 , Figure 15 ,

[0262] Figures 13-15 In the above-mentioned Figure 9 , Figure 9 , Figure 9 、 Figure 10As can be seen, when the resonant LLC switching power converter of the present application works in the large current load state 2, the drive voltages U(GS-Q3) and U(GS-Q4) of the auxiliary switching transistors Q3 and Q4 are both maintained at high level, the auxiliary switching transistors Q3 and Q4 are turned on and have very low on-resistance. At this time, the auxiliary capacitor C3 in series with the auxiliary switching transistor Q3 is equivalent to being connected in parallel between the drain D and the source S of the main switching transistor Q1, and the auxiliary capacitor C4 in series with the auxiliary switching transistor Q4 is equivalent to being connected in parallel between the drain D and the source S of the main switching transistor Q2.

[0263] As shown in FIG. 2, when the main switching transistor Q1 is in the on state, the drain D and the source S current I(DS-Q1) of the main switching transistor Q1 is about 500 mA, and the drain D and the source S voltage U(DS-Q1) of the main switching transistor Q1 is about 380 V. Figures 13-15 As shown in FIG. 2, when the main switching transistor Q1 is in the on state, the drain D and the source S current I(DS-Q1) of the main switching transistor Q1 is about 500 mA, and the drain D and the source S voltage U(DS-Q1) of the main switching transistor Q1 is about 380 V.

[0264] As shown in FIG. 2, when the main switching transistor Q1 is in the on state, the drain D and the source S current I(DS-Q1) of the main switching transistor Q1 is about 500 mA, and the drain D and the source S voltage U(DS-Q1) of the main switching transistor Q1 is about 380 V. Figures 13-15 As shown in FIG. 2, when the main switching transistor Q1 is in the on state, the drain D and the source S current I(DS-Q1) of the main switching transistor Q1 is about 500 mA, and the drain D and the source S voltage U(DS-Q1) of the main switching transistor Q1 is about 380 V.

[0265] In the half-bridge LLC switching power converter, the calculation formula of the turn-off loss of the NMOSFET main switching transistor Q1 is as follows:

[0266] Formula 5:

[0267]

[0268] Therefore, it can be known that in the fast turn-off process of the main switching transistor Q1, the resonant LLC switching power converter of the present application realizes the non-delayed conversion of the power converter resonant cavity current I(L1) path, I(L1) is non-delayed switched from the semiconductor on channel between the drain D and the source S of the main switching transistor Q1 (working mode 1) to the auxiliary capacitor C3 branch and the parasitic capacitor branch of the main switching transistor Q1 (working mode 2), so that the main switching transistor Q1 current I(DS-Q1) is non-delayed immediately and greatly reduced in its turn-off process.

[0269] T in Formula 5 ON-OFF-Q1 The parameter is the voltage U between the drain (D) and source (S) of the main switching transistor Q1 during the turn-off process. DS-Q1 With current value I DS-Q1 At the same time, the crossover time period with values ​​greater than zero, such as Figures 13-15 As shown and as mentioned above, it can be seen that T ON-OFF-Q1 ≈0nS. Therefore, according to Formula 5, E ON-OFF-Q1 Since the current is approximately 0J, the innovative resonant LLC switching power converter of this invention makes the turn-off loss of the main switching transistor Q1 very small, realizing the zero-current soft turn-off function of the main switching transistor Q1.

[0270] like Figures 13-15 As shown, during the transition of the main switching transistor Q2 from the on state to the off state in operating mode 4, the current I(DS-Q2) flowing through the drain (D) and source (S) of the main switching transistor Q2 immediately drops from approximately 500mA to less than 100mA. Since the main switching transistor Q2 is completely off at this time, this current of approximately 100mA is the current flowing through the parasitic capacitance between the drain (D) and source (S) of the main switching transistor Q2, and its loss is extremely low and can be ignored. At this time, the current I(C3) flowing through the auxiliary capacitor C3 immediately rises from approximately 50mA to a peak value of approximately 400mA.

[0271] like Figures 13-15 As shown, when the drive voltage U(GS-Q2) of the main switching transistor Q2 has dropped to near 0V, and the main switching transistor Q2 is completely turned off, the drain-source voltage U(DS-Q2) of the main switching transistor Q2 only rises from near 0V to about 380V at a relatively slow slope. Therefore, in operating mode 4, there is no crossover period where the drain-source current waveform I(DS-Q2) and voltage waveform U(GS-Q2) of the main switching transistor Q2 are both significantly greater than 0.

[0272] The hard turn-off loss of the NMOSFET auxiliary switching transistor Q4 in the half-bridge LLC switching power converter is calculated using the following formula (Formula 6):

[0273]

[0274] The symbols in Formula 5-6 have the following meanings: U VS-Q1 Input the instantaneous voltage value between the negative terminal of the DC voltage VI-GND of circuit node VS-Q1 and the reference point 0; U R* T is the instantaneous voltage across resistor R*. ON-OFF-Q* During the transition period when the switching transistor Q* changes from a fully on state to a fully off state, the voltage U between the drain (D) and source (S) of Q* is...DS-Q* the current I DS-Q* the time period when the voltage U W is the working switch frequency of the main switch transistors Q1 and Q2 in the resonant LLC switch power converter; E ON-OFF-Q* is the loss power value generated by the transition period of the switch tube Q* from the fully on state to the fully off state; U DS-Q* is the voltage instantaneous value of the drain D and the source S of the NMOSFET switch tube Q*; I DS-Q* is the current instantaneous value flowing through the semiconductor channel between the drain D and the source S of the NMOSFET switch tube Q*; I DS-MAX-Q* is the current peak value flowing through the semiconductor channel between the drain D and the source S of the NMOSFET switch tube Q*; * is a digital or letter label.

[0275] Therefore, it can be known that the resonant LLC switch power converter of the application realizes the path non-delay conversion of the resonant cavity current I(L1) of the power converter during the fast off process of the main switch transistor Q2, and I(L1) is non-delay switched from the semiconductor conduction channel (working mode 3) between the drain D and the source S of the main switch transistor Q2 to the auxiliary capacitor C4 branch and the parasitic capacitor branch (working mode 4) of the main switch transistor Q2, so that the current I(DS-Q2) of the main switch transistor Q2 immediately and greatly decreases non-delay during the off process of the main switch transistor Q2.

[0276] T ON-OFF-Q2 is the voltage value U DS-Q2 is the current value I DS-Q2 the crossing time period greater than zero, such as Figures 13-15 It can be known that T ON-OFF-Q2 ≈0nS. Therefore, according to formula 6, E ON-OFF-Q2 ≈0J, so that the resonant LLC switch power converter of the application makes the off loss of the main switch transistor Q2 very small, and realizes the soft off function of the main switch transistor Q2.

[0277] As shown in Figure 13 , when the drain D and the source S voltage U(DS-Q1) of the main switch transistor Q1 decreases to close to 0V, its driving voltage U(GS-Q1) rises from close to 0V to about 12V, so that the resonant LLC switch power converter of the application makes the on loss of the main switch transistor Q1 very small, and realizes the zero voltage soft on function of the main switch transistor Q1.

[0278] As shown in Figure 13As shown, when the voltage U(DS-Q2) between the drain D and source S of the main switching transistor Q2 drops to near 0V, its second driving voltage U(GS-Q2) only rises from near 0V to about 12V. Therefore, the innovative resonant LLC switching power converter of this invention makes the turn-on loss of the main switching transistor Q2 very small, realizing the zero-voltage soft-turn-on function of the main switching transistor Q2.

[0279] In summary, the innovative resonant LLC switching power converter of this invention achieves zero-voltage soft-turn-on and zero-current soft-turn-off functions for the main switching transistors Q1 and Q2, resulting in extremely low switching losses and further improving the efficiency of the power converter. Therefore, this invention meets the growing demands of today's socio-economic society for high efficiency, energy saving, and carbon reduction, and has the prospect of large-scale application, yielding broad socio-economic benefits.

[0280] like Figures 1-15 As shown, the half-bridge switching circuit in the innovative resonant LLC switching power converter of this invention is innovatively composed of two main switching transistors Q1 and Q2, two auxiliary switching transistors Q3 and Q4, and two auxiliary capacitors C4 and C5, and its innovative connection method.

[0281] In the high-current load state 2, the innovative resonant LLC switching power converter of this invention achieves a zero-delay switching of the resonant cavity current I(L1) path during the rapid turn-off process of the main switching transistor Q1 in the operating mode 2 of the high-current load state 2. I(L1) switches without delay from the semiconductor conduction channel between the drain D and source S of Q1 (operating mode 1) to the branch flowing into the auxiliary capacitor C3 and the parasitic capacitance branch of the main switching transistor Q1 (operating mode 2), thereby causing the current I(DS-Q1) of the main switching transistor Q1 to drop significantly without delay during its turn-off process.

[0282] T in Formula 5 ON-OFF-Q1 The parameter is the voltage U between the drain (D) and source (S) of the main switching transistor Q1 during the turn-off process. DS-Q1 With current value I DS-Q1 As mentioned above, the crossover time period with a value greater than zero in this innovative resonant LLC switching power converter is T... ON-OFF-Q1 ≈0nS, therefore according to formula 5, E ON-OFF-Q1 Since the current is approximately 0J, the innovative resonant LLC switching power converter of this invention makes the turn-off loss of its main switching transistor Q1 extremely small, thus realizing the zero-current soft turn-off function of the main switching transistor Q1.

[0283] In the working mode 4 period of the large current load state 2 of the resonant LLC switching power converter of the application, the path of the resonant cavity current I(L1) of the power converter is converted without delay in the fast turn-off process of the main switching transistor Q2, I(L1) is switched without delay from the semiconductor conduction channel between the drain D and the source S of the main switching transistor Q2 (working mode 3) to the auxiliary capacitor C4 branch and the parasitic capacitor branch of the main switching transistor Q2 (working mode 4), so that the current I(DS-Q2) of the main switching transistor Q2 immediately decreases without delay in its turn-off process.

[0284] T in formula 6 ON-OFF-Q2 The parameter is the voltage value U between the drain D and the source S of the main switching transistor Q2 in its turn-off process DS-Q2 The current value I DS-Q2 The cross period is greater than zero, as described above, it can be known that T in the resonant LLC switching power converter of the application ON-OFF-Q2 ≈0nS, therefore, according to formula 6, E ON-OFF-Q2 ≈0J. Therefore, the resonant LLC switching power converter of the application makes the turn-off loss of the main switching transistor Q2 very small, and realizes the zero current soft turn-off function of the main switching transistor Q2.

[0285] As shown in Figure 14 , when the voltage U(DS-Q1) between the drain D and the source S of the main switching transistor Q1 drops to close to 0V, its drive voltage U(GS-Q1) rises from close to 0V to about 12V, therefore, the resonant LLC switching power converter of the application makes the turn-on loss of Q1 very small, and realizes the zero voltage soft turn-on function of the main switching transistor Q1.

[0286] As shown in Figure 14 , when the voltage U(DS-Q2) between the drain D and the source S of the main switching transistor Q2 drops to close to 0V, its second drive voltage U(GS-Q2) rises from close to 0V to about 12V, therefore, the resonant LLC switching power converter of the application makes the turn-on loss of the main switching transistor Q2 very small, and realizes the zero voltage soft turn-on function of the main switching transistor Q2.

[0287] As described above, compared with Figure 1 the large current hard turn-off state of the switching module in the prior art shown in the figure, the resonant LLC switching power converter of the application realizes the zero voltage soft turn-on and zero current soft turn-off functions of the main switching transistors Q1 and Q2 in the large current load state 2, so that the switching losses of them are very small, and the efficiency of the power converter is further improved compared with the prior art.

[0288] As shown in Figure 10As shown, in the small current load state 1, the output drive signals U(GS-Q3) and U(GS-Q4) of the second control module 112 are both low, and the auxiliary switching transistors Q3 and Q4 are maintained in the off state, at which time the auxiliary capacitors C4 and C5 are disconnected from the half-bridge arm main switching transistors Q1 and Q2.

[0289] A prior art has the following problems: because a prior art directly connects a capacitor in parallel between the drain D and the source S of a bridge arm switching tube, when the driving pulse is a discontinuous pulse group output, the switching tube is in a hard-on state at the first pulse after the driving pulse is interrupted, at which time the increased auxiliary capacitor of the switching tube increases switching loss. Because in general, when the resonant LLC switching power supply converter is in an idle standby and a small load state, the control module adjusts the operating frequency to increase to the set maximum safe frequency value in order to stabilize the output voltage, and still cannot achieve the stable voltage requirement. At this time, the control module will enter the driving pulse group output state without interruption in order to stabilize the output voltage, and therefore the above-mentioned prior art method will increase the loss and reduce the efficiency of the LLC switching power supply converter in the standby and small load state.

[0290] As described above, the resonant LLC switching power supply converter of the present application does not have the problem of increased switching tube on loss of the prior art. Therefore, the present application not only realizes the zero-current off and zero-voltage on functions of the main switching transistor Q1 or Q2 in the large current load state 2, but also meets the requirements of not increasing the on power consumption of the main switching transistor Q1 or Q2 in the small current load state 1, and realizes low standby power consumption and small load high efficiency.

[0291] In summary, the resonant LLC switching power supply converter of the present application can further reduce loss and improve efficiency in the full load range and full operating state, and therefore can fully meet the higher requirements of energy saving, carbon reduction, low standby power consumption, and small load high efficiency in the social and economic development trend, and is conducive to the large-scale popularization and application of this technology in more emerging application scenarios and higher requirements.

[0292] As Figure 10 As shown, in the resonant LLC switching power supply converter of the present application, the input and output control logic of the second control module 112 is defined as follows: two necessary conditions are required for the output drive signal U(GS-Q3) to be converted from low to high or from high to low.

[0293] As Figure 10As shown, when the power converter enters the large current load state 2 from the small current load state 1, the auxiliary switch transistors Q3 and Q4 cannot be turned on immediately, at this time the second control module 112 also needs to judge the state of the main switch transistors Q1 and Q2, and waits for the main switch transistors Q1 and Q2 to enter the stable on state, and then triggers the auxiliary switch transistors Q3 and Q4 to be turned on immediately. When the auxiliary switch transistors Q3 and Q4 are triggered to be turned on, they will be locked in the on state. At this time, only when the conditions for triggering the auxiliary switch transistors Q3 and Q4 to be turned off are met, the auxiliary switch transistors Q3 and Q4 can be turned off. Therefore, the auxiliary switch transistors Q3 and Q4 both realize the high-efficiency zero-voltage soft on function, and the turning on of the auxiliary switch transistors Q3 and Q4 does not exist any interference phenomenon to the main circuit.

[0294] As shown, Figure 10 When the LLC power converter enters the small current load state 1 from the large current load state 2, the auxiliary switch transistors Q3 and Q4 cannot be turned off immediately, at this time the second control module 112 also needs to judge the state of the main switch transistors Q1 and Q2, and waits for the main switch transistors Q1 and Q2 to enter the stable on state, and then triggers the auxiliary switch transistors Q3 and Q4 to be turned off immediately. When the auxiliary switch transistors Q3 and Q4 are triggered to be turned off, they will be locked in the off state. At this time, only when the conditions for triggering the auxiliary switch transistors Q3 and Q4 to be turned on are met, the auxiliary switch transistors Q3 and Q4 can be turned on. Therefore, the auxiliary switch transistors Q3 and Q4 both realize the high-efficiency zero-voltage soft off function, and the turning off of the auxiliary switch transistors Q3 and Q4 does not exist any interference phenomenon to the main circuit.

[0295] Compared with the prior art as shown in Figure 1 The innovative resonant LLC switching power converter of the present application not only realizes the high-efficiency zero-voltage soft on and zero-current soft off functions of the main switch transistors Q1 and Q2 in the large current load state 2, but also realizes the high-efficiency zero-voltage on and zero-voltage off functions of the auxiliary switch transistors Q3 and Q4 in all working states.

[0296] The innovative second control module 112 and its circuit composition and connection mode in the innovative resonant LLC switching power converter of the present application are all independent additional circuits without changing the existing control circuit. The first control module 111 applied in the present application is compatible with the control module of the prior art, so the total research and development and updating cost of the power supply system can be reduced. Therefore, the existing resonant LLC switching power converter with low efficiency can be upgraded to the innovative resonant LLC switching power converter of the present application at extremely low cost, and the present application can realize high efficiency and energy saving in any working current state.

[0297] The half-bridge arm switching circuit of the present application is innovatively composed of two main switching transistors Q1 and Q2 and two auxiliary switching transistors Q3 and Q4, and two auxiliary capacitors C4 and C5, and the connection mode is also innovative.

[0298] In summary, the innovative resonant LLC switching power supply converter of the present application meets the increasing demand for large load high efficiency and small load high efficiency, energy saving, carbon reduction and low standby power consumption in today's society and economy, and is versatile, low in update cost, high in reliability and meets the technical needs of a large number of traditional and emerging application fields, so the present application has a large-scale application prospect and considerable social and economic benefits.

[0299] The above invention points of the innovative resonant LLC switching power supply converter of the present application can also be applied to an LLC switching power supply with a full-bridge topology, and the innovative resonant full-bridge LLC switching power supply converter of the present application is composed of the half-bridge arm switching circuit innovated by the above invention points.

[0300] The above invention points of the innovative resonant LLC switching power supply converter of the present application can also be applied to a full-bridge switching power supply with a phase-shift control mode, and the innovative phase-shift full-bridge switching power supply converter of the present application is composed of two groups of the half-bridge arm switching circuit innovated by the above invention points.

[0301] The innovative resonant LLC switching power supply converter, resonant full-bridge LLC switching power supply converter and phase-shift full-bridge switching power supply converter of the present application can be applied to a high-frequency switching combination power supply with any appearance and structure.

[0302] The innovative resonant LLC switching power supply converter, resonant full-bridge LLC switching power supply converter and phase-shift full-bridge switching power supply converter of the present application can be realized in any printed circuit board layout connection mode.

[0303] The components in the innovative resonant LLC switching power supply converter, resonant full-bridge LLC switching power supply converter and phase-shift full-bridge switching power supply converter of the present application can be any encapsulated components with the same principle and performance and their series and parallel combinations.

[0304] Embodiment Two

[0305] The embodiment two provides a control method of the resonant LLC switching power supply converter based on the embodiment one, as shown in the figure, the control method is applied to a control module and comprises the following steps. Figure 16 As shown in the figure, the control method is applied to a control module and comprises the following steps.

[0306] Step S10. Outputting a first control signal to the first switch module to control the on-off state of the first switch module, and outputting a second control signal to the second switch module to control the on-off state of the second switch module.

[0307] Step S20. Obtaining the output current of the resonant LLC switching power supply converter, comparing the output current with the preset current, and controlling the on-off state of the third switch module according to the comparison result and the state of the first control signal, and controlling the on-off state of the fourth switch module according to the comparison result and the state of the second control signal.

[0308] Further, the step S20 of obtaining the output current of the resonant LLC switching power supply converter comprises:

[0309] The output current of the resonant LLC switching power supply converter is obtained by detecting the effective value of the alternating component of the voltage across the second capacitor module.

[0310] Further, as shown in Figure 17 The step S20 of controlling the on-off state of the third switch module according to the comparison result and the state of the first control signal comprises:

[0311] Step S201. When the output current is greater than the preset current, and the first control signal is switched from the first level signal to the second level signal, delay control the third switch module from the first switch state to the second switch state, and maintain the second switch state.

[0312] Step S202. When the output current is not greater than the preset current, and the first control signal is switched from the first level signal to the second level signal, delay control the third switch module from the second switch state to the first switch state, and maintain the first switch state.

[0313] In step S201, when the output current is greater than the preset current, it is determined that the load is in a large current state, and at this time the third switch module needs to be controlled to adapt to the working requirement of high current. When the first control signal is switched from the first level signal to the second level signal, it indicates that the control needs to be switched, and in this case, the third switch module is switched from the first switch state (e.g. off state) to the second switch state (e.g. on state) through delay control. Once the third switch module enters the second switch state, it will maintain its on state to ensure that the current can flow stably and reach the predetermined working mode of the system to meet the demand under high load current.

[0314] In step S202, when the output current is not greater than the preset current, it is determined that the load is in a small current state, and at this time the third switch module does not need to maintain the on state to avoid unnecessary power loss.

[0315] When the first control signal is switched from the first level signal to the second level signal, the control system still needs to be adjusted in time. In this case, through time delay control, the third switch module will be switched from the second switch state (on state) back to the first switch state (off state), ensuring that the current flow is cut off, thereby reducing power loss under low load. When the third switch module is maintained in the first switch state, it ensures that the system is in a low-power consumption operation mode, which meets the high-efficiency working condition under small load.

[0316] The technical effect of the embodiment is that through the control strategies of steps S201 and S202, the third switch module can dynamically adjust its on-off state according to the size of the load current, realizing efficient power consumption management and system optimization. Under high-current load, through time delay control, it is switched to the on state and maintained in the on state, ensuring stable current flow and meeting the high-load working requirement, while realizing zero-voltage conduction and reducing switching loss. Under low-current load, through time delay control, it is switched to the off state, reducing power consumption and ensuring efficient operation of the system under small load condition. The overall design effectively improves the operating efficiency of the power converter, optimizes the power conversion performance, and significantly reduces the switching loss.

[0317] Further, as shown in Figure 18 controlling the on-off state of the fourth switch module according to the current comparison result and the state of the second control signal includes:

[0318] Step S203. When the output current is greater than the preset current, and the second control signal is switched from the first level signal to the second level signal, the fourth switch module is controlled by time delay to switch from the first switch state to the second switch state and maintain the second switch state.

[0319] Step S204. When the output current is not greater than the preset current, and the second control signal is switched from the first level signal to the second level signal, the fourth switch module is controlled by time delay to switch from the second switch state to the first switch state and maintain the first switch state.

[0320] The control processes of steps S203 and S204 are similar to those of steps S201 and S202, and will not be described here.

[0321] The above embodiments are only used to illustrate the technical solutions of the present application, but not to limit it; although the present application has been described in detail with reference to the foregoing embodiments, those skilled in the art should understand that they can still modify the technical solutions recorded in the foregoing embodiments, or make equivalent replacement for part of the technical features; and these modifications or replacements do not make the essence of the corresponding technical solutions deviate from the spirit and scope of the technical solutions of the embodiments of the present application, and should be included in the protection scope of the present application.

Claims

1. A resonant LLC switching power supply converter, characterized by, The application relates to a resonant LLC switching power converter. The resonant LLC switching power converter comprises a first switching module, a second switching module, a third switching module, a fourth switching module, a first capacitor module, a second capacitor module, a third capacitor module, an energy storage and isolation module and a control module. The first end of the first switching module is connected to the first end of the second capacitor module, the second end of the first switching module is connected to the first end of the second switching module, the second end of the third switching module, the first end of the third capacitor module and the first end of the energy storage and isolation module respectively, the second end of the second capacitor module is connected to the first end of the third switching module, the second end of the second switching module is connected to the second end of the fourth switching module and the first end of the first capacitor module respectively, the second end of the first capacitor module is connected to the second end of the energy storage and isolation module, the second end of the third capacitor module is connected to the first end of the fourth switching module, and the control module is connected to the control end and the second end of the first switching module, the control end and the second end of the second switching module, the control end and the second end of the third switching module and the control end and the second end of the fourth switching module respectively. The control module is used for outputting a first control signal to the first switching module to control the on-off state of the first switching module, and outputting a second control signal to the second switching module to control the on-off state of the second switching module. The control module is also used for acquiring the output current of the resonant LLC switching power converter, comparing the output current with a preset current, and controlling the on-off state of the third switching module according to the comparison result and the state of the first control signal, and controlling the on-off state of the fourth switching module according to the comparison result and the state of the second control signal.

2. The resonant LLC switching power supply converter of claim 1, wherein, The control module comprises: A first control module connected to the control end of the first switching module and the control end of the second switching module respectively, used for outputting a first control signal to the first switching module to control the on-off state of the first switching module, and outputting a second control signal to the second switching module to control the on-off state of the second switching module. A second control module connected to the control end of the first switching module, the control end of the second switching module, the control end of the third switching module and the control end of the fourth switching module respectively, used for acquiring the output current of the resonant LLC switching power converter, comparing the output current with a preset current, and controlling the on-off state of the third switching module according to the comparison result and the state of the first control signal, and controlling the on-off state of the fourth switching module according to the comparison result and the state of the second control signal.

3. The resonant LLC switching power supply converter of claim 2, wherein, The second control module comprises: A reference voltage module used for outputting a reference voltage. a current detection module, a first input end of which is connected with a second end of the first capacitor module, a second input end of which is connected with an output end of the reference voltage module, for collecting an output current of the resonant LLC switching power supply converter, converting the output current into a collection voltage, and comparing the collection voltage with the reference voltage to output a first comparison signal; a first control sub-module, a first input end of which receives the first control signal, a second input end of which is connected with an output end of the reference voltage module, a third input end of which is connected with an output end of the current detection module, and an output end of which is connected with a control end of the third switch module, for outputting a first on-off control signal to the third switch module according to the first control signal, the reference voltage and the first comparison signal; a second control sub-module, a first input end of which receives the second control signal, a second input end of which is connected with an output end of the reference voltage module, a third input end of which is connected with an output end of the current detection module, and an output end of which is connected with a control end of the fourth switch module, for outputting a second on-off control signal to the fourth switch module according to the second control signal, the reference voltage and the first comparison signal.

4. The resonant LLC switching power supply converter of claim 3, wherein, the first control sub-module comprises: a first delay trigger module, a first input end of which receives the first control signal, a second input end of which receives the reference voltage, and a third input end of which receives the first comparison signal, for outputting a first weak current on-off control signal according to the first control signal, the reference voltage and the first comparison signal; a first drive module, an input end of which is connected with an output end of the first delay trigger module, for outputting a first on-off control signal after amplifying and isolating the first weak current on-off control signal.

5. The resonant LLC switching power supply converter of claim 4, wherein, the first delay trigger module comprises: a first comparison module, a first input end of which receives the first control signal, and a second input end of which receives the reference voltage, for converting the first control signal into a first collection voltage after time delay, and outputting a first trigger signal when the first collection voltage is greater than the reference voltage; a first trigger module, a first input end of which receives the first comparison signal, and a second input end of which is connected with an output end of the first comparison module, for outputting the first weak current on-off control signal according to the first trigger signal.

6. The resonant LLC switching power supply converter of claim 3, wherein, the second control sub-module comprises: a second delay trigger module, a first input end of which receives the second control signal, a second input end of which receives the reference voltage, and a third input end of which receives the first comparison signal, for outputting a second weak current on-off control signal according to the second control signal, the reference voltage and the first comparison signal; a second drive module, an input end of which is connected with an output end of the second delay trigger module, for outputting a second on-off control signal after amplifying and isolating the output second weak current on-off control signal.

7. The resonant LLC switching power supply converter of claim 6, wherein, the second delay trigger module comprises: a second comparison module, a first input end of which receives the second control signal, a second input end of which receives the reference voltage, and which is configured to convert the second control signal into a second sampling voltage through time delay and output a second trigger signal when the second sampling voltage is greater than the reference voltage; a second trigger module, a first input end of which receives the first comparison signal, a second input end of which is connected to an output end of the second comparison module, and which is configured to output the second weak current on-off control signal according to the second trigger signal.

8. The resonant LLC switching power supply converter of claim 1, wherein, The first switch module is a first MOS tube, the second switch module is a second MOS tube, the first control signal includes a first voltage control signal and a second voltage control signal, and the second control signal includes a third voltage control signal and a fourth voltage control signal. The resonant LLC switching power supply converter further includes: a first voltage dividing unit, a first end of which receives the first voltage control signal, a second end of which is connected to a gate of the first MOS tube, and a third end of which receives the second voltage control signal after being connected to a source of the first MOS tube, and which is configured to control on-off of the first MOS tube according to the first voltage control signal and the second voltage control signal; a second voltage dividing unit, a first end of which receives the third voltage control signal, a second end of which is connected to a gate of the second MOS tube, and a third end of which receives the fourth voltage control signal after being connected to a source of the second MOS tube, and which is configured to control on-off of the second MOS tube according to the third voltage control signal and the fourth voltage control signal; The control module is configured to control the third switch module to start conduction when it is detected that the output current is greater than the preset current, and a high-level signal is formed between the first voltage control signal and the second voltage control signal or between the second voltage control signal and a ground end of the control module. The control module is further configured to control the fourth switch module to start conduction when the output current is greater than the preset current and a high-level signal is formed between the third voltage control signal and the fourth voltage control signal.

9. A control method of a resonant LLC switching power converter based on the resonant LLC switching power converter of claim 1, characterized in that, The control method is applied to the control module and includes: outputting a first control signal to the first switch module to control on-off state of the first switch module, and outputting a second control signal to the second switch module to control on-off state of the second switch module; obtaining an output current of the resonant LLC switching power supply converter, comparing the output current with a preset current, and controlling on-off state of the third switch module according to a comparison result and a state of the first control signal, and controlling on-off state of the fourth switch module according to the comparison result and a state of the second control signal.

10. The control method according to claim 9, characterized by, The obtaining of the output current of the resonant LLC switching power supply converter includes: obtaining the output current of the resonant LLC switching power supply converter by detecting an effective value of an alternating current component of a voltage across the second capacitor module.

11. The control method according to claim 9, characterized by, The control of the on-off state of the third switch module according to the comparison result and the state of the first control signal includes: When the output current is greater than the preset current, and the first control signal is switched from the first level signal to the second level signal, the third switch module is switched from the first switch state to the second switch state in a time-delay manner and maintained in the second switch state. When the output current is not greater than the preset current, and the first control signal is switched from the first level signal to the second level signal, the third switch module is switched from the second switch state to the first switch state in a time-delay manner and maintained in the first switch state.

12. The control method according to claim 9, characterized by, The control of the on-off state of the fourth switch module according to the comparison result and the state of the second control signal comprises: When the output current is greater than the preset current, and the second control signal is switched from the first level signal to the second level signal, the fourth switch module is switched from the first switch state to the second switch state in a time-delay manner and maintained in the second switch state. When the output current is not greater than the preset current, and the second control signal is switched from the first level signal to the second level signal, the fourth switch module is switched from the second switch state to the first switch state in a time-delay manner and maintained in the first switch state.

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