Control device for AC motor and electric power steering device

Abstract

A control device for an AC motor includes: a current detector; a first coordinate converter that converts a pre-conversion detection current, which is a detection result based on the current detector, into a two-axis detection current, which is a current on the two rotating axes; a two-axis current filter that reduces a noise component of the two-axis detection current; the second coordinate converter is used for converting the filtered two-axis current into a filtered three-phase current of a static coordinate system; the first controller is used for generating a first three-phase voltage instruction based on the two-axis detection current or the filtered two-axis current; a second controller that generates a second three-phase voltage command based on the filtered three-phase current; and an inverter that applies a voltage to the AC motor on the basis of the first three-phase voltage command and the second three-phase voltage command.

Description

Technical Field

[0001] The present disclosure relates to a control device for an alternating current motor and an electric power steering device. Background Art

[0002] In the current control of an alternating current motor, a vector control method that performs voltage command calculation on a rotating two-axis (for example, d-axis and q-axis) has been used in the past. As a specific example of the disclosure related to the vector control method, Patent Documents 1 and 2 below can be cited. In Figure 1 and Figure 4 of Patent Document 1, etc., the second voltage command calculation unit is used to generate a second voltage command without performing coordinate conversion on the detected current (iu, iv, iw) on the stationary coordinates. In Patent Document 2, the current flowing through the three-phase winding is detected, and the detected current is converted into values on the rotating two-axis (d-axis and q-axis). Further, it is disclosed that by adding the latest detected value on the dq-axis to the past detected value, the noise component of the resonance period included in the detected value is reduced. Prior Art Documents Patent Documents

[0003] Patent Document 1: Japanese Patent No. 5178768 Gazette Patent Document 2: International Publication No. 2021 / 144867 Summary of the Invention Technical Problem to be Solved by the Invention

[0004] When the alternating current motor rotates at high speed, the frequency of the detected current (iu, iv, iw) on the stationary coordinates as shown in Patent Document 1 also increases. Here, when the frequency of the detected current on the stationary coordinates is high, if a filter is used to remove the noise component included in the detected current, a problem of phase delay after the filtering process will occur. The phase delay is the main cause of reducing the control accuracy of the alternating current motor. In addition, for example, the method of reducing the noise component disclosed in Patent Document 2 is a process on the dq-axis, so it is difficult to combine with the structure of Patent Document 1.

[0005] The present disclosure is proposed to solve the above problems, and its purpose is to provide a control device for an alternating current motor and an electric power steering device that can suppress a decrease in the control accuracy of the alternating current motor during high-speed rotation and can reduce the noise component included in the detected current. Technical Solution for Solving the Technical Problem

[0006] The control device for an alternating current motor according to the present disclosure includes: a current detector that detects the currents of three phases to which power is supplied to the alternating current motor; a first coordinate converter that converts a pre-conversion detected current based on the detection result of the current detector into a two-axis detected current that is a current on a rotating two-axis; a two-axis current filter that reduces the noise component of the two-axis detected current; a second coordinate converter that converts the two-axis detected current, from which the noise component has been reduced by the two-axis current filter, i.e., the filtered two-axis current, into a filtered three-phase current in a stationary coordinate system; a first controller that generates a first three-phase voltage command based on the two-axis detected current or the filtered two-axis current; a second controller that generates a second three-phase voltage command based on the filtered three-phase current; and an inverter that applies a voltage to the alternating current motor based on the first three-phase voltage command and the second three-phase voltage command.

[0007] In addition, the electric power steering device according to the present disclosure includes the above-described control device for an alternating current motor and the alternating current motor that generates an assist torque during vehicle steering. Advantageous Effects of the Invention

[0008] According to the present disclosure, it is possible to provide a control device for an alternating current motor and an electric power steering device that can suppress a decrease in the control accuracy of the alternating current motor during high-speed rotation and can reduce the noise component included in the detected current. BRIEF DESCRIPTION OF THE DRAWINGS

[0009] Figure 1 is a diagram showing the configuration of the control device for an alternating current motor according to Embodiment 1. Figure 2 is a diagram for explaining the principle of generating a switching signal in Embodiment 1. Figure 3 is showing Figure 1 a structural example of the two-axis current filter. Figure 4 is showing Figure 1 a structural example of the d-axis current controller. Figure 5 is showing Figure 1 a structural example of the q-axis current controller. Figure 6 is showing Figure 1 a structural example of the second controller. Figure 7 is showing Figure 1 a first example of the operation of the selector shown. Figure 8 is showing Figure 1Diagram of the second example of the operation of the selector shown Figure 9 It represents Figure 1 Diagram of the third example of the operation of the selector shown Figure 10 It is a diagram showing an example of the sound - frequency characteristic of an AC motor Figure 11 It is a diagram showing the result of verifying the magnitude of the phase delay accompanying the filtering process Figure 12 It is a diagram showing the structure of the control device for an AC motor according to Embodiment 2 Figure 13 It represents Figure 12 Diagram of a structural example of the second controller Figure 14 It is a diagram showing the structure of the control device for an AC motor according to Embodiment 3 Figure 15 It is a graph showing an example of the waveform of the voltage across the shunt resistor Figure 16 It is a diagram showing the structure of the control device for an AC motor according to Embodiment 4 Figure 17 It is a diagram for explaining the operation conduction / cutoff signal and the weighted average gain in Embodiment 4 Figure 18 It is a diagram showing the structure of the electric power steering device according to Embodiment 5 Detailed implementation mode

[0010] Embodiment 1. Figure 1 It is a block diagram showing the schematic structure of the control device for an AC motor according to Embodiment 1 (hereinafter referred to as control device 1). As Figure 1 shown, the control device 1 includes a rotor position detector 11, an inverter 12, and a control unit 13. The control device 1 controls the AC motor 10 based on the current command values id_ref and iq_ref input from the outside of the control device 1. The current command values id_ref and iq_ref will be described later

[0011] The AC motor 10 is a three-phase AC motor having three-phase windings U, V, and W. In this specification, the coordinate system corresponding to the three-phase windings U, V, and W is referred to as a stationary coordinate system or a uvw coordinate system. Further, the AC motor 10 is an AC motor capable of being controlled based on two rotating axes. In this specification, the "two rotating axes" refer to two axes that rotate synchronously with the rotor of the AC motor 10 and are orthogonal to each other in a cross-sectional plane. The "cross-sectional plane" refers to a plane perpendicular to the central axis of the rotor. For example, the two rotating axes may be d-q axes. The d-axis is the axis connecting the central axis of the rotor and the magnetic pole. The q-axis is the axis orthogonal to both the d-axis and the above-mentioned central axis. Further, the two rotating axes may be γ-δ axes. The γ-axis is the axis offset in the rotational direction with respect to the d-axis. The δ-axis is the axis orthogonal to both the γ-axis and the above-mentioned central axis. One of the two rotating axes is referred to as the first axis, and the other is referred to as the second axis. For example, when the d-axis is referred to as the first axis, the q-axis is referred to as the second axis. Alternatively, the q-axis may be the first axis and the d-axis may be the second axis. Similarly, when the γ-axis is referred to as the first axis, the δ-axis is referred to as the second axis.

[0012] The case where the AC motor 10 is a permanent magnet synchronous AC motor and the two rotating axes are d-q axes will be described below. However, the AC motor 10 may be, for example, a wound field synchronous AC motor, an induction AC motor, a synchronous reluctance motor, etc. Further, the d-axis and the q-axis in the following disclosure may be replaced with the δ-axis and the γ-axis.

[0013] The rotor position detector 11 includes a resolver, an encoder, an MR (magnetoresistive) sensor, etc., and detects the rotor position θ using them. The rotor position θ is the position of the rotor of the AC motor 10 in the rotational direction. In the present embodiment, the rotor position detector 11 is used to detect the rotor position θ of the AC motor 10. However, a configuration may be adopted in which the rotor position θ of the AC motor 10 is estimated without using the rotor position detector 11. That is, in the present disclosure, the control device 1 may not include the rotor position detector 11.

[0014] The inverter 12 is a power converter that applies a voltage to the AC motor 10. Specifically, under the control of the control unit 13, the inverter 12 converts the direct current supplied from the DC power supply BT into an alternating current and supplies the converted alternating current to the AC motor 10. The DC power supply BT includes, in addition to a battery, devices for supplying direct current such as a DC-DC converter, a diode rectifier, and a PWM rectifier. In this specification, the output voltage (DC bus voltage) of the DC power supply BT is represented as Vdc.

[0015] The inverter 12 includes upper-arm switching elements Sup, Svp, Swp, lower-arm switching elements Sun, Svn, Swn, and shunt resistors Ru, Rv, Rw. The upper-arm switching elements Sup, Svp, Swp are connected to the positive electrode of the DC power supply BT. The lower-arm switching elements Sun, Svn, Swn are respectively connected to the upper-arm switching elements Sup, Svp, Swp, and are respectively connected to the negative electrode of the DC power supply BT via the shunt resistors Ru, Rv, Rw.

[0016] Here, the series circuit of phase U (power line 12u) is formed by the upper-arm switching element Sup, the lower-arm switching element Sun, and the shunt resistor Ru. In the series circuit of phase U, the connection point between the upper-arm switching element Sup and the lower-arm switching element Sun is connected to the winding U of the AC motor 10.

[0017] In addition, the series circuit of phase V (power line 12v) is formed by the upper-arm switching element Svp, the lower-arm switching element Svn, and the shunt resistor Rv. In the series circuit of phase V, the connection point between the upper-arm switching element Svp and the lower-arm switching element Svn is connected to the winding V of the AC motor 10.

[0018] In addition, the series circuit of phase W (power line 12w) is formed by the upper-arm switching element Swp, the lower-arm switching element Swn, and the shunt resistor Rw. In the series circuit of phase W, the connection point between the upper-arm switching element Swp and the lower-arm switching element Swn is connected to the winding W of the AC motor 10.

[0019] As the upper-arm switching elements Sup, Svp, Swp and the lower-arm switching elements Sun, Svn, Swn, semiconductor switches such as IGBT (Insulated Gate Bipolar Transistor), bipolar transistors, and MOSFET (Metal-Oxide-Semiconductor Field Effect Transistor) can be used, for example.

[0020] The switching signals Gup, Gvp, and Gwp output from the control unit 13 are respectively input to the upper-arm switching elements Sup, Svp, and Swp. The switching signals Gun, Gvn, and Gwn output from the control unit 13 are respectively input to the lower-arm switching elements Sun, Svn, and Swn. The upper-arm switching elements Sup, Svp, and Swp and the lower-arm switching elements Sun, Svn, and Swn are turned on or off by the switching signals Gup, Gvp, Gwp, Gun, Gvn, and Gwn output from the control unit 13. In this specification and the drawings, the switching signals Gup, Gvp, Gwp, Gun, Gvn, and Gwn are sometimes collectively referred to and labeled as "switching signals Gup to Gwn".

[0021] For example, when the switching signal Gup is "turn-on command (=1)", the upper-arm switching element Sup is turned on, and when the switching signal Gup is "turn-off command (=0)", the upper-arm switching element Sup is turned off. The same applies to the other switching elements (the upper-arm switching elements Svp, Swp and the lower-arm switching elements Sun, Svn, Swn). Thereby, the inverter 12 generates an alternating current for supplying to the alternating-current motor 10 from the direct current supplied from the direct-current power supply BT.

[0022] The shunt resistors Ru, Rv, and Rw are resistance elements for current detection. The shunt resistor Ru outputs a terminal voltage VRu (= -Ru × iu) proportional to the current (alternating-current motor current) iu flowing through the winding U of the alternating-current motor 10 to the control unit 13. The shunt resistor Rv outputs a terminal voltage VRv (= -Rv × iv) proportional to the current (alternating-current motor current) iv flowing through the winding V of the alternating-current motor 10 to the control unit 13. The shunt resistor Rw outputs a terminal voltage VRw (= -Rw × iw) proportional to the current (alternating-current motor current) iw flowing through the winding W of the alternating-current motor 10 to the control unit 13. In this specification and the drawings, the terminal voltages VRu, VRv, and VRw are sometimes collectively referred to and labeled as "terminal voltages VRu to VRw".

[0023] Here, the terminal voltages VRu, VRv, and VRw are values obtained by multiplying the alternating-current motor currents iu, iv, and iw by the resistance values of the shunt resistors Ru, Rv, and Rw, and are quantities proportional to the currents iu, iv, and iw. Therefore, it can be said that the terminal voltages VRu, VRv, and VRw are values for detecting the current (detection values of the alternating-current motor current). The inverter 12 in the present embodiment has current-carrying lines 12u, 12v, and 12w for supplying current to the alternating-current motor 10, and shunt resistors Ru, Rv, and Rw connected in series with the current-carrying lines 12u, 12v, and 12w. That is, the inverter 12 is a so-called shunt-resistor type inverter. In addition, the inverter 12 can be integrated with the AC motor 10. The integrated inverter 12 and AC motor 10 are referred to as a power unit.

[0024] The control unit 13 uses the current command values id_ref, iq_ref, the terminal voltages VRu, VRv, VRw, and the rotor position θ as input values. Based on these input values, the control unit 13 generates the switching signals Gup to Gwn for driving the inverter 12. The control unit 13 is, for example, a PWM controller implemented by a discrete-time arithmetic unit such as a microcomputer or a DSP (Digital Signal Processor). The control unit 13 includes a two-axis current filter 21, a current detector 22, a first coordinate converter 23, a first controller 24, a selector 26, a PWM signal generator 27, a second coordinate converter 28, a third coordinate converter 29, and a second controller 15.

[0025] The current command values id_ref, iq_ref are command values (target values) of the current supplied to the AC motor 10, which are input to the control device 1 from the outside. The current command value id_ref is also referred to as the "field weakening current command value", and the current command value iq_ref is also referred to as the "torque current command value". The current command values id_ref, iq_ref can be calculated by appropriately combining known MTPA (Maximum Torque Per Ampere) control, MTPV (Maximum Torque Per Voltage) control, and field weakening control for each operating range (range of speed-torque characteristics).

[0026] Next, the PWM signal generator 27 will be described. The PWM signal generator 27 outputs the switching signals Gup to Gwn modulated by PWM (Pulse Width Modulation) based on the final voltage commands vu, vv, vw output from the selector 26. The final voltage commands vu, vv, and vw will be described later.

[0027] Figure 2 It is a diagram for explaining the generation principle of the switching signal in Embodiment 1. The PWM signal generator 27 compares the final voltage commands vu, vv, vw with the carrier triangular wave (carrier) C of the period Tc (frequency fc) to generate the switching signals Gup to Gwn. The final voltage commands vu, vv, vw correspond to the U phase, V phase, and W phase, respectively.

[0028] Specifically, if the final voltage command vu is greater than the carrier triangular wave C, the PWM signal generator 27 sets the switch signal Gup to on ("1") and sets the switch signal Gun to off ("0"). Conversely, if the final voltage command vu is less than the carrier triangular wave C, the PWM signal generator 27 sets the switch signal Gup to off ("0") and sets the switch signal Gun to on ("1").

[0029] In addition, if the final voltage command vv is greater than the carrier triangular wave C, the PWM signal generator 27 sets the switch signal Gvp to on ("1") and sets the switch signal Gvn to off ("0"). Conversely, if the final voltage command vv is less than the carrier triangular wave C, the PWM signal generator 27 sets the switch signal Gvp to off ("0") and sets the switch signal Gvn to on ("1").

[0030] In addition, if the final voltage command vw is greater than the carrier triangular wave C, the PWM signal generator 27 sets the switch signal Gwp to on ("1") and sets the switch signal Gwn to off ("0"). Conversely, if the final voltage command vw is less than the carrier triangular wave C, the PWM signal generator 27 sets the switch signal Gwp to off ("0") and sets the switch signal Gwn to on ("1").

[0031] In addition, a short - circuit prevention time (dead - time) can be set for the switch signals Gup~Gwn so that the upper - arm switching elements Sup, Svp, Swp and the lower - arm switching elements Sun, Svn, Swn of the inverter 12 are not simultaneously in the on state.

[0032] The switch signals Gup~Gwn include a mode in which all the lower - arm switching elements Sun, Svn, Swn are in the on state within one electrical angular cycle of the AC motor 10. Specifically, as shown in the interval D in Figure 2 it includes a mode in which all the switch signals Gun, Gvn, Gwn are on (1).

[0033] Here, in the PWM - modulated voltage applied from the inverter 12 to the AC motor 10, in addition to the components of the final voltage commands vu, vv, vw, there are also components that are integer multiples of the period Tc of the carrier triangular wave C. Therefore, when a current with a component that is an integer multiple of the period Tc is passed through the AC motor 10, the AC motor 10 generates abnormal noise according to the value of the period Tc.

[0034] In order to prevent such abnormal noises from occurring, for example, when the AC motor 10 is used as a motor for power-assisted steering, the period Tc of the carrier triangular wave C can be set to 60 [μs] or less. By setting Tc = 60 [μs], the frequency fc of the abnormal noise (= 1 / Tc) is 16.6 kHz, and it is not easy to generate unpleasant noises. More preferably, the period Tc of the carrier triangular wave C can be set to about 50 [μs]. By setting Tc = 50 [μs], the frequency fc of the abnormal noise (= 1 / Tc) becomes about 20 kHz, which is hardly audible to humans. The audible frequency band for humans is about 20 Hz to 20 kHz. In addition, hereinafter, it is assumed that Tc = 50 [μs] for explanation.

[0035] Next, the Figure 1 current detector 22 shown will be described. The current detector 22 uses the terminal voltages VRu, VRv, VRw of the shunt resistors Ru, Rv, Rw and the switching signals Gup to Gwn output from the PWM signal generator 27 to output the pre-conversion detected currents ius, ivs, iws. Specifically, the current detector 22 obtains the terminal voltages VRu, VRv, VRw of the shunt resistors Ru, Rv, Rw at the timing "X" shown in Figure 2 . This timing "X" is the timing when the carrier triangular wave C reaches the maximum value (DC bus voltage Vdc).

[0036] As Figure 2 shown, at the timing "X", the switching signals Gun, Gvn, Gwn input to the lower-arm switching elements Sun, Svn, Swn are all turned on ("1"). Therefore, the current detector 22 obtains the values of the pre-conversion detected currents ius, ivs, iws by dividing the terminal voltages VRu, VRv, VRw of the shunt resistors Ru, Rv, Rw by -Ru, -Rv, -Rw, respectively.

[0037] The first coordinate converter 23 performs coordinate conversion based on the pre-conversion detected currents ius, ivs, iws detected by the current detector 22 and the rotor position θ detected by the rotor position detector 11. Thereby, the first coordinate converter 23 converts the pre-conversion detected currents ius, ivs, iws into two-axis detected currents id, iq (dq-axis currents) that are currents on the rotating two axes (d, q axes). In addition, the first coordinate converter 23 inputs the two-axis detected currents id, iq as the operation results to the two-axis current filter 21.

[0038] After reducing the noise components (ripples) included in the two-axis detected currents id, iq, the two-axis current filter 21 outputs the filtered two-axis currents idf, iqf. Hereinafter, the process based on the two-axis current filter 21 may sometimes be referred to as "filtering process". Hereinafter, Figure 3A detailed description of the two-axis current filter 21 is given. As Figure 3 shown, the two-axis current filter 21 has a first filter section 21d and a second filter section 21q.

[0039] The first filter section 21d uses the detected current id to obtain the filtered two-axis current idf. The first filter section 21d has a first delay unit 601, a second delay unit 602, a first adder 603, and a first multiplier 604. The first delay unit 601 calculates the current id_z1 one period before. The current id_z1 one period before is the detected current id at a time 1×Ts before the time when the detected current id is detected. Here, Ts is the current detection period. The first delay unit 601 outputs the current id_z1 one period before to the second delay unit 602.

[0040] Next, the second delay unit 602 calculates the current id_z2 two periods before based on the current id_z1 one period before. The current id_z2 two periods before is the detected current id at a time 2×Ts before the time when the detected current id is detected. Then, the first adder 603 adds the detected current id and the current id_z2 two periods before, and outputs the result to the first multiplier 604. The first multiplier 604 calculates the filtered two-axis current idf by multiplying the output from the first adder 603 by 0.5 times. As Figure 1 shown, the filtered two-axis current idf is output to the second coordinate converter 28 and the first deviation calculator 24a.

[0041] As Figure 3 shown, the second filter section 21q uses the detected current iq to obtain the filtered two-axis current iqf. The second filter section 21q has a third delay unit 605, a fourth delay unit 606, a second adder 607, and a second multiplier 608. The third delay unit 605 calculates the current iq_z1 one period before. The current iq_z1 one period before is the detected current iq at a time 1×Ts before the time when the detected current iq is detected. The third delay unit 605 outputs the current iq_z1 one period before to the fourth delay unit 606.

[0042] Next, the fourth delay unit 606 calculates the current iq_z2 two periods before based on the current iq_z1 one period before. The current iq_z2 two periods before is the detected current iq at a time 2×Ts before the time when the detected current iq is detected. Then, the second adder 607 adds the detected current iq and the current iq_z2 two periods before, and outputs the result to the second multiplier 608. The second multiplier 608 calculates the filtered two-axis current iqf by multiplying the output from the second adder 607 by 0.5 times. AsFigure 1 As shown, the filtered two-axis current \(i_{qf}\) is output to the second coordinate converter 28 and the second deviation calculator 24b.

[0043] Next, the principle of reducing the noise component by the two-axis current filter 21 will be described. When a certain signal contains a component with a period \(T\) and the signal is discretely detected, if the detection value at an arbitrary time is added to the detection value half a period before that time (the time before \(T / 2\)), the component with the period \(T\) is canceled out and becomes zero.

[0044] For example, if it is a signal with a period of \(2\pi\), the detection value \(S1\) at an arbitrary time can be presented as follows. Here, \(A\) is the amplitude of the signal and \(\alpha\) is the initial phase of the signal. \(S1 = A\cdot\sin(\theta+\alpha)\) In addition, the detection value \(S2\) half a period before the time of the detection value \(S1\) (the time before \(\pi\)) can be presented as follows. \(S2 = A\cdot\sin(\theta+\alpha-\pi)\) It can be seen from the above mathematical formula that \(S1 + S2 = 0\). This is the principle of reducing the noise component by the two-axis current filter 21.

[0045] Here, it is assumed that the noise components with the resonance period \(Tr\) are included in the two-axis detected currents \(id\) and \(iq\). Considering the case where the resonance period \(Tr\) and the current detection period \(Ts\) are not exactly integer multiples and the noise components cannot be completely eliminated. Even in this case, it is preferable to add the detection values at the time \(t\) when the two-axis detected currents \(id\) and \(iq\) are detected and the time \(t'\) closest to half a period before the resonance period \(Tr\) relative to the time \(t\) (the time before \(t - Tr / 2\)). Thus, the noise components with the resonance period \(Tr\) can be reduced. The above "closest time \(t'\)" can be expressed as \(t'=t - Ts\times n\). Where \(n\) is a natural number. That is, the time \(t'\) is the time after tracing a natural number multiple of the current detection period \(Ts\) from the time \(t\) and is the time closest to \(t - Tr / 2\).

[0046] In this embodiment, the current detection period \(Ts\) is determined such that the resonance period \(Tr\) and the current detection period \(Ts\) satisfy the relationship \(Tr\geq2\times Ts\). For example, when the mechanical resonance frequency of the AC motor 10 is about 2500 Hz, the resonance period \(Tr\) is about 400 μs. Therefore, the current detection period \(Ts\) can also be set to 100 μs to satisfy the relationship \(Tr\geq2\times Ts\).

[0047] In addition, the specific structure of the two-axis current filter 21 is not limited to Figure 3, other structures capable of reducing the noise component can be adopted. For example, the two-axis current filter 21 can be a low-pass filter with a cut-off frequency lower than the mechanical resonance frequency of the AC motor 10. Alternatively, the two-axis current filter 21 can also be a notch filter capable of reducing the gain transfer characteristic of the AC motor 10 with respect to the mechanical resonance frequency. As the two-axis current filter 21, an IIR (Infinite Impulse Response) filter or an FIR (Finite Impulse Response) filter can be used.

[0048] Here, the handling in the two-axis current filter 21 immediately after the control device 1 starts is described. Hereinafter, the first filter section 21d is taken as an example for description, but it is preferably the same handling is also performed in the second filter section 21q. As described above, the first filter section 21d obtains the filtered two-axis current idf based on the detected current id as the current detection value, and the current id_z1 one cycle before and the current id_z2 two cycles before as the past detection values. Immediately after starting, since there are no values corresponding to id_z1 and id_z2, the filtered two-axis current idf cannot be directly calculated. Therefore, in order to perform the calculation, a certain value needs to be set for id_z1 and id_z2.

[0049] Therefore, in the present embodiment, immediately after starting, id_z1 = id_z2 = idf = id. That is, when the control device 1 performs control, immediately after starting, for the "past input values" (id_z1, id_z2) and the "filter output value" (idf) of the two-axis current filter 21, the latest "input value" (the two-axis detected current id) is set to perform the calculation. Through this handling, it is possible to avoid torque shock of the AC motor 10 caused by the deviation of the values before and after the filtering process due to inappropriate values input as id_z1 and id_z2.

[0050] As Figure 1 shown, the first controller 24 includes a first deviation calculator 24a, a second deviation calculator 24b, a d-axis current controller 24d, a q-axis current controller 24q, and a control coordinate converter 25. The first controller 24 uses the above current command values id_ref, iq_ref, and the filtered two-axis currents idf, iqf output from the two-axis current filter 21 to calculate the first three-phase voltage commands vu1, vv1, vw1. Hereinafter, the first controller 24 will be described in detail.

[0051] The first deviation calculator 24a calculates a d-axis current deviation ed, which is the deviation between the current command value id_ref on the d-axis and the filtered two-axis current idf. The second deviation calculator 24b calculates a q-axis current deviation eq, which is the deviation between the current command value iq_ref on the q-axis and the filtered two-axis current iqf. The value of the d-axis current deviation ed calculated by the first deviation calculator 24a is input to the d-axis current controller 24d. The value of the q-axis current deviation eq calculated by the second deviation calculator 24b is input to the q-axis current controller 24q. The d-axis current controller 24d calculates a d-axis voltage command value vd using the d-axis current deviation ed. The q-axis current controller 24q calculates a q-axis voltage command value vq using the q-axis current deviation eq.

[0052] In this specification, sometimes the deviation between the current command value and the detected current value on the first axis of the rotating two axes is referred to as the first deviation. Similarly, sometimes the deviation between the current command value and the detected current value on the second axis of the rotating two axes is referred to as the second deviation. For example, when the d-axis is set as the first axis, the d-axis current deviation ed is the "first deviation", and the q-axis current deviation eq is the "second deviation". Similarly, when the q-axis is set as the first axis, the q-axis current deviation eq is the "first deviation", and the d-axis current deviation ed is the "second deviation".

[0053] Figure 4 shows the structure of the d-axis current controller 24d, Figure 5 shows the structure of the q-axis current controller 24q. As Figure 4 shown, the d-axis current controller 24d includes a d-axis proportional amplifier 101d, a d-axis integral amplifier 103d, an integrator 109d, and an adder 111d.

[0054] The d-axis proportional amplifier 101d calculates a d-axis proportional output Vdp by multiplying the d-axis current deviation ed by Kpd. That is, Vdp = ed × Kpd. "Kpd" is the d-axis proportional gain multiplied to make the actual AC motor current relative to the current command value id_ref a preferred response. For example, Kpd = ωcc × Ld. Here, ωcc is the response angular frequency for adjusting the frequency response of the AC motor current to the current command value within a preferred range (more specifically, the reciprocal of the time constant of the feedback control system), and Ld is the d-axis inductance of the AC motor 10. However, the value of Kpd is not limited to ωcc × Ld and can be appropriately adjusted by actually measuring the responsiveness of the actual AC motor current to the current command value id_ref, etc. The d-axis proportional output Vdp calculated by the d-axis proportional amplifier 101d is input to the adder 111d.

[0055] The d-axis integral amplifier 103d calculates the self-axis integral input Cdi (integral compensation control) by multiplying the d-axis current deviation ed by Kid. That is, Cdi = ed × Kid. "Kid" is the integral gain multiplied by the d-axis current deviation ed to set the steady-state value of the d-axis current deviation ed to 0. For example, Kid = ωcc × R. Here, R is the winding resistance value of the AC motor 10. However, the value of Kid is not limited to ωcc × R and can be adjusted appropriately according to actual measurement results, etc.

[0056] The self-axis integral input Cdi calculated by the d-axis integral amplifier 103d is input to the integrator 109d. The integrator 109d performs an integration operation on the d-axis integral input Cdi and outputs the result as the d-axis integral output Vdi to the adder 111d. The adder 111d adds the d-axis proportional output Vdp and the d-axis integral output Vdi to obtain the d-axis voltage command value vd. As Figure 1 shown, the obtained voltage command value vd is input to the control coordinate converter 25.

[0057] As Figure 5 shown, the q-axis current controller 24q includes a q-axis proportional amplifier 101q, a q-axis integral amplifier 103q, an integrator 109q, and an adder 111q. The q-axis proportional amplifier 101q calculates the q-axis proportional output Vqp by multiplying the q-axis current deviation eq by Kpq. That is, Vqp = eq × Kpq. "Kpd" is the q-axis proportional gain multiplied to make the actually flowing AC motor current a preferred response relative to the current command value iq_ref. For example, Kpq = ωcc × Lq. Here, ωcc is the response angular frequency for adjusting the frequency response of the AC motor current to the current command value within a preferred range (more specifically, the reciprocal of the time constant of the feedback control system), and Lq is the q-axis inductance of the AC motor 10. However, the value of Kpq is not limited to ωcc × Lq and can be adjusted appropriately by actually measuring the responsiveness of the actually flowing AC motor current to the current command value iq_ref, etc. The q-axis proportional output Vdq calculated by the q-axis proportional amplifier 101q is input to the adder 111q.

[0058] The q-axis integral amplifier 103q calculates the self-axis integral input Cqi (integral compensation control) by multiplying the q-axis current deviation eq by Kiq. That is, Cqi = eq × Kiq. "Kiq" is the integral gain multiplied by the q-axis current deviation eq to make the steady-state value of the q-axis current deviation eq 0. For example, Kiq = ωcc × R. However, the value of Kiq is not limited to ωcc × R and can be adjusted appropriately based on actual measurement results, etc.

[0059] The self-axis integration input Cqi calculated by the q-axis integration amplifier 103q is input to the integrator 109q. The integrator 109q performs an integration operation on the q-axis integration input Cqi and outputs the result as the q-axis integration output Vqi to the adder 111q. The adder 111q adds the q-axis proportional output Vqp and the q-axis integration output Vqi to obtain the q-axis voltage command value vq. As Figure 1 shown, the obtained voltage command value vq is input to the control coordinate converter 25.

[0060] As Figure 1 shown, the voltage command values vd, vq on the rotating two axes (d-axis, q-axis) and the rotor position θ are input to the control coordinate converter 25. The control coordinate converter 25 performs coordinate conversion on the voltage command values vd, vq based on the rotor position θ and calculates the first three-phase voltage commands vu1, vv1, vw1. The first three-phase voltage commands vu1, vv1, vw1 are input to the selector 26 and the second controller 15.

[0061] The second coordinate converter 28 performs coordinate conversion on the filtered two-axis currents idf, idq output from the two-axis current filter 21 based on the rotor position θ detected by the rotor position detector 11. The second coordinate converter 28 outputs the filtered three-phase currents iuf, ivf, iwf obtained through this operation to the second controller 15. That is, the filtered three-phase currents iuf, ivf, iwf are the results of converting the values on the rotating two axes, namely the filtered two-axis currents idf, idq, into values in the stationary coordinate system (uvw coordinate system).

[0062] The third coordinate converter 29 performs coordinate conversion on the current command values id_ref, iq_ref based on the rotor position θ detected by the rotor position detector 11, thereby calculating the three-phase current command values iu_ref, iv_ref, iw_ref. In addition, the third coordinate converter 29 outputs the calculation result to the second controller 15.

[0063] The second controller 15 calculates the second three-phase voltage commands vu2, vv2, vw2 based on the current command values id_ref, iq_ref and the filtered three-phase currents iuf, ivf, iwf. As Figure 6 shown, the second controller 15 includes a u-phase proportional amplifier 15u, a v-phase proportional amplifier 15v, a w-phase proportional amplifier 15w, a u-phase adder 200u, a v-phase adder 200v, a w-phase adder 200w, etc.

[0064] As Figure 6As shown, in the second controller 15, the stored values vu1z, vv1z, and vw1z are used as the input values for the operation. The stored value vu1z is the value of the first three-phase voltage command vu1 at a time ΔT before the control target time. The stored value vv1z is the value of the first three-phase voltage command vv1 at a time ΔT before the control target time. The stored value vw1z is the value of the first three-phase voltage command vw1 at a time ΔT before the control target time. ΔT is the control execution cycle of the control unit 13. The stored values vu1z, vv1z, and vw1z can be stored, for example, in the memory area inside the second controller 15.

[0065] The second controller 15 calculates the u-phase current deviation eu by subtracting the filtered three-phase current iuf from the current command value iu_ref. The u-phase current deviation eu is input to the u-phase proportional amplifier 15u. The stored value vu1z is added to the value obtained by multiplying the u-phase current deviation eu by Kpu by the u-phase adder 200u, thereby calculating the second three-phase voltage command vu2 related to the u-phase (proportional compensation control). Here, Kpu is the proportional gain multiplied to make the AC motor current iu have a desired response relative to the current command value iu_ref. For example, it is given by Kpu = ωcc × Lu. Lu is the u-phase inductance. However, the value of Kpu is not limited to ωcc × Lu and can be appropriately adjusted by actually measuring the responsiveness of the actually flowing AC motor current iu to the current command value iu_ref, etc.

[0066] The second controller 15 calculates the v-phase current deviation ev by subtracting the filtered three-phase current ivf from the current command value iv_ref. The v-phase current deviation ev is input to the v-phase proportional amplifier 15v. The stored value vv1z is added to the value obtained by multiplying the v-phase current deviation ev by Kpv by the v-phase adder 200v, thereby calculating the second three-phase voltage command vv2 related to the v-phase (proportional compensation control). Here, Kpv is the proportional gain multiplied to make the AC motor current iu have a desired response relative to the current command value iv_ref. For example, it is given by Kpv = ωcc × Lv. Lv is the v-phase inductance. However, the value of Kpv is not limited to the value of ωcc × Lv and can be appropriately adjusted by actually measuring the responsiveness of the actually flowing AC motor current iv to the current command value iv_ref, etc.

[0067] The second controller 15 calculates the w-phase current deviation ew by subtracting the filtered three-phase current iwf from the current command value iw_ref. The w-phase current deviation ew is input to the w-phase proportional amplifier 15w. The stored value vw1z and the value obtained by multiplying the w-phase current deviation ew by Kpw by the w-phase proportional amplifier 15w are added by the w-phase adder 200w, thereby calculating the second three-phase voltage command vw2 related to the w-phase (proportional compensation control). Here, Kpw is the proportional gain multiplied to make the AC motor current iw become the desired response with respect to the current command value iw_ref. For example, it is given by Kpw = ωcc × Lw. Lw is the w-phase inductance. However, the value of Kpw is not limited to the value of ωcc × Lw, and can be appropriately adjusted by actually measuring the responsiveness of the actually flowing AC motor current iw to the current command value iw_ref, etc.

[0068] As described above, the second controller 15 uses the stored values vu1z, vv1z, and vw1z and the current deviations eu, ev, and ew of each phase when calculating the second three-phase voltage commands vu2, vv2, and vw2. Therefore, the integral control terms included in the first three-phase voltage commands vu1, vv1, and vw1 generated by the first controller 24 can also be reflected in the second three-phase voltage commands vu2, vv2, and vw2. Therefore, it is possible to suppress the jitter that occurs when the input to the PWM signal generator 27 switches between the first three-phase voltage commands vu1, vv1, and vw1 and the second three-phase voltage commands vu2, vv2, and vw2. Furthermore, compared with the case where an integrator is provided in the second controller 15 for integral control, the calculation amount can be reduced.

[0069] As Figure 1 shown, the second three-phase voltage commands vu2, vv2, and vw2 calculated by the second controller 15 are output to the selector 26. The selector 26 selects one of the first three-phase voltage commands vu1, vv1, vw1 and the second three-phase voltage commands vu2, vv2, vw2, and outputs it as the final voltage commands vu, vv, vw to the PWM signal generator 27.

[0070] Use Figure 7 to illustrate an operation example of the selector 26. Figure 7 "ΔT" in Figure 7 is the execution period based on the control of the control unit 13 as described above. The execution period ΔT is determined according to the responsiveness to the current command values id_ref, iq_ref, the carrier period (frequency) of the inverter 12, etc. The case where ΔT = 100 μs will be described below. In

[0071] As Figure 7 shown, the control unit 13 causes one of the first controller 24 and the second controller 15 to execute. The selector 26 selects the first three-phase voltage commands vu1, vv1, vw1 when the control unit 13 causes the first controller 24 to execute, and selects the second three-phase voltage commands vu2, vv2, vw2 when the control unit 13 causes the second controller 15 to execute.

[0072] In the Figure 7 example, the filter 21 performs filtering processing for each ΔT. However, the period for performing the filtering processing may not be the same as ΔT. The period for performing the filtering processing is the same as or shorter than the period in which the selector 26 switches from the first three-phase voltage commands vu1, vv1, vw1 to the second three-phase voltage commands vu2, vv2, vw2, or the period in which the selector 26 switches from the second three-phase voltage commands vu2, vv2, vw2 to the first three-phase voltage commands vu1, vv1, vw1. By doing so, it is possible to suppress a performance degradation due to discretization of the output from the filter 21.

[0073] In the Figure 7 example, the first controller 24 is selected every 2×ΔT. However, the selection of the first controller 24 and the second controller 15 is not limited to the Figure 7 example, and for example, it may be selected as Figure 8 , Figure 9 for example. In the Figure 8 example, the first controller 24 is selected every 3×ΔT. In the Figure 9 example, the first controller 24 is selected every 4×ΔT. It is possible to determine at what ratio the first controller 24 is selected based on the period for executing the integral compensation control. For example, in a case where it is desired to further improve the steady-state deviation, the ratio of selecting the first controller 24 can be increased, and otherwise, the ratio of selecting the first controller 24 can be decreased.

[0074] Next, the operation of the control device 1 according to the present embodiment will be described. According to the control device 1, the following operations (1) to (3) can be obtained.

[0075] Operation (1) Among the first controller 24 and the second controller 15, the amounts of computation are compared. As Figure 4 , Figure 5 shown, the first controller 24 performs proportional and integral control on the dq coordinates as the rotating two axes. In contrast, the second controller 15 performs proportional control in the stationary coordinate system (uvw coordinate system). Therefore, the amount of computation of the second controller 15 becomes smaller than that of the first controller 24. Therefore, compared with the case where the control device 1 executes the processing performed by the first controller 24 every ΔT, asFigures 7 - 9 As shown, the amount of calculation during the processing performed by the second controller 15 is also small. Therefore, the overall processing load of the CPU included in the control device 1 can be reduced.

[0076] Function (2) When the detection noise is included in the detected currents ius, ivs, and iws before conversion, it is preferable to use the results from which the detection noise has been removed and perform the processing by the first controller 24 and the second controller 15. For example, Figure 10 An example of the sound-frequency characteristics of the AC motor 10 is shown. Figure 10 The vertical axis represents sound, and the horizontal axis represents frequency. In this example, the detection noise becomes larger near 2500 Hz, which is the mechanical resonance frequency of the AC motor 10. According to the control device 1, by using the filtered two-axis currents idf and idq in which the noise components have been reduced by the two-axis current filter 21, the influence of such detection noise can be suppressed.

[0077] Function (3) Figure 11 The results of verifying the magnitude of the phase delay accompanying the filtering process are shown. Figure 11 In (a1) to (a3) in, it represents the case where the AC motor 10 rotates at a low speed (fundamental wave 50 Hz), and (b1) to (b3) represent the case where the AC motor 10 rotates at a high speed (fundamental wave 250 Hz). In each graph, the horizontal axis is time, and the vertical axis is the current value. The current values corresponding to the U phase, V phase, and W phase are represented by thin lines, thick lines, and dotted lines. (a1) and (b1) are the waveforms of the detected currents ius, ivs, and iws before conversion. (a2) and (b2) are the graphs of the comparative examples. Specifically, in the comparative examples shown in (a2) and (b2), they represent the waveforms when the detected currents ius, ivs, and iws before conversion are directly filtered (the same processing as the two-axis current filter 21).

[0078] According to Figure 11 the comparison between (a1) and (a2) in, it can be seen that in the case of low-speed rotation, even if the detected currents ius, ivs, and iws before conversion are filtered, no phase delay will occur. However, according to Figure 11 the comparison between (b1) and (b2) in, it can be seen that in the case of high-speed rotation, if the detected currents ius, ivs, and iws before conversion are filtered, a phase delay will occur. That is, if the current in the stationary coordinate system is filtered, a phase delay will occur especially in the case of high-speed rotation, which may reduce the control accuracy of the AC motor 10.

[0079] Figure 11(a3) and (b3) are the curve graphs of the embodiments corresponding to this embodiment. Specifically, in the embodiments shown in (a3) and (b3), filtering processing is performed on the result (two-axis detected currents id and iq) after coordinate conversion of the detected currents ius, ivs, and iws before conversion by the first coordinate converter 23. As a result, the filtered two-axis currents idf and iqf are obtained. In Figure 11 (a3) and (b3), the waveforms of the result (filtered three-phase currents iuf, ivf, and iwf) after coordinate conversion of the filtered two-axis currents idf and iqf by the second coordinate converter 28 are shown.

[0080] According to Figure 11 the comparison between (a1) and (a3), in the case of low-speed rotation, no phase delay occurs, and the noise components included in the waveform of (a1) are removed. In addition, according to Figure 11 the comparison between (b1) and (b3), in the case of high-speed rotation, no phase delay occurs either, and the noise components included in the waveform of (b1) are removed. That is, it is confirmed that according to the structure of the control device 1, the phase delay accompanied by the filtering process, especially in the case of high-speed rotation, can be improved. In this way, by improving the phase delay, the control accuracy of the AC motor 10 can be increased.

[0081] Through the above functions (1) to (3), the control device 1 according to this embodiment can reduce the operation load, can also suppress the phase delay during high-speed rotation, and can contribute to the quieting of the AC motor 10 based on the removal of noise components.

[0082] In addition, in this embodiment, in the operation of the first controller 24, the filtered two-axis currents idf and iqf are used. However, instead of the filtered two-axis currents idf and iqf, the first controller 24 can also perform operations using the two-axis detected currents id and iq. More specifically, the first coordinate converter 23 can output the two-axis detected currents id and iq to the first deviation calculator 24a and the second deviation calculator 24b, and calculate the deviations ed and eq based on the two-axis detected currents id and iq. In this case, by adjusting the proportional gains Kpd and Kpq in the proportional compensation control or the proportional gains Kid and Kiq in the integral compensation control, quieting can also be achieved.

[0083] As described above, the control device 1 according to the present embodiment includes: a current detector 22 that detects three-phase currents iu, iv, and iw supplied to the AC motor 10; a first coordinate converter 23 that converts the detection results of the current detector 22, i.e., the pre-conversion detection currents ius, ivs, and iws, into currents on the rotating two axes, i.e., the two-axis detection currents id and iq; a two-axis current filter 21 that reduces the noise components of the two-axis detection currents id and iq; a second coordinate converter 28 that converts the filtered two-axis currents idf and idq, whose noise components have been reduced by the two-axis current filter 21, into the filtered three-phase currents iuf, ivf, and iwf in the stationary coordinate system; a first controller 24 that generates first three-phase voltage commands vu1, vv1, and vw1 based on the two-axis detection currents id and iq or the filtered two-axis currents idf and idq so that the two-axis detection currents id and iq become desired values; a second controller 15 that generates second three-phase voltage commands vu2, vv2, and vw2 based on the filtered three-phase currents iuf, ivf, and iwf; and an inverter 12 that applies voltages to the AC motor 10 based on the first three-phase voltage commands vu1, vv1, and vw1 and the second three-phase voltage commands vu2, vv2, and vw2.

[0084] According to such a control device 1, during the generation of the second three-phase voltage commands vu2, vv2, and vw2, the noise components are reduced, so that the vibration or noise of the AC motor 10 can be reduced. In addition, compared with the case of filtering the pre-conversion detection currents ius, ivs, and iws, for example, by filtering the two-axis detection currents id and iq, the phase delay during high-speed rotation can be reduced.

[0085] In addition, the two-axis current filter 21 is configured to calculate the filtered two-axis currents idf and iqf using the past two-axis detection currents (the current id_z1 one cycle ago and the current id_z2 two cycles ago). Then, the two-axis current filter 21 substitutes the latest two-axis detection currents id and iq as the past two-axis detection currents and performs the calculation immediately after the control device 1 of the AC motor 10 starts. Thereby, even immediately after starting the control device 1, appropriate control can be performed. For example, the torque shock caused by substituting inappropriate values as the past two-axis detection currents immediately after starting can be reduced.

[0086] In addition, in the present embodiment, the second controller 15 generates the second three-phase voltage commands vu2, vv2, and vw2 using the values obtained by multiplying the current deviations eu, ev, and ew in the stationary coordinate system (U-phase, V-phase, W-phase) by the proportional gains Kpu, Kpv, and Kpw. In such proportional control in the stationary coordinate system, the amount of calculation can be reduced compared to the proportional-integral control in the rotating two-axis coordinate system. Therefore, by using not only the first controller 24 but also partially using the calculation results of the second controller 15 (the second three-phase voltage commands vu2, vv2, and vw2), the amount of calculation of the entire device can be reduced.

[0087] In addition, the two-axis current filter 21 may also be a notch filter that reduces the mechanical resonance frequency of the AC motor 10, or a low-pass filter whose cut-off frequency is below the mechanical resonance frequency. In this case, by reducing the noise components that appear based on the mechanical resonance frequency, it is possible to further contribute to quieting.

[0088] Embodiment 2. Next, Embodiment 2 in the present disclosure will be described. Its basic structure is the same as that of Embodiment 1. Therefore, the description of the parts that overlap with Embodiment 1 will be omitted, and the description will focus on the differences. As Figure 12 shown, the control device 2 according to the present embodiment does not include the third coordinate converter 29 described in Embodiment 1. In addition, instead of the second controller 15 in Embodiment 1, a second controller 15b is provided.

[0089] Use Figure 13 to describe the second controller 15b. In the second controller 15b, the stored values iufz, ivfz, and iwfz are used as the input values for the calculation. The stored values iufz, ivfz, and iwfz are the values of the filtered three-phase currents iuf, ivf, and iwf at a time ΔT before the control target time. The stored values iufz, ivfz, and iwfz can be stored, for example, in a memory area inside the second controller 15b.

[0090] The second controller 15b calculates the current deviations eu, ev, and ew by subtracting the filtered three-phase currents iuf, ivf, and iwf from the stored values iufz, ivfz, and iwfz. The U-phase proportional amplifier 15u, the V-phase proportional amplifier 15v, and the W-phase proportional amplifier 15w multiply the gains Kpu, Kpv, and Kpw by the current deviations eu, ev, and ew respectively, and output them to the U-phase adder 200u, the V-phase adder 200v, and the W-phase adder 200w. For these outputs, the U-phase adder 200u, the V-phase adder 200v, and the W-phase adder 200w add the stored values vu1z, vv1z, and vw1z respectively. Thereby, the second three-phase voltage commands vu2, vv2, and vw2 are calculated.

[0091] Here, taking the case where control is performed as shown in Figure 7 as an example, the arithmetic function of the second controller 15b will be described. In Figure 7 this example, the control unit 13 performs a process of discretizing time every natural multiple of the execution cycle ΔT. Hereinafter, the time sequence of the control by the control unit 13 will be generalized by a natural number K (K = 1, 2, 3...). For example, the (K - 1)th process represents the process at a moment ΔT before the Kth process.

[0092] First, the "difference between the current iu flowing through the winding U of the AC motor 10 and the current command value iu_ref" is defined as the u-phase current deviation eu. According to this definition, the Kth u-phase current deviation eu(K) is represented by the following formula (2-1). eu(K) = iu_ref(K) - iu(K)...(2 - 1) The (K - 1)th u-phase current deviation eu(K - 1) is represented by the following formula (2-2). eu(K - 1) = iu_ref(K - 1) - iu(K - 1)...(2 - 2) Here, it is considered that the changes in the three-phase current command values iu_ref, iv_ref, and iw_ref occur with a delay compared to the changes in the actual currents iu, iv, and iw of the AC motor 10. Therefore, in formula (2-1), it is assumed that iu_ref(K) = iu_ref(K - 1). On this basis, if formula (2-2) is substituted into formula (2-1), the following formula (2-3) can be obtained. eu(K) = eu(K - 1) + (iu(K - 1) - iu(K))...(2 - 3)

[0093] If both sides of formula (2-3) are multiplied by Kpu, the following formula (2-4) is obtained. Kpu·eu(K) = Kpu·eu(K - 1) + Kpu·(iu(K - 1) - iu(K))...(2 - 4) If both sides of formula (2-4) are multiplied by the integral term I, the following formula (2-5) is obtained. Kpu·eu(K) + I = {Kpu·eu(K - 1) + I} + Kpu·(iu(K - 1) - iu(K))...(2 - 5) The function of the integral term I is to improve the steady-state deviation and has a small contribution to the responsiveness. If it is considered that the change in the integral term I during the execution cycle ΔT is small enough, the term enclosed by {} on the right side of formula (2-5) is equal to vu1(K - 1) (= vu1z). Therefore, formula (2-5) is represented as the following formula (2-6). Kpu·eu(K)+I = vu1(K - 1)+Kpu·(iu(K - 1)-iu(K))…(2-6)

[0094] The left side of Equation (2-6) is identical to the second three-phase voltage command vu2(K) related to U to be obtained at the Kth step. Therefore, the following equation (2-7) is obtained. vu2(K)=vu1(K - 1)+Kpu·(iu(K - 1)-iu(K))…(2-7) Therefore, Equation (2-7) is expressed as the following equation (2-8). vu2 = vu1z+Kpu·(iufz - iuf)…(2-8) Equation (2-8) is consistent with Figure 13 the arithmetic processing for obtaining the second three-phase voltage command vu2. The same applies to the V phase and the W phase.

[0095] As described above, in the present embodiment, the second controller 15 generates the second three-phase voltage commands vu2, vv2, and vw2 using the values obtained by multiplying the current deviations eu, ev, and ew in the stationary coordinate system (U phase, V phase, W phase) by the proportional gains Kpu, Kpv, and Kpw. In such proportional control in the stationary coordinate system, the amount of computation can be reduced compared to the proportional-integral control in the rotating two-axis coordinate system. Therefore, by utilizing not only the first controller 24 but also partially the arithmetic results (the second three-phase voltage commands vu2, vv2, and vw2) of the second controller 15, the overall computation amount of the device can be reduced. Furthermore, in the present embodiment, the same effect as in Embodiment 1 can be obtained while omitting the third coordinate converter 29 related to the generation of the current command values iu_ref, iv_ref, and iw_ref. Therefore, the processing load of the CPU can be reduced more effectively.

[0096] Embodiment 3. Next, Embodiment 3 in the present disclosure will be described. Its basic structure is the same as that of Embodiment 1. Therefore, the description of the parts overlapping with Embodiment 1 will be omitted, and the description will focus on the differences. In the present embodiment, an example of the processing after applying the starting control device 1 described in Embodiment 1 will be described.

[0097] Figure 14This is a diagram showing the structure of the control device 3 according to the present embodiment. Descriptions of the same parts as in Embodiments 1 and 2 are omitted. The control device 3 includes an amplitude calculator 100, a first auxiliary selector 102, and a second auxiliary selector 101. The first auxiliary selector 102 switches the current input to the first controller 24 between the two-axis detected currents id and iq and the filtered two-axis currents idf and iqf based on Vamp_z which is the previous value of the voltage amplitude Vamp that is the output of the amplitude calculator 100. The second auxiliary selector 101 switches the current input to the second controller 15b between the pre-conversion detected currents ius, ivs, iws and the filtered three-phase currents iuf, ivf, iwf based on Vamp_z. Hereinafter, a detailed description will be given.

[0098] In the amplitude calculator 100, the voltage amplitude Vamp of the first three-phase voltage commands vu1, vv1, vw1 is calculated according to the following formula (3-1). Vamp = {(Vu1 2 + Vv1 2 + Vw1 2 ) / (Vdc 2 / 2)} 0.5 …(3-1) The value of the output voltage Vdc of the DC power supply BT in formula (3-1) may also be a value detected by a voltage detection sensor (not shown). Alternatively, for the purpose of reducing costs by reducing the number of sensors, the output voltage Vdc may be set to a fixed value (for example, Vdc = 12 [V]).

[0099] In the second auxiliary selector 101, the previous value Vamp_z of the voltage amplitude Vamp is compared with the amplitude threshold Vth, and the current input to the second controller 15b is selected as follows. When Vamp_z > Vth: Input iuf, ivf, iwf to the second controller 15b…(3-2) When Vamp_z ≤ Vth: Input ius, ivs, iws to the second controller 15b…(3-3) That is, when the previous value Vamp_z is greater than the amplitude threshold Vth, the filtered three-phase currents iuf, ivf, iwf are input to the second controller 15b. In addition, when the previous value Vamp_z is less than the amplitude threshold Vth, the pre-conversion detected currents ius, ivs, iws that have not undergone filtering processing are input to the second controller 15b. However, when the previous value Vamp_z is the same as the amplitude threshold Vth, the filtered three-phase currents iuf, ivf, iwf may be input to the second controller 15b. The setting of the amplitude threshold Vth will be described later.

[0100] The first auxiliary selector 102 compares the previous value Vamp_z of the voltage amplitude Vamp with the amplitude threshold Vth and selects the current input to the first controller 24 as follows. When Vamp_z > Vth: Input idf and iqf to the first controller 24... (3-4) When Vamp_z ≤ Vth: Input id and iq to the first controller 24... (3-5) That is, when the previous value Vamp_z is greater than the amplitude threshold Vth, the filtered two-axis currents idf and iqf are input to the first controller 24. In addition, when the previous value Vamp_z is less than the amplitude threshold Vth, the unfiltered two-axis detected currents id and iq are input to the first controller 24. However, when the previous value Vamp_z is the same as the amplitude threshold Vth, the filtered two-axis currents idf and iqf can be input to the first controller 24.

[0101] Next, the amplitude threshold Vth will be described. Figure 14 The shown inverter 12 is a so-called shunt resistor type inverter. However, the following description can also be applied to other types of inverters. If a resistor is inserted into the live wires 12u, 12v, and 12w of the main circuit and the three-phase currents iu, iv, and iw are detected based on the voltage across this resistor, the structure of the inverter 12 can be changed.

[0102] Figure 15 The Gxn shown represents an example of the waveform of any one of the switching signals Gun, Gvn, and Gwn of the lower arm switching elements Sun, Svn, and Swn. In addition, VRx represents an example of the waveform of any one of the voltages VRu, VRv, and VRw across the shunt resistors Ru, Rv, and Rw. In Figure 15 In the shown example, after the signal Gxn changes from 0 to 1, the voltage VRx across the shunt resistor rings within a few μs. Ringing is a phenomenon in which the voltage across the shunt resistor changes within a certain period of time when switching is performed in the inverter 12. If the current detector 22 obtains the pre-conversion detected currents ius, ivs, and iws based on the voltages VRu, VRv, and VRw across the shunt resistor including this ringing, the detection result will contain errors. If the pre-conversion detected currents ius, ivs, and iws contain errors, the two-axis detected currents id and iq after coordinate conversion will also contain errors.

[0103] In order to accurately obtain the detected currents ius, ivs, and iws before conversion, the conduction times of the corresponding lower-arm switching elements Sun, Svn, and Swn are preferably longer than the time threshold Tmin set according to the ringing convergence time. In order to make the conduction times of the lower-arm switching elements Sun, Svn, and Swn longer than the time threshold Tmin, the voltage input to the PWM signal generator 27 can be below Vdc×(Tc - Tmin) / Tc indicated by the dash line in Figure 2 Below Vdc×(Tc - Tmin) / Tc indicated by the dash line in

[0104] In Figure 2 the final voltage commands vu, vv, and vw (i.e., the voltages input to the PWM signal generator 27) are all below Vdc×(Tc - Tmin) / Tc. Therefore, the error caused by ringing can be excluded, and the detected currents ius, ivs, and iws before conversion can be accurately obtained based on the voltages VRu, VRv, and VRw across the two ends of the shunt resistors. Thus, the detection method of obtaining the detected currents ius, ivs, and iws before conversion based on the voltages VRu, VRv, and VRw across the two ends of the shunt resistors corresponding to the three phases respectively is called "three-phase detection".

[0105] However, if the voltage amplitude Vamp increases due to an increase in the rotational speed of the AC motor 10 or the like, a part of the final voltage commands vu, vv, and vw sometimes takes a larger value close to the maximum value Vdc of the carrier triangular wave C. When the value of the final voltage command vu is greater than the value of Vdc×(Tc - Tmin) / Tc (hereinafter also referred to as the "upper limit value"), the difference between the moment when Gun switches from 0 to 1 and the moment of timing X becomes smaller. Therefore, the influence of ringing is included in the voltage VRu across the two ends of the u-phase shunt resistor obtained at timing X.

[0106] Therefore, the detected current before conversion related to the phase in which the conduction times of the switching signals Gun, Gvn, and Gwn of the lower-arm switching elements are shorter than the time threshold Tmin can be generated from the other two phases. Thus, the detection method of obtaining the detected current before conversion of one phase in the three phases based on the detected currents before conversion of the remaining two phases in the three phases is called "two-phase detection". For example, when the conduction time of the switching signal Gun is shorter than the time threshold Tmin, the detected current ius of the U phase can be calculated by ius = -ivs - iws. Similarly, when the conduction time of the switching signal Gvn is shorter than the time threshold Tmin, it can be calculated as ivs = -ius - iws, or when the conduction time of the switching signal Gwn is shorter than the time threshold Tmin, it can be calculated as iws = -ius - ivs.

[0107] The relationship between the number of phases for current detection and the voltage amplitude Vamp is as follows. That is, when the voltage amplitude Vamp is low and the instantaneous values of the final voltage commands vu, vv, and vw of the three phases are all below the upper limit value (Vdc × (Tc - Tmin) / Tc), "three-phase detection" is used. Or, when the voltage amplitude Vamp is high and the instantaneous value of any one of the final voltage commands vu, vv, and vw of the three phases becomes above the upper limit value (Vdc × (Tc - Tmin) / Tc), "two-phase detection" is used. Here, if "three-phase detection" and "two-phase detection" are compared, the accuracy of "three-phase detection" is better. Therefore, in order to improve the control accuracy of the AC motor 10, it is preferable to use "three-phase detection" as much as possible. However, from the viewpoint of increasing the output of the AC motor 10, it is sometimes preferable to use "two-phase detection" by increasing the voltage amplitude Vamp.

[0108] Therefore, in the present embodiment, the value of the amplitude threshold Vth is set to "(Tc - Tmin) / Tc". It can be said that "(Tc - Tmin) / Tc" is a value normalized by dividing the above upper limit value "Vdc × (Tc - Tmin) / Tc" by Vdc. By setting the amplitude threshold Vth in this way, when performing "two-phase detection", the current value after passing through the filter 21 is selected in the auxiliary selectors 101 and 102. Thereby, the influence caused by the reduction of the current detection accuracy can be reduced, and the increase in vibration / noise generated from the AC motor 10 can be prevented. In addition, in the case of performing "three-phase detection", the current detection accuracy is good. In three-phase detection, by selecting the current value that has not passed through the filter 21, the responsiveness of the AC motor 10 can be improved.

[0109] In the above description, the operation of the voltage amplitude Vamp using the first three-phase voltage commands vu1, vv1, and vw1 has been described. However, the control can also be performed based on the comparison between the voltage amplitude of the second three-phase voltage commands vu2, vv2, and vw2 and the amplitude threshold Vth. Or, the voltage amplitudes of both the first three-phase voltage commands vu1, vv1, and vw1 and the second three-phase voltage commands vu2, vv2, and vw2 can be compared with the amplitude threshold Vth, and the respective comparison results can be combined to perform the control.

[0110] In addition, generally, including the AC motor 10 of the present embodiment, the applied voltage to the AC motor is approximately proportional to the rotational speed of the AC motor. Therefore, as the physical quantity input to the auxiliary selectors 101 and 102, a value related to the rotational speed can also be used instead of the voltage amplitude Vamp. For example, the rotational speed when the voltage amplitude Vamp of the AC motor 10 is equal to the amplitude threshold Vth is set as the rotational speed threshold Nth. Also, the switching based on the amplitude threshold Vth in the present embodiment can be replaced with switching based on the rotational speed threshold Nth. Additionally, switching can be performed using both the amplitude threshold Vth and the rotational speed threshold Nth.

[0111] In this specification, a concept including the voltage amplitude Vamp and physical quantities (such as rotational speed) proportional thereto is defined as an amplitude-related physical quantity. The physical threshold (amplitude threshold Vth) is used to determine whether to control based on the output from the two-axis current filter 21. Additionally, the concept corresponding to the "amplitude threshold Vth" for the voltage amplitude Vamp is defined as the "physical threshold" for the amplitude-related physical quantity. That is, if the amplitude-related physical quantity is above the physical threshold, the current value after passing through the filter 21 is selected; otherwise, the current value not passing through the filter 21 can also be selected.

[0112] In addition, in Figure 14 it is also possible to adopt a structure that uses the second auxiliary selector 101 and omits the first auxiliary selector 102. That is, in the operation of the first controller 24, the two-axis detected currents id, iq or the filtered two-axis currents idf, idq can always be used. Or, it is also possible to adopt a structure that uses the first auxiliary selector 102 and omits the second auxiliary selector 101. That is, in the operation of the second controller 15b, any of the detected currents ius, ivs, iws before conversion and the filtered three-phase currents iuf, ivf, iwf can always be used.

[0113] Next, the processing of the two-axis current filter 21 will be described. In the following description, a state where the voltage amplitude Vamp is lower than the amplitude threshold Vth is referred to as the "first state", and a state where the voltage amplitude Vamp is equal to or higher than the amplitude threshold Vth is referred to as the "second state". In the present embodiment, immediately after switching from the first state to the second state, the same problem as that of the two-axis current filter 21 immediately after startup described in the first embodiment occurs. That is, in the first state, the operation based on the two-axis current filter 21 is not performed. When switching from this state to the second state, if the operation based on the two-axis current filter 21 is to be performed, there are no values corresponding to the current id_z1 one cycle before and the current id_z2 two cycles before. Therefore, the filtered two-axis current idf cannot be calculated. Or, even if the calculation can be performed, it may be directly performed while maintaining the value when changing from the second state to the first state last time. Therefore, in order to perform the operation appropriately, appropriate values need to be set for id_z1 and id_z2.

[0114] Therefore, in the present embodiment, immediately after switching from the first state (Vamp < Vth) to the second state (Vamp ≥ Vth), id_z1 = id_z2 = idf = id. That is, for the "past input values" (id_z1, id_z2) and the "filter output value" (idf) of the two-axis current filter 21, the latest input value is set to perform the operation. By this processing, it is possible to avoid torque shock of the AC motor 10 caused by inappropriate values being input as id_z1 and id_z2 and deviation between the values before and after filtering.

[0115] As described above, in the control device 3 according to the present embodiment, when the amplitude-related physical quantity related to the voltage amplitude of the first three-phase voltage commands vu1, vv1, vw1 or the voltage amplitude of the second three-phase voltage commands vu2, vv2, vw2 is equal to or higher than the physical threshold, the first three-phase voltage commands vu1, vv1, vw1 are generated based on the filtered two-axis currents idf, iqf, and the second three-phase voltage commands vu2, vv2, vw2 are generated based on the filtered three-phase currents iuf, ivf, iwf. In addition, when the amplitude-related physical quantity is less than the physical threshold, the first three-phase voltage commands vu1, vv1, vw1 are generated based on the two-axis detected currents id, iq, and the second three-phase voltage commands vu2, vv2, vw2 are generated based on the detected currents ius, ivs, iws before conversion.

[0116] According to such a control device 3, when the amplitude-related physical quantity exceeds the physical threshold, the current value after the filtering process based on the two-axis current filter 21 is used. When the amplitude-related physical quantity exceeds the physical threshold, the influence of the noise component can be reduced by performing the filtering process. In addition, when the amplitude-related physical quantity is less than the physical threshold, the processing load of the operation can be reduced by not performing the filtering process.

[0117] In addition, in the control device 3, the two-axis current filter 21 is configured to calculate the filtered two-axis currents idf and iqf using the past two-axis detected currents (the current id_z1 one cycle ago and the current id_z2 two cycles ago). When the two-axis current filter 21 switches from the first state where the amplitude-related physical quantity is less than the physical threshold to the second state where the amplitude-related physical quantity is equal to or greater than the physical threshold, the latest two-axis detected currents id and iq are substituted as the past two-axis detected currents for calculation. Thus, even when switching from the first state to the second state, appropriate control can be performed. For example, when switching from the first state to the second state, torque shock caused by substituting inappropriate values as the past two-axis detected currents can be reduced.

[0118] In addition, the inverter 12 has current-carrying lines 12u, 12v, and 12w for supplying current to the AC motor 10 and shunt resistors Ru, Rv, and Rw connected in series with the current-carrying lines 12u, 12v, and 12w. The current detector 22 detects the pre-conversion detected currents ius, ivs, and iws based on the voltages VRu, VRv, and VRw across the shunt resistors Ru, Rv, and Rw. Then, the physical threshold (for example, the amplitude threshold Vth) is determined based on the period Tc of the carrier triangular wave C and the time threshold Tmin. More specifically, for example, if the physical threshold is the amplitude threshold Vth, then Vth = (Tc - Tmin) / Tc. Thus, in the control device using the shunt resistor type inverter 12, the influence of the current detection accuracy caused by ringing can also be reduced.

[0119] Embodiment 4. Next, Embodiment 4 in the present disclosure will be described. Its basic structure is the same as that of Embodiment 3. Therefore, the description of the parts repeated with Embodiment 3 is omitted, and the description will be centered on the differences. Figure 16 The structure of the control device 4 according to this embodiment is shown. The control device 4 is provided with a weighted average processing unit 401 on the basis of the control device 3 according to Embodiment 3.

[0120] As Figure 16As shown, the two-axis detection currents id and iq are input from the first coordinate converter 23 to the weighted average processing unit 401. In addition, the filtered two-axis currents idf and iqf are input from the two-axis current filter 21 to the weighted average processing unit 401. In addition, the previous value Vamp_z of the voltage amplitude Vamp, which is the output of the amplitude calculator 100, is input to the weighted average processing unit 401. The weighted average processing unit 401 calculates the weighted average outputs idf' and iqf' according to the following equations (4-1) and (4-2). idf'=(1-Kf)×id+Kf×idf…(4-1) iqf'=(1-Kf)×id+Kf×idf…(4-2) In equations (4-1) and (4-2), Kf is the weighted average gain. The weighted average gain Kf is as follows: Figure 17 As shown in FIG. 1 , the weighted average gain Kf is determined based on the previous value Vamp_z of the voltage amplitude Vamp. The specific numerical value of the weighted average gain Kf varies within a range of 0 to 1 based on the previous value Vamp_z.

[0121] exist Figure 17 The upper part of shows the relationship between the calculation on / off command of the two-axis current filter 21 and the previous value Vamp_z of the voltage amplitude Vamp. Figure 17 The lower part of represents the relationship between the weighted average gain Kf of the two-axis current filter 21 and the previous value Vamp_z. The operation on / off instruction is an instruction to switch whether to perform filtering processing based on the two-axis current filter 21. When the operation on / off instruction is on, filtering processing is performed, and when it is off, filtering processing is not performed. The operation on / off instruction can be generated by the amplitude operator 100, for example, or by other parts that can perform calculations.

[0122] like Figure 17 As shown in the upper part of , if the previous value Vamp_z exceeds the second amplitude threshold value Vth2, the calculation on / off instruction is set to on, otherwise it is set to off. The second amplitude threshold value Vth2 is a physical quantity having the same dimension as the amplitude threshold value Vth. The second amplitude threshold value Vth2 is set to a value lower than the amplitude threshold value Vth.

[0123] like Figure 17 As shown in the lower part of , when the previous value Vamp_z is less than the second amplitude threshold value Vth2, the weighted average gain Kf is set to 0. When the previous value Vamp_z is greater than the amplitude threshold value Vth, the weighted average gain Kf is set to 1. When the previous value Vamp_z is between the second amplitude threshold value Vth2 and the amplitude threshold value Vth, the weighted average gain Kf changes linearly between 0 and 1 in conjunction with the value of the previous value Vamp_z.

[0124] According to the operation of the weighted average processing unit 401 described above, when the previous value Vamp_z of the voltage amplitude Vamp is less than or equal to the second amplitude threshold Vth2, the weighted average outputs idf' and iqf' are the same as the inputs of the two-axis current filter 21 (the two-axis detected currents id and iq). When the previous value Vamp_z is greater than or equal to the amplitude threshold Vth, the weighted average outputs idf' and iqf' are the same as the outputs of the two-axis current filter 21 (the filtered two-axis currents idf and iqf). When the previous value Vamp_z is between the second amplitude threshold Vth2 and the amplitude threshold Vth, the weighted average outputs idf' and iqf' are the values obtained by weighted averaging the inputs (the two-axis detected currents id and iq) and the outputs (the filtered two-axis currents idf and iqf) of the two-axis current filter 21 based on the previous value Vamp_z.

[0125] The weighted average processing unit 401 inputs the weighted average outputs idf′ and iqf′ to the first auxiliary selector 102. The first auxiliary selector 102 according to the present embodiment performs a process of switching the replacement between the two-axis detected currents id and iq and the filtered two-axis currents idf and iqf described in Embodiment 3 to the switching between the two-axis detected currents id and iq and the weighted average outputs idf' and iqf'. That is, the first auxiliary selector 102 according to the present embodiment switches the current input to the first controller 24 between the two-axis detected currents id and iq and the weighted average outputs idf' and iqf' based on the previous value Vamp_z of the voltage amplitude Vamp.

[0126] In this specification, the concept of the "second amplitude threshold Vth2" corresponding to the voltage amplitude Vamp is defined as the "second physical threshold" for the amplitude-related physical quantity. That is, in the control device 4 according to the present embodiment, when the previous value of the amplitude-related physical quantity exceeds the second physical threshold smaller than the physical threshold, the two-axis current filter 21 performs an operation (filtering process). In addition, when the previous value of the amplitude-related physical quantity is between the second physical threshold and the physical threshold, the weighted average processing unit 401 performs a weighted average process on the two-axis detected currents id and iq and the filtered two-axis currents idf and iqf. Then, when the weighted average processing unit 401 performs the weighted average process, the greater the amplitude-related physical quantity, the higher the proportion of the filtered two-axis currents idf and iqf.

[0127] According to the control device 4 according to the present embodiment, by including the weighted average processing unit 401, it is possible to reduce the jitter of the input from the first auxiliary selector 102 to the first controller 24. That is, compared with switching the input to the first controller 24 between the two-axis detected currents id, iq and the filtered two-axis currents idf, iqf, the numerical change is smaller when switching the input to the first controller 24 between the two-axis detected currents id, iq and the weighted average outputs idf′, iqf′. Therefore, it is possible to suppress the generation of noise, vibration, etc. from the AC motor 10 due to the switching of the first auxiliary selector 102.

[0128] Embodiment 5. Next, Embodiment 4 in the present disclosure will be described. Its basic structure is the same as that of Embodiment 1. Therefore, the description of the parts repeated with Embodiment 1 will be omitted, and the description will be centered on the differences. In the present embodiment, the case where the techniques described in Embodiments 1 to 4 are applied to the control of the AC motor included in the electric power steering device will be described.

[0129] As Figure 18 shown, the electric power steering device 900 according to the present embodiment includes a control device 5, a steering wheel 901, an AC motor 10, a torque detector 903, etc. The electric power steering device 900 is mounted on a vehicle. The steering wheel 901 is operated by the driver. By operating the steering wheel 901, the front wheels 902 of the vehicle are driven. Since the basic structure of the control device 5 is the same as that of the control device 1 in Embodiment 1, the detailed description is omitted, and the description will be centered on the differences.

[0130] The torque detector 903 detects the steering torque Tst of the driver on the steering wheel 901 and outputs the detection result to the control device 5. The driving force of the AC motor 10 is transmitted to the front wheels 902 etc. via the driving force transmission mechanism 904. The electric power steering device 900 uses the driving force generated by the AC motor 10 as an auxiliary torque in vehicle steering to assist the vehicle steering based on the driver.

[0131] The control device 5 has a current command value calculator 501. The steering torque Tst, the traveling speed S of the vehicle, etc. are input to the current command value calculator 501. The current command value calculator 501 calculates the torque current command value iq_ref and the field weakening current command value id_ref based on these inputs. In the calculation of the current command value calculator 501, known vibration suppression control, viscous compensation control, inertia compensation control, etc. can also be used in combination.

[0132] As the AC motor 10 for the electric power steering device 900, a brushless motor, a permanent magnet synchronous motor, etc. are used. The resonance period Tr of the frame of the AC motor 10, or the power unit in which the AC motor 10 and the inverter 12 are integrated, or the entire electric power steering device 900 is, for example, 200 μs or more and 500 μs or less. That is, the resonance frequency fr of these structures can also be 2 kHz or more and 5 kHz or less. In addition, the resonance period Tr is more preferably 300 μs or more and 400 μs or less. That is, the resonance frequency fr can also be 2.5 kHz or more and 3.3 kHz or less. The current control response (cut-off frequency) of the AC motor 10 is, for example, 100 Hz or more and 1250 Hz or less, or 200 Hz or more and 800 Hz or less. The period Tc of the carrier triangular wave C is 50 μs or more and 60 μs or less.

[0133] According to the control devices 1 to 5 or the electric power steering device 900 according to Embodiments 1 to 5, even when having the resonance period Tr and the like as described above, it is possible to highly accurately remove the resonance frequency component included in the detected current. The structure according to the present disclosure is suitable for a vehicle that requires a high-quality steering feeling, or a vehicle in which a driver requires quietness when rapidly steering the steering wheel 901 (the AC motor 10 rotates at a high speed).

[0134] The above has described Embodiments 1 to 5, but the present disclosure is not limited to the above embodiments and can be freely changed without departing from the gist of the present disclosure. In addition, the above Embodiments 1 to 5 can be appropriately combined. For example, the techniques described in Embodiments 1 to 4 can also be applied to the electric power steering device 900 described in Embodiment 5. Or, the control devices 1 to 5 can be applied to the control of the AC motor 10 other than the electric power steering device.

[0135] Each structure included in the above-described AC motor control devices 1 to 5 and the electric power steering device 900 has a computer system inside. Moreover, a program for realizing the functions of each structure included in the above-described AC motor control devices 1 to 5 and the electric power steering device 900 can be recorded in a computer-readable recording medium, and the program recorded in the recording medium is read and executed by the computer system, thereby performing processing in each structure included in the above-described AC motor control devices 1 to 5 and the electric power steering device 900. Here, "the program recorded in the recording medium is read and executed by the computer system" includes installing the program in the computer system. The "computer system" mentioned here includes hardware such as an OS and peripheral devices.

[0136] In addition, a "computer system" may include a plurality of computer devices connected via a network including communication lines such as the Internet, WAN, LAN, dedicated lines, etc. Further, a "computer-readable recording medium" refers to portable media such as floppy disks, magneto-optical disks, ROMs, CD-ROMs, and storage devices such as hard disks built into a computer system. Thus, a recording medium storing a program may also be a non-transitory recording medium such as a CD-ROM.

[0137] In addition, the recording medium further includes an internal or external recording medium that can be accessed from a distribution server to distribute the program. Further, it may be a structure in which the program is divided into a plurality of parts, downloaded at different timings, and then combined by each structure included in the control devices 1 to 5 of the AC motor and the electric power steering device 900, and the distribution servers for distributing the divided programs may also be different. Further, a "computer-readable recording medium" also includes a medium that holds a program for a certain period of time, such as a volatile memory (RAM) inside a computer system that becomes a server or a client when sending a program through a network. Further, the above program may also be a program for implementing a part of the above functions. And it may also be a so-called differential file (differential program) that can implement the above functions through combination with a program already recorded in a computer system. Reference Numeral Explanation

[0138] 1 - 5 Control Devices 10 AC Motor 12 Inverter 12u, 12v, 12w Conductive Lines 15, 15b Second Controllers 21 Two-Axis Current Filter 22 Current Detector 23 First Coordinate Converter 24 First Controller 28 Second Coordinate Converter 401 Weighted Average Processing Unit 900 Electric Power Steering Device.

Claims

1. A control device for an AC motor, characterized in that, it includes: a current detector that detects the currents of three phases supplied to the AC motor; a first coordinate converter that converts the pre-conversion detected current based on the detection result of the current detector into a two-axis detected current that is the current on the rotating two axes; a two-axis current filter that reduces the noise component of the two-axis detected current; a second coordinate converter that converts the two-axis detected current after the noise component is reduced by the two-axis current filter, i.e., the filtered two-axis current, into the filtered three-phase current in the stationary coordinate system; a first controller that generates a first three-phase voltage command based on the two-axis detected current or the filtered two-axis current; a second controller that generates a second three-phase voltage command based on the filtered three-phase current; and an inverter that applies a voltage to the AC motor based on the first three-phase voltage command and the second three-phase voltage command.

2. The control device for an AC motor according to claim 1, characterized in that, the two-axis current filter is configured to calculate the filtered two-axis current using the past two-axis detected current, and when the control device of the AC motor is just started, the two-axis current filter substitutes the latest two-axis detected current as the past two-axis detected current for calculation.

3. The control device for an AC motor according to claim 1, characterized in that, when the amplitude-related physical quantity related to the voltage amplitude of the first three-phase voltage command or the voltage amplitude of the second three-phase voltage command is above the physical threshold, the first three-phase voltage command is generated based on the filtered two-axis current, and the second three-phase voltage command is generated based on the filtered three-phase current, when the amplitude-related physical quantity is less than the physical threshold, the first three-phase voltage command is generated based on the two-axis detected current, and the second three-phase voltage command is generated based on the pre-conversion detected current.

4. The control device for an AC motor according to claim 3, characterized in that, the two-axis current filter is configured to calculate the filtered two-axis current using the past two-axis detected current, and when switching from the first state where the amplitude-related physical quantity is less than the physical threshold to the second state where the amplitude-related physical quantity is above the physical threshold, the two-axis current filter substitutes the latest two-axis detected current as the past two-axis detected current for calculation.

5. The control device for an AC motor according to claim 3 or 4, characterized in that, it further includes a weighted average processing unit, and when the previous value of the amplitude-related physical quantity exceeds a second physical threshold smaller than the physical threshold, the two-axis current filter performs the calculation. When the last value of the amplitude-related physical quantity is between the second physical threshold and the physical threshold, the weighted average processing unit performs weighted average processing on the two-axis detection current and the filtered two-axis current. When the weighted average processing unit performs the weighted average processing, the greater the amplitude-related physical quantity, the higher the proportion of the filtered two-axis current is increased.

6. The control device for an AC motor according to any one of claims 1 to 5, characterized in that the second controller calculates the second three-phase voltage command using a value obtained by multiplying the current deviation in the stationary coordinate system by a proportional gain.

7. The control device for an AC motor according to any one of claims 1 to 6, characterized in that the inverter has a current-carrying wire for supplying current to the AC motor and a shunt resistor connected in series with the current-carrying wire, the current detector detects the pre-conversion detection current based on the voltage across the shunt resistor, The physical threshold is determined based on the period of the carrier triangular wave and the time threshold.

8. The control device for an AC motor according to any one of claims 1 to 7, characterized in that the two-axis current filter is a notch filter that reduces the mechanical resonance frequency of the AC motor, or a low-pass filter with a cut-off frequency below the mechanical resonance frequency.

9. An electric power steering device, characterized in that it includes: the control device for an AC motor according to any one of claims 1 to 8; and the AC motor that generates an assist torque during vehicle steering.

Images