Band-gap reference circuit, chip, and electronic device

Abstract

The embodiment of the invention provides a band-gap reference circuit, a chip and electronic equipment, the band-gap reference circuit comprises a reference module, the reference module is used for generating a first positive temperature coefficient current, a second positive temperature coefficient current and a negative temperature coefficient voltage, and the ratio of the first positive temperature coefficient current to the second positive temperature coefficient current is equal to a preset ratio; and the current control module is used for adjusting the first positive temperature coefficient current and the second positive temperature coefficient current so as to calibrate the reference voltage. The reference voltage can be calibrated by adjusting the current, and compared with a traditional resistor adjusting mode, the problem that the calibration precision of the reference voltage is limited due to the fact that the minimum adjusting resistance value of a resistor is limited by the technology can be solved.

Description

Technical Field

[0001] The present application relates to the field of integrated circuit technology, and particularly to a bandgap reference circuit, a chip and an electronic device. Background Art

[0002] A bandgap voltage reference (BGR) can output a reference voltage that is substantially independent of the power supply voltage, process, and temperature. Due to its unique advantages, this circuit is widely used in various systems (such as voltage regulators, analog-to-digital converters, data acquisition systems, digital-to-analog converters, MCUs) to provide a stable reference voltage for the system.

[0003] In the related art, a bandgap reference circuit usually generates a positive temperature coefficient current and a negative temperature coefficient voltage. The positive temperature coefficient current flows through an adjustable resistor to generate a positive temperature coefficient voltage. After superimposing the positive temperature coefficient voltage and the negative temperature coefficient voltage, a reference voltage with a temperature coefficient close to 0 can be generated by adjusting the resistance value of the adjustable resistor. However, since the calibration accuracy of the reference voltage is affected by the minimum adjustment resistance value of the adjustable resistor, and the minimum adjustment resistance value of the adjustable resistor is limited by process factors such as process mismatch and process corner variation, this leads to the problem of limited calibration accuracy of the current reference voltage. Summary of the Invention

[0004] In view of the above problems, embodiments of the present application provide a bandgap reference circuit, a chip and an electronic device to solve the above technical problems.

[0005] In a first aspect, an embodiment of the present application provides a bandgap reference circuit. The bandgap reference circuit generates a reference voltage based on a negative temperature coefficient voltage and a positive temperature coefficient current. The bandgap reference circuit includes:

[0006] A reference module for generating a first positive temperature coefficient current, a second positive temperature coefficient current, and a negative temperature coefficient voltage, where the ratio of the first positive temperature coefficient current to the second positive temperature coefficient current is equal to a preset ratio;

[0007] A current control module for adjusting the magnitudes of the first positive temperature coefficient current and the second positive temperature coefficient current to calibrate the reference voltage;

[0008] Wherein, after the current control module adjusts the first positive temperature coefficient current and the second positive temperature coefficient current, the ratio of the first positive temperature coefficient current to the second positive temperature coefficient current remains unchanged at the preset ratio.

[0009] In a second aspect, an embodiment of the present application further provides a chip, including the above bandgap reference circuit.

[0010] In a third aspect, an embodiment of the present application further provides an electronic device, including the above-mentioned chip or bandgap reference circuit.

[0011] In the present application, a first positive temperature coefficient current, a second positive temperature coefficient current, and a negative temperature coefficient voltage are generated by a reference module. Since the bandgap reference circuit generates a reference voltage based on the negative temperature coefficient voltage and the positive temperature coefficient current, when the magnitudes of the first positive temperature coefficient current and the second positive temperature coefficient current are adjusted, the purpose of calibrating the reference voltage by adjusting the current magnitude can be achieved. Compared with the traditional method of adjusting the resistance magnitude, the present application can avoid the problem that the calibration accuracy of the reference voltage is limited due to the minimum adjustable resistance value of the resistor being restricted by the process.

[0012] These aspects or other aspects of the present application will be more clearly understood in the following description of the embodiments. BRIEF DESCRIPTION OF THE DRAWINGS

[0013] In order to more clearly illustrate the technical solutions in the embodiments of the present application, the following will briefly introduce the drawings required for the description of the embodiments. Obviously, the drawings in the following description are only some embodiments of the present application. For those skilled in the art, without creative efforts, other drawings can be obtained based on these drawings.

[0014] Figure 1 Shows a schematic diagram of a bandgap reference circuit in the related art.

[0015] Figure 2 Shows a schematic diagram of a bandgap reference circuit in an embodiment of the present application.

[0016] Figure 3 Shows another schematic diagram of a bandgap reference circuit in an embodiment of the present application.

[0017] Figure 4 Shows another schematic diagram of a bandgap reference circuit in an embodiment of the present application.

[0018] Figure 5 Shows another schematic diagram of a bandgap reference circuit in an embodiment of the present application.

[0019] Figure 6 Shows another schematic diagram of a bandgap reference circuit in an embodiment of the present application.

[0020] Figure 7 Shows another schematic diagram of a bandgap reference circuit in an embodiment of the present application.

[0021] Figure 8 Shows another schematic diagram of a bandgap reference circuit in an embodiment of the present application.

[0022] Figure 9 Another schematic diagram of the bandgap reference circuit in an embodiment of the present application is shown.

[0023] Figure 10 Another schematic diagram of the bandgap reference circuit in an embodiment of the present application is shown.

[0024] Figure 11 Another schematic diagram of the bandgap reference circuit in an embodiment of the present application is shown.

[0025] Figure 12 Another schematic diagram of the bandgap reference circuit in an embodiment of the present application is shown.

[0026] Figure 13 Another schematic diagram of the bandgap reference circuit in an embodiment of the present application is shown.

[0027] Figure 14 A schematic diagram of the first current source in an embodiment of the present application is shown.

[0028] Figure 15 Another schematic diagram of the bandgap reference circuit in an embodiment of the present application is shown.

[0029] Figure 16 A schematic diagram of the transconductance amplifier in an embodiment of the present application is shown.

[0030] Wherein, 10 is the reference module, 11 is the current output unit, 20 is the current control module, and 21 is the current source unit;

[0031] The first positive temperature coefficient current I1, the second positive temperature coefficient current I2, the first negative temperature coefficient voltage VBE1, the second negative temperature coefficient voltage VBE2, the first node A, the second node B, the third node C, the fourth node D, the reference voltage VBG, the power supply terminal VDD, the first compensation current △I1, and the second compensation current △I2;

[0032] The first resistor R1, the second resistor R2, the third resistor R3, the fourth resistor R4, the fifth resistor R5, the sixth resistor R6, the seventh resistor R7, the first triode Q1, the second triode Q2, the third triode Q3, the fourth triode Q4, the first transistor M1, the second transistor M2, the third transistor M3, the fourth transistor M4, the operational amplifier OP, the transconductance amplifier OTA, the first current source IS1, and the second current source IS2. Detailed implementation manners

[0033] The following details the implementation manners of the present application. The examples of the implementation manners are shown in the drawings, where the same or similar reference numerals denote the same or similar elements or elements having the same or similar functions throughout. The implementation manners described below with reference to the drawings are exemplary only for explaining the present application and should not be construed as limiting the present application.

[0034] To enable those skilled in the art to better understand the solutions of this application, the technical solutions in the embodiments of this application will be clearly and completely described below in conjunction with the accompanying drawings in the embodiments of this application. Obviously, the described embodiments are only a part of the embodiments of this application, rather than all the embodiments. Based on the embodiments in this application, all other embodiments obtained by those skilled in the art without creative efforts fall within the scope of protection of this application.

[0035] In the embodiments of this application, it should be noted that in this text, relational terms such as first and second are only used to distinguish one entity or operation from another entity or operation, and do not necessarily require or imply any actual relationship or order between these entities or operations.

[0036] Moreover, the term "comprising", "including" or any other variant thereof is intended to cover non-exclusive inclusion, so that a process, method, article or device comprising a series of elements not only includes those elements, but also includes other elements not expressly listed, or also includes elements inherent to such process, method, article or device. Without further limitation, an element defined by the statement "comprising one..." does not exclude the existence of additional identical elements in the process, method, article or device comprising the element.

[0037] In the description of the embodiments of this application, words such as "example" or "for example" are used to represent examples, explanations or descriptions. Any embodiment or design solution described as "for example" or "for instance" in the embodiments of this application is not construed as being more preferred or having more advantages than another embodiment or design solution. The use of words such as "example" or "for example" is intended to present relative concepts in a clear manner.

[0038] In addition, "a plurality of" in the embodiments of this application means two or more. In view of this, "a plurality of" in the embodiments of this application can also be understood as "at least two". "At least one" can be understood as one or more, for example, understood as one, two or more. For example, including at least one means including one, two or more, and does not limit which ones are included. For example, including at least one of A, B, and C, then what can be included is A, B, C, A and B, A and C, B and C, or A, B, and C.

[0039] It should be noted that in the embodiments of this application, "and / or" describes the association relationship of associated objects, indicating that there can be three relationships. For example, A and / or B can represent: A exists alone, A and B exist simultaneously, and B exists alone. In addition, the character " / ", unless otherwise specified, generally represents an "or" relationship between the associated objects before and after.

[0040] It should be noted that "connection" in the embodiments of the present application can be understood as electrical connection, and the connection between two electrical components can be a direct or indirect connection between the two electrical components. For example, when A is connected to B, it can be either a direct connection between A and B or an indirect connection between A and B through one or more other electrical components.

[0041] In the embodiments of the present application, the first pole / first end of each transistor is one of the source and the drain, and the second pole / second end of each transistor is the other of the source and the drain. Since the source and the drain of a transistor can be symmetric in structure, there may be no difference between the source and the drain in terms of structure. That is to say, there may be no difference between the first pole / first end and the second pole / second end of the transistors in the embodiments of the present application in terms of structure. Exemplarily, when the transistor is a P-type transistor, the first pole / first end of the transistor is the source, and the second pole / second end is the drain; Exemplarily, when the transistor is an N-type transistor, the first pole / first end of the transistor is the drain, and the second pole / second end is the source.

[0042] In the circuit structure provided by the embodiments of the present application, nodes such as the first node and the second node do not represent actually existing components, but represent the convergence points of relevant couplings in the circuit diagram. That is to say, these nodes are nodes equivalent to the convergence points of relevant couplings in the circuit diagram.

[0043] Currently, a bandgap reference circuit usually generates a reference voltage with a temperature coefficient close to 0 by using a positive temperature coefficient current and a negative temperature coefficient voltage. Refer to Figure 1 , Figure 1 shows a schematic diagram of a bandgap reference circuit in the related art. Among them, the bandgap reference circuit includes a transistor M0, a trimming resistor Rtrim, a resistor R01, a resistor R02, a resistor R03, a triode Q1, a triode Q2, and an operational amplifier AMP.

[0044] Since the inverting input terminal of the operational amplifier AMP is connected to node A, the non-inverting input terminal of the operational amplifier is connected to node B, and the output terminal of the operational amplifier AMP is connected to the gate of the transistor M0, according to the virtual short and virtual open characteristics of the operational amplifier AMP, it can be known that the voltages of node A and node B are equal. At this time, the voltages across the resistor R02 and the resistor R03 are equal. Therefore, the ratio of the magnitudes of the currents flowing through the resistor R02 (i.e., the triode Q1) and the resistor R03 (i.e., the triode Q2) is:

[0045] I C1 / I C2 = R03 / R02

[0046] Since the voltage difference between the emitter and the base of the triode and the collector current of the triode satisfy the following formula:

[0047]

[0048] where V BE is the voltage between the emitter and the base of the triode, V T is the thermal voltage, I C is the collector current of the triode, and I S is the saturation current of the triode.

[0049] It can be known that the voltage differences V BE1 between the emitter and the base of the triode Q1 and V BE2 between the emitter and the base of the triode Q2 are respectively:

[0050]

[0051] where I C1 is the collector current of the triode Q1, I S1 is the saturation current of the triode Q1, I C2 is the collector current of the triode Q2, and I S2 is the saturation current of the triode Q2.

[0052] Then, the voltage difference across the resistor R01 is:

[0053]

[0054] Assume that R03 / R02 = 1 / 4 (i.e., I C1 / I C2 = 1 / 4), and I S1 / I S2 = 8. It can be known that the voltage difference across the resistor R01 is:

[0055] ΔV BE = V T ln32

[0056] Therefore, the magnitudes of the currents flowing through the resistors R01 and R03 are respectively:

[0057] I C1 = V T ln32 / R01

[0058] I C2 = 4*V T ln32 / R01

[0059] Since the thermal voltage V Tis a positive temperature coefficient voltage. Therefore, the currents flowing through resistors R01 and R03 are both positive temperature coefficient currents. At this time, the reference voltage VBG output from the source of transistor M0 can be calculated by the following formula:

[0060] VBG = V EB2 + I C2 * R03 + (I C1 + I C2 ) * Rtrim

[0061] It can be seen from the above formula that the V BE2 term is a negative temperature coefficient voltage, the I C2 * R03 term and the (I C1 + I C2 ) * Rtrim term are positive temperature coefficient voltages. Therefore, by adjusting the resistance value of the adjustable resistor Rtrim, a reference voltage with a temperature coefficient close to 0 can be finally output.

[0062] However, since the calibration accuracy of the reference voltage is affected by the minimum adjustment resistance value of the adjustable resistor, and the minimum adjustment resistance value of the adjustable resistor is limited by process factors such as process mismatch and process corner variation, this leads to the problem of limited calibration accuracy of the current reference voltage.

[0063] For this reason, the present application provides a bandgap reference circuit, a chip and an electronic device, which will be described in detail below.

[0064] First, refer to Figure 2 , Figure 2 shows a schematic diagram of a bandgap reference circuit in an embodiment of the present application. Among them, the bandgap reference circuit includes a reference module 10 and a current control module 20.

[0065] Specifically, the reference module 10 can generate a first positive temperature coefficient current I1, a second positive temperature coefficient current I2, and a negative temperature coefficient voltage, so that the bandgap reference circuit can generate a reference voltage based on the negative temperature coefficient voltage and the positive temperature coefficient current. Generally, the negative temperature coefficient voltage includes a first negative temperature coefficient voltage VBE1 and a second negative temperature coefficient voltage VBE2. The reference module 10 can include two triodes. Since the base-emitter voltage of the triode is a negative temperature coefficient voltage, the reference module 10 can generate a first negative temperature coefficient voltage VBE1 and a second negative temperature coefficient voltage VBE2.

[0066] In some embodiments of the present application, refer to Figure 3 , Figure 3Another schematic diagram of the bandgap reference circuit in the embodiment of the present application is shown. The reference module 10 includes a current output unit 11, a first resistor R1, a first triode Q1, and a second triode Q2. The first connection end of the current output unit 11 is connected to the first end of the first resistor R1. The second end of the first resistor R1 is connected to the emitter of the first triode Q1. The collector and base of the first triode Q1 are connected to the ground terminal. The second connection end of the current output unit 11 is connected to the emitter of the second triode Q2. The collector and base of the second triode Q2 are connected to the ground terminal.

[0067] It should be noted that assuming that the current control module 20 is not working, the first connection end of the current output unit 11 outputs a first positive temperature coefficient current I1 flowing through the first resistor and the first triode Q. The second connection end of the current output unit 11 outputs a second positive temperature coefficient current I2 flowing through the second triode Q2. According to the voltage difference between the emitter and the base of the triode and the collector current formula of the triode, the voltage differences between the emitters and the bases of the first triode Q1 and the second triode Q2 are as follows:

[0068]

[0069] Among them, V BE1 is the voltage difference between the emitter and the base of the first triode Q1, I S1 is the saturation current of the first triode Q1, V BE2 is the voltage difference between the emitter and the base of the second triode Q2, V T is the thermal voltage, and I S2 is the saturation current of the second triode Q2.

[0070] When the current output unit 11 is working, the voltages at the first connection end and the second connection end of the current output unit 11 are equal. It can be known that the magnitude of the first positive temperature coefficient current I1 can be calculated according to the following formula:

[0071]

[0072] Assuming I S1 :I S2 = 8, and the preset ratio of the first positive temperature coefficient current I1 to the second positive temperature coefficient current I2 is I1:I2 = 1:4. It can be known that:

[0073] I1 = (V BE2 -V BE1 ) / R1 = V T ln32 / R1

[0074] I2 = 4*I1 = 4*V T ln32 / R1

[0075] It can be seen that due to the thermal voltage V T being a positive temperature coefficient voltage, the first positive temperature coefficient current I1 and the second positive temperature coefficient current I2 are both positive temperature coefficient currents. At the same time, according to the fact that the voltage difference between the emitter and the base of the triode is a negative temperature coefficient voltage, it can be known that the voltage difference V BE1 between the emitter and the base of the first triode Q1 (i.e., the first negative temperature coefficient voltage VBE1) and the voltage difference V BE2 between the emitter and the base of the second triode Q2 (i.e., the second negative temperature coefficient voltage VBE2) are both negative temperature coefficient voltages.

[0076] As an exemplary embodiment of the current output unit 11, refer to Figure 4 , Figure 4 which shows another schematic diagram of the bandgap reference circuit in the embodiment of the present application. Among them, the current output unit 11 includes a first transistor M1, a fourth resistor R4, a fifth resistor R5, a sixth resistor R6, and an operational amplifier OP. The first end of the first transistor M1 is connected to the power supply terminal VDD, and the second end of the first transistor M1 is connected to the first end of the fourth resistor R4. The first end of the fifth resistor R5 is connected to the second end of the fourth resistor R4, and the second end of the fifth resistor R5 is connected to the first end of the first resistor R1. The first end of the sixth resistor R6 is connected to the second end of the fourth resistor R4, and the second end of the sixth resistor R6 is connected to the emitter of the second triode Q2. The first input terminal of the operational amplifier OP is connected to the first node A between the fifth resistor R5 and the first resistor R1, the second input terminal of the operational amplifier OP is connected to the second node B between the sixth resistor R6 and the second triode Q2, and the output terminal of the operational amplifier OP is connected to the control terminal of the first transistor M1.

[0077] It should be noted that in Figure 4 , the first node A serves as the first connection end of the current output unit 11, the second node B serves as the second connection end of the current output unit 11. The first positive temperature coefficient current I1 generated by the reference module 10 refers to the current flowing through the fifth resistor R5, and the second positive temperature coefficient current I2 generated by the reference module 10 refers to the current flowing through the sixth resistor R6. According to the virtual short and virtual open characteristics of the operational amplifier OP, it can be known that the voltages of the first node A and the second node B are equal. Since both the fifth resistor R5 and the sixth resistor R6 are connected to the fourth node D, it can be known that the voltage differences across the fifth resistor R5 and the sixth resistor R6 are equal. Therefore, the first positive temperature coefficient current I1 output through the fifth resistor R5 and the second positive temperature coefficient current I2 output through the sixth resistor R6 satisfy the following relationship:

[0078] I1 = U / R5

[0079] I2 = U / R6

[0080] I1:I2 = R6:R5

[0081] Wherein, U is the voltage difference across the fifth resistor R5 and the sixth resistor R6.

[0082] Therefore, when the resistance ratio of the sixth resistor R6 to the fifth resistor R5 is equal to a preset ratio, the ratio of the first positive temperature coefficient current I1 to the second positive temperature coefficient current I2 can be made equal to the preset ratio. For example, taking the preset ratio of 1:4 as an example, if R6:R5 = 1:4, then I1:I2 = 1:4.

[0083] As another exemplary embodiment of the current output unit 11, refer to Figure 5 , Figure 5 which shows another schematic diagram of the bandgap reference circuit in the embodiment of the present application. Among them, the current output unit 11 includes a first transistor M1, a second transistor M2, and an operational amplifier OP; the first end of the first transistor M1 is connected to the power supply terminal VDD, the second end of the first transistor M1 is connected to the first end of the first resistor R1, and the control end of the first transistor M1 is connected to the output end of the operational amplifier OP; the first end of the second transistor M2 is connected to the power supply terminal VDD, the second end of the second transistor M2 is connected to the emitter of the second triode Q2, and the control end of the second transistor M2 is connected to the output end of the operational amplifier OP; the first input terminal of the operational amplifier OP is connected to the first node A between the first transistor M1 and the first resistor R1, and the second input terminal of the operational amplifier OP is connected to the second node B between the second transistor M2 and the second triode Q2.

[0084] It should be noted that in Figure 5 , the first node A serves as the first connection end of the current output unit 11, the second node B serves as the second connection end of the current output unit 11, the first positive temperature coefficient current I1 generated by the reference module 10 refers to the current flowing through the first transistor M1, and the second positive temperature coefficient current I2 generated by the reference module 10 refers to the current flowing through the second transistor M2. According to the virtual short and virtual open characteristics of the operational amplifier OP, it can be known that the voltages of the first node A and the second node B are equal. At the same time, since the output end of the operational amplifier OP is connected to the control ends of the first transistor M1 and the second transistor M2, the source voltages and gate voltages of the first transistor M1 and the second transistor M2 are equal. According to the MOS transistor current calculation formula, the magnitudes of the first positive temperature coefficient current I1 output by the first transistor M1 and the second positive temperature coefficient current I2 output by the second transistor M2 satisfy the following relationship:

[0085] I1:I2 = (W1 / L1):(W2 / L2)

[0086] Wherein, W1 / L1 is the aspect ratio of the first transistor M1, and W2 / L2 is the aspect ratio of the second transistor M2.

[0087] It can be seen that when the ratio between the aspect ratio of the first transistor M1 and the aspect ratio of the second transistor M2 is equal to a preset ratio, the ratio between the first positive temperature coefficient current I1 and the second positive temperature coefficient current I2 can also be made equal to the preset ratio.

[0088] Those skilled in the art can understand that the implementation manner of the bandgap reference circuit is not limited to this. For example, referring to Figure 6 , Figure 6 FIG. shows another schematic diagram of the bandgap reference circuit in the embodiment of the present application. The current output unit 11 may also adopt the first transistor M1 and the second transistor M2 that are current mirrors of each other, and the third transistor M3 and the fourth transistor M4 that are current mirrors of each other. The voltages of the first node A and the second node B are clamped to be equal through two current mirrors, and at the same time, by controlling the mirror ratio of the two current mirrors, the ratio between the first positive temperature coefficient current I1 and the second positive temperature coefficient current I2 is made equal to the preset ratio; for another example, referring to Figure 7 , Figure 7 FIG. shows another schematic diagram of the bandgap reference circuit in the embodiment of the present application. The first triode Q1 and the second triode Q2 may also be respectively replaced by the diode D1 and the diode D2; for another example, the first resistor R1 may be replaced by a plurality of resistors connected in series or in parallel.

[0089] The current control module 20 is used to adjust the magnitudes of the first positive temperature coefficient current I1 and the second positive temperature coefficient current I2. After the current control module 20 adjusts the first positive temperature coefficient current I1 and the second positive temperature coefficient current I2, the ratio between the first positive temperature coefficient current I1 and the second positive temperature coefficient current I2 remains unchanged as the preset ratio. For example, the ratio between the first positive temperature coefficient current I1 and the second positive temperature coefficient current I2 is fixed at 1:4 and remains unchanged. Since the bandgap reference circuit generates the reference voltage VBG based on the negative temperature coefficient voltage and the positive temperature coefficient current, the reference voltage VBG can be calibrated through the current control module 20.

[0090] For example, in the bandgap reference circuit of the present application, the calculation formula of the reference voltage VBG may be:

[0091] VBG = VBE + IPTAT * R

[0092] In the above formula, VBE is the first negative temperature coefficient voltage VBE1 or the second negative temperature coefficient voltage VBE2, IPTAT is the positive temperature coefficient current, and the magnitude of the IPTAT current is determined based on the magnitudes of the first positive temperature coefficient current I1 and the second positive temperature coefficient current I2. It can be seen that when the current control module 20 adjusts the magnitudes of the first positive temperature coefficient current I1 and the second positive temperature coefficient current I2, the reference voltage VBG output by the bandgap reference circuit can be calibrated.

[0093] In the embodiments of the present application, the present application generates the first positive temperature coefficient current I1, the second positive temperature coefficient current I2, the first negative temperature coefficient voltage VBE1, and the second negative temperature coefficient voltage VBE2 through the reference module 10. Since the bandgap reference circuit generates the reference voltage VBG based on the negative temperature coefficient voltage and the positive temperature coefficient current, after the current control module 20 adjusts the magnitudes of the first positive temperature coefficient current I1 and the second positive temperature coefficient current I2, the purpose of calibrating the reference voltage VBG by adjusting the current magnitude can be achieved. Compared with the traditional method of adjusting the resistor magnitude, it is beneficial to avoid the problem that the calibration accuracy of the reference voltage is limited due to the minimum adjustment resistance value of the resistor being restricted by the process.

[0094] In some embodiments of the present application, the current control module 20 is configured to input or extract the first compensation current △I1 and the second compensation current △I2 to the reference module 10. The first negative temperature coefficient voltage VBE1 is generated based on the first positive temperature coefficient current I1 and the first compensation current △I1, and the second negative temperature coefficient voltage VBE2 is generated based on the second positive temperature coefficient current I2 and the second compensation current △I2; wherein, the ratio of the first compensation current △I1 to the second compensation current △I2 is equal to a preset ratio.

[0095] For example, referring to Figure 8 , Figure 8 shows another schematic diagram of the bandgap reference circuit in the embodiments of the present application. Among them, the current control module 20 inputs the first compensation current △I1 to the first node A and inputs the second compensation current △I2 to the second node B. Then, the total current I01 flowing through the first triode Q1 is: I01 = I1 + △I1, and the total current I02 flowing through the second triode Q2 is: I02 = I2 + △I2. Since the ratio of the first compensation current △I1 to the second compensation current △I2 is equal to the preset ratio, and the ratio of the first positive temperature coefficient current I1 to the second positive temperature coefficient current I2 is equal to the preset ratio, it can be known that:

[0096] I01:I02 = △I1:△I2 = I1:I2 = K

[0097] wherein, K is the preset ratio.

[0098] According to the foregoing, the total current I01 flowing through the first triode Q1 and the total current I02 flowing through the second triode Q2 can be calculated by the following formula:

[0099]

[0100] It can be seen that the magnitudes of the total current I01 flowing through the first triode Q1 and the total current I02 flowing through the second triode Q2 are related to the ratio of I02:I01. Since the ratio of the first compensation current △I1 to the second compensation current △I2 is equal to the preset ratio, and the ratio of the first positive temperature coefficient current I1 to the second positive temperature coefficient current I2 is equal to the preset ratio, this can make the ratio of I02:I01 equal to the preset ratio and remain unchanged. Therefore, the total current I01 flowing through the first triode Q1 and the total current I02 flowing through the second triode Q2 remain unchanged.

[0101] For example, assume I1:I2 = 1:4 (i.e., the preset ratio is 1:4), I S1 :I S2 = 8, it can be known that:

[0102] I01 = (V BE1 - V BE2 ) / R1 = V T ln32 / R1

[0103] I02 = 4 * I1 = 4 * V T ln32 / R1

[0104] Therefore, the first positive temperature coefficient current I1 and the second positive temperature coefficient current I2 can be calculated by the following formula:

[0105] I1 = V T ln32 / R1 - △I1

[0106] I2 = 4 * V T ln32 / R1 - △I2

[0107] It can be seen that when the ratio of the first compensation current △I1 to the second compensation current △I2 input or extracted by the current control module 20 is also equal to the preset ratio, when the current control module 20 adjusts the magnitudes of the first compensation current △I1 and the second compensation current △I2, the magnitudes of the first positive temperature coefficient current I1 and the second positive temperature coefficient current I2 can be adjusted, and finally the calibration of the bandgap reference voltage VBG can be achieved by adjusting the current magnitude.

[0108] Another example, refer to Figure 9 , Figure 9Another schematic diagram of the bandgap reference circuit in the embodiment of the present application is shown. After the current control module 20 inputs a current to the fourth node D through a current source, the magnitudes of the first compensation current △I1 input to the first triode Q1 and the first compensation current △I2 input to the second triode Q2 are as follows:

[0109] △I1 = △I * R6 / (R5 + R6)

[0110] △I2 = △I * R5 / (R5 + R6)

[0111] Therefore, the ratio between the first compensation current △I1 and the first compensation current △I2 is as follows:

[0112] △I1:△I2 = R6:R5

[0113] Combined with I1:I2 = R6:R5, I01 = I1 + △I1, and I02 = I2 + △I2, it can be known that:

[0114] I01:I02 = (I1 + △I1):(I2 + △I2) = R6:R5

[0115] It can be seen that by inputting a current to the fourth node D through a current source, the ratio between the total current I01 flowing through the first triode Q1 and the total current I02 flowing through the second triode Q2 can also be made equal to a preset ratio. Thus, after the current control module 20 adjusts the magnitudes of the first compensation current △I1 and the second compensation current △I2, the magnitudes of the first positive temperature coefficient current I1 and the second positive temperature coefficient current I2 can be adjusted, and finally, the calibration of the bandgap reference voltage VBG can be achieved by adjusting the current magnitude.

[0116] In the embodiment of the present application, since the current control module 20 can adjust the magnitudes of the first positive temperature coefficient current I1 and the second positive temperature coefficient current I2, the calibration of the reference voltage VBG can be achieved by adjusting the current.

[0117] For example, referring to Figure 8 , according to the resistor voltage division, the reference voltage VBG can be calculated by the following formula:

[0118] VBG = V BE2 + I2 * R6 + (I1 + I2) * R4

[0119] Substituting the foregoing calculation formulas of I1 and I2 into the above formula, we can obtain:

[0120] VBG = V BE2 +(4 * V T ln32 / R1 - △I2) * R6 + (5 * V T ln32 / R1 - △I1 - △I2) * R4

[0121] Assuming that the ΔI1 and ΔI2 currents output by the current control module 20 are zero temperature coefficient currents or negative temperature coefficient currents, it can be seen that at this time, the V BE2 term is a negative temperature coefficient voltage, the I2*R6 term and the (I1 + I2)*R4 term are positive temperature coefficient voltages. Since the magnitudes of ΔI1 and ΔI2 are controlled by the current control module 20, the reference voltage VBG can be calibrated by adjusting the current.

[0122] For another example, refer to Figure 9 , according to the resistor voltage division, the reference voltage VBG can be calculated by the following formula:

[0123] VBG = V BE2 +(I2 + ΔI2)*R6+(I1 + I2)*R4

[0124] Substituting the aforementioned calculation formulas of I1, I2, and I02 into the above formula, we can obtain:

[0125] VBG = V BE2 +(4*V T ln32 / R1)*R6+(5*V T ln32 / R1 - ΔI)*R4

[0126] Assuming that the ΔI current output by the current control module 20 is a zero temperature coefficient current or a negative temperature coefficient current, it can be seen that at this time, the VBE2 term is a negative temperature coefficient voltage, the (I2 + ΔI2)*R6 term and the (I1 + I2)*R4 term are positive temperature coefficient voltages. Since the magnitude of ΔI is controlled by the current control module 20, the reference voltage VBG can be calibrated by adjusting the current.

[0127] For yet another example, refer to Figure 10 , Figure 10 shows another schematic diagram of the bandgap reference circuit in the embodiment of the present application. The bandgap reference circuit further includes a third transistor M3, a seventh resistor R7, and a fourth triode Q4; the first end of the third transistor M3 is connected to the power supply terminal VDD, the second end of the third transistor M3 is connected to the first end of the seventh resistor R7, and the control end of the third transistor M3 is connected to the output end of the operational amplifier OP; the first end of the fourth triode Q4 is connected to the second end of the seventh resistor R7, the second end of the fourth triode Q4 is connected to the ground terminal, and the base of the fourth triode Q4 is connected to the ground terminal.

[0128] In Figure 10 , if the ratio between the total current I01 flowing through the first triode Q1 and the total current I02 flowing through the second triode Q2 is equal to 1:4, and assuming I S1 :I S2= 8, it can be known that:

[0129] I01 = (V BE1 - V BE2 ) / R1 = V T ln32 / R1

[0130] I02 = 4 * I1 = 4 * V T ln32 / R1

[0131] I1 = I01 - △I1 = V T ln32 / R1 - △I1

[0132] I2 = I02 - △I2 = 4 * V T ln32 / R1 - △I2

[0133] Assume that the ratio of the aspect ratio of the first transistor M1 to the aspect ratio of the second transistor M2 is 1:4, and the ratio of the aspect ratio of the third transistor M3 to the aspect ratio of the second transistor M2 is 1:1. It can be known that:

[0134] I3 = I2 = 4 * V T ln32 / R1 - △I2

[0135] Wherein, I3 is the current flowing through the third transistor M3.

[0136] Therefore, according to the resistor voltage division, it can be known that the reference voltage VBG can be calculated by the following formula:

[0137] VBG = V BE4 + I3 * R7

[0138] VBG = V BE4 +(4 * V T ln32 / R1 - △I2) * R7

[0139] Wherein, VBE4 is the voltage difference between the emitter and the base of the fourth triode Q4.

[0140] Assume that the △I2 current output by the current control module 20 is a zero temperature coefficient current or a negative temperature coefficient current. It can be seen that at this time, the VBE4 term is a negative temperature coefficient voltage, and the I3 * R7 term is a positive temperature coefficient voltage. Since the magnitude of △I2 is controlled by the current control module 20, the reference voltage VBG can be calibrated by adjusting the current.

[0141] In summary, when the magnitudes of the first compensation current ΔI1 and the second compensation current ΔI2 are equal to the preset ratio, the total current I01 flowing through the first triode Q1 and the total current I02 flowing through the second triode Q2 remain unchanged at this time. Since the current control module 20 can control the magnitudes of the first compensation current ΔI1 and the second compensation current ΔI2, the current control module 20 can indirectly change the magnitudes of the first positive temperature coefficient current I1 and the second positive temperature coefficient current I2, thereby ultimately achieving the purpose of calibrating the reference voltage VBG by adjusting the current.

[0142] As an exemplary embodiment of the current control module 20, refer to Figure 11 , Figure 11 which shows another schematic diagram of the bandgap reference circuit in the embodiment of the present application. Among them, the current control module 20 includes a first current source IS1 and a second current source IS2. The first current source IS1 is connected to the reference module 10 (the first node A). Therefore, the first current source IS1 can input or extract the first compensation current ΔI1 to the first triode Q1; the second current source IS2 is connected to the reference module 10 (the second node B). Therefore, the second current source IS2 can input or extract the second compensation current ΔI2 to the second triode Q2.

[0143] As another exemplary embodiment of the current control module 20, refer to Figure 12 , Figure 12 which shows another schematic diagram of the bandgap reference circuit in the embodiment of the present application. Among them, the current control module 20 includes a second resistor R2, a third resistor R3, and a current source unit 21; the current source unit 21 is connected to the first end of the second resistor R2, and the second end of the second resistor R2 is connected to the reference module 10 (the first node A) to input or extract the first compensation current ΔI1 to the reference module 10 (the first node A) through the second resistor R2; the current source unit 21 is connected to the first end of the third resistor R3, and the second end of the third resistor R3 is connected to the reference module (the second node B) to input or extract the second compensation current ΔI2 to the reference module 10 through the third resistor R3; the ratio of the third resistor R3 to the second resistor R2 is equal to the preset ratio.

[0144] It should be noted that since the operational amplifier OP clamps the voltages of the first node A and the second node B to be equal, and both the second resistor R2 and the third resistor R3 are connected to the third node C, the voltages across the second resistor R2 and the third resistor R3 are equal. Therefore, the magnitude of the first compensation current ΔI1 flowing through the second resistor R2 and the magnitude of the second compensation current ΔI2 flowing through the third resistor R3 satisfy the following relationship:

[0145] ΔI1:ΔI2 = R3:R2

[0146] It can be seen that when the ratio of the third resistor R3 to the second resistor R2 is equal to a preset ratio, the ratio of the first compensation current △I1 to the second compensation current △I2 can also be made equal to the preset ratio.

[0147] In some embodiments of the present application, referring to Figure 13 , Figure 13 shows another schematic diagram of the bandgap reference circuit in the embodiments of the present application. Among them, the current source unit 21 includes a third triode Q3 and a first current source IS1; the first end of the third triode Q3 is connected to the third node C of the second resistor R2 and the third resistor R3, the second end of the third triode Q3 is connected to the ground terminal, and the base of the third triode Q3 is connected to the ground terminal; the first current source IS1 is connected to the first end of the third triode Q3 to inject current into the third triode Q3 and change the voltage of the third node C.

[0148] Specifically, when the first current source IS1 injects current into the third triode Q3, the voltage of the third node C will be changed. If the voltage of the third node C is greater than the voltages of the first node A and the second node B, then the third node C will input the first compensation current △I1 and the second compensation current △I2 to the first node A and the second node B respectively; conversely, if the voltage of the third node C is less than the voltages of the first node A and the second node B, then the third node C will extract the first compensation current △I1 and the second compensation current △I2 from the first node A and the second node B respectively, so as to change the input current IBG of the current output unit 11.

[0149] As an exemplary embodiment of the first current source IS1, referring to Figure 14 , Figure 14 shows a schematic diagram of the first current source IS1 in the embodiments of the present application. Among them, the first current source IS1 includes a first operational amplifier OP0, a standard resistor R0, an NMOS transistor MN1, a PMOS transistor MP1, and a plurality of PMOS transistors MP2. The output terminal of the first operational amplifier OP0 is connected to the gate of the NMOS transistor MN1, the inverting input terminal of the first operational amplifier OP0 is connected to the source of the NMOS transistor MN1, and the non-inverting input terminal of the first operational amplifier OP0 is connected to the reference voltage VBG.

[0150] According to the virtual short and virtual open characteristics of the operational amplifier OP, it can be known that the source voltage of the NMOS transistor MN1 is the reference voltage VBG. Therefore, the magnitude of the current flowing through the standard resistor R0 is:

[0151] I = VBG / R0

[0152] Meanwhile, since PMOS transistor MP1 and PMOS transistor MP2 are current mirrors of each other, assuming that the sizes of PMOS transistor MP1 and each PMOS transistor MP2 are equal, the output current Itrim of multiple PMOS transistors MP2 is as follows:

[0153] Itrim = N * VBG / R0

[0154] Where N is the number of PMOS transistors MP2 connected to the circuit.

[0155] It can be seen that by changing the number of PMOS transistors MP2, the magnitude of the current output by the first current source IS1 can be changed. Therefore, the present application can calibrate the bandgap reference circuit by changing the magnitude of the current output by the first current source IS1.

[0156] In some embodiments of the present application, referring to Figure 15 , Figure 15 shows another schematic diagram of the bandgap reference circuit in the embodiments of the present application. Among them, the bandgap reference circuit further includes a transconductance amplifier OTA; the first input terminal of the transconductance amplifier OTA is connected to the third node C, the second input terminal of the transconductance amplifier OTA is connected to the second node B, and the output terminal of the transconductance amplifier OTA is connected to the first end of the first transistor M1.

[0157] It should be noted that the transconductance amplifier OTA can output a current signal according to the voltage signals of the second node B and the third node C, and the magnitude of this current signal satisfies the following relational expression:

[0158] I0 = △I1 + △I2

[0159] According to Kirchhoff's current law, it can be known that the input current IBG of the current output unit in Figure 15 satisfies the following relational expression:

[0160] IBG = I01 + I02 + △I1 + △I2

[0161] IBG = Iin + I0

[0162] Where Iin is the magnitude of the current output by the power supply terminal VDD, and I0 is the magnitude of the current output by the transconductance amplifier OTA.

[0163] Therefore, the magnitude of the current output by the power supply terminal VDD satisfies the following relational expression:

[0164] Iin = IBG - I0 = I01 + I02

[0165] Since the ratio between the total current I01 flowing through the first triode Q1 and the total current I02 flowing through the second triode Q2 is equal to a preset ratio, and the magnitudes of the total current I01 flowing through the first triode Q1 and the total current I02 flowing through the second triode Q2 are unchanged, the magnitude of the current Iin output by the power supply terminal VDD also remains unchanged.

[0166] That is to say, although the magnitude of the input current IBG of the current output unit 11 is changed by the current control module 20, the transconductance amplifier OTA can make the magnitude of the current output by the power supply terminal VDD be a constant positive temperature coefficient current (such as 5 * V T ln32 / R1). Therefore, other circuit structures (such as a current mirror circuit) can also be arranged between the power supply terminal VDD and the first end of the first transistor M1 to output a positive temperature coefficient current in a scenario where the positive temperature coefficient current is required.

[0167] As an exemplary embodiment of a transconductance amplifier OTA, refer to Figure 16 , Figure 16 FIG. shows a schematic diagram of a transconductance amplifier OTA in an embodiment of the present application. The transconductance amplifier OTA includes a second operational amplifier OP1, a third operational amplifier OP2, transistors M01 to M08, and a resistor RL. Among them, transistor M01 and transistor M02 are current mirrors of each other. Transistors M03 and M04 are connected to a bias voltage VBP to bias the currents of input transistors M05 and M06; transistors M07 and M08 are connected to a bias voltage VBN to bias the currents flowing through transistors M05 and M06.

[0168] The output terminal of the second operational amplifier OP1 is connected to the gate of transistor M05. The inverting input terminal of the second operational amplifier OP1 is connected to the m1 node. The non-inverting input terminal of the second operational amplifier OP1 is connected to the VP voltage or the VM voltage under the control of the control signals Φ1 and Φ2. One of the VP voltage and the VM voltage is the voltage of the second node B, and the other is the voltage of the third node C; the output terminal of the third operational amplifier OP2 is connected to the gate of transistor M06. The inverting input terminal of the third operational amplifier OP2 is connected to the m2 node. The non-inverting input terminal of the third operational amplifier OP is connected to the VP voltage or the VM voltage under the control of the control signals Φ1 and Φ2. One end of the resistor RL is connected to the m1 node, and the other end is connected to the m2 node.

[0169] According to the virtual short and virtual open characteristics of the operational amplifier OP, during the operation of the transconductance amplifier OTA, the second operational amplifier OP1 clamps the voltage of the m1 node to the VP voltage or the VM voltage, and the third operational amplifier OP2 clamps the voltage of the m2 node to the VP voltage or the VM voltage. Assuming that the voltage of the m1 node is equal to the voltage of the third node C and the voltage of the m2 node is equal to the voltage of the second node B, the current flowing from the m1 node through the resistor RL into the m2 node is:

[0170] IL = (VC - VB) / RL

[0171] Combined with the fact that the voltage of the second node B and the voltage of the third node C satisfy the formula VC - VB = △I2 * R3, it can be known that:

[0172] IL = △I2 * R3 / RL

[0173] At the same time, since the transistor M01 and the transistor M02 are current mirrors of each other and the current magnitudes of the branches where the transistor M01 and the transistor M02 are located are equal, the current magnitude output by the transconductance amplifier OTA is:

[0174] I0 = 2 * IL = 2 * △I2 * R3 / RL

[0175] Assuming that △I1:△I2 = k (i.e., the preset ratio is k), when RL = 2 * R3 / (k + 1), the above formula can be converted to:

[0176] I0 = (k + 1) * △I2 = k * △I2 + △I2

[0177] I0 = △I1 + △I2

[0178] It can be seen that when RL = 2 * R3 / (k + 1), the current output by the transconductance amplifier OTA can be made equal to the sum of the first compensation current △I1 and the second compensation current △I2, so as to output the current through the transconductance amplifier OTA and control the current magnitude output by the power supply terminal VDD to be a constant positive temperature coefficient current.

[0179] It should be noted that the above takes the current of the m1 node flowing through the resistor RL into the m2 node as an example for illustration. At this time, the transconductance amplifier inputs current to the first end of the first transistor M1. In some possible embodiments, by controlling the signals Φ1 and Φ2 to change the input of the VP voltage and the VM voltage, it is also possible to make the current of the m2 node flow through the resistor RL into the m1 node. At this time, the transconductance amplifier extracts current from the first end of the first transistor M1; at the same time, a chopping circuit can also be added to the transconductance amplifier OTA to eliminate the offset error that may be introduced to the greatest extent.

[0180] Next, in order to better illustrate the implementation verification process of the present application, Figure 13Illustratively explain the derivation process of the specific calculation formula of the reference voltage VBG in this application. For the convenience of explaining the calculation process, in Figure 13 it is set that R1 = 7R, R2 = 10R, R3 = 2.5R, R4 = 3.5R, R5 = 12R, R6 = 3R, where R is the resistance value of the unit resistor. The above specific resistance values will be directly substituted into the calculation process and will not be elaborated later.

[0181] First, assume that after the first current source IS1 inputs current, the voltage of the third node C is higher than the voltages of the first node A and the second node B. Since △I1:△I2 = 1:4, according to Kirchhoff's current law, at this time, the current of the third triode Q3 and the current output by the first current source IS1 satisfy the following relational expression:

[0182]

[0183] Among them, I PNP3 is the current of the third triode Q3, and I trim is the current output by the first current source IS1.

[0184] Assume that the size ratio of the second triode Q2 to the third triode Q3 is 1:1, then the above formula can be transformed into:

[0185]

[0186] Among them, I S is the saturation current of the second triode Q2 and the third triode Q3.

[0187] Further simplification gives:

[0188]

[0189] Since the order of magnitude of 2R / V T *I trim -2R / V T *I PNP3 is relatively small, here ex≈x + 1 is used for equivalent substitution, and the above formula can be converted to

[0190]

[0191] According to Kirchhoff's current law, it can be known that the difference between the current of the third triode Q3 and the current output by the first current source IS1 is the change amount of the input current IBG of the current output unit 11. Therefore, the input current IBG of the current output unit 11 and the change amount of the input current satisfy the following relational expression:

[0192]

[0193]

[0194] Among them, △IBG is the change in the input current IBG of the current output unit 11, and ΔV BE is the difference between V BE2 and V BE1 .

[0195] Substituting the relationship formula between the current of the third triode Q3 and the current output by the first current source IS1 into the above formula, we can get:

[0196]

[0197] Meanwhile, in Figure 13 the reference voltage VBG can be calculated according to the following formula:

[0198]

[0199] Substituting the calculation formula of the input current IBG of the current output unit 11 into the above formula, we can get:

[0200]

[0201] The above derivation process assumes that after the first current source IS1 inputs current, the voltage of the third node C is higher than the voltages of the first node A and the second node B. Now, we will derive the calculation formula of the reference voltage VBG when the voltage of the third node C is lower than the voltages of the first node A and the second node B after the first current source IS1 inputs current:

[0202] When the voltage of the third node C is lower than the voltages of the first node A and the second node B after the first current source IS1 inputs current, the relationship between the current of the third triode Q3 and the current output by the first current source IS1 satisfies the following formula:

[0203]

[0204]

[0205] Simplifying the above formula in the same way, we can get

[0206]

[0207] Using the quadratic formula, we can get:

[0208]

[0209] Since in this case I PNP3 is greater than Itrim and is positive, so we discard a negative root, and the current of the third triode Q3 satisfies the following formula:

[0210]

[0211] Similarly, according to Kirchhoff's current law, the change in the input current IBG of the current output unit 11 satisfies the following relational expression:

[0212]

[0213] Substituting into the calculation formula of the reference voltage VBG, we can obtain:

[0214]

[0215]

[0216] Therefore, combining the results of the two cases, the reference voltage VBG satisfies the following relational expression:

[0217]

[0218] It can be seen that the first term on the right side of the above formula is a negative temperature coefficient voltage, and the second term is a positive temperature coefficient voltage. Therefore, by adjusting the magnitude of the current Itrim output by the first current source IS1, the calibration of the reference voltage VBG can be achieved, thereby ensuring the zero temperature drift characteristic of the reference voltage VBG. At the same time, for the individual differences caused by the chip manufacturing process, the magnitudes of the first term and the second term on the right side of the above formula will change due to chip individual differences. Therefore, for the bandgap reference circuits of different chips, the reference voltage VBG output by the bandgap reference circuit is not a fixed value. Therefore, the present application can also achieve the purpose of adjusting the magnitude of the reference voltage VBG within a certain range.

[0219] The embodiment of the present application also provides a chip, which includes the above-mentioned bandgap reference circuit. A chip (Integrated Circuit, IC) is also called a chip, and this chip can be, but is not limited to, a SOC (System on Chip, chip-level system) chip, a SIP (system in package, system-level package) chip. Since the chip of the present application has the bandgap reference circuit described in the above embodiment, it has all the beneficial effects of the bandgap reference circuit in the above embodiment, and will not be elaborated here.

[0220] The embodiments of the present application further provide an electronic device, which includes a device main body and the above-mentioned chip disposed in the device main body. The electronic device may be, but is not limited to, a weighing scale, a body fat scale, a nutrition scale, an infrared electronic thermometer, a pulse oximeter, a body composition analyzer, a mobile power supply, a wireless charger, a fast charger, a vehicle charger, an adapter, a display, a USB (Universal Serial Bus) expansion dock, a stylus, a true wireless earphone, a car center console screen, a car, a smart wearable device, a mobile terminal, a smart home device. The smart wearable device includes, but is not limited to, a smart watch, a smart bracelet, and a cervical massager. The mobile terminal includes, but is not limited to, a smart phone, a laptop, a tablet computer, and a POS (point of sales terminal) machine. The smart home device includes, but is not limited to, a smart socket, a smart rice cooker, a smart sweeper, and a smart light.

[0221] The above are only the preferred embodiments of the present application, and do not impose any form of limitation on the present application. Although the present application has been disclosed above with preferred embodiments, it is not intended to limit the present application. Any person skilled in the art can make some changes or modifications to equivalent embodiments by using the above-disclosed technical content within the scope of the technical solution of the present application. However, as long as it does not depart from the content of the technical solution of the present application, any brief modifications, equivalent changes, and modifications made to the above embodiments based on the technical essence of the present application still fall within the scope of the technical solution of the present application.

Claims

1. A bandgap reference circuit, wherein the bandgap reference circuit generates a reference voltage based on a negative temperature coefficient voltage and a positive temperature coefficient current, characterized in that: The bandgap reference circuit comprises: A reference module, the reference module is used to generate a first positive temperature coefficient current, a second positive temperature coefficient current and a negative temperature coefficient voltage, wherein a ratio of the first positive temperature coefficient current to the second positive temperature coefficient current is equal to a preset ratio; A current control module, the current control module is used to adjust the magnitude of the first positive temperature coefficient current and the second positive temperature coefficient current to calibrate the reference voltage; After the current control module adjusts the first positive temperature coefficient current and the second positive temperature coefficient current, the ratio of the first positive temperature coefficient current to the second positive temperature coefficient current remains unchanged at the preset ratio.

2. The bandgap reference circuit according to claim 1, wherein: The negative temperature coefficient voltage includes a first negative temperature coefficient voltage and a second negative temperature coefficient voltage, and the current control module is used to input or extract a first compensation current and a second compensation current to the reference module; The first negative temperature coefficient voltage is generated based on the first positive temperature coefficient current and the first compensation current, and the second negative temperature coefficient voltage is generated based on the second positive temperature coefficient current and the second compensation current; Wherein, a ratio of the first compensation current to the second compensation current is equal to the preset ratio.

3. The bandgap reference circuit according to claim 2, wherein: The current control module includes a first current source and a second current source; The first current source is connected to the reference module to input or extract a first compensation current to the reference module, and the second current source is connected to the reference module to input or extract a second compensation current to the reference module.

4. The bandgap reference circuit according to claim 2, wherein: The current control module includes a second resistor, a third resistor and a current source unit; The current source unit is connected to a first end of the second resistor, and a second end of the second resistor is connected to the reference module, so as to input or extract a first compensation current to the reference module through the second resistor; The current source unit is connected to a first end of the third resistor, and a second end of the third resistor is connected to the reference module, so as to input or extract a second compensation current to the reference module through the third resistor; Wherein, the ratio of the third resistor to the second resistor is equal to the preset ratio.

5. The bandgap reference circuit according to claim 4, characterized in that: The current source unit includes a third triode and a first current source; The first end of the third transistor is connected to the third node of the second resistor and the third resistor, the second end of the third transistor is connected to the ground, and the base of the third transistor is connected to the ground; The first current source is connected to the first end of the third transistor to inject current into the third transistor and change the voltage of the third node.

6. The bandgap reference circuit according to claim 5, characterized in that: The reference module includes a current output unit, a first resistor, a first transistor and a second transistor; The first connection end of the current output unit is connected to the first end of the first resistor, the second end of the first resistor is connected to the emitter of the first transistor, and the collector and base of the first transistor are connected to the ground end; The second connection end of the current output unit is connected to the emitter of the second transistor, and the collector and the base of the second transistor are connected to the ground end.

7. The bandgap reference circuit according to claim 6, wherein: The current output unit includes a first transistor, a fourth resistor, a fifth resistor, a sixth resistor and an operational amplifier; The first end of the first transistor is connected to the power supply end, and the second end of the first transistor is connected to the first end of the fourth resistor; The first end of the fifth resistor is connected to the second end of the fourth resistor, and the second end of the fifth resistor is connected to the first end of the first resistor; The first end of the sixth resistor is connected to the second end of the fourth resistor, and the second end of the sixth resistor is connected to the emitter of the second transistor; The first input terminal of the operational amplifier is connected to a first node between the fifth resistor and the first resistor, the second input terminal of the operational amplifier is connected to a second node between the sixth resistor and the second transistor, and the output terminal of the operational amplifier is connected to the control terminal of the first transistor; Wherein, a resistance ratio of the sixth resistor to the fifth resistor is equal to the preset ratio.

8. The bandgap reference circuit according to claim 7, characterized in that: The bandgap reference circuit also includes a transconductance amplifier; The first input terminal of the transconductance amplifier is connected to the third node, the second input terminal of the transconductance amplifier is connected to the second node, and the output terminal of the transconductance amplifier is connected to the first terminal of the first transistor.

9. The bandgap reference circuit according to claim 6, wherein: The current output unit includes a first transistor, a second transistor and an operational amplifier; The first end of the first transistor is connected to the power supply end, the second end of the first transistor is connected to the first end of the first resistor, and the control end of the first transistor is connected to the output end of the operational amplifier; The first end of the second transistor is connected to the power supply end, the second end of the second transistor is connected to the emitter of the second triode, and the control end of the second transistor is connected to the output end of the operational amplifier; The first input terminal of the operational amplifier is connected to a first node between the first transistor and the first resistor, and the second input terminal of the operational amplifier is connected to a second node between the second transistor and the second triode; The ratio between the width-to-length ratio of the first transistor and the width-to-length ratio of the second transistor is equal to the preset ratio.

10. The bandgap reference circuit according to claim 9, characterized in that: The bandgap reference circuit also includes a third transistor, a seventh resistor and a fourth triode; The first end of the third transistor is connected to the power supply end, the second end of the third transistor is connected to the first end of the seventh resistor, and the control end of the third transistor is connected to the output end of the operational amplifier; The first end of the fourth transistor is connected to the second end of the seventh resistor, the second end of the fourth transistor is connected to the ground, and the base of the fourth transistor is connected to the ground.

11. A chip, characterized in that: Comprising the bandgap reference circuit as claimed in any one of claims 1 to 10.

12. An electronic device, characterized in that: The invention comprises a device body and the chip as claimed in claim 11 arranged in the device body.

Images