T-type energy storage converter off-grid control method based on improved sliding mode active disturbance rejection

Abstract

The invention discloses an off-grid control method for a T-type energy storage converter based on improved sliding mode active disturbance rejection, and the method comprises the steps: obtaining a mathematical model under a two-phase rotating coordinate system through a main topological structure of a T-type three-level energy storage converter, and designing a current PI inner ring and a droop outer ring; a mathematical model corresponding to the voltage loop is converted into a first-order active-disturbance-rejection normal form, an order-increasing linear expansion observer with a disturbance differential item as a new state quantity is designed and is in adaptive weighted parallel connection with a traditional observer, noise in the steady state is reduced, and the disturbance rejection capacity in the transient state is reserved. And then designing a state error feedback control law by adopting a self-adaptive improved super-spiral sliding mode control law, so as to further improve the anti-interference capability of the active-disturbance-rejection controller and weaken the inherent buffeting of the active-disturbance-rejection controller. And finally, the power voltage and current are output to the SVPWM control T-type energy storage converter through three-loop control. A simulation result shows that compared with a traditional method, the method has better dynamic response performance and anti-interference performance.

Description

Technical Field

[0001] The present invention relates to an off-grid control method for a T-type energy storage converter based on improved sliding mode active disturbance rejection, belonging to the technical field of off-grid control methods for T-type energy storage converters. Background Art

[0002] The power conversion system (PCS) of energy storage has the advantages of high-efficiency energy conversion, bidirectional energy flow, and enhanced power grid stability. It plays a core role in solving new energy power generation and constructing microgrids today, and can assist in outputting smooth renewable energy to achieve peak shaving, valley filling, and intelligent control. Among them, the T-type three-level converter is gradually becoming a key component in new energy power generation, distributed energy systems, and microgrids due to its low harmonics and low switching losses.

[0003] In off-grid operation, because droop control is an equivalent control with the characteristics of plug-and-play, it is widely used. However, the inner loop of droop control often adopts the control technology of voltage-current PI double closed-loop. When facing the uncertain switching of the microgrid load, it will cause a large output voltage drop, and its transient process is relatively slow when the power voltage is given, which is not conducive to the frequently changing system environment.

[0004] Active disturbance rejection control has the advantages of strong robustness and is widely used in industry for disturbance suppression. However, for traditional active disturbance rejection control, especially when facing fast and high-frequency disturbances, it is difficult to balance the disturbance observation speed and observation noise, and it is difficult to guarantee its own observation accuracy. Moreover, it is difficult for traditional linear control laws to achieve optimal control. Summary of the Invention

[0005] The technical problem to be solved by the present invention is to overcome the defects of the prior art and provide an off-grid control method for a T-type energy storage converter based on improved sliding mode active disturbance rejection, which improves the traditional active disturbance rejection controller to enhance the anti-disturbance ability and robustness of the overall energy storage converter control.

[0006] Preferably, the present invention provides an off-grid control method for a T-type energy storage converter based on improved sliding mode active disturbance rejection, including:

[0007] Step 1: Use the main topological structure of the T-type three-level converter to obtain the mathematical model in the two-phase rotating coordinate system; obtain the proportional coefficient and integral coefficient of the current inner loop based on the typical type I system, and obtain the power outer loop according to the droop formula;

[0008] Step 2: Transform the formula corresponding to the voltage loop in the mathematical model into the first-order active disturbance rejection normal form to obtain a second-order linear extended state observer;

[0009] Step 3: Add a perturbation differential term to the second-order linear extended observer obtained in Step 2 as a new state variable, construct a new order-increased LESO with a perturbation differential term, and add a lag factor at the total perturbation output for preliminary noise suppression;

[0010] Step 4: Connect the second-order linear extended state observer obtained in Step 2 and the order-increased LESO obtained in Step 3 in parallel with weights to obtain an LESO with adaptive weighted parallel connection;

[0011] Step 5: Introduce an improved adaptive super-twisting sliding mode control law into the state error feedback control law, use the hyperbolic tangent function to weaken the inherent chattering of the sliding mode, and construct a voltage loop based on the information of the lumped perturbation observation value provided by the LESO with adaptive weighted parallel connection;

[0012] Step 6: Based on the power outer loop, voltage loop, and current inner loop, obtain the output modulation signal; transmit the output modulation signal to the SVPWM module with neutral point balance to control the main topology of the T-type three-level converter.

[0013] Preferably, Step 1 includes:

[0014] Using the main topology of the T-type three-level converter, obtain the mathematical model in the two-phase rotating coordinate system:

[0015]

[0016] where C f is the filter capacitor of the AC-side LC filter, L f is the filter inductor of the LC filter, i 1d and i 1q are the d-axis component and q-axis component of the filter inductor current of the main topology of the T-type three-level converter respectively; i 2d and i 2q are the d-axis component and q-axis component of the load current on the load side respectively; u cd and u cq are the d-axis component and q-axis component of the output voltage respectively, u d is the output d-axis voltage of the main topology of the T-type three-level converter, u q is the output q-axis voltage of the main topology of the T-type three-level converter, R 1 is the parasitic resistance of the filter inductor of the AC-side LC filter, ω is the angular frequency corresponding to the known power frequency; the formulas corresponding to the voltage loop are the third and fourth formulas from top to bottom in Formula (1);

[0017] Based on the requirement that the change in the voltage angular frequency of the power grid should be less than or equal to 1% and the change in amplitude should be less than or equal to 5%, determine the droop coefficient of the power outer loop as:

[0018]

[0019] Wherein, P max is the maximum active power output by the main topology of the T-type three-level converter when the frequency drops, ω 0 is the rated angular frequency output by the main topology of the T-type three-level converter, P 0 is the rated active power output by the main topology of the T-type three-level converter, E 0 is the rated voltage output by the main topology of the T-type three-level converter, Q 0 is the rated reactive power output by the main topology of the T-type three-level converter, ω min is the minimum angular frequency allowed when the main topology of the T-type three-level converter outputs the maximum active power, Q max is the maximum reactive power output by the main topology of the T-type three-level converter when the voltage drops to the maximum allowable value, E min is the minimum voltage amplitude allowed when the main topology of the T-type three-level converter outputs the maximum reactive power;

[0020] Construct a power outer loop, and the specific formula of the power outer loop is as follows:

[0021]

[0022] Wherein, P is the active power output by the main topology of the T-type three-level converter when the frequency drops, ω 0 is the rated angular frequency output by the main topology of the T-type three-level converter, P 0 is the rated active power output by the main topology of the T-type three-level converter, E 0 is the rated voltage output by the main topology of the T-type three-level converter, Q 0 is the rated reactive power output by the main topology of the T-type three-level converter, Q is the reactive power output by the main topology of the T-type three-level converter, ω c is the minimum voltage amplitude allowed when the main topology of the T-type three-level converter outputs the maximum reactive power, G LP (s) is a filter used to filter out the high-order harmonics calculated, s is the complex frequency variable of the Laplace transform, ω c represents the cut-off frequency of the filter. The droop equation includes the first-line formula and the second-line formula in the above formula, and the power calculation equation includes the third-line formula, the fourth-line formula and the fifth-line formula in the above formula;

[0023] Based on the typical type-I system, obtain the proportional coefficient and integral coefficient of the current inner loop, and obtain the power outer loop according to the droop formula, including:

[0024] The open-loop transfer function of the current inner loop in the S domain is:

[0025]

[0026] In the formula, G oi (s) represents the numerical value of the open-loop transfer function of the inner current loop, s is the complex frequency variable of the Laplace transform, K iP represents the proportional coefficient, K PWM represents the modulation gain, T i =L / R represents the constant pole, T s represents the switching period, L represents the inductance value of the filter, and R represents the parasitic resistance of the inductor of the LC filter;

[0027] Based on the typical type-I system, the proportional coefficient K ip and the integral coefficient K iI of the inner current loop are obtained as follows:

[0028]

[0029] Preferably, in step 2, the formula corresponding to the voltage loop in the mathematical model is transformed into the first-order active disturbance rejection normal form to obtain a second-order linear extended state observer, including:

[0030] Perform first-order active disturbance rejection normalization on the capacitor voltage state quantity in step 1:

[0031]

[0032] In the formula, b 0 =1 / C f is the control gain, C f is the filtering capacitor of the AC-side LC filter, i 1d and i 1q are respectively the d-axis component and the q-axis component of the filtering inductor current of the main topology structure filter of the T-type three-level converter; f d represents the total disturbance on the d-axis, f q represents the total disturbance on the q-axis:

[0033]

[0034] Among them, w d represents the unknown disturbance on the d-axis, w q represents the unknown disturbance on the q-axis, the coupling term is regarded as the known disturbance on the dq axis; i 2d and i 2q are respectively the d-axis component and the q-axis component of the load current on the load side; u cd and u cq are respectively the d-axis component and the q-axis component of the output voltage;

[0035] Let x 1 =u c 、x 2= f, the state - space equation of the system is:

[0036]

[0037] In the formula, is the differential of the system state variable, y is the system output variable, b 0 is the control gain, is the disturbance differential term, u = i;

[0038] Since the inner - loop gain of the current loop is often much larger than the outer - loop, so i ref = i;

[0039] According to Equation (7), a second - order linear extended state observer is established:

[0040]

[0041] In the formula, x 1 and x 2 are the estimated values of the capacitor voltage and the total disturbance respectively; β 1 and β 2 are the LESO feedback gain coefficients of the voltage loop; Through pole placement, the observer feedback gain coefficients are configured at the observer bandwidth ω o , and β 1 = 2ω o ,

[0042]

[0043] Preferably, in step 3, add the disturbance differential term of the second - order linear extended observer obtained in step 2 as a new state variable, construct a new extended - order LESO with the disturbance differential term, and add a lag factor at the total disturbance output for preliminary noise suppression, including:

[0044] Expand the estimated value x 2 of the total disturbance and the disturbance differential term into a new state variable x 3 , and obtain the corresponding observed value By observing the change trend of the lumped disturbance, obtain a new extended - order LESO with the disturbance differential term:

[0045]

[0046] Among them, β 1 = 3ω o , b 0 is the control gain, u = i;

[0047] Add a lag factor on the total disturbance output side:

[0048]

[0049] Wherein, is the lumped disturbance initially filtered by the lag factor, T e is the lag filtering time factor, and s is the complex frequency variable of the Laplace transform.

[0050] Preferably, in step 4, the second-order linear extended state observer obtained in step 2 and the order-increased LESO obtained in step 3 are weighted in parallel to obtain an adaptive weighted parallel LESO, including:

[0051] The second-order linear extended state observer obtained in step 2 and the order-increased LESO obtained in step 3 are weighted in parallel to obtain an adaptive weighted parallel LESO:

[0052]

[0053] Wherein, z 1 is the estimated capacitor voltage output by the entire adaptive parallel weighted observer, z 2 is the estimated lumped disturbance output by the entire adaptive parallel weighted observer, is the capacitor voltage estimated by the order-increased LESO, is the capacitor voltage estimated by the second-order linear extended state observer, is the estimated lumped disturbance of the order-increased LESO initially filtered by the lag factor, is the lumped disturbance estimated by the second-order linear extended state observer, and δ is the adaptive weighting factor:

[0054]

[0055] Wherein, a, b, c, d, and f are all adjustment factors and are greater than 0, x ref is the capacitor voltage reference value, x 1 is the actual capacitor voltage value.

[0056] Preferably, in step 5, an improved adaptive super-twisting sliding mode control law is introduced into the state error feedback control law, the hyperbolic tangent function is used to weaken the inherent chattering of the sliding mode, and based on the information of the lumped disturbance observation value provided by the adaptive weighted parallel LESO, its disturbance rejection ability and transient performance are further improved. The disturbances in the mathematical modeling in step 2 include:

[0057] Design the sliding mode surface according to equation (7) in step 2:

[0058]

[0059] Wherein, s is the sliding mode surface function, c > 0 is the sliding mode integral coefficient, x ref is the given value of the capacitor voltage, z 1is the estimated value of the capacitor voltage output by the adaptive weighted observer, and e is the voltage reference and the estimation error;

[0060] When on the sliding surface, s = 0, let We get:

[0061]

[0062] Substitute the system state values x 1 and x 2 in Equation (7) of Step 2 with the observed values z 1 and z 2 , and substitute them into Equation (14) to obtain the equivalent control quantity u eq :

[0063]

[0064] When outside the sliding surface, the reaching law is the improved super-twisting reaching law:

[0065]

[0066] Where k 1 , k 2 , k 3 are the parameters to be designed for the sliding mode controller, all greater than 0, r is the coefficient to be designed, tanh(s) is the hyperbolic tangent function, and sgn(s) is the sign function;

[0067] Adaptively adjust the above reaching law to increase the reaching speed when the reaching law is far from the sliding surface and decrease the reaching speed when approaching:

[0068]

[0069] Where a and b are adjustment factors;

[0070] According to Equation (14) and Equation (17), calculate the switching control quantity u sw :

[0071] u sw = k 1 ρ 1 (|s|)tanh(s) + k 2 ∫sgn(s)dt + k 3 ρ 2 (|e|)s(18),

[0072] Based on Equation (18), calculate the control u output by the voltage sliding mode auto-disturbance rejection control:

[0073] u = u eq + u sw (19).

[0074] Preferably, in step 6, based on the power outer loop, voltage loop and current inner loop, an output modulation signal is obtained; the output modulation signal is transmitted to the SVPWM module with neutral point balance to control the main topology of the T-type three-level converter, including:

[0075] The control voltage output by the power-voltage-current three-loop calculated in steps 1 to 5 is sent to the inverse Park transformation to obtain three-phase voltage modulation waves; the modulation waves are sent to the SVPWM modulator and adjusted by the balance factor method of the neutral point charge conservation principle to output PWM waves; the operation of the T-type converter is controlled by 12-way PWM control signals.

[0076] Preferably, the present invention provides an electronic device, including a memory, a processor, and a computer program stored on the memory and executable on the processor. When the processor executes the program, the steps of the method according to any one of the first aspects are implemented.

[0077] Preferably, the present invention provides a computer-readable storage medium, on which a computer program is stored. When the computer program is executed by a processor, the steps of the method according to any one of the first aspects are implemented.

[0078] The beneficial effects achieved by the present invention:

[0079] 1. In step 1, the power loop and current loop in the power-voltage-current three-loop are initially constructed, and then in step 2, a first-order active disturbance rejection control is built to improve its transient performance and disturbance rejection ability. Then, in step 3, a new state variable with an added disturbance differential component is built to increase the order, further improving the transient performance and disturbance rejection, and the observer bandwidth is increased through a lag factor.

[0080] 2. By adopting the parallel connection of the increased-order LESO and the traditional LESO adaptively in step 4, the proportion of the increased-order LESO at steady state is reduced, and the proportion during transient and disturbance is increased, achieving a smaller noise at steady state and good transient performance and disturbance rejection effect.

[0081] 3. By adopting a sliding mode control rate to replace the traditional linear control rate in step 5, the advantages of nonlinear control are exerted, further improving the disturbance rejection performance of the active disturbance rejection, and enhancing the robustness of the system. The disturbance differential increased-order LESO is fully utilized to improve the disturbance observation ability, and the noise generated by the increased-order LESO is reduced by weighted parallel connection with the traditional LESO. Then, an adaptive weight is adopted to increase the proportion of the increased-order LESO during large disturbances and transient processes, improving the response speed and robustness of the control, and increasing the traditional proportion at steady state to reduce the influence of noise, giving full play to the advantages of the observer and compensating for the disadvantages.

[0082] 4. Adopt a sliding mode control law to replace the traditional linear control law, give play to the advantages of non-linear control, further improve its disturbance rejection performance of active disturbance rejection, and enhance the robustness of the system. Description of the Drawings

[0083] To more clearly illustrate the technical solutions of the present application, the drawings required for use in the embodiments will be briefly introduced below. Obviously, for those of ordinary skill in the art, without creative efforts, other drawings can also be obtained based on these drawings.

[0084] Figure 1 It is the overall control block diagram of the main topology structure of the T-type three-level converter proposed by the present invention;

[0085] Figure 2 It is the main circuit topology diagram of the main topology structure of the T-type three-level converter used by the present invention;

[0086] Figure 3 It is the adaptive improved super-twisting sliding mode control block diagram proposed by the present invention;

[0087] Figure 4 It is the adaptive weighted parallel LESO control block diagram proposed by the present invention;

[0088] Figure 5 It is the comparison diagram of the output voltage amplitudes of the improved adaptive sliding mode active disturbance rejection control and PI, LADRC control under sudden load addition and voltage reference change proposed by the present invention;

[0089] Figure 6 It is the comparison diagram of the output active powers of the improved adaptive sliding mode active disturbance rejection control and PI, LADRC control under sudden load addition and voltage reference change proposed by the present invention;

[0090] Figure 7 It is the comparison diagram of the output reactive powers of the improved adaptive sliding mode active disturbance rejection control and PI, LADRC control under sudden load addition and voltage reference change proposed by the present invention. Detailed Embodiments

[0091] Refer to Figure 1 , the present application discloses a grid-connected control method for a T-type energy storage converter based on improved sliding mode active disturbance rejection, including:

[0092] Step 1, use the main topology structure of the T-type three-level converter to obtain the mathematical model in the two-phase rotating coordinate system; obtain the proportional coefficient and integral coefficient of the current inner loop based on the typical type I system, and obtain the power outer loop according to the droop formula;

[0093] Step 2, according to Figure 3 and Figure 4For the structural design, prepare to design the voltage loop, transform the formula corresponding to the voltage loop in the mathematical model into the first-order active disturbance rejection normal form to obtain a second-order linear extended state observer;

[0094] Step 3: Add a disturbance differential term to the second-order linear extended observer obtained in Step 2 as a new state variable, construct a new augmented-order LESO with a disturbance differential term, and add a lag factor at the total disturbance output for preliminary noise suppression;

[0095] Step 4: Connect the second-order linear extended state observer obtained in Step 2 and the augmented-order LESO obtained in Step 3 in parallel with weights to obtain an adaptive weighted parallel LESO, further reducing the noise amplification phenomenon caused by the augmented-order LESO, and perform adaptive processing on the weighting factor so that the proportion of the augmented-order LESO increases during the disturbance transient state and decreases during the steady state;

[0096] Step 5: Introduce an improved adaptive super-twisting sliding mode (STSM) control law into the state error feedback control law (SEF), use the hyperbolic tangent function to weaken the inherent chattering of the sliding mode, and construct a voltage loop based on the information of the lumped disturbance observation value provided by the adaptive weighted parallel LESO to suppress the disturbance in the mathematical modeling of the active disturbance rejection normal form in Step 2 and achieve fast and accurate control of the system state;

[0097] Step 6: Adjust the power outer loop, voltage loop, and current inner loop to obtain the output modulation signal; transmit the output modulation signal to the SVPWM module with neutral point balance to control the main topology of the T-type three-level converter.

[0098] Step 1 includes:

[0099] Using the main topology of the T-type three-level converter, obtain the mathematical model in the two-phase rotating coordinate system:

[0100]

[0101] where C f is the filter capacitor of the AC-side LC filter, L f is the filter inductor of the LC filter, i 1d and i 1q are the d-axis component and q-axis component of the filter inductor current of the main topology of the T-type three-level converter respectively; i 2d and i 2q are the d-axis component and q-axis component of the load current on the load side respectively; u cd and u cq are the d-axis component and q-axis component of the output voltage respectively, u d is the output d-axis voltage of the main topology of the T-type three-level converter, uq For the q - axis voltage output by the main topology of the T - type three - level converter, R 1 is the parasitic resistance of the filtering inductor of the AC - side LC filter, and ω is the angular frequency corresponding to the known power frequency; the formulas corresponding to the voltage loop are the third and fourth formulas from top to bottom in formula (1).

[0102] According to Article 3.1 of "Power Quality - Frequency and Magnitude Deviations of Power Systems" established by the state: Based on the requirement that the change in the voltage angular frequency of the power grid should be less than or equal to 1% and the magnitude change should be less than or equal to 5%, the droop coefficient is determined as:

[0103]

[0104] In the formula, P max is the maximum active power output by the main topology of the T - type three - level converter when the frequency drops, ω 0 is the rated angular frequency output by the main topology of the T - type three - level converter, P 0 is the rated active power output by the main topology of the T - type three - level converter, E 0 is the rated voltage output by the main topology of the T - type three - level converter, Q 0 is the rated reactive power output by the main topology of the T - type three - level converter, ω min is the minimum angular frequency allowed when the main topology of the T - type three - level converter outputs the maximum active power, Q max is the maximum reactive power output by the main topology of the T - type three - level converter when the voltage drops to the maximum allowable value, E min is the minimum voltage amplitude allowed when the main topology of the T - type three - level converter outputs the maximum reactive power;

[0105] Construct the power outer loop, and the specific formula of the power outer loop is as follows:

[0106]

[0107] In the formula, P is the active power output by the main topology of the T - type three - level converter when the frequency drops, ω 0 is the rated angular frequency output by the main topology of the T - type three - level converter, P 0 is the rated active power output by the main topology of the T - type three - level converter, E 0 is the rated voltage output by the main topology of the T - type three - level converter, Q 0 is the rated reactive power output by the main topology of the T - type three - level converter, Q is the reactive power output by the main topology of the T - type three - level converter, ω c is the minimum voltage amplitude allowed when the main topology of the T - type three - level converter outputs the maximum reactive power, G LP(s) is a filter for filtering out high-order harmonics in the calculation, where s is the complex frequency variable of the Laplace transform, and ω c represents the cut-off frequency of the filter. The droop equation includes the first-line formula and the second-line formula in the above formula, and the power calculation equation includes the third-line formula, the fourth-line formula, and the fifth-line formula in the above formula;

[0108] Based on the typical type-I system, the proportional coefficient and integral coefficient of the current inner loop are obtained, and the power outer loop is obtained according to the droop formula, including:

[0109] The open-loop transfer function of the current inner loop in the S domain is:

[0110]

[0111] In the formula, G oi (s) represents the numerical value of the open-loop transfer function of the current inner loop, s is the complex frequency variable of the Laplace transform, K iP represents the proportional coefficient, K PWM represents the modulation gain, T i =L / R represents the constant pole, T s represents the switching period, L represents the filter inductor value, and R represents the parasitic resistance of the inductor of the LC filter;

[0112] Based on the typical type-I system, the proportional coefficient K ip and the integral coefficient K iI are obtained as follows:

[0113]

[0114] Step 2, according to Figure 3 and Figure 4 structural design, the formula corresponding to the voltage loop in the mathematical model is transformed into the first-order active disturbance rejection normal form to obtain a second-order linear extended state observer, including:

[0115] Perform first-order active disturbance rejection normalization on the capacitor voltage state quantity in Step 1:

[0116]

[0117] In the formula, b 0 =1 / C f is the control gain, C f is the filtering capacitor of the AC-side LC filter, i 1d and i 1q are the d-axis component and q-axis component of the filtering inductor current of the main topology structure filter of the T-type three-level converter respectively; f d represents the total disturbance on the d axis, and f q represents the total disturbance on the q axis:

[0118]

[0119] Among them, w d represents the unknown disturbance of the d-axis, and w q represents the unknown disturbance of the q-axis. The coupling term is considered as the known disturbance of the dq-axis; i 2d and i 2q are the d-axis component and the q-axis component of the load current on the load side respectively; u cd and u cq are the d-axis component and the q-axis component of the output voltage respectively; Therefore, the dq structures are the same, and the same design can be adopted separately, and no special distinction will be made hereinafter;

[0120] Let x 1 = u c and x 2 = f. The space state equation of the system is:

[0121]

[0122] In the formula, is the differential of the system state quantity, y is the system output quantity, b 0 is the control gain, is the disturbance differential term, and u = i;

[0123] Since the inner-loop gain of the current loop is often much larger than the outer loop, it can be considered that i ref = i;

[0124] According to Equation (7), a second-order linear extended state observer is established:

[0125]

[0126] In the formula, x 1 and x 2 are the estimated value of the capacitor voltage and the estimated value of the total disturbance respectively; β 1 and β 2 are the LESO feedback gain coefficients of the voltage loop; Through pole placement, the observer feedback gain coefficients are configured at the observer bandwidth ω o to obtain β 1 = 2ω o and

[0127]

[0128] Step 3: Add the disturbance differential term obtained in Step 2 as a new state quantity to the second-order linear extended observer, construct a new augmented-order LESO with the disturbance differential term, and add a lag factor at the total disturbance output for preliminary noise suppression, including:

[0129] To further improve the observation performance and control bandwidth of the LESO observer, the disturbance differential term of the estimated total disturbance x 2 is expanded into a new state variable x 3 , and the corresponding observed value is obtained. By observing the change trend of the lumped disturbance, a new augmented-order LESO with a disturbance differential term is obtained:

[0130]

[0131] where β 1 = 3ω o , b 0 is the control gain, u = i;

[0132] To broaden the estimation range of the LESO for the total disturbance and effectively solve the system instability caused by too large ω ο , a lag factor is added to the output side of the total disturbance:

[0133]

[0134] In the formula, is the lumped disturbance preliminarily filtered by the lag factor, T e is the lag filtering time factor, and s is the complex frequency variable of the Laplace transform.

[0135] Step 4, the second-order linear extended state observer obtained in Step 2 and the augmented-order LESO obtained in Step 3 are weighted and paralleled to obtain an adaptive weighted parallel LESO, including:

[0136] To further solve the noise amplification caused by the augmented differential term, the second-order linear extended state observer obtained in Step 2 and the augmented-order LESO obtained in Step 3 are weighted and paralleled to obtain an adaptive weighted parallel LESO:

[0137]

[0138] In the formula, z 1 is the estimated capacitor voltage output by the entire adaptive parallel weighted observer, z 2 is the estimated lumped disturbance output by the entire adaptive parallel weighted observer, is the capacitor voltage estimated by the augmented-order LESO, is the capacitor voltage estimated by the second-order linear extended state observer, is the estimated lumped disturbance of the augmented-order LESO preliminarily filtered by the lag factor, is the lumped disturbance estimated by the second-order linear extended state observer, and δ is the adaptive weighting factor:

[0139]

[0140] Wherein, a, b, c, d, and f are all adjustment factors and are greater than 0, and x ref is the reference value of the capacitor voltage, and x 1 is the actual value of the capacitor voltage; through this adaptive factor δ, when in the steady state, the proportion of the augmented LESO is about 0.5, and when in a large disturbance and transient process, the proportion of the augmented LESO rapidly increases to 0.8.

[0141] Step 5: Introduce an improved adaptive super-twisting sliding mode control law into the state error feedback control law, use the hyperbolic tangent function to weaken the inherent chattering of the sliding mode, and further improve its disturbance rejection ability and transient performance based on the information of the lumped disturbance observation value provided by the LESO based on adaptive weighted parallel. The disturbances in the mathematical modeling in Step 2 include:

[0142] Design the sliding mode surface according to Equation (7) in Step 2:

[0143]

[0144] Wherein, s is the sliding mode surface function, c > 0 is the sliding mode integral coefficient, x ref is the given value of the capacitor voltage, z 1 is the estimated value of the capacitor voltage output by the adaptive weighted observer, and e is the voltage given and estimation error;

[0145] When on the sliding mode surface, s = 0, let to obtain:

[0146]

[0147] Substitute the system state values x 1 and x 2 in Equation (7) in Step 2 with the observed values z 1 and z 2 and substitute them into Equation (14) to obtain the equivalent control quantity u eq :

[0148]

[0149] When outside the sliding mode surface, the reaching law is the improved super-twisting reaching law:

[0150]

[0151] Wherein, k 1 and k 2 and k 3 are the parameters to be designed of the sliding mode controller and are all greater than 0, r is the coefficient to be designed, tanh(s) is the hyperbolic tangent function, and sgn(s) is the sign function;

[0152] The above approaching rate is adaptively adjusted to increase the approaching speed when the approaching rate is far from the sliding surface and decrease the approaching speed when it is close to the sliding surface:

[0153]

[0154] where a and b are adjustment factors;

[0155] According to Equations (14) and (17), the switching control quantity u is calculated as sw :

[0156] u sw = k 1 ρ 1 (|s|)tanh(s) + k 2 ∫sgn(s)dt + k 3 ρ 2 (|e|)s(18),

[0157] Based on Equation (18), the control u output by the voltage sliding mode auto-disturbance rejection control is calculated as:

[0158] u = u eq + u sw (19).

[0159] In step 6, the three-loop control outputs of power, voltage, and current are given to the SVPWM module with neutral point balance to control the main topology of the T-type three-level converter, including:

[0160] The control voltage output by the three-loop of power-voltage-current calculated in steps 1 to 5 is sent to the inverse Park transformation to obtain the three-phase voltage modulation wave; the modulation wave is sent to the SVPWM modulator after per-unit processing and adjusted by the balance factor method of the neutral point charge conservation principle, and then the PWM wave is output; the operation of the T-type converter is controlled through 12 PWM control signals to achieve the overall control strategy.

[0161] The control voltage output by the three-loop of power-voltage-current designed in steps 1 to 5 is sent to the inverse Park transformation to obtain the three-phase voltage modulation wave; the modulation wave is sent to the SVPWM modulator after per-unit processing and adjusted by the balance factor method of the neutral point charge conservation principle, and then the PWM wave is output; the operation of the T-type converter is controlled through 12 PWM control signals to achieve the overall control strategy. As Figure 1 shown, the present invention includes a T-type energy storage converter, an LC filter, a first line impedance R C , a second line impedance L C ,

[0162] The T-type energy storage converter includes IGBTs and diodes. A diode is reversely connected in parallel between the C pole and the E pole of the IGBT. The specific structure is as Figure 2As shown, the current \(i\) output by the T-type energy storage converter 1 is sequentially input into the LC filter, the first line impedance \(R\) c , the second line impedance \(L\) c and the load. Based on the voltage \(v\) of the filter capacitor \(C\) of the LC filter f , and the output current \(i\) flowing through the line impedance cabc , power calculation is performed, and the voltage \(v\) of the filter capacitor \(C\) of the LC filter 2abc is input into the adaptive weighted parallel LESO; the output current \(i\) f and the voltage of the filter capacitor \(C\) of the LC filter cabc are subjected to power calculation after dq transformation, and the active power \(P\) and reactive power \(Q\) output therefrom, together with the rated active power \(P\) 2abc , the rated reactive power \(Q\) f , the rated frequency \(\omega\) 0 , and the rated voltage \(E\) 0 are input into the droop equation, and the parameters \(\omega\) and \(E\) output by the droop equation are used to generate a reference voltage to obtain \(V\) 0 ; the outputs \(Z\) o and \(Z\) oref of the adaptive weighted parallel LESO and \(V\) 1 are subjected to adaptive improved super-twisting sliding mode processing, and the output current \(i\) of the adaptive improved super-twisting sliding mode 2 , and the current \(i\) oref are subjected to current PI control and then output to the SVPWM. ref , and the current \(i\) labc is output to the SVPWM after current PI control.

[0163] Figure 2 is the main circuit topology diagram of the T-type three-level converter main topology used in the present invention, including:

[0164] Power supply \(U\) dc , the first capacitor \(C\) 1 , the second capacitor \(C\) 2 are connected in series. The power supply \(U\) dc , the first insulated gate bipolar transistor \(Q\) a1 , the third insulated gate bipolar transistor \(Q\) a3 , the second insulated gate bipolar transistor \(Q\) a2 and the second capacitor \(C\) 2 are connected in series. The first insulated gate bipolar transistor \(Q\) a1 is anti-parallelly connected with the first diode \(D\) a1 , the second insulated gate bipolar transistor \(Q\) a2 is anti-parallelly connected with the second diode \(D\) a2 , the third insulated gate bipolar transistor \(Q\) a3 is anti-parallelly connected with the third diode \(D\) a3 , the power supply \(U\) dcThe upper fourth insulated gate bipolar transistor Q b1 and the sixth insulated gate bipolar transistor Q b3 and the fifth insulated gate bipolar transistor Q b2 and the second capacitor C 2 are connected in series. The fourth insulated gate bipolar transistor Q b1 is anti-parallel connected with the fourth diode D b1 , and the fifth insulated gate bipolar transistor Q b2 is anti-parallel connected with the fifth diode D b2 , and the sixth insulated gate bipolar transistor Q b3 is anti-parallel connected with the sixth diode D b3 , and the power supply U dc and the upper seventh insulated gate bipolar transistor Q c1 and the ninth insulated gate bipolar transistor Q c3 and the eighth insulated gate bipolar transistor Q c2 and the second capacitor C 2 are connected in series. The seventh insulated gate bipolar transistor Q c1 is anti-parallel connected with the seventh diode D c1 , and the eighth insulated gate bipolar transistor Q c2 is anti-parallel connected with the eighth diode D c2 , and the ninth insulated gate bipolar transistor Q c3 is anti-parallel connected with the ninth diode D c3 , and the voltage output connection point u a1 between the first insulated gate bipolar transistor Q a3 and the third insulated gate bipolar transistor Q a and the power supply U dc and the second capacitor C 2 is connected in series with the fourth insulated gate bipolar transistor Q a4 , and the tenth insulated gate bipolar transistor Q a4 is anti-parallel connected with the tenth diode D a4 , and the voltage output connection point u b1 between the fourth insulated gate bipolar transistor Q b3 and the sixth insulated gate bipolar transistor Q b and the power supply U dc and the second capacitor C 2 is connected in series with the eleventh insulated gate bipolar transistor Q b4 , and the eleventh insulated gate bipolar transistor Q b4 is anti-parallel connected with the eleventh diode D b4 , and the voltage output connection point u c1 between the seventh insulated gate bipolar transistor Q c3 and the ninth insulated gate bipolar transistor Q c and the power supply U dc and the second capacitor C2 A twelfth insulated gate bipolar transistor Q is connected in series at the connection point N therebetween c4 , and for the twelfth insulated gate bipolar transistor Q c4 , a twelfth diode D is anti-parallel connected thereon c4 , and the voltage output connection points u a 、u b 、u c are all connected to the filter inductor L of the LC filter f , and for the parasitic resistance R of the filter inductor f , the filter capacitor C of the LC filter is Y-connected between the three lines f and intersects at the n c point, and the connection points u ca 、u cb 、u cc of the three pairs of filter inductors and filter capacitors are connected to the load.

[0165] In Matlab / Simulink, the present invention and the retrograde verification are carried out. According to the derivation of the above steps, its simulation model is built. Initially, a 15kW + 9kVar load is taken, and a 15kW + 9kVar load is cut in again at 0.4s. At 0.8s, the voltage setpoint drops by 50V, and the voltage setpoint is restored at 1s.

[0166] Figure 5 FIG. is a comparison diagram of the output voltage amplitudes of the improved adaptive sliding mode active disturbance rejection control and PI, LADRC control proposed by the present invention under sudden load addition and voltage setpoint change; for the improved scheme of the present invention, the voltage amplitude changes smoothly during the startup process, there will be no sudden drop, and the adjustment time is small. When the load is cut in, the voltage drop is small and the recovery speed is fast. When the voltage setpoint changes, the tracking is the fastest, and the ripple change of the voltage amplitude in each steady state stage is smaller.

[0167] Figure 6 FIG. is a comparison diagram of the output active power of the improved adaptive sliding mode active disturbance rejection control and PI, LADRC control proposed by the present invention under sudden load addition and voltage setpoint change, Figure 7 FIG. is a comparison diagram of the output reactive power of the improved adaptive sliding mode active disturbance rejection control and PI, LADRC control proposed by the present invention under sudden load addition and voltage setpoint change. It can be seen from the above two figures that for the improved scheme of the present invention, during the startup process and the disturbance process, the rise time is the smallest and the transition is the smoothest.

[0168] In an embodiment of the present application, the present invention provides an electronic device, including a memory, a processor, and a computer program stored on the memory and executable on the processor. When the processor executes the program, the steps of the method described in any one of the above are implemented.

[0169] In an embodiment of the present application, the present invention provides a computer-readable storage medium, on which a computer program is stored, and when the computer program is executed by a processor, the steps of the method described in any one of the above are implemented.

[0170] Each embodiment in this specification is described in a progressive manner. For the same or similar parts among the embodiments, reference can be made to each other. Each embodiment focuses on the differences from other embodiments.

[0171] After considering the specification and practicing the invention herein, those skilled in the art will readily conceive of other embodiments of the present invention. This application is intended to cover any variations, uses, or adaptations of the present invention, which follow the general principles of the present invention and include common general knowledge or conventional technical means in the technical field not invented by the present invention. The specification and embodiments are only regarded as exemplary.

[0172] The above specific implementation manners further elaborate on the purpose, technical solution, and beneficial effects of the present application. It should be understood that the above are only specific implementation manners of the present application and are not used to limit the protection scope of the present application. Any modifications, equivalent replacements, improvements, etc. made on the basis of the technical solution of the present application shall be included in the protection scope of the present application.

Claims

1. A T-type energy storage converter off-grid control method based on improved sliding mode anti-disturbance control is characterized in that: include: Step 1, using the main topology of a T-type three-level converter, obtaining a mathematical model in a two-phase rotating coordinate system; Based on the typical type I system, the proportional coefficient and integral coefficient of the current inner loop are obtained, and the power outer loop is obtained according to the droop formula; Step 2, transform the formula corresponding to the voltage loop in the mathematical model into a first-order auto-disturbance rejection paradigm to obtain a second-order linear extended state observer; Step 3: Add the disturbance differential term to the second-order linear extended observer obtained in step 2 as a new state variable, construct a new increased-order LESO with disturbance differential term, and add a lag factor to the total disturbance output for preliminary noise suppression; Step 4, weighted parallel connection of the second-order linear extended state observer obtained in step 2 and the increased-order LESO obtained in step 3 to obtain an adaptive weighted parallel LESO; Step 5, introducing an improved adaptive super-helical sliding mode control law into the state error feedback control law, using the hyperbolic tangent function to weaken the inherent chattering of the sliding mode, and constructing a voltage loop based on the information of the lumped disturbance observation value provided by the adaptive weighted parallel LESO; Step 6: Based on the power outer loop, the voltage loop and the current inner loop, an output modulation signal is obtained; and the output modulation signal is transmitted to the SVPWM module with center point balance to control the main topology structure of the T-type three-level converter.

2. The off-grid control method of a T-type energy storage converter based on improved sliding mode anti-disturbance control according to claim 1 is characterized in that: Step 1 includes: Using the main topology of the T-type three-level converter, the mathematical model in the two-phase rotating coordinate system is obtained: In the formula, C f is the filter capacitor of the AC side LC filter, L f is the filter inductance of the LC filter, i 1d and i 1q are the d-axis component and q-axis component of the filter inductor current of the main topology filter of the T-type three-level converter respectively; i 2d and i 2q are the d-axis component and q-axis component of the load current on the load side respectively; u cd and u cq are the d-axis component and q-axis component of the output voltage, u d The d-axis voltage is outputted by the main topology of the T-type three-level converter, u q is the q-axis voltage outputted by the main topology of the T-type three-level converter, R1 is the parasitic resistance of the filter inductor of the LC filter on the AC side, and ω is the angular frequency corresponding to the known power frequency; the formula corresponding to the voltage loop is the formula in the third and fourth rows from top to bottom in formula (1); Based on the requirement that the voltage angular frequency change of the power grid should be less than or equal to 1% and the amplitude change should be less than or equal to 5%, the droop coefficient is determined as: Where P max is the maximum active power output by the main topology of the T-type three-level converter when the frequency drops, ω0 is the rated angular frequency output by the main topology of the T-type three-level converter, P0 is the rated active power output by the main topology of the T-type three-level converter, E0 is the rated voltage output by the main topology of the T-type three-level converter, Q0 is the rated reactive power output by the main topology of the T-type three-level converter, ω min Q is the minimum angular frequency allowed when the main topology of the T-type three-level converter outputs maximum active power. max is the maximum reactive power output by the main topology of the T-type three-level converter when the voltage drops to the maximum allowable value, E min The minimum voltage amplitude allowed when the main topology of the T-type three-level converter outputs maximum reactive power; Construct a power outer loop. The specific formula of the power outer loop is as follows: Wherein, P is the active power output by the main topology of the T-type three-level converter when the frequency decreases, ω0 is the rated angular frequency output by the main topology of the T-type three-level converter, P0 is the rated active power output by the main topology of the T-type three-level converter, E0 is the rated voltage output by the main topology of the T-type three-level converter, Q0 is the rated reactive power output by the main topology of the T-type three-level converter, Q is the reactive power output by the main topology of the T-type three-level converter, ω c G is the minimum voltage amplitude allowed when the main topology of the T-type three-level converter outputs the maximum reactive power. LP (s) is a filter used to filter out the higher harmonics of the calculation, s is the complex frequency variable of the Laplace transform, ω c represents the cutoff frequency of the filter, wherein the droop equation includes the first and second row formulas in the above formula, and the power calculation equation includes the third, fourth and fifth row formulas in the above formula; Based on the typical type I system, the proportional coefficient and integral coefficient of the current inner loop are obtained, and the power outer loop is obtained according to the droop formula, including: The open-loop transfer function of the inner current loop in the S domain is: In the formula, G oi (s) represents the value of the open-loop transfer function of the current inner loop, s is the complex frequency variable of the Laplace transform, K iP Represents the proportionality coefficient, K PWM represents the modulation gain, T i =L / R represents a constant pole, T s represents the switching cycle, L represents the filter inductance value, and R represents the parasitic resistance of the LC filter inductor; Based on the typical type I system, the proportional coefficient K of the current inner loop is obtained ip and the integral coefficient K iI :

3. The off-grid control method of a T-type energy storage converter based on improved sliding mode anti-disturbance control according to claim 1 is characterized in that: Step 2: Transform the formula corresponding to the voltage loop in the mathematical model into a first-order auto-disturbance rejection paradigm to obtain a second-order linear extended state observer, including: The capacitor voltage state quantity in step 1 is normalized into the first-order auto-disturbance rejection formula: Where b0 = 1 / C f is the control gain, C f is the filter capacitor of the AC side LC filter, i 1d and i 1q are the d-axis component and q-axis component of the filter inductor current of the main topology filter of the T-type three-level converter respectively; d represents the total disturbance on the d-axis, f q represents the total disturbance on the q-axis: Among them, w d represents the unknown disturbance on the d-axis, w q represents the unknown disturbance of the q axis, and the coupling term is considered as the known disturbance of the dq axis; i 2d and i 2q are the d-axis component and q-axis component of the load current on the load side respectively; u cd and u cq They are the d-axis component and q-axis component of the output voltage respectively; Let x1 = u c , x2=f, the spatial state equation of the system is: In the formula, and is the differential of the system state quantity, y is the system output, b0 is the control gain, is the disturbance differential term, u=i; Since the inner loop gain of the current loop is often much greater than the outer loop gain, i ref =i; According to formula (7), a second-order linear extended state observer is established: Where x1 and x2 are the estimated values ​​of the capacitor voltage and the total disturbance, respectively; β1 and β2 are the LESO feedback gain coefficients of the voltage loop; through pole configuration, the observer feedback gain coefficient is configured within the observer bandwidth ω o At, we get β1=2ω o , 4. The off-grid control method of a T-type energy storage converter based on improved sliding mode anti-disturbance control according to claim 1 is characterized in that: Step 3: Add the disturbance differential term to the second-order linear expansion observer obtained in step 2 as a new state variable, construct a new increased-order LESO with disturbance differential term, and add a hysteresis factor to the total disturbance output for preliminary noise suppression, including: Expand the estimated value of the total disturbance x2 into a new state variable x3 and get the corresponding observation value By observing the changing trend of the lumped disturbance, a new increased-order LESO with a disturbance differential term is obtained: Where β1 = 3ω o , b0 is the control gain, u=i; Add a hysteresis factor to the total disturbance output side: In the formula, is the lumped disturbance after preliminary filtering by the hysteresis factor, T e is the lag filter time factor, and s is the complex frequency variable of the Laplace transform.

5. The off-grid control method of T-type energy storage converter based on improved sliding mode anti-disturbance control according to claim 1 is characterized in that: Step 4, weighted parallel connection of the second-order linear extended state observer obtained in step 2 and the increased-order LESO obtained in step 3 to obtain an adaptive weighted parallel LESO, including: The second-order linear extended state observer obtained in step 2 is weighted in parallel with the enhanced-order LESO obtained in step 3 to obtain an adaptive weighted parallel LESO: Where z1 is the estimated capacitor voltage output by the entire adaptive parallel weighted observer, z2 is the estimated lumped disturbance output by the entire adaptive parallel weighted observer, is the capacitor voltage estimated by the enhanced-order LESO, is the capacitor voltage estimated by the second-order linear extended state observer, is the estimated lumped disturbance of the increased-order LESO after preliminary filtering with the hysteresis factor, is the lumped disturbance estimated by the second-order linear extended state observer, and δ is the adaptive weighting factor: In the formula, a, b, c, d, and f are adjustment factors and are greater than 0, x ref is the capacitor voltage reference value, and x1 is the capacitor voltage actual value.

6. The off-grid control method of a T-type energy storage converter based on improved sliding mode anti-disturbance control according to claim 3 is characterized in that: Step 5: introduce an improved adaptive super-helical sliding mode control law into the state error feedback control law, use the hyperbolic tangent function to weaken the inherent chattering of the sliding mode, and further improve its anti-disturbance capability and transient performance based on the information of the lumped disturbance observation value provided by the adaptive weighted parallel LESO. The disturbance in the mathematical modeling of step 2 includes: Design the sliding surface according to equation (7) in step 2: Where s is the sliding surface function, c>0 is the sliding integral coefficient, and x ref is the given value of capacitor voltage, z1 is the estimated value of capacitor voltage output by adaptive weighted observer, and e is the error between given voltage and estimated voltage; When on the sliding surface, s = 0, let get: Replace the system state values ​​x1 and x2 in equation (7) in step 2 with the observed values ​​z1 and z2, and substitute them into equation (14) to obtain the equivalent control quantity u eq : When outside the sliding surface, the approach rate For the improved superhelical approach rate: Where, k1, k2, k3 are the parameters to be designed for the sliding mode controller and are all greater than 0, r is the coefficient to be designed, tanh(s) is the hyperbolic tangent function, and sgn(s) is the sign function; The above approach rate is adaptively adjusted so that the approach speed increases when the approach rate is far away from the sliding surface and decreases when it is close to the sliding surface: In the formula, a and b are adjustment factors; According to equations (14) and (17), the switching control amount u is calculated sw : u sw =k1ρ1(|s|)tanh(s)+k2∫sgn(s)dt+k3ρ2(|e|)s(18), Based on formula (18), the control u of the voltage sliding mode ADRC output is calculated as: in=in eq +in sw (19).

7. The off-grid control method of a T-type energy storage converter based on improved sliding mode anti-disturbance control according to claim 1 is characterized in that: In step 6, based on the power outer loop, the voltage loop and the current inner loop, an output modulation signal is obtained; the output modulation signal is transmitted to the SVPWM module with center point balance to control the main topology structure of the T-type three-level converter, including: The control voltage output by the power-voltage-current three-loop calculated in step 1 to step 5 is sent to the inverse Park transformation to obtain a three-phase voltage modulation wave; the modulation wave is sent to the SVPWM modulator after per-unit processing and the balance factor method of the midpoint charge conservation principle is used to adjust the output PWM wave; the operation of the T-type converter is controlled by 12 PWM control signals.

8. An electronic device comprising a memory, a processor, and a computer program stored in the memory and executable on the processor, characterized in that: When the processor executes the program, the steps of the method according to any one of claims 1 to 7 are implemented.

9. A computer-readable storage medium having a computer program stored thereon, characterized in that: When the computer program is executed by a processor, the steps of the method according to any one of claims 1 to 7 are implemented.

Images