General vector voltage control method and system for network-forming converter

Abstract

The invention discloses a general vector voltage control method and system for a network-constructed converter, relates to the technical field of converter control, and aims to solve the problem of long voltage control and regulation time of a high-voltage network-constructed converter. The general vector voltage control system comprises a current inner ring and a voltage outer ring; pIF control is adopted in the voltage outer loop, outer loop voltage control is conducted on the high-voltage grid-forming type current converter through voltage proportional-integral negative feedback and current feedforward, current converter output is composed of proportional-integral negative feedback and current feedforward, and the proportional-integral negative feedback comprises a proportional element and an integral element; the alternating current feedforward link is added in the voltage outer loop control without the alternating current capacitor, so that the adjusting time can be shortened, and meanwhile, the stability of the high-voltage grid-forming type converter system without the alternating current capacitor is improved.

Description

Technical Field

[0001] The present invention relates to the technical field of converter control, and particularly relates to a general vector voltage control method and system for a grid-forming converter. Background Art

[0002] The grid-forming converter has the independent operation ability without relying on traditional synchronous generators and has become a key equipment for constructing a 100% renewable energy power grid. Vector control has been widely applied to the control of grid-connected converters. Traditional two-level and three-level converters are mainly used in medium and low voltage scenarios. Since the high-frequency harmonic content of their output voltage is relatively large, they need to be connected to the grid through a low-pass filter composed of a series inductor and a parallel capacitor. The grid-forming converter operates as a stable voltage source. The traditional vector control is composed of a cascaded current inner loop and a voltage outer loop, where the voltage outer loop is constructed based on the charge and discharge relationship of the AC parallel capacitor. The virtual synchronous machine control simulates the inertia and damping characteristics of the synchronous generator by adjusting the frequency command value of the voltage outer loop of the converter. The rated AC voltage of the high-voltage grid-forming converter can reach dozens to hundreds of kilovolts and has been applied in flexible AC-DC power transmission and directly-connected energy storage systems. The modular multilevel converter (MMC) has become the mainstream high-voltage grid-forming converter. Due to the large number of levels, the harmonic content of its output voltage has been significantly reduced, and there is no need to use an AC parallel capacitor anymore. In addition, for cost reasons, high-voltage parallel capacitors are generally not installed in high-voltage grid-forming converters either. Therefore, the traditional voltage outer loop control constructed based on the charge and discharge relationship of the parallel capacitor cannot be directly applied to high-voltage grid-forming converters without AC capacitors, and the voltage outer loop control without AC capacitors has gradually received attention. Currently, there is a voltage outer loop control method without AC capacitors based on AC voltage proportional integral negative feedback in the existing technology.

[0003] For example, there is a Chinese patent with the publication number CN115051404A, which relates to an AC voltage control method for a high-voltage grid-forming converter. The present invention discloses an AC voltage control method for a high-voltage grid-forming converter. The present invention introduces the voltage-current proportional relationship to construct a new outer loop voltage control of the converter, which constitutes a control architecture for a high-voltage grid-forming converter together with the inner loop current control. The voltage-current proportional outer loop control does not depend on the shunt capacitive filter on the AC side of the converter and is suitable for high-voltage large-capacity multilevel converters with high output waveform quality. The shunt capacitive filter can be cancelled, effectively reducing the system cost; during a grid fault, the grid-forming converter can automatically and quickly enter the positive-sequence current limiting and negative-sequence current suppression states under the action of the voltage-current proportional control, avoiding overcurrent blocking and damage of the converter, and automatically recovering to normal operation after the grid fault is cleared; however, the integral link final value regulation of the Chinese patent with the publication number CN115051404A has higher requirements and its control adjustment time is longer. Summary of the Invention

[0004] To solve the problem of long voltage control adjustment time of the high-voltage network-forming converter, the present invention proposes a general vector voltage control method and system for the network-forming converter. By adding an AC current feedforward link to the voltage outer loop control without AC capacitors, the adjustment time can be shortened, and at the same time, the complexity and stability of the high-voltage network-forming converter system without AC capacitors can be improved.

[0005] To achieve the above object, the present invention adopts the following technical solutions: A general vector voltage control method for a network-forming converter, including a current inner loop and a voltage outer loop; the voltage outer loop adopts PIF control: the outer loop voltage control of the high-voltage network-forming converter is carried out through voltage proportional-integral negative feedback and current feedforward. The output of the converter is composed of proportional-integral negative feedback and current feedforward, and the proportional-integral negative feedback includes a proportional link and an integral link.

[0006] In this technical solution, the voltage outer loop of the high-voltage network-forming converter without AC capacitors adopts PIF control, which not only includes a voltage proportional-integral negative feedback link, but also includes a current feedforward link. By adding an AC current feedforward link, the control adjustment time of the high-voltage network-forming converter system without AC capacitors can be shortened.

[0007] Preferably, the dq-axis current command values of the current inner loop of the converter are composed of the proportional-integral negative feedback of the dq-axis voltage of the common electrical connection point and the dq-axis current feedforward of the converter output. The inputs of the proportional link and the integral link are obtained by subtracting the dq-axis voltage component at the common connection point from the dq-axis voltage command value at the common connection point.

[0008] Preferably, by canceling the current feedforward link in the PIF control, PI control is obtained, that is, voltage proportional-integral control.

[0009] Preferably, under PI control, the dq-axis current command value of the converter is obtained by multiplying the transfer function by the first difference, and the first difference is obtained by subtracting the dq-axis voltage component at the common connection point from the dq-axis voltage command value at the common connection point.

[0010] Preferably, in the steady state, the final values of the integral links in the PIF control are all 0.

[0011] Preferably, by canceling the integral link in the PIF control, PF control is obtained, and the PF control includes current feedforward and a voltage proportional link.

[0012] Preferably, under PF control, the dq-axis current command value of the converter is obtained by adding the dq-axis current components of the converter and a proportional part, and the proportional part is obtained by multiplying a proportional gain by a first difference, where the first difference is equal to the difference between the dq-axis voltage command value at the point of common coupling (PCC) and the dq-axis voltage components at the PCC.

[0013] Preferably, the dq-axis current components at the PCC are equal to the dq-axis current components output by the converter.

[0014] The present invention also adopts the following technical solution: a high-voltage network-forming converter system without AC capacitors, which adopts the general vector voltage control method of a network-forming converter as claimed above, and is characterized in that it includes a modular multilevel converter, the modular multilevel converter is connected with a reactor, the reactor is connected with a transformer, and the transformer is connected with a load.

[0015] Preferably, there is no need to set up a shunt capacitive filter at the PCC.

[0016] The beneficial effects of the present invention are as follows: it can shorten the control adjustment time of the high-voltage network-forming converter system without AC capacitors, and improve the complexity and stability of the high-voltage network-forming converter system without AC capacitors. Description of the Drawings

[0017] Figure 1 is the vector control structure diagram of the general vector voltage control method of a network-forming converter of the present invention.

[0018] Figure 2 is the structure diagram of the high-voltage network-forming converter system without AC capacitors of the present invention.

[0019] Figure 3 is the trajectory diagram of the change of all eigenvalues of the system when ki2 increases in Embodiment 1 of the present invention.

[0020] Figure 4 is the trajectory diagram of the change of the dominant eigenvalues of the system when ki2 increases in Embodiment 1 of the present invention.

[0021] Figure 5 is the trajectory diagram of the change of all eigenvalues of the system when kp2 increases in Embodiment 1 of the present invention.

[0022] Figure 6 is the trajectory diagram of the change of the dominant eigenvalues of the system when kp2 increases in Embodiment 1 of the present invention.

[0023] Figure 7 is the trajectory diagram of the change of all eigenvalues of the system when RL increases in Embodiment 1 of the present invention.

[0024] Figure 8This is the trajectory diagram of the dominant eigenvalue change of the system when RL increases in Embodiment 1 of the present invention.

[0025] Figure 9 This is the trajectory diagram of all eigenvalue changes of the system when LL increases in Embodiment 1 of the present invention.

[0026] Figure 10 This is the trajectory diagram of the dominant eigenvalue change of the system when LL increases in Embodiment 1 of the present invention.

[0027] Figure 11 This is the voltage command tracking response diagram of the grid-forming MMC under three controls in Embodiment 1 of the present invention.

[0028] Figure 12 This is the load change response diagram of the grid-forming MMC under two controls in Embodiment 1 of the present invention.

[0029] Figure 13 This is the active power and reactive power diagram output by the grid-forming MMC at the PCC in Embodiment 1 of the present invention.

[0030] Reference numerals: Modular multilevel converter 1; Reactor 2; Transformer 3; Load 4. Detailed implementation manners

[0031] To make the objectives, technical solutions and advantages of the present invention clearer and more understandable, the present invention will be further described in detail below with reference to the accompanying drawings and embodiments. It should be understood that the specific implementation manners described herein are only the best embodiments of the present invention, which are only used to explain the present invention and do not limit the protection scope of the present invention. All other embodiments obtained by those of ordinary skill in the art without creative efforts fall within the protection scope of the present invention.

[0032] Embodiment 1 This embodiment provides a general vector voltage control method for a grid-forming converter, which introduces current feedforward on the basis of voltage proportional-integral control, and can shorten the control adjustment time of a high-voltage grid-forming converter system without AC capacitors.

[0033] In the traditional medium- and low-voltage grid-forming converter structure, a voltage source converter (VSC) is connected to a load through a reactor, and the inductance and resistance of the reactor are represented by L and R respectively.

[0034] Since the high-frequency harmonic content of the output voltage of traditional two-level and three-level converters is relatively large, a shunt capacitive filter C is installed at the point of common coupling (PCC) to provide a path for high-frequency harmonics to enter the ground.

[0035] The total inductance and total resistance of the power line and load are represented by L L and R L respectively.

[0036] The three-phase AC voltages and currents output by the converter are represented by vectors v and i respectively, and the three-phase AC voltages and currents output at the PCC are represented by vectors u and i L respectively.

[0037] In a traditional network-forming converter, the output current of the converter is determined by the voltage drop across the reactor, and the voltage at the PCC is determined by the charging and discharging of the shunt capacitor.

[0038] In the per-unit value and dq coordinate systems, the voltage and current relationships of a traditional converter are as follows: Lsi d / ω0 = v d -u d -Ri d +ωLi q (1a) Lsi q / ω0 = v q -u q -Ri q -ωLi d (1b) Csu d / ω0 = i d -i Ld +ωCu q (2a) Csu q / ω0 = i q -i Lq -ωCu d (2b) where i d and i q are the dq-axis current components of the converter, i Ld and i Lq are the dq-axis current components at the PCC; v d and v q are the dq-axis voltage components of the converter, u d and u q are the dq-axis voltage components at the PCC; ω and ω0 are the per-unit values of the actual angular frequency and the rated angular frequency of the system, and s is the Laplace operator.

[0039] The synchronous signal for the dq-axis transformation takes the value of where the initial phase angle θ0 is taken such that the PCC voltage direction coincides with the d-axis, so that the dq-axis voltage components at the PCC are the voltage amplitude and 0 respectively.

[0040] Under the per-unit value system, the traditional current inner loop and voltage outer loop controls are represented by Equations (4) and (5) respectively: where i d * and i q * are the dq-axis current command values of the converter respectively, U * and 0 are the dq-axis voltage command values at the PCC; k p1 and k i1 , k p2 and k i2 are two groups of PI control parameters respectively.

[0041] The overall structure of the vector control is as Figure 1 shown, and its output is the three-phase voltage command value v abc of the converter. The current inner loop control can effectively avoid overcurrent of the converter by limiting the current command value. The voltage outer loop control is achieved by adjusting the current command value of the current inner loop control.

[0042] The structure of the high-voltage grid-forming MMC is as Figure 2 shown, where L is the sum of half of the converter arm reactance value and the transformer leakage reactance value, and R is the sum of half of the arm resistance value and the transformer equivalent resistance value.

[0043] Since the MMC uses a large number of sub-modules in cascade to reach a high voltage level, the number of step levels of its output AC voltage is also large, and the harmonic content is significantly reduced, so the shunt filter capacitor on the AC side can be cancelled.

[0044] On the other hand, the cost of the high-voltage shunt filter capacitor is very high, and the parallel filter capacitor is generally not installed in the actual high-voltage converter system.

[0045] The output current of the MMC can also be determined by Equation (1), so the traditional current inner loop control is still applicable. However, the voltage at the PPC is no longer determined by the charging and discharging of the parallel capacitor, but by the load state as follows: u d = L L si d / ω0 + R L i d - ωL L i q (6a) u q = L L si q / ω0 + R L i q + ωL L i d (6b).

[0046] Because the traditional voltage outer-loop control is designed according to the charge and discharge relationship of the shunt capacitor in Equation (2), it cannot be directly applied to the network-forming converter without an AC capacitor.

[0047] Based on Equation (2) and Equation (5), the voltage outer-loop control without an AC capacitor can be derived. By setting the shunt AC capacitor C in Equation (5) to 0 and equating it to an infinite impedance, the voltage outer-loop control without an AC capacitor, i.e., PIF control, can be obtained as follows: where the dq-axis current components at the PCC are equal to the dq-axis current components output by the converter.

[0048] The voltage outer-loop control without an AC capacitor in Equation (7) consists of voltage proportional-integral negative feedback and current feedforward. Therefore, PIF control can also be regarded as a special case where the value of the shunt AC capacitor in the network-forming converter in the traditional voltage outer-loop control is set to 0.

[0049] The two types of voltage outer-loop control without an AC capacitor, namely PI control and PF control, can be obtained by simplifying PIF control.

[0050] By canceling the current feedforward link in PIF control, PI control, i.e., voltage proportional-integral control, can be obtained:

[0051] Due to the role of the integral link in PIF and PI control, the final steady-state error is 0, and the voltage can track the command value as follows: u d (t = ∞) = U * (9a) u q (t = ∞) = 0 (9b).

[0052] Due to the role of the integral link in the current inner-loop control, the final steady-state error is also 0, and the current can track the command value as follows:

[0053] Substituting Equation (9) and Equation (10) into Equation (7) of PIF control gives

[0054] Substituting Equation (9) and Equation (10) into Equation (8) of PI control gives

[0055] Under steady state, the final values of the integral links in the PIF control of Equation (7) are all 0, while the final values of the integral links in the PI control of Equation (8) are the dq-axis currents respectively.

[0056] Since the final value regulation requirement of the integral link in PI control is higher, its control regulation time will be longer than that of PIF control regulation.

[0057] After canceling the integral link in PIF control, PF control can be obtained:

[0058] Substituting Equation (10) into Equation (13) shows that even without the integral link, the dq-axis voltage components can still track their command values under steady state.

[0059] Based on the above steady-state analysis, after introducing the current feedforward link, the control difficulty of the voltage outer loop control is reduced, and accordingly, the voltage control regulation time can be reduced. The regulation times of PIF and PF controls are less than that of PI control.

[0060] Next, a dynamic modeling and stability analysis are carried out on the general vector voltage control method of a network-forming converter of the present invention.

[0061] According to Equation (6), the control effect of the voltage outer loop control without AC capacitance is easily affected by the load state. To analyze the performance of the voltage control itself and obtain the large-signal dynamic model of the converter system, the converter is connected to a passive network.

[0062] Based on Figure 2 the high-voltage network-forming MMC system, the system performance under PIF and PF controls is modeled, where PF control can be regarded as obtained by taking the integral parameter k i2 in PIF control as 0. The current inner loop control of Equation (4) can be expressed as where The PIF voltage outer loop control of Equation (7) can be expressed as where M od =k i2 (U * -u d ) / s(17a) M oq =k i2 (-u q ) / s(17b) Substituting Equation (14) and Equation (16) into Equation (1) gives Lsi d / ω0=-Ri d +Mid +k p1 M od -k p1 k p2 u d +k p1 k p2 U * (18a) Lsi q / ω0=-Ri q +M iq +k p1 M oq -k p1 k p2 u q (18b) The state variables are selected as follows X=[i d i q M id M iq M od M oq T (19) Substituting Equation (6) into Equation (18) gives the differential equations of the dq-axis current components i d and i q of the converter. Substituting Equation (6), Equation (16) and Equation (18) into Equation (15) gives the differential equations of M id and M iq Substituting Equation (6) and Equation (18) into Equation (17) gives the differential equations of M od and M oq Let the grid-forming converter operate at the rated frequency, and its large-signal linearization model can be expressed by Equation (20):

[0063] The specific parameters are shown in Equations (21) to (25): Where

[0064] Next, eigenvalue analysis is performed on the general vector voltage control method of a grid-forming converter of the present invention.

[0065] Analyze the characteristics of the eigenvalues changing with the change of system parameters. The system parameters and control parameters of the grid-forming MMC system are as follows Table 1 System parameters Parameter Value Rated AC Voltage / kV 36.5 Rated AC Frequency / Hz 50 Converter Topology MMC Converter Rated Capacity / MVA 20 Converter Rated AC Voltage / kV 31 Converter Rated DC Voltage / kV + / -30 Number of Sub - modules per Arm / piece 49 Arm Inductance / mH 53 Arm Resistance / Sub - module Capacitance / uF 4650 Transformer Rated Capacity / MVA 20 Transformer Structure Y Transformer Turns Ratio / (kV / kV) 36.5 / 31 Transformer Leakage Reactance / pu 0.08 Grid Equivalent Resistance / Grid Equivalent Inductance / mH 110 Table 2 Control parameters​ The eigenvalues of PIF control and PF control are represented by black 'x' and red '+', respectively. The control parameter k i2 varies from 0 to 5, and the variation of the system eigenvalues is as shown Figure 3 below, where the enlarged dominant eigenvalues are as shown Figure 4 below.

[0066] Under PIF control, when k i2 takes the value of 3, the relationship between the state variables and the eigenvalues is described by the participation factor matrix of Equation (26), where the upper eigenvalue sorting corresponds to Figure 3 the order from left to right. According to Equations (19) and (26), the first group of eigenvalues λ 11 and λ 12 are mainly related to the dq-axis currents i d and i q . The second group of eigenvalues λ 13 and λ 14 are mainly related to the integral values M id and M iq of the current inner-loop control; the third group of eigenvalues λ 15 and λ 16 are the dominant eigenvalues and are mainly related to the integral values M od and M oq of the voltage outer-loop control. As the value of k i2 increases, the dominant eigenvalues will move away from the imaginary axis, and the system stability is enhanced.

[0067] Under PF control, the integral coefficient k i2 is 0, and the integral values M od and M oq of the voltage outer-loop control in Equation (17) are always 0, and the state variables and eigenvalues are reduced to 4. The relationship between the state variables and the eigenvalues is described by the participation factor matrix of Equation (27), where the upper eigenvalue sorting corresponds to Figure 3 the order from left to right. According to Equation (27), the first group of eigenvalues λ 11 and λ 12 are mainly related to the dq-axis currents i d and i q . The second group of eigenvalues λ 13 and λ 14 are mainly related to the integral values M id and M iq of the current inner-loop control. As the value of k i2 increases, the dominant eigenvalues will move away from the imaginary axis, and the system stability is enhanced.

[0068] Under the two controls, the control parameter k p2 increases from 0.5 to 5, and the changes in the system eigenvalues are as Figure 5 shown, where the enlarged dominant eigenvalue is as Figure 6 shown. It can be seen that the system can maintain stability under both controls, but the dominant eigenvalue of the PIF control is closer to the imaginary axis than that of the PF control.

[0069] Under the two controls, when the grid parameter R L increases from 0.85 to 3.25 pu, the changes in the system eigenvalues are as Figure 7 shown, where the enlarged dominant eigenvalue is as Figure 8 shown. When the grid parameter L L increases from 0.4 to 2 pu, the changes in the system eigenvalues are as Figure 9 shown, where the enlarged dominant eigenvalue is as Figure 10 shown.

[0070] Based on the large-signal analysis, the systems under PIF and PF controls can both maintain stability within a large load change range. Therefore, it is feasible to apply the two controls to the high-voltage grid-forming MMC. A set of left eigenvalues related to the dq-axis currents i d and i q under the two controls are relatively close, indicating the similarity of the current responses under the two controls; so according to Equation (6), the voltage responses under the two controls also have similarities. The structure of the PF control is simpler and more stable than that of the PIF control.

[0071] Through time-domain simulation, a comparative study is carried out on the system responses of the high-voltage grid-forming MMC under three voltage outer-loop controls without AC capacitors.

[0072] Regarding the voltage command tracking response, initially, the grid-forming MMC supplies power to the load normally. At 1.6 s, the d-axis voltage command increases from 0.85 pu to 1.15 pu, and the dq-axis current components and dq-axis voltage components at the PCC are as Figure 11 (a), Figure 11 (b), Figure 11 (c) and Figure 11 (d) shown.

[0073] After the voltage command changes, the d-axis voltage at the PCC can track the voltage command under all three controls, but there are differences in the control adjustment time. Although the integral coefficient in the PI control is 6 times that in the PIF control, the adjustment time of the PI control is about 0.15 s, while the adjustment times of the PIF and PF controls are only about 0.04 s. It can be seen that the current feed-forward link can significantly reduce the adjustment time of the voltage outer-loop control, which is consistent with the steady-state analysis results.

[0074] Regarding the response to load changes, the responses of the grid-forming MMC under PIF and PF control to load changes are simulated and compared here. At 1.6 s, the resistance value and inductance value of the load increase from 71.5 Ω and 110 mH to 113.2 Ω and 118 mH respectively, and the dq-axis current components and dq-axis voltage components at the PCC are as Figure 12 (a), Figure 12 (b), Figure 12 (c) and Figure 12 (d) shown. The active and reactive powers output at the PCC are as Figure 13 (a) and Figure 13 (b) shown.

[0075] After the load changes, the regulation time of both voltage outer-loop controls is less than 0.03 s. Due to the decrease in load, both the active and reactive powers output at the PCC decrease. Therefore, the grid-forming MMC can normally respond to load changes under both controls.

[0076] Embodiment 2 This embodiment provides a high-voltage grid-forming converter system without AC capacitors, which adopts the above-mentioned general vector voltage control method of a grid-forming converter, referring to Figure 2 , and includes a modular multilevel converter 1, a reactor 2, a transformer 3, and a load 4. There is no need to set up a shunt capacitive filter at the common connection point.

[0077] The modular multilevel converter is composed of cascading a plurality of sub-modules (Sub-module, SM) with the same structure. The structure of the sub-module can be divided into three types: half H-bridge type, full H-bridge type, and double-clamped type sub-module type. The modular multilevel converter is connected with a reactor, the reactor is connected with a transformer, and the transformer is connected with a load.

[0078] In traditional medium- and low-voltage grid-forming converters, the voltage source converter is connected to the load through a reactor. The reactor includes an inductor L and a resistor R. Since the high-frequency harmonic content of the output voltage of traditional two-level and three-level converters is relatively large, it is necessary to install a shunt capacitive filter C at the common connection point to provide a path for high-frequency harmonics to enter the ground.

[0079] In a high-voltage grid-forming converter system without AC capacitors in this embodiment, L is the sum of half of the converter arm reactance value and the transformer leakage reactance value, and R is the sum of half of the arm resistance value and the equivalent resistance of the transformer. Since the MMC uses a large number of sub-modules in cascade to reach a high voltage level, the number of stepped levels of its output AC voltage is also large, and the harmonic content is significantly reduced. Therefore, there is no need to shunt a filter capacitor on the AC side.

[0080] Since the cost of high-voltage shunt filter capacitors is very high, in a high-voltage network-forming converter system without AC capacitors in this embodiment, there is no need to install shunt filter capacitors, which can greatly reduce the cost.

[0081] The output current of a traditional medium- and low-voltage network-forming converter is determined by the voltage drop across the reactor, and the voltage at the common connection point is determined by the charging and discharging of the shunt capacitor.

[0082] In a high-voltage network-forming converter system without AC capacitors in this embodiment, the output current of the modular multilevel converter can also be determined by the voltage drop across the reactor, and its current inner-loop control method is the same as that of the traditional medium- and low-voltage network-forming converter.

[0083] However, the difference is that in a high-voltage network-forming converter system without AC capacitors, the voltage at the common connection point is no longer determined by the charging and discharging of the shunt capacitor, but by the load state. Therefore, the traditional voltage outer-loop control designed according to the charging and discharging relationship of the shunt capacitor cannot be directly applied to the network-forming converter without AC capacitors.

[0084] A high-voltage network-forming converter system without AC capacitors in this embodiment adopts the above-mentioned general vector voltage control method of a network-forming converter. The voltage outer-loop of the modular multilevel converter adopts PIF control, and a current feed-forward link is introduced on the basis of voltage proportional-integral control, which can not only shorten the control adjustment time of the high-voltage network-forming converter system without AC capacitors, but also improve the complexity and stability of the high-voltage network-forming converter system without AC capacitors.

Claims

1. A general vector voltage control method for a grid-type converter, characterized in that: It includes a current inner loop and a voltage outer loop; the voltage outer loop adopts PIF control: the outer loop voltage of the high-voltage grid-type converter is controlled by voltage proportional integral negative feedback and current feedforward, and the inner loop of the converter current is dq The shaft current command value is composed of proportional integral negative feedback and current feedforward, and the proportional integral negative feedback includes a proportional link and an integral link.

2. The universal vector voltage control method for a grid-type converter according to claim 1, characterized in that: Inverter current inner loop dq The axis current command value is determined by the common connection point dq Proportional integral negative feedback of shaft voltage and converter output dq The shaft current feedforward is added, and the input of the proportional link and the integral link is the common connection point. dq The shaft voltage command value and the common connection point dq The shaft voltage components are obtained by taking the difference.

3. The universal vector voltage control method for a grid-type converter according to claim 1, characterized in that: By canceling the current feedforward link in the PIF control, PI control, namely voltage proportional integral control, is obtained.

4. The universal vector voltage control method for a grid-type converter according to claim 3, characterized in that: Under PI control, the inner loop of the converter current dq The axis current command value is determined by the common connection point dq The proportional integral negative feedback of the shaft voltage is obtained, and the proportional integral negative feedback includes a proportional link and an integral link, and the input of the proportional link and the integral link is at the common connection point dq The shaft voltage command value and the common connection point dq The shaft voltage components are obtained by taking the difference.

5. A universal vector voltage control method for a grid-type converter according to any one of claims 1 to 4, characterized in that: In steady state, the final value of the integral input in PIF control is 0.

6. The universal vector voltage control method for a grid-type converter according to claim 1, characterized in that: By canceling the integral link in the PIF control, the PF control is obtained, and the PF control includes a current feedforward and a voltage proportional negative feedback link.

7. A universal vector voltage control method for a grid-type converter according to claim 6, characterized in that: Under PF control, the inner loop of the converter current dq The axis current command value is determined by the common connection point dq Proportional negative feedback of the shaft voltage and the converter output dq The proportional negative feedback is obtained by multiplying the proportional gain by the first difference, and the first difference is equal to the common connection point. dq The shaft voltage command value and the common connection point dq The difference between the shaft voltage components.

8. A high voltage grid-type converter system without AC capacitor, using a universal vector voltage control method for a grid-type converter according to any one of claims 1 to 7, characterized in that: The invention comprises a converter, wherein the converter is connected to a reactor, the reactor is connected to a transformer, and the transformer is connected to a load.

9. A universal vector voltage control method for a grid-type converter according to claim 8, characterized in that: No parallel capacitive filter is required at the common connection point.

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