Single-stage bidirectional DC / AC converter and control method thereof

Abstract

The invention relates to the technical field of power electronics, and discloses a single-stage bidirectional DC / AC converter and a control method thereof. The single-stage bidirectional DC / AC converter comprises a pre-stage circuit and a post-stage circuit which are multiplexed by a switching tube. Wherein the pre-stage circuit is an L-LLC resonant converter or a CLLC resonant converter, and the post-stage circuit is an interleaved Buck converter. And an output inverter bridge at the secondary side of the transformer is replaced by a bridgeless structure. According to the single-stage bidirectional DC / AC converter, the number of switching tubes is reduced through a multiplexing technology, the topological structure can be simplified, the stability is improved by reducing the damage probability, and the cost is reduced; furthermore, by utilizing a soft switching technology of a pre-stage circuit, ZVS of four primary-side switching tubes and half secondary-side switching tubes can be realized, the switching loss is greatly reduced, and the working efficiency is improved; an output inverter bridge is omitted, the number of switching tubes can be further reduced, and the power density and the working efficiency are improved.

Description

Technical Field

[0001] The present invention relates to the field of power electronics technology, and particularly to a single-stage bidirectional DC / AC converter and its control method. Background Art

[0002] In V2G technology, the bidirectional DC / AC converter is the hardware core and the instruction execution device. To achieve the functions of V2G, it is required that it can not only achieve bidirectional power flow, but also execute the scheduling instructions of the smart grid for rapid adjustment and control the input and output power quality. In addition, for the bidirectional OBC (On Board Charger), it is also required to have the advantages of high efficiency, high reliability and high power density. Therefore, it is difficult to design an on-vehicle bidirectional converter applicable to V2G technology.

[0003] At present, the research on on-board bidirectional chargers for electric vehicles mainly focuses on the AC / DC perspective, and there is less discussion from the DC / AC perspective. U. Sharma and B. Singh proposed a bidirectional on-board charger with multi-stage constant current charging ability, considering the voltage and current fluctuations on the grid side, but did not consider the impact of voltage fluctuations on the DC side. M. Q. Chen, B. Liu and L. Jing proposed a simplified control strategy using a corrected current reference phase, but it is mainly applied to the control of reactive power for bidirectional power flow. P. Nayak, S. K. Pramanick and K. Rajashekara proposed a single-stage DC / AC converter topology with fewer switching devices and further improved the converter efficiency using soft-switching technology. However, this topology cannot achieve bidirectional power flow and has limited application scenarios. H. Heydari-doostabad and T. O’Donnell proposed a new wide-range bidirectional DC / DC converter with high efficiency, but its topology is relatively complex and it needs to be combined with a post-stage DC / AC converter to achieve the V2G function. Y. Chen, Z. Jiang, L. Wei, Y. Zhang and J. Jiang proposed an asymmetric full-bridge DC / AC converter, which realizes single-stage conversion through a multi-way switch and suppresses common-mode current from the topological principle, but did not consider the impact of input-side voltage fluctuations. L. Zhu, H. Wu, T. Mu, F. Yang and X. Ma proposed an asymmetric three-level dual-input bidirectional DC / AC converter, which considers the variation of the electric vehicle battery voltage with the battery capacity. However, the input and output of this converter are not isolated, and it is difficult to ensure safety in high-power scenarios. In addition, the converter has a low operating frequency and it is difficult to improve the power density. S. Taghizadeh, M. J. Hossain and J. Lu proposed a two-stage bidirectional DC / AC converter, which uses an isolated structure and soft-switching technology to reduce switching losses. However, the two-stage topology has a large number of switching devices and it is difficult to improve the power density.

[0004] It can be seen that in terms of converter topology, with the continuous development of electric vehicles in recent years, higher requirements have been put forward for the charging speed and portability of bidirectional OBCs. This requires bidirectional OBCs to have a higher power density, that is, the charger has a smaller volume under the same power condition; and a smaller volume also means less heat dissipation, which also poses certain requirements for the efficiency of bidirectional OBCs. Traditional DC / AC converters use transformers for isolation and voltage level conversion, but they often operate under power frequency conditions. Since the volume of magnetic components such as transformers and inductors decreases with the increase in frequency, the volume of transformers and inductors and other magnetic components is often large under power frequency conditions and cannot be applied to bidirectional OBCs for electric vehicles. In recent years, the directly driven single-pole DC / AC converter studied is not suitable for high-power bidirectional OBCs due to its complex control strategy and lack of isolation function. Although the two-stage bidirectional DC / AC converter has both isolation functions and a simple two-stage conversion control strategy, it uses a large number of switching tubes, has large losses, and it is difficult to improve the power density.

[0005] In terms of control strategy, traditional control strategies often have only one degree of freedom. For example, traditional PWM (Pulse width modulation) control realizes the control of the output by changing the duty cycle, but it also means that it cannot adjust the voltage of the intermediate bus capacitor. Converters using traditional control strategies often can only operate under specific designed working conditions. When the working conditions change and the input voltage changes, the performance of the converter will decline, resulting in the loss of ZVS, reduced efficiency, or even inability to work. The voltage characteristic of the electric vehicle battery is that the voltage decreases during the discharge process, so traditional control strategies are difficult to apply to bidirectional OBCs for electric vehicles.

[0006] In terms of modeling and analysis, most of the existing parameter design and analysis methods for LLC resonant converters are based on frequency domain analysis methods. However, the accuracy of frequency domain analysis methods is low, resulting in large analysis errors, poor results of parameter design, and the need for multiple iterations. Summary of the Invention

[0007] To solve the above technical problems existing in the prior art, the present invention aims to provide a cascaded single-stage topology, which has simple control, isolation function, few switching tubes, can achieve soft switching, has high efficiency, and can realize a DC / AC converter with high power density and the corresponding control method.

[0008] Specifically, in the first aspect, the present invention provides a single-stage bidirectional DC / AC converter, including: Switching tubes Q1, Q2, Q3, and Q4 form a full bridge; the drains of switching tubes Q1 and Q3 are both connected to the positive pole of the DC power supply, and the sources of switching tubes Q2 and Q4 are both connected to the negative pole of the DC power supply; The two output terminals of the full bridge are respectively connected to both ends of the primary side of the transformer; The same-name terminals of the secondary side of the transformer are sequentially connected to the different-name terminal of the secondary side of the transformer via inductor L r , inductor L rm_a and capacitor C r ; Further included are: The output terminal of inductor L B1 is connected to the same-name terminal of the secondary side of the transformer via inductor L r , and the output terminal of inductor L B2 is connected to the different-name terminal of the secondary side of the transformer via capacitor C r ; The output terminals of inductor L B2 and inductor L B1 are connected to each other; The source electrode of switching transistor Q5 and the drain electrode of switching transistor Q6 are both connected to the input terminal of inductor L B1 , and the source electrode of switching transistor Q7 and the drain electrode of switching transistor Q8 are both connected to the input terminal of inductor L B2 ; The drain electrodes of switching transistor Q H , switching transistor Q5 and switching transistor Q7 are all connected to the positive electrode of bus capacitor C bus , and the source electrodes of switching transistor Q L , switching transistor Q6 and switching transistor Q8 are all connected to the negative electrode of bus capacitor C bus ; In the single-stage bidirectional DC / AC converter, the output terminal of inductor L B1 is the first output terminal, and the source electrode of switching transistor Q H is the second output terminal.

[0009] In a second aspect, the present invention provides another single-stage bidirectional DC / AC converter, including a CLLC resonant converter; in the CLLC resonant converter, switching transistors Q1, Q2, Q3 and Q4 form a full bridge; the drain electrodes of switching transistors Q1 and Q3 are both connected to the positive electrode of the DC power supply, and the source electrodes of switching transistors Q2 and Q4 are both connected to the negative electrode of the DC power supply; the output terminal A of the full bridge is sequentially connected to the same-name terminal of the primary side of the transformer via capacitor C r1 and inductor L r1 , and the output terminal B is connected to the different-name terminal of the primary side of the transformer; the output terminal A is the midpoint between switching transistors Q1 and Q2, and the output terminal B is the midpoint between switching transistors Q3 and Q4; the same-name terminals of the secondary side of the transformer are sequentially connected to inductor L r2 and capacitor C r2 and then connected to inductor L B1 , and the different-name terminal is connected to inductor L B2 ; Further included are: The output terminals of inductor L B2 and inductor L B1 are connected to each other; The source electrode of switching transistor Q5 and the drain electrode of switching transistor Q6 are both connected to inductor L B1 at the input terminal, and the source electrode of switching transistor Q7 and the drain electrode of switching transistor Q8 are both connected to inductor L B2 at the input terminal; the drain electrodes of switching transistors Q H , Q5 and Q7 are all connected to the positive electrode of bus capacitor C bus , and the source electrodes of switching transistors Q L , Q6 and Q8 are all connected to the negative electrode of bus capacitor C bus ; In the single-stage bidirectional DC / AC converter, the output terminal of inductor L B1 is the first output terminal, and the source electrode of switching transistor Q H is the second output terminal.

[0010] Thirdly, this aspect also provides a control method for the above two single-stage bidirectional DC / AC converters, including the following steps: Step S1: Sample the bus voltage v bus , calculate the difference between the bus voltage v bus and the bus reference voltage V ref_bus ; Perform proportional-integral control on the difference to obtain the initial voltage control signal V fs* , and obtain the voltage control signal V fs after amplitude limiting. Input it into the PFM (Pulse Frequency Modulation) module to generate a triangular carrier wave v saw2_fs ; Shift the triangular carrier wave v saw2_fs by 180° to obtain a triangular carrier wave v saw1_fs ; Sample the output current i o of the single-stage bidirectional DC / AC converter and the grid voltage V AC , calculate the difference between K times the output current i o and the grid voltage V AC as the error reference signal; Perform proportional-integral control on the error reference signal to obtain the initial current control signal i con* , and obtain the current control signal i con after amplitude limiting; Step S2: Input the triangular carrier wave v saw2_fs and the current control signal i con into the negative input terminal and the positive input terminal of the first comparator respectively to obtain the control signal v gs2* ; Input the control signal v gs2* and the grid voltage V ACPerform logical operations to generate the control signal v of the switch tube Q1 gs1 and the control signal v of the switch tube Q2 gs2 ; The triangular carrier v saw1_fs and the current control signal i con Corresponding to the negative input terminal and the positive input terminal of the second comparator, the control signal v is obtained gs1* ; The control signal v gs1* and grid voltage V AC Perform logical operations to generate the control signal v of the switch tube Q3 gs3 and the control signal v of the switch tube Q4 gs4 ; The grid voltage V AC The positive input terminal of the third comparator and the negative input terminal of the fourth comparator are input respectively, and the inputs of the negative input terminal of the third comparator and the positive input terminal of the fourth comparator are both 0; the third comparator generates a switch tube Q L The control signal v gsL , the fourth comparator generates the switch tube Q H The control signal v gsH ; The control signal v gs2 and control signal v gs3 Perform an AND operation to generate the control signal v of the switch tube Q6 gs6 and the control signal v of the switch tube Q7 gs7 ; The control signal v gs1 and control signal v gs4 Perform an AND operation to generate the control signal v of the switch tube Q5 gs5 And the control signal v of the switch tube Q8 gs8 ; Logical operations satisfy: If the grid voltage V AC is positive and the control signal v gs* is 1, or if the grid voltage V AC is negative and the control signal v gs* is 0, the control signal vgsA is 1 and the control signal vgsB is 0; otherwise, the control signal vgA is 0 and the control signal vgsB is 1; wherein, If the control signal v gs* is the control signal v gs1* , then the control signal v gsA is the control signal v gs1 , control signal v gsB is the control signal v gs2 ; If the control signal v gs* is the control signal v gs2* , then the control signal vgsA is the control signal v gs3 , the control signal v gsB is the control signal v gs4 .

[0011] It can be seen that in the technical solution provided by the present invention, on the one hand, the single-stage bidirectional DC / AC converter adopts a single-stage structure, and the number of switching tubes is reduced through multiplexing technology, which can simplify the topology structure, reduce the probability of damage, improve stability, and reduce costs; further, the resonant converter can achieve safety isolation through a transformer, and by using the soft-switching technology of the L-LLC resonant converter or the CLLC resonant converter, ZVS of 4 switching tubes on the primary side and ZVS of half of the switching tubes on the secondary side can be achieved in the full working range, greatly reducing the switching loss and further improving the working efficiency; by replacing the output inverter bridge with the bridge-less technology, the number of switching tubes can be further reduced, and the power density and working efficiency can be improved. BRIEF DESCRIPTION OF THE DRAWINGS

[0012] Figure 1 is the topology diagram of the single-stage bidirectional DC / AC converter in the present invention.

[0013] Figure 2 is the key waveform diagram of the single-stage bidirectional DC / AC converter in the present invention.

[0014] Figure 3 is the equivalent circuit diagram of the working mode of the single-stage bidirectional DC / AC converter in the present invention from t0 to t1 during the positive half power frequency period.

[0015] Figure 4 is the equivalent circuit diagram of the working mode of the single-stage bidirectional DC / AC converter in the present invention from t1 to t2 during the positive half power frequency period.

[0016] Figure 5 is the equivalent circuit diagram of the working mode of the single-stage bidirectional DC / AC converter in the present invention from t2 to t3 during the positive half power frequency period.

[0017] Figure 6 is the equivalent circuit diagram of the working mode of the single-stage bidirectional DC / AC converter in the present invention from t3 to t4 during the positive half power frequency period.

[0018] Figure 7 is the equivalent circuit diagram of the working mode of the single-stage bidirectional DC / AC converter in the present invention from t4 to t5 during the positive half power frequency period.

[0019] Figure 8 is the equivalent circuit diagram of the working mode of the single-stage bidirectional DC / AC converter in the present invention from t5 to t6 during the positive half power frequency period.

[0020] Figure 9It is the equivalent circuit diagram of the working mode from t0 to t1 within the negative half-power frequency period of the single-stage bidirectional DC / AC converter in the present invention.

[0021] Figure 10 It is the equivalent circuit diagram of the working mode from t1 to t2 within the negative half-power frequency period of the single-stage bidirectional DC / AC converter in the present invention.

[0022] Figure 11 It is the equivalent circuit diagram of the working mode from t2 to t3 within the negative half-power frequency period of the single-stage bidirectional DC / AC converter in the present invention.

[0023] Figure 12 It is the equivalent circuit diagram of the working mode from t3 to t4 within the negative half-power frequency period of the single-stage bidirectional DC / AC converter in the present invention.

[0024] Figure 13 It is the equivalent circuit diagram of the working mode from t4 to t5 within the negative half-power frequency period of the single-stage bidirectional DC / AC converter in the present invention.

[0025] Figure 14 It is the equivalent circuit diagram of the working mode from t5 to t6 within the negative half-power frequency period of the single-stage bidirectional DC / AC converter in the present invention.

[0026] Figure 15 It is the topology diagram of an alternative scheme of the single-stage bidirectional DC / AC converter in the present invention.

[0027] Figure 16 It is the curve diagram of different battery voltages changing with time.

[0028] Figure 17 It is the gain curve diagram of the resonant converter.

[0029] Figure 18 It is the schematic diagram of the PWM / PFM hybrid control process of the single-stage bidirectional DC / AC converter in the present invention.

[0030] Figure 19 It is the schematic diagram of the structure of the D-PI module in the present invention.

[0031] Figure 20 It is the schematic diagram of the control process of the PFM module of the single-stage bidirectional DC / AC converter in the present invention.

[0032] Figure 21 It is the schematic diagram of the control process of the logic operation module of the single-stage bidirectional DC / AC converter in the present invention.

[0033] Figure 22 They are the 3 basic equivalent circuit diagrams of the converter under PWM control.

[0034] Figure 23It is the key waveform diagram of each mode of the converter under PWM control.

[0035] Figure 24 It is the schematic diagram of the solution process of the time-stepping iterative method model.

[0036] Figure 25 It is the simulation result and time-domain modeling diagram of the single-stage bidirectional DC / AC converter in the present invention under the input of 250V and output of 150V.

[0037] Figure 26 It is the simulation result and time-domain modeling diagram of the single-stage bidirectional DC / AC converter in the present invention under the input of 250V and output of 80V.

[0038] Figure 27 It is the schematic diagram of the relationship between the switching frequency of the single-stage bidirectional DC / AC converter in the present invention and the input voltage. Specific Embodiment

[0039] Hereinafter, the technical solution provided by the present invention will be further elaborated in detail with reference to the accompanying drawings.

[0040] 1. Topology of the single-stage bidirectional DC / AC converter: From the perspectives of safety and reliability, etc., it is usually required that high-power converters have the function of electrical isolation. In power electronic converters, transformers are usually used to achieve the function of electrical isolation. Traditional DC / AC converters use transformers to achieve electrical isolation and generally operate under low-frequency conditions. The transformers are large in volume and not suitable for distributed micro-inverters and electric vehicle V2G systems. The volume of magnetic components such as transformers and inductors will decrease with the increase of frequency. Therefore, high-frequencyization is an important development direction of power electronics and an important means to make power electronic converters better meet many basic requirements such as volume and power density.

[0041] High frequency operation is the main method to achieve high power density in switching power supplies. However, it often leads to an increase in the switching losses of power devices, which is not conducive to achieving high efficiency in switching power supplies. The use of soft-switching technology can reduce the switching losses of switching devices during turn-on and turn-off instants, increase the overlapping time of the voltage across the switching devices and the current passing through them, thereby reducing the switching losses of the switching devices, improving the efficiency of the converter, and enhancing stability by reducing the probability of damage. The resonant converter is a typical soft-switching power electronic converter topology. In order to reduce the volume of magnetic components such as transformers and improve power density, the operating frequency of the single-stage bidirectional DC / AC converter is increased while reducing the switching losses. In the present invention, the single-stage bidirectional DC / AC converter adopts an L-LLC resonant converter, and by utilizing the periodic zero-crossing of the resonant inductor current, zero-voltage switching (ZVS) of the primary-side switching tube and some of the secondary-side switching tubes is achieved, reducing the switching losses of the converter. The subsequent interleaved parallel Buck converter can achieve step-down and reduce the output current ripple function, and power factor correction can also be achieved when the converter operates in the reverse direction.

[0042] Furthermore, by using the switching tube multiplexing technology, the switching tubes of the latter stage of the L-LLC resonant converter and the interleaved parallel Buck converter are multiplexed. At the same time, by using the bridge-less technology, the secondary-side inverter bridge is cancelled and replaced with a bridge-less topology, thus obtaining the single-stage bidirectional DC / AC converter as shown in Figure 1 The specific single-stage bidirectional DC / AC converter is composed of an L-LLC DC / DC front end and an interleaved parallel Buck DC / AC rear end. The inductor L B1 is connected to the midpoint of the bridge arm Q5-Q6 (i.e., the bridge arm formed by the switching tubes Q5 and Q6) to form the first step-down converter, while the inductor L B2 is connected to the midpoint of the bridge arm Q7-Q8 (i.e., the bridge arm formed by the switching tubes Q7 and Q8) to form the second step-down converter. The two step-down converters, together with the switching tubes Q H and the switching tube Q L further form an interleaved totem-pole type DC / AC converter. The active rectification network of the L-LLC converter shares the full bridge formed by the switching tubes Q5-Q8 with the interleaved totem-pole type DC / AC converter. In addition, passive components, including the inductor L r (resonant inductor), the capacitor C r (resonant capacitor), the transformer, and the inductor L m_a (auxiliary inductor), form a resonant circuit, and the full bridge formed by the switching tubes Q5-Q8 is used as an inverter network. Among them, the inductor L m is the equivalent magnetizing inductor.

[0043] The characteristics of the single-stage bidirectional DC / AC converter are as follows: 1) It adopts a single-stage structure, with advantages such as low cost, small volume, and simple control; 2) The bridge-less structure eliminates the power loss caused by the inverter bridge; 3) Interleaved parallel Buck reduces the ripple of the output current; 4) LLC topology enables zero-voltage switching of the primary and secondary switching transistors within a wide range; 5) It can be used in bidirectional converters.

[0044] The key waveforms of the single-stage bidirectional DC / AC converter are as Figure 2 shown. During the positive half power frequency cycle, the switching transistor Q L (pilot switch) is always in the on state, and the duty cycle D determines the conduction time of switching transistors Q5 and Q7. Switching transistors Q5 and Q7 are the main switches of two interleaved buck converters. During the negative half power frequency cycle, the switching transistor Q H (pilot switch) is always in the on state, and the duty cycle D determines the conduction time of switching transistors Q6 and Q8. Switching transistors Q6 and Q8 are the main switches of two interleaved buck converters. The equivalent circuits of the operating modes at different time intervals during the positive half power frequency cycle of the single-stage bidirectional DC / AC converter are respectively as Figures 3 to 8 shown. The equivalent circuits of the operating modes at different time intervals during the negative half power frequency cycle of the single-stage bidirectional DC / AC converter are respectively as Figures 9 to 14 shown. During the positive half power frequency cycle, the equivalent circuit of the operating mode during the positive half switching cycle of the single-stage bidirectional DC / AC converter is as Figures 3 to 5 shown, and the equivalent circuit of the operating mode during the negative half switching cycle is as Figures 6 to 8 shown. During the negative half power frequency cycle, the equivalent circuit of the operating mode during the positive half switching cycle of the single-stage bidirectional DC / AC converter is as Figures 9 to 11 shown, and the equivalent circuit of the operating mode during the negative half switching cycle is as Figures 12 to 14 shown.

[0045] Optionally, as Figure 15 shown, the front-stage L-LLC resonant converter can be replaced with a CLLC resonant converter. The source of switching transistor Q1 is connected to the same-named side of the primary of the transformer through capacitor C r1 and inductor L r1 . The same-named side of the secondary of the transformer is connected to inductor L r1 and capacitor C r1 in sequence and then connected to inductor L B1 . The source of switching transistor Q3 is connected to the opposite-named side of the primary of the transformer, and the opposite-named side of the secondary of the transformer is connected to inductor L B2 .

[0046] Optionally, the rear-stage interleaved parallel Buck converter can be replaced with a Buck converter, and the output is changed to an inverter bridge output. All switching transistors are silicon carbide MOSFETs. Further, as Figure 1 shown, at the output end, an inductor L fand capacitor C f The LC filter circuit composed of it filters out high - order harmonics, making the output waveform more stable.

[0047] 2. PWM / PFM hybrid control strategy: In terms of the control strategy, if traditional PWM control is adopted, since the duty cycle of the subsequent - stage Buck structure is less than 0.5, the Bus capacitor voltage (bus capacitor voltage) must be greater than 2 times the maximum grid voltage, and it is preferably set to be slightly greater than 2 times the maximum grid voltage. Otherwise, the converter efficiency will decrease. However, the grid voltage of an electric - vehicle battery is often not a fixed value. The relationship curves of several different battery voltages with the discharge time are as Figure 16 shown. Different colors represent different battery models. The solid line is for a brand - new battery, and the dashed line is for the old battery after 300 cycles of the corresponding new battery. It can be seen that as the battery discharges continuously, its grid voltage will decrease. At the same time, with the use of the battery, battery aging will make the battery voltage drop more significantly. Considering factors such as deep discharge of electric vehicles, ambient temperature, and battery aging, the discharge voltage of an electric - vehicle battery often has a large variation range. When the battery voltage decreases with discharge, the traditional PWM control does not change the switching frequency, and the resonant - cavity gain remains unchanged. When the input voltage drops to a certain value, the traditional PWM control will not be able to work properly. If a relatively high Bus capacitor voltage is forcibly set for the working range of the converter, it will cause the efficiency of the converter during normal operation to decrease significantly. Therefore, traditional PWM control is not suitable for the electric - vehicle V2G system.

[0048] Therefore, the present invention proposes a PWM / PFM hybrid control strategy that can adapt to a wide - range voltage input. By adjusting the switching frequency to adjust the resonant - cavity gain of the DC / DC converter, the Bus capacitor voltage of the single - stage bidirectional DC / AC converter can be stabilized at a certain value under the condition of input - voltage variation; since the Bus capacitor voltage is fixed and the change of the switching frequency does not affect the subsequent - stage DC / AC control, stable AC output is achieved by controlling the duty cycle.

[0049] The gain expression of the L - LLC resonant converter is: ; In the formula, l is the inductance ratio, Q is the quality factor of the circuit, f n is the normalized frequency, and their specific definitions are as shown in the following formulas: ; ; ; ; In the formula, R ac is the AC equivalent load, f sis the switching frequency of the L-LLC converter, f r is the first resonance frequency.

[0050] The gain curve of the L-LLC resonant converter is as Figure 17 shown, where M is the gain of the resonant cavity, that is, the ratio of the grid voltage of the resonant cavity to the input voltage of the resonant cavity. Specifically in the present invention, it refers to the ratio of the bus voltage v bus to the grid voltage of the transformer, and the resonant converter operates in the over-resonance mode. As Figure 17 can be seen, within the operating range of the L-LLC resonant converter, the gain of the L-LLC resonant converter decreases monotonically with the increase of the switching frequency. When the input voltage of the L-LLC resonant converter drops, the gain can be increased by reducing the switching frequency to maintain the stability of the bus voltage v bus .

[0051] The PWM / PFM hybrid control strategy is as Figure 18 shown, including inner-loop control and outer-loop control. The outer-loop control adjusts the output current by adopting PWM control, and the inner-loop adopts PFM control to adjust the bus voltage v bus . As can be seen from the foregoing, the gain of the L-LLC converter can be controlled by adjusting the switching frequency f s , and then the regulation of the grid voltage can be realized. The inner-loop control first samples the bus voltage v bus and digitizes it through the ADC. By feeding back the error signal between the digital quantity of the bus voltage v bus and the reference voltage V ref_bus to the controller D-PI bus , the bus voltage control signal is obtained to determine the frequencies of the digital sawtooth carriers v saw1_fs and v saw2_fs , and the bus voltage is stabilized by changing the frequencies. In the outer-loop, the sampled output current i o is digitized through the ADC. By transmitting the error signal between the amplified K-fold digital quantity of the output current and the power-frequency sinusoidal reference voltage VAC to the controller D-PI io , the output current control signal i con is obtained. By comparing it with the digital sawtooth carriers v saw1_fs and v saw2_fs , the duty cycle of the control signal is obtained, that is, the PWM control is realized, and the AC output is adjusted through the duty cycle. Among them, the controller D-PI bus refers to a PI (Proportional Integral) controller for the digital quantity of the bus voltage v bus , and the controller D-PI io refers to a PI controller for the output current i oDigital PI (Proportional Integral) controller.

[0052] The specific process is as follows: Inner loop control: (1) The bus capacitor C is voltage sampled through the ADC (Analog to Digital Converter) module bus to obtain the bus voltage v bus and passed to the D-PI bus controller.

[0053] (2) The structure of the D-PI bus is as shown in Figure 19 Using the set bus reference voltage V ref_bus (designed value is 380V) minus the bus voltage Vbus to obtain the error reference signal, and after performing P (proportional control link) and I (integral control link) on this error signal respectively and adding them together to obtain the initial voltage control signal v fs* , after performing a limiting operation on the initial voltage control signal, the final voltage control signal V fs is obtained. The limiting operation ensures that the frequency of the converter always remains within a reasonable working range, that is, when v fs* is greater than the voltage control signal v fsmax corresponding to the maximum operating frequency, limit v fs to v fsmax ; when v fs* is less than the voltage control signal v fsmin corresponding to the minimum operating frequency (resonant frequency point), limit v fs to v fsmin . The structure of the D-PI iod is the same as that of the D-PI bu controller.

[0054] (3) Carrier generation: The PFM module generates a triangular carrier with a corresponding frequency according to the input voltage control signal v fs . The PFM module is as shown in Figure 20 , mainly composed of a comparator, a monostable flip-flop, a bidirectional switch, a current source, and a capacitor. The voltage control signal v fs is compared with the capacitor voltage through the comparator and then input to the monostable flip-flop; under the action of the trigger signal, the monostable flip-flop generates a pulse signal with a fixed width to control the short-term conduction of the bidirectional switch, and the capacitor discharges through the loop formed by the bidirectional switch and then recharges, and the capacitor voltage increases linearly from 0. Through this process, v fs can control the single charging time of the capacitor, thereby generating a triangular carrier v saw2_fs, a triangular carrier wave v can be obtained by shifting the phase by 180° (180°shifting). saw1_fs .

[0055] Outer loop control: (1) The output current is sampled by the ADC module to obtain the output current i. o After being amplified by K times, it is transmitted to the D-PI io controller; the grid voltage is sampled by the ADC module to obtain the grid voltage VAC and transmitted to the D-PI io controller.

[0056] (2) The structure of the D-PIio controller is also as Figure 19 shown. Using the grid voltage V AC minus the output current K*io amplified by K times to obtain an error reference signal. After performing P (proportional control link) and I (integral control link) on this error signal respectively and then adding them together to obtain the initial current control signal i con* , after performing a limiting operation on the initial current control signal, the final current control signal i con is obtained. The limiting operation ensures that the duty cycle of the converter is within a reasonable range and prevents the converter from short-circuiting due to the simultaneous conduction of the upper and lower switching tubes of the same bridge arm. When i con* is greater than the current control signal i conmax corresponding to the maximum duty cycle, limit i con to i conmax ; when i con* is less than the current control signal i conmin corresponding to the minimum duty cycle, limit i con to i conmin .

[0057] (3) The current control signal i con and the triangular carrier waves v saw1_fs , v saw2_fs obtained from the inner loop control are compared by a comparator to obtain the control signals V gs1* and V gs2* corresponding to the duty cycle. The duty cycle control signal and the sampled grid voltage pass through a logic operation module to obtain the control signals v gs1 , control signal v gs2 , control signal v gs3 and control signal v gs4 corresponding to the switching tubes Q1, Q2, Q3, and Q4 to control (Controlling) the corresponding switching tubes. The structure of the logic control module is as Figure 21 shown, and its main function is to determine the main switching tubes of the primary full-bridge according to the positive and negative of the grid voltage.

[0058] The control signal corresponding to the switching transistor is obtained through the logic control module, amplified, and then used to control the corresponding switching transistor. When the grid voltage is greater than 0 and in the positive half power frequency cycle, switching transistors Q1 and Q3 serve as the main switching transistors. At this time, the control signal v gs1 is equal to the control signal v gs2* , the control signal v gs3 is equal to the control signal v gs1* , and the control signal v gs2 is complementary to the control signal v gs1 , the control signal v gs4 is complementary to the control signal v gs3 ; when the grid voltage is less than 0 and in the negative half power frequency cycle, switching transistors Q2 and Q4 serve as the main switching transistors. At this time, the control signal v gs2 is equal to the control signal v gs2* , the control signal vgs4 is equal to the control signal vgs1*, and the control signal v gs1 is complementary to the control signal v gs2 , the control signal v gs3 is complementary to the control signal v gs4 .

[0059] As shown in Table 1, when V AC is a positive voltage: V gsA is equal to V gs* serving as the main switching transistor, V gsB is complementary to V gs* . Specifically, when V gs* is 1, V gsA is also 1, and V gsB is 0; when V gs* is 0, V gsA is 0, and V gsB is 1; When V AC is a negative voltage: V gsB is equal to V gs* serving as the main switching transistor, V gsA is complementary to V gs* . Specifically, when V gs* is 1, V gsB is also 1, and V gsA is 0; when V gs* is 0, V gsB is 0, and V gsA is 1. Among them, 1 and 0 respectively represent high and low level signals.

[0060] Table 1 Truth Table of Logic Control Module .

[0061] Specifically, the logical operations include: Take the grid voltage V AC as the input, and input it into the positive input terminal of the fifth comparator and the negative input terminal of the sixth comparator respectively. The inputs of the negative input terminal of the fifth comparator and the positive input terminal of the sixth comparator are both 0; the fifth comparator generates a control signal v gs5* , and the sixth comparator generates a control signal v gs6* ; Perform a NAND operation on the control signal v gs5* and the control signal vgs* to generate a control signal v1; perform a NAND operation on the control signal v gs5* and the inverted signal of the control signal vgs* to generate a control signal v2; Perform a NAND operation on the control signal v gs6* and the control signal vgs* to generate a control signal v3; perform a NAND operation on the control signal v gs6* and the inverted signal of the control signal vgs* to generate a control signal v4; Perform a NAND operation on the control signal v1 and the control signal v4 to generate a control signal v gsA ; perform a NAND operation on the control signal v2 and the control signal v3 to generate a control signal v gsB ; Among them, if the control signal vgs* is the control signal v gs1* , then the control signal v gsA is the control signal v gs3 , and the control signal v gsB is the control signal v gs4 ; if the control signal vgs* is the control signal v gs2* , then the control signal vgsA is the control signal v gs1 , and the control signal v gsB is the control signal v gs2 .

[0062] (4)Performing an AND operation on the control signal v gs1 and the control signal v gs4 can obtain the control signals v gs5 and v gs8 of the switching transistors Q5 and Q8. Similarly, performing an AND operation on the control signal v gs2 and the control signal v gs3 can obtain the control signals vgs6 and V gs7 of the switching transistors Q6 and Q7.

[0063] (5)Comparing the sampled grid voltage V AC with 0 through a comparator can obtain the control signals v L and v H of the switching transistors Q gsL and vgsH , when the grid voltage V AC is greater than 0, Q L turns on. When the grid voltage V AC is less than 0, the switching transistor Q H conducts.

[0064] It should be noted that the above control signals v gs1 to control signal v gs8 , control signal v gsH and control signal v gsL are all input to the corresponding switching transistors through the drive circuit.

[0065] 3. Time-domain modeling of the single-stage bidirectional DC / AC converter: The traditional First Harmonic Approximation has a high analysis accuracy when the switching frequency is close to the series resonance frequency. However, when the switching frequency is far from the series resonance frequency, the proportion of odd harmonics increases significantly, resulting in an increase in analysis error. In addition, the instantaneous grid voltage and current of the single-stage bidirectional DC-AC converter change constantly, which is a typical time-varying system. The traditional modeling theory of resonant converters is based on steady-state analysis and cannot be applied to this time-varying system. The time-domain analysis method is more suitable for the analysis of time-varying systems and has higher accuracy compared with the traditional first-harmonic analysis method.

[0066] Due to the effect of PWM control, the input voltage V AB of the resonant cavity of the full-bridge LLC resonant converter is no longer a square wave under PFM control, and the traditional analysis method is no longer applicable. In this paper, a time-domain analysis method is adopted, that is, a time-domain model is established by numerical calculation methods. The three basic equivalent circuits of the converter under PWM control are as Figure 22 shown, namely the P mode, P w mode, and Z mode. Among them, i s is the output current of the resonant cavity, and v Cr is the voltage across the resonant capacitor Cr. The three-mode equivalent circuits only focus on the resonant cavity and equivalently analyze the input part and output part of the resonant cavity in different situations.

[0067] Taking the P mode as an example, with the capacitor voltage as the unknown, according to the relationship between the voltage and current at both ends of the capacitor and inductor and Kirchhoff's voltage law, a second-order differential equation can be obtained and the general solution can be found: ; ; In the formula, u Cr (t) is the voltage of the resonant capacitor, which is the function to be solved in the second-order differential equation. w r is the angular frequency (w r = 2πfr ), V in is the input voltage of the resonant cavity, V o is the grid voltage of the resonant cavity. A and B are two undetermined coefficients of the general solution. After substituting the resonant inductor current and resonant capacitor voltage at the initial moment as known conditions below, A and B can be obtained. A = U Cr0 -V in +V o , B = Z r IL r0 . Among them, U Cr0 is the resonant inductor voltage at time 0, Z r is the characteristic impedance, I Lr0 is the resonant inductor current at time 0.

[0068] Substitute the resonant inductor current and resonant capacitor voltage at the initial moment as known conditions into the general solution. The resonant inductor current and resonant capacitor voltage in this mode can be expressed by the initial values of the inductor current and capacitor voltage. The solution processes for the expressions of the resonant inductor current and resonant capacitor voltage in other modes are the same: ; ; In the formula, U Cr0 is the resonant inductor voltage at time 0, Z r is the characteristic impedance, I Lr0 is the resonant inductor current at time 0, i Lr (t) is the resonant inductor current at time t.

[0069] The working modes of the single-stage bidirectional DC / AC converter under normal conditions are P mode, Pw mode, and Z mode. Under normal conditions, ZVS of the primary side switch tube can be achieved, and ZVS of switch tubes Q6 and Q8 can be achieved in the positive half power frequency period, and ZVS of switch tubes Q5 and Q7 can be achieved in the negative half power frequency period. The key waveforms in this mode are as Figure 23 shown.

[0070] The time period from t0 to t1 is P mode. At time t0, switch tubes Q1, Q4, Q5, and Q8 are turned on. At this time, V Lm1 is clamped by the input voltage V IN / N of the resonant cavity. At the same time, due to the bus capacitor C bus being large enough, the bus voltage v bus can be regarded as a constant value within half a resonant period. Therefore, the grid voltage V Lm of the resonant cavity is clamped by the bus capacitor voltage, and the auxiliary excitation inductor current i Lm rises linearly. Therefore, the resonant capacitor voltage, resonant inductor current, and auxiliary excitation inductor current in P mode can be expressed as: ; In the formula, u Cr0 (t) is the resonant capacitor voltage at time t during the period from t0 to t1, U Cr0 is the resonant capacitor voltage at time t0, i Lr0 (t) is the resonant capacitor current at time t during the period from t0 to t1, I Lr0 is the resonant capacitor current at time t0, i m0 (t) is the auxiliary excitation inductor current during the period from t0 to t1, I m is the opposite of the auxiliary excitation inductor current at time t0. i Lr Reaches the maximum value I n at time t1, and the duration of the P mode corresponds to the duty cycle δ of the control signal.

[0071] The period from t1 to t2 is the P w mode. At time t1, the control signals V gs1 , control signal V gs5 and control signal V gs8 drop to 0, the switch Q1 turns off, and the switches Q2 and Q4 turn on. At this time, the input voltage is clamped by 0; since the currents of the switches Q5 and Q8 are less than 0 at time t1, the switches Q5 and Q8 continue to conduct through the body diodes for freewheeling. Therefore, the grid voltage V Lm of the resonant cavity is still clamped by the bus capacitor voltage, and the auxiliary excitation inductor current i Lm rises linearly. Therefore, the resonant capacitor voltage, resonant inductor current, and auxiliary excitation inductor current in the P w mode can be expressed as follows: ; In the formula, u Cr1 (t) is the resonant capacitor voltage at time t during the period from t1 to t2, U Cr1 is the resonant capacitor voltage at time t1, i Lr1 (t) is the resonant capacitor current at time t during the period from t1 to t2, I Lr1 is the resonant capacitor current at time t1, i m1 (t) is the auxiliary excitation inductor current during the period from t1 to t2.

[0072] The period from t2 to t3 is the Z mode. At this time, the primary side switches Q1 and Q3 are still off, and the switches Q2 and Q4 are on. The input voltage V in is clamped by 0; for the secondary side, the reverse current of the switch Q5 just drops to 0 at time t2, the switch Q5 turns off, and the switches Q6 and Q8 conduct through the body diodes for freewheeling. Therefore, the grid voltage V Lm of the resonant cavity is clamped by 0, and the auxiliary excitation inductor current i LmRemain unchanged. Therefore, the resonant capacitor voltage, resonant inductor current, and auxiliary excitation inductor current in the Z mode can be expressed as follows: ; where u Cr2 (t) is the resonant capacitor voltage at time t during the period from t2 to t3, U Cr2 is the resonant capacitor voltage at t2, i Lr2 (t) is the resonant capacitor current at time t during the period from t2 to t3, I Lr2 is the resonant capacitor current at t2, i m2 (t) is the auxiliary excitation inductor current during the period from t2 to t3.

[0073] The sum of the times of the above three modes is T s / 2, and there is a mode time constraint: ; where Δt1 is the duration of the P mode, Δt2 is the duration of the P w mode, Δt3 is the duration of the Z mode; T s is the switching period.

[0074] When the L-LLC converter switches between the above modes, since the inductor current and capacitor voltage do not change abruptly, the continuity constraints of the resonant inductor current and resonant capacitor voltage can be expressed as: ; ; where i Lr0 (t1) is the resonant capacitor current at t1 during the period from t0 to t1, i Lr1 (t1) is the resonant capacitor current at t1 during the period from t1 to t2, i Lr1 (t2) is the resonant capacitor current at t2 during the period from t1 to t1, i Lr2 (t2) is the resonant capacitor current at t2 during the period from t2 to t3; u Cr0 (t1) is the resonant capacitor voltage at t1 during the period from t0 to t1, u Cr1 (t1) is the resonant capacitor voltage at t1 during the period from t1 to t2, u Cr1 (t2) is the resonant capacitor voltage at t2 during the period from t1 to t2, u Cr2 (t2) is the resonant capacitor voltage at t2 during the period from t2 to t3.

[0075] According to the symmetry of the L-LLC converter, in a half switching period, the initial value and the ending value of the auxiliary excitation inductor current, resonant inductor current, and resonant capacitor voltage are opposite in a half switching period. Therefore, its symmetry constraint can be expressed as: ; Wherein, i Lm0 (t1) is the auxiliary excitation inductor current at time t0 during the period t0 - t1, i Lm2 (t3) is the auxiliary excitation inductor current at time t3 during the period t2 - t3, i Lr0 (t0) is the resonant capacitor current at time t0 during the period t0 - t1, i Lr2 (t3) is the resonant capacitor current at time t3 during the period t2 - t3, u Lr0 (t0) is the resonant capacitor voltage at time t0 during the period t0 - t1, u Lr2 (t3) is the resonant capacitor voltage at time t3 during the period t2 - t3.

[0076] Since the reason for the conversion from the P w mode to the Z mode is that the reverse freewheeling current of the switching transistor Q5 just drops to 0, there is a mode conversion condition constraint, and the formula is as follows: ; Wherein, i Q5 (t2) is the reverse freewheeling current of the switching transistor Q5 at time t2, i m2 (t2) is the auxiliary excitation inductor current at time t2 during the period t2 - t3, i LB2 (t2) is the Buck inductor current at time t2 during the period t2 - t3.

[0077] Only the P mode of the single-stage bidirectional DC / AC converter has energy input, so the average input power P in can be expressed as: ; Assuming that the efficiency of the converter is 100%, considering that the bus capacitor C bus has energy input and output within one power frequency period, there is a power balance constraint: ; Wherein, P bus is the power consumed by the bus, P out is the power output by the bus, v bus is the bus voltage, Δv bus is the voltage change of the bus capacitor in half a switching period, v out is the grid voltage.

[0078] The calculation formula for the voltage change of the bus capacitor in half a switching period is as follows: .

[0079] The secondary side switching tube is reused in two levels. Considering the duty cycle constraint of the interleaved parallel Buck structure, the formula is as follows: ; In the formula, Δt1 is the duration of the P mode, and Δt2 is the duration of the P w mode.

[0080] Taking Δt1, Δt2 and f s and the initial values of the resonant inductor current and resonant capacitor voltage of each mode as unknowns, V o , v out as known quantities, substituting the expressions of the resonant inductor current, resonant capacitor voltage and auxiliary excitation inductor current of the above three modes into each constraint condition, 9 transcendental equations can be obtained. Solving them can obtain the above unknowns and the values of the resonant inductor current, resonant capacitor voltage and auxiliary excitation inductor current at the initial moment of each mode. Finally, using the time-stepping iteration method, starting from an accurate initial value to solve the model at this moment, and then using the solution as the initial value of the next moment to solve the model of the next moment, and continuously iterating, the visualization of the resonant inductor current and resonant capacitor voltage within the power frequency period can be realized. The solution process is as Figure 24 shown.

[0081] 4. Simulation verification: Considering the deep discharge, battery aging and temperature factors of electric vehicles, the converter of the present invention has a DC input of 200 - 400V and an output of 110 effective value of power frequency sinusoidal alternating current. Since the duty cycle of the subsequent interleaved parallel Buck converter cannot be greater than 0.5, the designed Bus capacitor voltage must be greater than twice the maximum value of the grid voltage. The formula is as follows: ; In the formula, v out_max is the maximum value of the grid voltage.

[0082] Taking the bus voltage as 380V, at the same time, the single-stage bidirectional DC / AC converter operates in the over-resonant state, that is, the switching frequency is greater than the resonant frequency, and there is a maximum gain of 1 when the switching frequency is close to the resonant frequency. To ensure that the single-stage bidirectional DC / AC converter can work normally at the lowest input voltage, the turns ratio N needs to satisfy the following relational expression: ; In the formula, V IN is the DC input voltage, and M max is the maximum gain. Among them, N = 0.5 is taken.

[0083] To verify the feasibility of the proposed time-domain modeling analysis theory and PWM / PFM hybrid control strategy, the present invention builds a simulation model based on the PWM / PFM hybrid control strategy with the help of the PSIM simulation software. The comparison diagram of the simulation results and the time-domain modeling is as shown in Figure 25 , Figure 26 . In the figure, the blue curve is the resonant inductor current within half a switching period, and the red curve is the auxiliary excitation inductor current within half a switching period. The solid line and the dashed line respectively represent the simulation results and the time-domain analysis results. It can be seen from the comparison results that the time-domain analysis model established by the present invention has high accuracy. The relationship between the operating frequency of the single-stage bidirectional DC / AC converter and the input voltage is as shown in Figure 27 . It can be seen that as the input voltage decreases, the operating frequency of the converter also continuously decreases. By decreasing the frequency to increase the resonant cavity gain so as to maintain the intermediate bus capacitor voltage, the correctness of the control strategy proposed by the present invention is further verified.

[0084] It can be seen that in the technical solution provided by the present invention, on the one hand, the single-stage bidirectional DC / AC converter adopts a single-stage structure, and reduces the number of switching tubes through multiplexing technology, which can simplify the topology structure, improve the stability by reducing the damage probability, and reduce the cost; further, the resonant converter can achieve safety isolation through the transformer. By using the soft-switching technology of the L-LLC resonant converter or the CLLC resonant converter, it is possible to achieve ZVS for 4 switching tubes on the primary side and ZVS for half of the switching tubes on the secondary side within the full operating range, greatly reducing the switching loss and further improving the operating efficiency; by replacing the output inverter bridge with the bridge-less technology, the number of switching tubes can be further reduced, and the power density and operating efficiency can be improved.

[0085] On the other hand, the PWM / PFM hybrid control method is suitable for wide-range voltage input, can support deep discharge of electric vehicle batteries, and enables the converter to have high stability.

[0086] Further, the LC filter circuit can filter out high-order harmonics, making the output waveform more stable.

Claims

1. A single-stage bidirectional DC / AC converter, characterized in that, Comprising: The switching transistors Q1, Q2, Q3, and Q4 form a full bridge; the drains of the switching transistors Q1 and Q3 are both connected to the positive pole of the DC power supply, and the sources of the switching transistors Q2 and Q4 are both connected to the negative pole of the DC power supply; The two output terminals of the full bridge are respectively connected to both ends of the primary side of the transformer; The corresponding ends of the secondary side of the transformer are successively connected to the opposite ends of the secondary side of the transformer through inductors L r , inductor L rm_a and capacitor C r ; Also comprising: Inductor L B1 Through inductor L r Connected to the same-named terminal of the secondary side of the transformer. Inductor L B2 Through capacitor C r Connected to the opposite-named terminal of the secondary side of the transformer; Inductor L B2 And inductor L B1 The output terminals are connected to each other; The source of switch Q5 and the drain of switch Q6 are both connected to the input of inductor L B1 ; the source of switch Q7 and the drain of switch Q8 are both connected to the input of inductor L B2 ; the drains of switches Q H , Q5 and Q7 are all connected to the positive electrode of bus capacitor C bus ; the sources of switches Q L , Q6 and Q8 are all connected to the negative electrode of bus capacitor C bus . In the single-stage bidirectional DC / AC converter, the output terminal of the inductor L B1 is the first output terminal, and the source electrode of the switching transistor Q H is the second output terminal.

2. A single-stage bidirectional DC / AC converter, characterized in that, Comprising a CLLC resonant converter; In the CLLC resonant converter, the switching transistors Q1, Q2, Q3, and Q4 form a full bridge; the drains of the switching transistors Q1 and Q3 are both connected to the positive pole of the DC power supply, and the sources of the switching transistors Q2 and Q4 are both connected to the negative pole of the DC power supply; the output terminal A of the full bridge is sequentially connected to the same-name terminal of the primary side of the transformer through the capacitor C r1 and the inductor L r1 and is connected to the same-name terminal of the primary side of the transformer, and the output terminal B is connected to the opposite-name terminal of the primary side of the transformer; the output terminal A is the midpoint between the switching transistors Q1 and Q2, and the output terminal B is the midpoint between the switching transistors Q3 and Q4; the same-name terminal of the secondary side of the transformer is sequentially connected to the inductor L r2 and the capacitor C r2 and is connected to the inductor L B1 , and the opposite-name terminal is connected to the inductor L B2 ; Also comprising: Inductor L B2 and the inductor L B1 have their output terminals connected to each other; The source of switch Q5 and the drain of switch Q6 are both connected to the input terminal of inductor L B1 ; the source of switch Q7 and the drain of switch Q8 are both connected to the input terminal of inductor L B2 ; the drains of switch Q H , switch Q5 and switch Q7 are all connected to the positive pole of bus capacitor C bus ; the sources of switch Q L , switch Q6 and switch Q8 are all connected to the negative pole of bus capacitor C bus . In the single-stage bidirectional DC / AC converter, the output terminal of the inductor L B1 is the first output terminal, and the source electrode of the switching transistor Q H is the second output terminal.

3. A single-stage bidirectional DC / AC converter according to claim 1 or 2, characterized in that, Also comprising an LC filter circuit disposed between the first output terminal and the second output terminal.

4. A single-stage bidirectional DC / AC converter according to claim 1 or 2, characterized in that All the switching transistors are silicon carbide MOSFETs.

5. A control method for a single-stage bidirectional DC / AC converter, which is used for the single-stage bidirectional DC / AC converter according to claim 1 or 2, characterized in that, Comprising the following steps: Step S1: Sample the bus voltage v bus , and calculate the difference between the bus voltage v bus and the bus reference voltage V ref_bus . Perform proportional-integral control on the difference value to obtain an initial voltage control signal V fs* , and obtain a voltage control signal V after amplitude limiting fs , and input it into the PFM module to generate a triangular carrier wave v saw2_fs ; Shift the triangular carrier wave v saw2_fs by 180° to obtain the triangular carrier wave v saw1_fs ; Sample the output current i of the single-stage bidirectional DC / AC converter o and the grid voltage V AC , calculate the difference between K times the output current i o and the grid voltage V AC as the error reference signal; Perform proportional-integral control on the error reference signal to obtain an initial current control signal i con* , and obtain the current control signal i after amplitude limiting con ; Step S2: Input the triangular carrier wave v saw2_fs and the current control signal i con to the negative input terminal and the positive input terminal of the first comparator respectively to obtain the control signal v gs2* ; Perform a logical operation on the control signal v gs2* and the grid voltage V AC to generate the control signal v gs1 of the switching transistor Q1 and the control signal v gs2 of the switching transistor Q2; Apply the triangular carrier wave v saw1_fs and the current control signal i con to the negative input terminal and the positive input terminal of the second comparator respectively, to obtain the control signal v gs1* ; Perform a logical operation on the control signal v gs1* and the grid voltage V AC to generate the control signal v gs3 for the switching transistor Q3 and the control signal v gs4 ; Input the grid voltage V AC into the positive input terminal of the third comparator and the negative input terminal of the fourth comparator respectively. The inputs of the negative input terminal of the third comparator and the positive input terminal of the fourth comparator are both 0. The third comparator generates the control signal v L of the switching transistor Q gsL , and the fourth comparator generates the control signal v H of the switching transistor Q gsH ; Perform an AND operation on control signal v gs2 and control signal v gs3 to generate the control signal v gs6 for switch Q6 and the control signal v gs7 for switch Q7; perform an AND operation on control signal v gs1 and control signal v gs4 to generate the control signal v gs5 for switch Q5 and the control signal v gs8 for switch Q8; The logical operation satisfies: If the grid voltage V AC is positive and the control signal v gs* is 1, or if the grid voltage V AC is negative and the control signal v gs* is 0, then the control signal v gsA is 1 and the control signal v gsB is 0; otherwise, the control signal v gA is 0 and the control signal v gsB is 1; Wherein, If the control signal v gs* is the control signal v gs1* , then the control signal v gsA is the control signal v gs1 , the control signal v gsB is the control signal v gs2 ; If the control signal v gs* is the control signal v gs2* , then the control signal v gsA is the control signal v gs3 , the control signal v gsB is the control signal v gs4 .

6. The control method of a single-stage bidirectional DC / AC converter according to claim 5, characterized in that The logical operation comprises: Taking the grid voltage V AC as the input, it is respectively input into the positive input terminal of the fifth comparator and the negative input terminal of the sixth comparator. The inputs of the negative input terminal of the fifth comparator and the positive input terminal of the sixth comparator are both 0. The fifth comparator generates a control signal v gs5* , and the sixth comparator generates a control signal v gs6* ; Perform a NAND operation on the control signal v gs5* and the control signal vgs* to generate the control signal v1; perform a NAND operation on the control signal v gs5* and the inverted signal of the control signal vgs* to generate the control signal v2; Perform a NAND operation on the control signal v gs6* and the control signal vgs* to generate the control signal v3; perform a NAND operation on the control signal v gs6* and the inverted signal of the control signal vgs* to generate the control signal v4; Perform a NAND operation on control signal v1 and control signal v4 to generate control signal v gsA ; perform a NAND operation on control signal v2 and control signal v3 to generate control signal v gsB .

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