Induction motor low-speed section stator flux linkage observation improvement method based on pure voltage model

Abstract

The invention discloses an induction motor low-speed section stator flux linkage observation improvement method based on a pure voltage model. The method comprises the following steps: firstly, by analyzing a direct current offset saturation problem caused by an integration link in a pure voltage model, proposing to adopt a low-pass filter with dynamic cut-off frequency to replace a traditional integrator; aiming at the problems of flux linkage amplitude attenuation and phase lag in a low-speed section, a self-adaptive cut-off frequency regulation mechanism based on rotating speed feedback is constructed, the expression of the self-adaptive cut-off frequency regulation mechanism is omegac = k omegar + omegamin, and decoupling optimization of amplitude error compensation and noise suppression is realized; in combination with a double-integrator scheme of coordinate transformation, integral saturation is suppressed by setting a threshold value, and a closed-loop phase compensation strategy is designed based on slip ratio characteristics. The flux linkage amplitude attenuation of the low-speed section is eliminated by dynamically adjusting the parameters of the filter, the observation precision and the anti-interference capability of the high-speed section and the low-speed section are considered by adopting a variable cut-off frequency strategy, the direct current offset accumulation is inhibited by utilizing the threshold integrator, the complexity of the control system is effectively reduced, and the dynamic response speed is improved.

Description

Technical Field

[0001] The present invention relates to the technical field of motor drive control, and in particular to an improved method for observing the stator flux linkage of an induction motor at a low speed section based on a pure voltage model. Background Art

[0002] In induction motor control systems, stator flux observation is the core link for achieving high-precision field-oriented control, and its accuracy directly determines the dynamic response and steady-state performance of the motor. Traditional stator flux observation methods mainly rely on pure voltage models and directly calculate the flux by integrating the stator voltage. However, in the low-speed operating range, this method has significant defects: Integration drift problem: The pure voltage model relies on the open-loop integration operation of the stator voltage, but the proportion of the stator resistance voltage drop increases at low speeds. Small deviations or noise in the voltage sensor will accumulate over the integration time, resulting in serious distortion of the flux amplitude and phase. Parameter sensitivity: The temperature drift characteristics of the stator resistance in the voltage model will significantly affect the accuracy of low-speed flux observation, and in actual engineering, the resistance value is difficult to accurately compensate in real time. Back-EMF attenuation: The back-EMF amplitude is extremely low at low speeds, and the voltage signal-to-noise ratio decreases, further exacerbating the integration error.

[0003] To alleviate these issues, existing technologies typically use a current model or a hybrid model (voltage-current model switching) for flux observation. The current model is based on the rotor flux equation and calculates the flux using the rotor time constant and motor parameters. However, its accuracy is heavily dependent on the accuracy of the rotor resistance and inductance parameters. Parameter changes (such as temperature rise and magnetic saturation) can lead to model mismatch, with significant errors under low-speed and heavy-load conditions. The hybrid model uses a voltage model at high speeds and switches to a current model at low speeds. However, this requires complex switching logic and parameter identification algorithms, increasing system complexity. Furthermore, observation jumps near the switching point can cause control instability.

[0004] Furthermore, traditional voltage models rely on high-precision voltage and current sensors for implementation, leading to the following limitations: Multi-sensor reliance: Traditional solutions require simultaneous acquisition of stator voltage and current signals. Using multiple sensors not only increases hardware costs but also introduces errors due to asynchrony between the multi-channel signals. Poor low-speed current signal quality: Current ripple increases at low speeds. Insufficient current sampling window time or noise interference can lead to current reconstruction errors, further degrading flux observation accuracy.

[0005] To address these issues, an improved flux observation method based on a pure voltage model is urgently needed. This method can autonomously suppress integral drift in the low-speed range while avoiding reliance on complex hybrid models or multi-sensor redundant designs. Such an approach must address the following core challenges: Integrator anti-interference design: How to eliminate integral accumulation errors through adaptive compensation of the voltage signal without the assistance of a current model. Low-speed signal reconstruction optimization: How to use algorithms to enhance the availability of the voltage signal and ensure the continuity of flux observation under low signal-to-noise ratio conditions.

[0006] The improved method proposed in this patent uses an innovative dynamic error compensation mechanism and adaptive filtering technology to significantly improve the flux observation accuracy of the pure voltage model in the low-speed range without increasing the number of sensors or the complexity of the switching model, providing a reliable foundation for high-performance control of induction motors in the full-speed range. Summary of the Invention

[0007] This invention addresses the shortcomings of existing technologies by providing an improved method for observing the stator flux in the low-speed range of an induction motor based on a pure voltage model. This method reduces the structural complexity of the control system, improves the dynamic response speed of the drive system, enhances the waveform quality of the low-speed stator flux observation, and reduces the amplitude and phase errors of the flux.

[0008] To achieve the above object, the present invention adopts the following technical solutions:

[0009] In a first aspect, the present invention proposes an improved method for observing the stator flux linkage of an induction motor at a low speed section based on a pure voltage model, the method comprising the following steps:

[0010] S1: The problems existing in observing the stator flux of an induction motor using a pure voltage model are analyzed.

[0011] S2: The improved scheme of using low-pass filter in observing the stator flux of induction motor is analyzed.

[0012] S3: The stator flux observation scheme of the double integrator with fixed threshold based on coordinate transformation is analyzed.

[0013] S4: Analyzed the improvement scheme for the problem of stator flux observation in low speed section.

[0014] Furthermore, in step S1, the following problems exist when observing the stator flux of the induction motor using a pure voltage model:

[0015] First, the stator voltage equation of the induction motor in the αβ coordinate system is:

[0016]

[0017] where u sα and u sβ is the stator voltage of the motor in the αβ coordinate system, isα and i sβ is the stator current of the motor in the αβ coordinate system, R s is the stator resistance, ψ sα and ψ sβ is the stator flux component in the αβ coordinate system.

[0018] Converting it into a magnetic flux equation can yield the magnetic flux calculation equation:

[0019]

[0020] It can be seen that the stator flux observation based on the pure voltage model has an integral link, which will cause the observation results to be inconsistent with expectations. The problem is analyzed by integrating the signal x(t) with DC quantity. The signal x(t) equation is:

[0021] x(t)=A m sin(ω e t+θ0)+x dc

[0022] Among them A m is the amplitude of the signal, ω e is the motor stator angular frequency, θ0 is the initial phase of the signal, x dc is the DC signal present in the signal.

[0023] By performing Laplace transform on the signal x(t), we can get the equation:

[0024]

[0025] Integrating the above equation, the output signal equation is:

[0026]

[0027] Performing Laplace inverse transform on it can obtain the output signal equation under time domain conditions:

[0028]

[0029] It can be seen from the equation that the output signal consists of three parts, one part is the sinusoidal AC quantity, and the remaining two parts are DC offsets. If there is a DC signal x in the input signal dc , the output signal will be x dc As t increases, the final offset will cause the integrator to saturate as time increases. There is also a DC offset A m cosθ0 / ω e , its value is 0 only when θ0=kπ+π / 2, and it is non-zero in other cases.

[0030] Furthermore, in step S2, an improved solution for observing the stator flux of an induction motor using a low-pass filter is analyzed:

[0031] Based on the problem caused by the pure integrator in step S1, a low-pass filter is used to replace the pure integrator. After passing through the first-order low-pass filter, the output signal equation is:

[0032]

[0033] Among them, ω c is the cutoff frequency of the low-pass filter.

[0034] Performing Laplace inverse transform on the above output signal equation yields the output equation in the time domain:

[0035]

[0036] Where θ = arctanω e / ω c .

[0037] It can be seen from the formula that As time t increases, the term tends to And the cutoff frequency ω c The larger the value, the smaller the final value will be. The term will eventually approach 0 as time t increases, and the cutoff frequency ω c The larger it is, the faster it decays to 0.

[0038] Therefore, for the selection of the low-pass filter cutoff frequency, when ω c When ω is larger, DC offset and low-frequency signals can be suppressed well, but larger amplitude and phase errors will be introduced. On the contrary, when ω is c The smaller the value, the smaller the amplitude and phase errors will be, but the ability to suppress interference signals will be worse.

[0039] Furthermore, in step S3, the stator flux observation scheme of the double integrator with fixed threshold based on coordinate transformation is analyzed:

[0040] Based on the deficiencies in step S2, the amplitude and phase errors introduced by the low-pass filter are compensated, and the overall system equation is:

[0041]

[0042] Where y1=x / (s+ω c ) is the forward channel signal, y2=ω c z=(s+ω c ) is the feedback channel signal, and z is the compensation signal.

[0043] The feedback component mainly suppresses DC drift and saturation at low frequencies. During operation, if the output signal y is not saturated, then z = y. After calculation and integration, the output equation can be obtained as follows:

[0044]

[0045] At this time, the integrator is equivalent to a pure integrator, suppressing the amplitude and phase errors caused by the low-pass filter at low frequencies. If the output signal y reaches the saturation value, the output signal equation is:

[0046]

[0047] At this time, if the signal x contains DC, the output equation can be obtained by performing Laplace transform on the DC in the signal and the threshold L:

[0048]

[0049] Using the final value theorem, we can get the first and second equations in the above formula as:

[0050]

[0051] Therefore, it can be finally obtained that the DC bias output is independent of time and will not saturate. The expression is:

[0052]

[0053] Therefore, when |y(t)|<L, the output equation is:

[0054]

[0055] Entering the saturation region, that is, |y(t)|≥L, the output equation is:

[0056]

[0057] Among them, θ L is the initial phase of the input signal when the signal enters the saturation region.

[0058] Furthermore, in step S4, an improvement plan for the problem of observing the stator flux in the low-speed section is analyzed:

[0059] Based on step S3, it can be seen that in the high-speed stage, due to ω e >>ω c , the amplitude hardly changes, but at low speed, there will be a large error in the amplitude, so we need to compensate for it.

[0060] Compared with the traditional fixed cutoff frequency design, it is changed to be based on the rotor speed ω rDynamically adjust the cutoff frequency. The specific expression is:

[0061] ω c =kω r +ω min

[0062] Among them, k is the adjustment coefficient, ω r is the rotor speed, ω min For the minimum limit.

[0063] Regarding the assignment decay problem: The ideal magnetic flux amplitude is: A m / ω e .

[0064] Before improvement: at low speed (ω e <<ω c ) The approximate expression for amplitude comparison is:

[0065]

[0066] It can be seen that the amplitude is lower than the ideal flux amplitude.

[0067] At high speed (ω e >>ω c ), the flux amplitude is approximately A m / ω e .

[0068] After improvement: At low speed (ω r →0), at this time ω c ≈ω min , the magnetic flux amplitude expression can be expressed as: At this time, if you set ω min <<ω e , then the flux amplitude can be approximated to the ideal flux amplitude: A m / ω e , which solves the problem of magnetic flux amplitude attenuation in the low-speed section.

[0069] At high speed (ω r >>0), at this time ω c ≈kω r , which can effectively suppress noise and maintain amplitude accuracy.

[0070] The slip expression of the induction motor is: s=(ω syn -ω r ) / ω syn , where ω syn is the synchronous mechanical angular velocity, and its relationship with the synchronous electrical angular velocity is: ω e =pω syn , where p is the number of pole pairs of the motor.

[0071] For phase lag: θ = arctanω e / ω c .

[0072] Before improvement: at low speed (ω e <<ω c ): θ≈0°, the phase lag is small, but the flux amplitude is severely attenuated. e >>ω c ): θ≈90°, the phase lag is large, but the amplitude attenuation is small and the noise suppression is strong.

[0073] After improvement: Due to ω e =pω r / (1-s), the hysteresis angle expression can be obtained as:

[0074]

[0075] At low speed (ω r →0) hysteresis angle expression is:

[0076]

[0077] It can be seen that although the angle has increased, it can be compensated by closing the loop. If ω r <<ω min , θ→0. At medium and high speeds (ω r >>0), at this time ω c ≈kω r , the lag angle expression is:

[0078]

[0079] It can be seen that if the slip rate s is small, then θ≈arctan(p / k), which is independent of the speed and its value is determined by k and the pole pair number p.

[0080] Through improvements, it can be found that the improved strategy focuses on amplitude compensation at low speed and noise suppression at high speed, achieving decoupling optimization and significantly improving the overall performance of the system.

[0081] The beneficial effects of the present invention are:

[0082] 1. The dynamic cutoff frequency adaptive adjustment strategy proposed in this invention simplifies the system structure of the traditional hybrid observation model and significantly improves the dynamic response performance of low-speed flux observation.

[0083] 2. The threshold saturation feedback suppression mechanism proposed in the present invention significantly improves the accuracy of the flux amplitude by setting a dynamic saturation threshold to limit the DC offset accumulation of the integrator.

[0084] 3. The amplitude-phase decoupling optimization principle proposed in the present invention focuses on amplitude compensation at low speeds and noise suppression at high speeds, thereby improving the stability and anti-interference ability of flux observation. BRIEF DESCRIPTION OF THE DRAWINGS

[0085] Figure 1 The figure is a block diagram of a speed sensorless vector control system for an induction motor stator field orientation according to an embodiment of the present invention.

[0086] Figure 2 This is a diagram of an improved stator flux observation structure of an induction motor based on a pure voltage model according to an embodiment of the present invention.

[0087] Figure 3 1 is a pure integrator stator flux observation waveform diagram of an embodiment of the present invention.

[0088] Figure 4 This is a stator flux observation waveform diagram of a dual integrator with a fixed threshold based on coordinate transformation according to an embodiment of the present invention.

[0089] Figure 5 This is a stator flux observation waveform diagram of the dynamic cutoff frequency adjustment strategy according to an embodiment of the present invention. Specific implementation plan

[0090] The present invention will be further described in detail below with reference to the accompanying drawings and specific embodiments.

[0091] Example 1

[0092] Figure 1 It is the block diagram of the induction motor stator field oriented speed sensorless vector control system. Figure 2 This is a diagram of an improved stator flux observation structure for an induction motor based on a pure voltage model according to an embodiment of the present invention. This embodiment proposes an improved method for observing the stator flux in the low-speed section of an induction motor based on a pure voltage model, the method comprising the following steps:

[0093] S1: The problems existing in observing the stator flux of an induction motor using a pure voltage model are analyzed.

[0094] S2: The improved scheme of using low-pass filter in observing the stator flux of induction motor is analyzed.

[0095] S3: The stator flux observation scheme of the double integrator with fixed threshold based on coordinate transformation is analyzed.

[0096] S4: Analyzed the improvement scheme for the problem of stator flux observation in low speed section.

[0097] 1. Problems in observing the stator flux of an induction motor using a pure voltage model

[0098] First, the stator voltage equation of the induction motor in the αβ coordinate system is:

[0099]

[0100] where u sα and u sβ is the stator voltage of the motor in the αβ coordinate system, i sα and i sβ is the stator current of the motor in the αβ coordinate system, R s is the stator resistance, ψ sα and ψ sβ is the stator flux component in the αβ coordinate system.

[0101] Converting it into a magnetic flux equation can yield the magnetic flux calculation equation:

[0102]

[0103] It can be seen that the stator flux observation based on the pure voltage model has an integral link, which will cause the observation results to be inconsistent with expectations. The problem is analyzed by integrating the signal x(t) with DC quantity. The signal x(t) equation is:

[0104] x(t)=A m sin(ω e t+θ0)+x dc

[0105] Among them A m is the amplitude of the signal, ω e is the motor stator angular frequency, θ0 is the initial phase of the signal, x dc is the DC signal present in the signal.

[0106] By performing Laplace transform on the signal x(t), we can get the equation:

[0107]

[0108] Integrating the above equation, the output signal equation is:

[0109]

[0110] Performing Laplace inverse transform on it can obtain the output signal equation under time domain conditions:

[0111]

[0112] It can be seen from the equation that the output signal consists of three parts, one part is the sinusoidal AC quantity, and the remaining two parts are DC offsets. If there is a DC signal x in the input signal dc , the output signal will be x dcAs t increases, the final offset will cause the integrator to saturate as time increases. There is also a DC offset A m cosθ0 / ω e , its value is 0 only when θ0=kπ+π / 2, and it is non-zero in other cases.

[0113] In view of the limitations of pure integration, the low-pass filter is used to improve the stator flux observation of the induction motor. The low-pass filter is used to replace the pure integrator. After the first-order low-pass filter, the output signal equation is:

[0114]

[0115] Among them, ω c is the cutoff frequency of the low-pass filter.

[0116] Performing Laplace inverse transform on the above output signal equation yields the output equation in the time domain:

[0117]

[0118] Where θ = arctanω e / ω c .

[0119] It can be seen from the formula that As time t increases, the term tends to And the cutoff frequency ω c The larger the value, the smaller the final value will be. The term will eventually approach 0 as time t increases, and the cutoff frequency ω c The larger it is, the faster it decays to 0.

[0120] Therefore, for the selection of the low-pass filter cutoff frequency, when ω c When ω is larger, DC offset and low-frequency signals can be suppressed well, but larger amplitude and phase errors will be introduced. On the contrary, when ω is c The smaller the value, the smaller the amplitude and phase errors will be, but the ability to suppress interference signals will be worse.

[0121] 2. Stator flux observation scheme based on fixed threshold dual integrator of coordinate transformation

[0122] Compensating for the amplitude and phase errors introduced by the low-pass filter, the overall system equation is:

[0123]

[0124] Where y1=x / (s+ω c ) is the forward channel signal, y2=ω cz / (s+ω c ) is the feedback channel signal, and z is the compensation signal.

[0125] The feedback component mainly suppresses DC drift and saturation at low frequencies. During operation, if the output signal y is not saturated, then z = y. After calculation and integration, the output equation can be obtained as follows:

[0126]

[0127] At this time, the integrator is equivalent to a pure integrator, suppressing the amplitude and phase errors caused by the low-pass filter at low frequencies. If the output signal y reaches the saturation value, the output signal equation is:

[0128]

[0129] At this time, if the signal x contains DC, the output equation can be obtained by performing Laplace transform on the DC in the signal and the threshold L:

[0130]

[0131] Using the final value theorem, we can get the first and second equations in the above formula as:

[0132]

[0133] Therefore, it can be finally obtained that the DC bias output is independent of time and will not saturate. The expression is:

[0134]

[0135] Therefore, when |y(t)|<L, the output equation is:

[0136]

[0137] Entering the saturation region, that is, |y(t)|≥L, the output equation is:

[0138]

[0139] Among them, θ L is the initial phase of the input signal when the signal enters the saturation region.

[0140] 3. Improvement plan for the problem of stator flux observation in low-speed section

[0141] Compared with the traditional fixed cutoff frequency design, it is changed to be based on the rotor speed ω r Dynamically adjust the cutoff frequency. The specific expression is:

[0142] ω c =kω r+ω min

[0143] Among them, k is the adjustment coefficient, ω r is the rotor speed, ω min For the minimum limit.

[0144] Regarding the assignment decay problem: The ideal magnetic flux amplitude is: A m / ω e .

[0145] Before improvement: at low speed (ω e <<ω c ) The approximate expression for amplitude comparison is:

[0146]

[0147] It can be seen that the amplitude is lower than the ideal flux amplitude.

[0148] At high speed (ω e >>ω c ), the flux amplitude is approximately A m / ω e .

[0149] After improvement: At low speed (ω r →0), at this time ω c ≈ω min , the magnetic flux amplitude expression can be expressed as: At this time, if you set ω min << e , then the flux amplitude can be approximated to the ideal flux amplitude: A m / ω e , which solves the problem of magnetic flux amplitude attenuation in the low-speed section.

[0150] At high speed (ω r >>0), at this time ω c ≈kω r , which can effectively suppress noise and maintain amplitude accuracy.

[0151] The slip expression of the induction motor is: s=(ω syn -ω r ) / ω syn , where ω syn is the synchronous mechanical angular velocity, and its relationship with the synchronous electrical angular velocity is: ω e =pω syn , where p is the number of pole pairs of the motor.

[0152] For phase lag: θ = arctanω e / ω c .

[0153] Before improvement: at low speed (ω e <<ω c ): θ≈0°, the phase lag is small, but the flux amplitude is severely attenuated. e >>ω c ): θ≈90°, the phase lag is large, but the amplitude attenuation is small and the noise suppression is strong.

[0154] After improvement: Due to ω e =pω r / (1-s), the hysteresis angle expression can be obtained as:

[0155]

[0156] At low speed (ω r →0) The hysteresis angle expression is:

[0157]

[0158] It can be seen that although the angle has increased, it can be compensated by closing the loop. If ω r <<ω min , θ→0. At medium and high speeds (ω r >>0), at this time ω c ≈kω r , the lag angle expression is:

[0159]

[0160] It can be seen that if the slip rate s is small, then θ≈arctan(p / k), which is independent of the speed and its value is determined by k and the pole pair number p.

[0161] Through simulation, by giving a speed of 500 rpm and running for 3 seconds, the speed is reduced to 50 rpm. The data before and after the improvement are analyzed and plotted. It can be seen that the stator flux observation waveform achieved by pure integration before the improvement is as follows Figure 3 As shown in the figure, it can be seen that the components of the stator flux in the stationary coordinate system are obviously asymmetric, and the fluctuation of the flux amplitude in the steady state is also very large, and the observation effect is very poor; by changing the integration method, that is, the stator flux observation method of the double integrator with a fixed threshold based on coordinate transformation, the cutoff frequency of the low-pass filter is fixed at this time, and the stator flux observation waveform is as follows Figure 4 As shown in the figure, it can be seen that the components of the stator flux in the stationary coordinate system are symmetrical, and the fluctuation of the flux amplitude in the steady state is also very small. Compared with the pure integrator, the observation effect is very good, but when the speed decreases, the stator flux amplitude will become smaller. To solve this problem, by changing the traditional idea of fixed cutoff frequency, it is changed to dynamically adjust the cutoff frequency according to the rotor speed. The stator flux observation waveform is as follows Figure 5 As shown in the figure, it can be seen that the components of the stator flux in the stationary coordinate system are symmetrical, and the fluctuation of the flux amplitude in the steady state is also very small. The observation effect is very good. Regardless of whether the speed is high or low, the stator flux amplitude is a normal value, especially the stator flux amplitude in the low-speed section is well compensated. Through improvements, it can be found that the improved strategy focuses on amplitude compensation at low speed and noise suppression at high speed, realizing decoupling optimization and significantly improving the overall performance of the system.

Claims

1. An improved method for observing the stator flux of an induction motor in the low-speed section based on a pure voltage model, comprising the following steps: S1: Analyze the problems existing in observing the stator flux of an induction motor using a pure voltage model. S2: Analyze the improved scheme of using a low-pass filter in observing the stator flux of an induction motor. S3: Analyze the stator flux observation scheme of a double integrator with a fixed threshold based on coordinate transformation. S4: Analyze the improved scheme for the problems existing in observing the stator flux in the low-speed section.

2. The control method according to claim 1, characterized in that: In step S1, analyze the problems existing in observing the stator flux of an induction motor using a pure voltage model: First, the stator voltage equation of an induction motor in the αβ coordinate system is: where u sα and u sβ is the stator voltage of the motor in the αβ coordinate system, i sα and i sβ is the stator current of the motor in the αβ coordinate system, R s is the stator resistance, ψ sα and ψ sβ is the stator flux component in the αβ coordinate system. Converting it into a flux equation, the flux calculation equation can be obtained as: It can be seen that for the stator flux observation based on a pure voltage model, there is an integration link, which will cause the observation result not to meet the expectation. Analyze the problems existing through the integration of the signal x(t) with a DC component. The equation of the signal x(t) is: x(t)=A m sin(ω e t+θ0)+x dc Among them A m is the amplitude of the signal, ω e is the motor stator angular frequency, θ0 is the initial phase of the signal, x dc is the DC signal present in the signal. The equation can be obtained by performing a Laplace transform on the signal x(t) as: The equation of the output signal after integrating the above formula is: Performing an inverse Laplace transform on it, the equation of the output signal under time domain conditions can be obtained as: It can be seen from the equation that the output signal consists of three parts, one part is the sinusoidal AC quantity, and the remaining two parts are DC offsets. If there is a DC signal x in the input signal dc , the output signal will be x dc As t increases, the final offset will cause the integrator to saturate as time increases. There is also a DC offset A m cosθ0 / ω e , its value is 0 only when θ0=kπ+π / 2, and it is non-zero in other cases.

3. The control method according to claim 1, wherein: In step S2, analyze the improved scheme of using a low-pass filter in observing the stator flux of an induction motor: Based on the problems brought by the pure integrator in step S1, a low-pass filter is used to replace the pure integrator. After passing through a first-order low-pass filter, the equation of the output signal is: Among them, ω c is the cutoff frequency of the low-pass filter. Performing an inverse Laplace transform on the above output signal equation to obtain the output equation in the time domain as: Where θ = arctanω e / ω c . It can be seen from the formula that As time t increases, the term tends to And the cutoff frequency ω c The larger the value, the smaller the final value will be. The term will eventually approach 0 as time t increases, and the cutoff frequency ω c The larger it is, the faster it decays to 0. Therefore, for the selection of the low-pass filter cutoff frequency, when ω c When ω is larger, DC offset and low-frequency signals can be suppressed well, but larger amplitude and phase errors will be introduced. On the contrary, when ω is c The smaller the value, the smaller the amplitude and phase errors will be, but the ability to suppress interference signals will be worse.

4. The control method according to claim 1, wherein: In step S3, analyze the stator flux observation scheme of a threshold double integrator based on coordinate transformation: Based on the deficiencies in step S2, compensate for the amplitude and phase errors introduced by the low-pass filter. The total equation of the system is: Where y1=x / (s+ω c ) is the forward channel signal, y2=ω c z / (s+ω c ) is the feedback channel signal, and z is the compensation signal. The feedback component mainly suppresses the DC drift and saturation effect at low frequencies. During operation, if the output signal y is not saturated, then z = y. After calculation and integration, the output equation can be obtained as: At this time, this integrator is equivalent to a pure integrator, suppressing the amplitude and phase errors caused by the low-pass filter at low frequencies. If the output signal y reaches the saturation value, the equation of the output signal is: At this time, if the signal x contains a DC component, after performing a Laplace transform on the DC component in the signal and the threshold L, the output equation can be obtained as: Using the final value theorem, the equations of the first term and the second term in the above formula can be obtained as: Therefore, finally, it can be obtained that the DC bias output is independent of time and there will be no saturation phenomenon. The expression is: Therefore, when |y(t)| < L, the output equation is: Entering the saturation region, that is, |y(t)| ≥ L, the output equation is: Among them, θ L is the initial phase of the input signal when the signal enters the saturation region.

5. The control method according to claim 1, characterized in that: In step S4, analyze the improved scheme for the problems existing in observing the stator flux in the low-speed section: Based on step S3, it can be seen that in the high-speed stage, due to ω e >>ω c , the amplitude hardly changes, but at low speed, there will be a large error in the amplitude, so we need to compensate for it. Compared with the traditional fixed cutoff frequency design, it is changed to be based on the rotor speed ω r Dynamically adjust the cutoff frequency. The specific expression is: oh c =kω r +oh min in, k is the adjustment coefficient, ω r is the rotor speed, ω min For the minimum limit. Regarding the assignment decay problem: The ideal magnetic flux amplitude is: A m / ω e . Before improvement: at low speed (ω e <<ω c ) The approximate expression for amplitude comparison is: It can be seen that the amplitude is reduced compared to the ideal flux amplitude. At high speed (ω e >>ω c ), the flux amplitude is approximately A m / ω e . After improvement: At low speed (ω r →0), at this time ω c ≈ω min , the magnetic flux amplitude expression can be expressed as: At this time, if you set ω min <<ω e , then the flux amplitude can be approximated to the ideal flux amplitude: A m / ω e , which solves the problem of magnetic flux amplitude attenuation in the low-speed section. At high speed (ω r >>0), at this time ω c ≈kω r , which can effectively suppress noise and maintain amplitude accuracy. The slip expression of the induction motor is: s=(ω syn -ω r ) / ω syn , where ω syn is the synchronous mechanical angular velocity, and its relationship with the synchronous electrical angular velocity is: ω e =pω syn , where p is the number of pole pairs of the motor. For phase lag: θ = arctanω e / ω c . Before improvement: at low speed (ω e <<ω c ): θ≈0°, the phase lag is small, but the flux amplitude is severely attenuated. e >>ω c ): θ≈90°, the phase lag is large, but the amplitude attenuation is small and the noise suppression is strong. After improvement: Due to ω e =pω r / (1-s), the hysteresis angle expression can be obtained as: At low speed (ω r →0) The hysteresis angle expression is: It can be seen that although the angle has increased, it can be compensated by closing the loop. If ω r <<ω min , θ→0. At medium and high speeds (ω r >>0), at this time ω c ≈kω r , the lag angle expression is: It can be seen that if the slip ratio s is small, then θ ≈ arctan(p / k), which is independent of the speed magnitude and its value is determined by k and the number of pole pairs p. Through improvement, it can be found that the improvement strategy focuses on amplitude compensation at low speeds and noise suppression at high speeds, achieving decoupling optimization and significantly improving the overall performance of the system.

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