Method for performing decoding, communication device, processing device, and storage medium

Abstract

The communication apparatus may: generate P second bit sequences by cyclically shifting a first bit sequence obtained from a received signal based on P different shift parameters; and determining a codeword by performing continuous erase decoding of a list size L on each of the P second bit sequences. Each of the P different shift parameters satisfies the following. A predefined metric for the plurality of groups is determined based on an index i of a fragile bit among N bits within the first bit sequence, greater than a predetermined threshold. The predefined metric is obtained by adding the difference between the maximum operation value and the minimum operation value of each of the plurality of groups.

Description

Technical Field

[0001] The present disclosure relates to wireless communication systems. Background Art

[0002] Various technologies, such as machine-to-machine (M2M) communication, machine-type communication (MTC), and various devices requiring high data throughput, such as smartphones and tablet personal computers (PCs), have emerged and become widespread. Consequently, the data throughput required to be processed in cellular networks has rapidly increased. To meet this rapidly increasing data throughput, carrier aggregation technology or cognitive radio technology have been developed to efficiently utilize more frequency bands, as well as multiple-input multiple-output (MIMO) technology or multi-base station (BS) cooperation technology to increase the data capacity transmitted on limited frequency resources.

[0003] As more and more communication devices require greater communication capacity, enhanced mobile broadband (eMBB) communications, compared to traditional radio access technologies (RATs), are becoming increasingly popular. Furthermore, massive machine-type communications (mMTC), which connects multiple devices and objects to provide a variety of services anytime, anywhere, is a major consideration in next-generation communications.

[0004] Discussions are also underway to design communication systems that take into account services / user equipment (UE) that are sensitive to reliability and latency. The introduction of next-generation RATs is under discussion, taking into account eMBB communications, mMTC, and ultra-reliable low-latency communications (URLLC). Summary of the Invention

[0005] Technical issues

[0006] With the introduction of new radio communication technologies, the number of UEs that a BS should provide services to within a specified resource area is increasing, and the amount of data and control information that a BS transmits and receives to and from the UEs provided by the BS is also increasing. Because the amount of resources available for a BS to communicate with UEs is limited, a new method is needed for the BS to efficiently transmit and receive uplink / downlink data and / or uplink / downlink control information using limited radio resources. In other words, as the density of nodes and / or the density of UEs increases, a method is needed to efficiently utilize high-density nodes or high-density UEs for communication.

[0007] Furthermore, there is a need for methods to further improve the performance of polar codes, an alternative to existing channel codes. For example, the list-based continuous cancellation decoder used in polar code decoding performs a sorting operation on each bit, which can lead to significant implementation complexity and latency. Therefore, methods are needed to address these issues.

[0008] The objects to be achieved by the present disclosure are not limited to the contents specifically described above, and those skilled in the art will more clearly understand other objects not described herein from the following detailed description.

[0009] Technical Solution

[0010] In one aspect of the present disclosure, a method for decoding a received signal by a communication device in a wireless communication system is provided.

[0011] In another aspect of the present disclosure, a communication device is provided herein that is configured to decode a received signal in a wireless communication system. The communication device includes: at least one transceiver; at least one processor; and at least one computer memory operatively connected to the at least one processor and storing instructions that, when executed, cause the at least one processor to perform operations.

[0012] In another aspect of the present disclosure, a processing device in a wireless communication system is provided. The processing device includes: at least one processor; and at least one computer memory operatively connected to the at least one processor and storing instructions that, when executed, cause the at least one processor to perform operations.

[0013] In another aspect of the present disclosure, a computer-readable storage medium is provided, wherein the computer-readable storage medium is configured to store at least one program code, the program code including instructions that, when executed, cause at least one processor to perform operations.

[0014] The method or operation may include: receiving a signal associated with a first bit sequence having a length of N from another communication device, where N is an integer greater than 1; determining P different shift parameters for the first bit sequence, where P is an integer greater than 1; generating P second bit sequences by cyclically shifting the first bit sequence based on the P different shift parameters; and determining a codeword by performing successive cancellation decoding with a list size of L on each of the P second bit sequences, where L is an integer greater than 0. Determining the P different shift parameters for the first bit sequence includes determining a shift parameter s that satisfies the following: determining an index i of a vulnerable bit among the N bits of the first bit sequence; grouping operation values (e.g., channel log-likelihood ratio (LLR) values) for the first bit sequence into a plurality of groups based on the index i of the vulnerable bit; and a predefined metric (e.g., a sum of differences between minimum and maximum values) for the plurality of groups exceeds a predetermined threshold, wherein the predefined metric is obtained by summing the differences between the maximum operation value and the minimum operation value for each of the plurality of groups. In various aspects of the present disclosure, grouping operation values into multiple groups based on the index i of the vulnerable bit may include: determining to apply a boxplus operation to b_j=0 and to apply a sum operation to b_j=0 based on the binary representation of the index i of the vulnerable bit (b_1, b_2, ..., b_(n-1)); and determining the indices of the operation values combined by the boxplus operation among these operation values as a group.

[0015] In various aspects of the present disclosure, the first bit sequence may include bits obtained by another communication device by encoding information bits having a length of K using an error correction code having a size of N.

[0016] In various aspects of the present disclosure, the method or operation may include: determining a combination of candidates for the number of permutation paths and a candidate list size based on a fixed complexity for a parallelism of N / 2; and determining a combination with the best performance based on decoding performance of each of the combinations. The number of candidates for the permutation paths and the candidate list size for the combination with the best performance may be used as P and L, respectively.

[0017] The above solutions are only some examples of the present disclosure, and those skilled in the art can deduce and understand various examples into which the technical features of the present disclosure are incorporated from the following detailed description.

[0018] Beneficial effects

[0019] According to some implementations of the present disclosure, wireless communication signals can be efficiently transmitted / received, thereby improving the overall throughput of the wireless communication system.

[0020] According to some embodiments of the present disclosure, parallel decoding of polar codes can be feasible, thereby overcoming the disadvantage of increased complexity in traditional polar code decoding methods due to metric sorting.

[0021] Effects according to the present disclosure are not limited to those specifically described above, and other effects not described herein will be more clearly understood by those skilled in the art to which the present disclosure relates from the following detailed description. BRIEF DESCRIPTION OF THE DRAWINGS

[0022] The accompanying drawings are included to provide a further understanding of the present disclosure and illustrate examples of implementations of the present disclosure and together with the detailed description serve to explain the implementations of the present disclosure:

[0023] Figure 1 An example of a communication system 1 to which an implementation of the present disclosure is applied is shown;

[0024] Figure 2 is a block diagram illustrating an example of a communication device capable of performing the method according to the present disclosure;

[0025] Figure 3 Another example of a wireless device capable of performing implementations of the present disclosure is shown;

[0026] Figure 4 An example of a frame structure used in a wireless communication system based on the 3rd Generation Partnership Project (3GPP) is shown;

[0027] Figure 5 shows the process of processing a transport block (TB) on the sending side;

[0028] Figure 6 is an exemplary block diagram of a polar encoder;

[0029] Figure 7 The concepts of channel splitting and channel combining showing channel polarization;

[0030] Figure 8 shows the Nth level channel combination of polar code;

[0031] Figure 9 shows the evolution of the decoding path during the decoding of List L;

[0032] Figure 10 is a diagram illustrating the concept of selecting a position to which information bits are allocated in a polar code;

[0033] Figure 11 shows the puncturing and information bit allocation of polar codes;

[0034] Figure 12 An example of a permutation decoder is shown;

[0035] Figure 13 Figure 1 shows the correlation between decoding success or failure and metrics based on permutation (shift) parameters;

[0036] Figure 14 illustrates a decoding process according to some embodiments of the present disclosure;

[0037] Figure 15 is a diagram for explaining a process of partitioning channels according to some embodiments of the present disclosure;

[0038] Figure 16 Graphs the decoding success or failure based on the sum of minimum-maximum differences of a log-likelihood ratio (LLR) group according to a permutation (shift) parameter;

[0039] Figure 17 FIGURE 1 compares the performance of a decoding method for selecting a shift parameter according to some embodiments of the present disclosure with other decoding methods;

[0040] Figure 18 FIGURES compare the performance of a selected shift parameter decoding method and a consecutive cancellation list (SCL) decoding method according to some embodiments of the present disclosure;

[0041] Figure 19 Another example of a permutation decoder is shown;

[0042] Figure 20 Figure 1 shows the performance of permutation-based SCL decoding depending on the number of permutation paths and the list size;

[0043] Figure 21 Illustration of the required areas for the sorter and SCL decoder depending on the list size;

[0044] Figure 22 FIGURE 2 shows the comparison of normalized word error rate (WER) performance based on a fixed computational complexity.

[0045] Figure 23 illustrates physical channels used in a 3GPP-based communication system as an example of a wireless communication system and a signal transmission / reception process using the physical channels; and

[0046] Figure 24 The figure illustrates a random access procedure applicable to the embodiments of the present disclosure. DETAILED DESCRIPTION

[0047] Hereinafter, implementations according to the present disclosure will be described in detail with reference to the accompanying drawings. The detailed description given below with reference to the accompanying drawings is intended to illustrate exemplary implementations of the present disclosure, rather than to illustrate the only implementations that can be implemented according to the present disclosure. The following detailed description includes specific details in order to provide a thorough understanding of the present disclosure. However, it will be apparent to those skilled in the art that the present disclosure can be practiced without these specific details.

[0048] In some cases, known structures and devices may be omitted or shown in block diagram form to focus on important features of the structures and devices so as not to obscure the concepts of the present disclosure. The same reference numerals will be used throughout this disclosure to refer to the same or similar parts.

[0049] The following techniques, devices, and systems can be applied to various wireless multiple access systems. For example, multiple access systems may include code division multiple access (CDMA) systems, frequency division multiple access (FDMA) systems, time division multiple access (TDMA) systems, orthogonal frequency division multiple access (OFDMA) systems, single carrier frequency division multiple access (SC-FDMA) systems, multi-carrier frequency division multiple access (MC-FDMA) systems, and the like. CDMA can be implemented using radio technologies such as Universal Terrestrial Radio Access (UTRA) or CDMA2000. TDMA can be implemented using radio technologies such as Global System for Mobile Communications (GSM), General Packet Radio Service (GPRS), Enhanced Data Rates for GSM Evolution (EDGE) (i.e., GERAN), and the like. OFDMA can be specifically implemented using radio technologies such as Institute of Electrical and Electronics Engineers (IEEE) 802.11 (Wi-Fi), IEEE 802.16 (WiMAX), IEEE 802.20, and Evolved UTRA (E-UTRA). UTRA is part of the Universal Mobile Telecommunications System (UMTS), and 3rd Generation Partnership Project (3GPP) Long Term Evolution (LTE) is part of E-UMTS using E-UTRA. 3GPP LTE uses OFDMA on the downlink (DL) and SC-FDMA on the uplink (UL). LTE-Advanced (LTE-A) is an evolved version of 3GPP LTE.

[0050] For convenience of description, the present disclosure will be described assuming that it is applied to LTE and / or New RAT (NR). However, the technical features of the present disclosure are not limited thereto. For example, although the following detailed description is given based on a mobile communication system corresponding to the 3GPP LTE / NR system, the mobile communication system is applicable to any other mobile communication system except for matters specific to the 3GPP LTE / NR system.

[0051] For the terms and techniques used in this disclosure that are not described in detail, reference may be made to standard specifications based on 3GPP (for example, 3GPP TS 36.211, 3GPP TS 36.212, 3GPP TS 36.213, 3GPP TS36.321, 3GPP TS 36.300, 3GPP TS 36.331, 3GPP TS 37.213, 3GPP TS 38.211, 3GPP TS38.212, 3GPP TS38.213, 3GPP TS 38.214, 3GPP TS 38.300, 3GPP TS 38.331, etc.).

[0052] In an example of the present disclosure described later, if a device "assumes" something, this may mean that a channel transmission entity transmits the channel in accordance with the corresponding "assumption." This may also mean that a channel reception entity receives or decodes the channel in a form that conforms to the "assumption" provided that the channel is transmitted in accordance with the "assumption."

[0053] In the present disclosure, a user equipment (UE) may be fixed or mobile. Each of the various devices that transmit and / or receive user data and / or control information by communicating with a base station (BS) may be a UE. The term UE may be referred to as a terminal device, mobile station (MS), mobile terminal (MT), user terminal (UT), subscriber station (SS), wireless device, personal digital assistant (PDA), wireless modem, handheld device, etc. In the present disclosure, a BS refers to a fixed station that communicates with a UE and / or another BS and exchanges data and control information with the UE and another BS. The term BS may be referred to as an advanced base station (ABS), node B (NB), evolved node B (eNB), base transceiver system (BTS), access point (AP), processing server (PS), etc. Specifically, a BS of a universal terrestrial radio access (UTRAN) is referred to as an NB, a BS of an evolved UTRAN (E-UTRAN) is referred to as an eNB, and a BS of a new radio access technology network is referred to as a gNB. In the following, for convenience of description, NB, eNB, or gNB will be referred to as a BS regardless of the type or version of the communication technology.

[0054] In the present disclosure, a node refers to a fixed point that can send / receive radio signals to / from a UE by communicating with the UE. Regardless of its name, various types of BSs can be used as nodes. For example, a BS, NB, eNB, picocell eNB (PeNB), home eNB (HeNB), repeater, transponder, etc. can be a node. In addition, a node may not be a BS. For example, a radio remote head (RRH) or a radio remote unit (RRU) can be a node. Typically, RRH and RRU have a power level lower than that of the BS. Since the RRH or RRU (hereinafter, RRH / RRU) is typically connected to the BS via a dedicated line such as an optical cable, the collaborative communication between the RRH / RRU and the BS can be performed smoothly compared to the collaborative communication between the BS connected via a wireless link. At least one antenna is installed on each node. The antenna may refer to a physical antenna port or a virtual antenna or an antenna group. A node may also be referred to as a point.

[0055] In the present disclosure, a cell refers to a specific geographical area where one or more nodes provide communication services. Therefore, in the present disclosure, communication with a specific cell may mean communication with a BS or node that provides communication services to the specific cell. The DL / UL signal of a specific cell refers to the DL / UL signal from / to the BS or node that provides communication services to the specific cell. A cell that provides UL / DL communication services to a UE is specifically referred to as a serving cell. In addition, the channel state / quality of a specific cell refers to the channel state / quality of a channel or communication link generated between the BS or node that provides communication services to the specific cell and the UE. In a 3GPP-based communication system, the UE can use a CRS sent on a cell-specific reference signal (CRS) resource and / or a CSI-RS sent on a channel state information reference signal (CSI-RS) resource (allocated to a specific node by the antenna port of the specific node) to measure the DL channel state from a specific node.

[0056] The 3GPP-based communication system uses the concept of cells in order to manage radio resources, and distinguishes cells related to radio resources from cells of geographical areas.

[0057] A "cell" of a geographical area can be understood as a coverage area where a node can use a carrier to provide services, and a "cell" of radio resources is associated with a bandwidth (BW) that is a frequency range configured by the carrier. Since the DL coverage (the range in which a node can send a valid signal) and the UL coverage (the range in which a node can receive a valid signal from a UE) depend on the carrier that carries the signal, the coverage of a node can also be associated with the coverage of the "cell" of the radio resources used by the node. Therefore, the term "cell" can be used to sometimes indicate the service coverage area of a node, at other times to indicate a radio resource, or at other times to indicate a range that a signal using a radio resource can reach with effective strength.

[0058] In the 3GPP communications standard, the concept of cells is used to manage radio resources. A "cell" associated with radio resources is defined by a combination of DL resources and UL resources (i.e., a combination of DL component carriers (CCs) and UL CCs). A cell can be configured with only DL resources or a combination of DL and UL resources. If carrier aggregation is supported, the linkage between the carrier frequency of the DL resources (or DL CCs) and the carrier frequency of the UL resources (or UL CCs) can be indicated by system information. For example, the combination of DL and UL resources can be indicated by a System Information Block Type 2 (SIB2) linkage. In this case, the carrier frequency can be equal to or different from the center frequency of each cell or CC. When carrier aggregation (CA) is configured, the UE has only one radio resource control (RRC) connection with the network. During RRC connection establishment / reestablishment / handover, one serving cell provides non-access stratum (NAS) mobility information. During RRC connection reestablishment / handover, one serving cell provides security input. This cell is called a primary cell (Pcell). A Pcell refers to the cell operating on the primary frequency on which the UE performs the initial connection establishment procedure or initiates the connection reestablishment procedure. Depending on the UE's capabilities, a secondary cell (Scell) can be configured to form a set of serving cells along with the PCell. The Scell can be configured after the RRC connection is established and is used to provide additional radio resources in addition to the resources of the specific cell (SpCell). The carrier corresponding to the PCell on the DL is called the downlink primary CC (DL PCC), and the carrier corresponding to the PCell on the UL is called the uplink primary CC (UL PCC). The carrier corresponding to the Scell on the DL is called the downlink secondary CC (DL SCC), and the carrier corresponding to the Scell on the UL is called the uplink secondary CC (UL SCC).

[0059] As more and more communication devices require greater communication capacity, eMBB communication relative to traditional radio access technology (RAT) is needed. In addition, large-scale MTC, which provides various services anytime and anywhere by connecting multiple devices and objects to each other, is a major issue to be considered in the next generation of communications. In addition, the design of communication systems that take into account services / UEs that are sensitive to reliability and delay is also under discussion. Considering eMBB communication, large-scale MTC, ultra-reliable low-latency communication (URLLC), etc., the introduction of the next generation RAT is under discussion. Currently, in 3GPP, research on the next generation mobile communication system after EPC is underway. In this disclosure, for convenience, the corresponding technology is referred to as new RAT (NR) or fifth generation (5G) RAT, and a system using NR or supporting NR is referred to as an NR system.

[0060] Figure 1An example of a communication system 1 to which an implementation of the present disclosure is applied is shown. Figure 1 The communication system 1 applied to the present disclosure includes a wireless device, a base station, and a network. Here, the wireless device refers to a device that performs communication using a RAT (e.g., 5G NR or LTE (e.g., E-UTRA)) and may be referred to as a communication / radio / 5G device. The wireless device may include (but is not limited to) a robot 100a, vehicles 100b-1 and 100b-2, an extended reality (XR) device 100c, a handheld device 100d, a home appliance 100e, an Internet of Things (IoT) device 100f, and an artificial intelligence (AI) device / server 400. For example, the vehicle may include a vehicle with wireless communication capabilities, an autonomous vehicle, and a vehicle capable of vehicle-to-vehicle communication. Here, the vehicle may include an unmanned aerial vehicle (UAV) (e.g., a drone). The XR device may include an augmented reality (AR) / virtual reality (VR) / mixed reality (MR) device and may be implemented in the form of a head-mounted device (HMD), a head-up display (HUD) installed in a vehicle, a television, a smartphone, a computer, a wearable device, a home appliance device, a digital signage, a vehicle, a robot, etc. Handheld devices may include smartphones, smart tablets, wearable devices (e.g., smart watches or smart glasses), and computers (e.g., notebooks). Home appliances may include TVs, refrigerators, and washing machines. IoT devices may include sensors and smart meters. For example, a base station (BS) and a network may also be implemented as wireless devices, and a particular wireless device may operate as a base station (BS) or a network node relative to another wireless device.

[0061] Wireless devices 100a to 100f can connect to network 300 via BS 200. AI technology can be applied to wireless devices 100a to 100f, and wireless devices 100a to 100f can connect to AI server 400 via network 300. Network 300 can be configured using a 3G network, a 4G (e.g., LTE) network, or a 5G (e.g., NR) network. Although wireless devices 100a to 100f can communicate with each other via BS 200 / network 300, wireless devices 100a to 100f can perform direct communication (e.g., sidelink communication) with each other without going through the BS / network. For example, vehicles 100b-1 and 100b-2 can perform direct communication (e.g., vehicle-to-vehicle (V2V) / vehicle-to-everything (V2X) communication). IoT devices (e.g., sensors) can perform direct communication with other IoT devices (e.g., sensors) or other wireless devices 100a to 100f.

[0062] Wireless communication / connections 150a and 150b may be established between wireless devices 100a to 100f and BS 200, and between wireless devices 100a to 100f. Wireless communication / connections such as UL / DL communication 150a and sidelink communication 150b (or device-to-device (D2D) communication) may be established via various RATs (e.g., 5G NR). The wireless devices and BS / wireless devices may transmit / receive radio signals to / from each other via wireless communication / connections 150a and 150b. To this end, various configuration information configuration processes, various signal processing processes (e.g., channel coding / decoding, modulation / demodulation, and resource mapping / demapping), and at least a portion of resource allocation processes for transmitting / receiving radio signals may be performed based on various proposals of the present disclosure.

[0063] Figure 2 is a block diagram illustrating an example of a communication device capable of executing the method according to the present disclosure. Figure 2 , the first wireless device 100 and the second wireless device 200 may transmit and / or receive radio signals via various RATs (e.g., LTE and NR). Here, {the first wireless device 100 and the second wireless device 200} may correspond to Figure 1 {wireless device 100x and BS 200} and / or {wireless device 100x and wireless device 100x}.

[0064] The first wireless device 100 may include one or more processors 102 and one or more memories 104, and may further include one or more transceivers 106 and / or one or more antennas 108. The processor 102 may control the memory 104 and / or the transceiver 106 and may be configured to implement the functions, processes, and / or methods described / proposed below. For example, the processor 102 may process information in the memory 104 to generate first information / signals, and then transmit a radio signal including the first information / signals via the transceiver 106. The processor 102 may receive a radio signal including second information / signals via the transceiver 106, and then store information obtained by processing the second information / signals in the memory 104. The memory 104 may be connected to the processor 102 and may store various information related to the operation of the processor 102. For example, the memory 104 may execute some or all of the processes controlled by the processor 102 or store software code including commands for executing the processes and / or methods described / proposed below. Here, the processor 102 and the memory 104 may be part of a communication modem / circuit / chip designed to implement a RAT (e.g., LTE or NR). The transceiver 106 may be connected to the processor 102 and transmit and / or receive radio signals via one or more antennas 108. Each transceiver 106 may include a transmitter and / or a receiver. The transceiver 106 may be used interchangeably with a radio frequency (RF) unit. In the present disclosure, a wireless device may refer to a communication modem / circuit / chip.

[0065] The second wireless device 200 may include one or more processors 202 and one or more memories 204, and may also include one or more transceivers 206 and / or one or more antennas 208. The processor 202 may control the memory 204 and / or the transceiver 206 and may be configured to implement the functions, processes, and / or methods described / proposed below. For example, the processor 202 may process the information in the memory 204 to generate third information / signals, and then transmit a radio signal including the third information / signals via the transceiver 206. The processor 202 may receive a radio signal including fourth information / signals via the transceiver 206, and then store information obtained by processing the fourth information / signals in the memory 204. The memory 204 may be connected to the processor 202 and may store various information related to the operation of the processor 202. For example, the memory 204 may execute some or all of the processes controlled by the processor 202 or store software code including commands for executing the processes and / or methods described / proposed below. Here, the processor 202 and the memory 204 may be part of a communication modem / circuit / chip designed to implement a RAT (e.g., LTE or NR). The transceiver 206 may be connected to the processor 202 and transmit and / or receive radio signals via one or more antennas 208. Each transceiver 206 may include a transmitter and / or a receiver. The transceiver 206 may be used interchangeably with an RF unit. In this disclosure, a wireless device may refer to a communication modem / circuit / chip.

[0066] The wireless communication technologies implemented in the wireless devices 100 and 200 of the present disclosure may include narrowband IoT for low-power communication, as well as LTE, NR, and 6G. For example, NB-IoT technology may be an example of low-power wide area network (LPWAN) technology and may be implemented in standards such as LTE Cat. NB1 and / or LTE Cat. NB-IoT technology, however, is not limited to the aforementioned designations. Additionally or alternatively, the wireless communication technologies implemented in the wireless devices XXX and YYY of the present disclosure may perform communication based on LTE-M technology. For example, LTE-M technology may be an example of LPWAN technology and may be referred to by various names including enhanced machine-type communication (eMTC). For example, LTE-M technology may be implemented in accordance with at least one of the following standards: 1) LTE Cat. 0, 2) LTE Cat. M1, 3) LTE Cat. M2, 4) LTE non-bandwidth limited (non-BL), 5) LTE-MTC, 6) LTE machine-type communication, and / or 7) LTE M, etc., but LTE-M technology is not limited to the aforementioned designations. Additionally or alternatively, considering low-power communication, the wireless communication technology implemented in the wireless devices XXX and YYY of the present disclosure may include at least one of ZigBee, Bluetooth, and LPWAN, but the wireless communication technology is not limited to the above names. For example, ZigBee technology can create a personal area network (PAN) related to low / low-power digital communication based on various standards such as IEEE 802.15.4, and ZigBee technology can be referred to by various names.

[0067] In the following, the hardware elements of the wireless devices 100 and 200 will be described in more detail. One or more protocol layers may be implemented by (but not limited to) one or more processors 102 and 202. For example, the one or more processors 102 and 202 may implement one or more layers (e.g., functional layers such as a physical (PHY) layer, a medium access control (MAC) layer, a radio link control (RLC) layer, a packet data convergence protocol (PDCP) layer, a radio resource control (RRC) layer, and a service data adaptation protocol (SDAP) layer). The one or more processors 102 and 202 may generate one or more protocol data units (PDUs) and / or one or more service data units (SDUs) according to the functions, procedures, proposals, and / or methods disclosed in the present disclosure. The one or more processors 102 and 202 may generate messages, control information, data, or information according to the functions, procedures, proposals, and / or methods disclosed in the present disclosure. One or more processors 102 and 202 may generate a signal (e.g., a baseband signal) including a PDU, an SDU, a message, control information, data, or information according to the functions, procedures, proposals, and / or methods disclosed in the present disclosure, and provide the generated signal to one or more transceivers 106 and 206. One or more processors 102 and 202 may receive a signal (e.g., a baseband signal) from one or more transceivers 106 and 206 according to the functions, procedures, proposals, and / or methods disclosed in the present disclosure and obtain the PDU, SDU, message, control information, data, or information.

[0068] The one or more processors 102 and 202 may be referred to as controllers, microcontrollers, microprocessors, or microcomputers. The one or more processors 102 and 202 may be implemented by hardware, firmware, software, or a combination thereof. As an example, one or more application-specific integrated circuits (ASICs), one or more digital signal processors (DSPs), one or more digital signal processing devices (DSPDs), one or more programmable logic devices (PLDs), or one or more field-programmable gate arrays (FPGAs) may be included in the one or more processors 102 and 202. The functions, processes, proposals, and / or methods disclosed in this disclosure may be implemented using firmware or software, and the firmware or software may be configured to include modules, processes, or functions. The firmware or software configured to perform the functions, processes, proposals, and / or methods disclosed in this disclosure may be included in the one or more processors 102 and 202 or stored in the one or more memories 104 and 204 to be driven by the one or more processors 102 and 202. The functions, processes, proposals, and / or methods disclosed in this disclosure may be implemented using firmware or software in the form of code, commands, and / or command sets.

[0069] One or more memories 104 and 204 may be connected to one or more processors 102 and 202 and store various types of data, signals, messages, information, programs, codes, commands, and / or instructions. One or more memories 104 and 204 may be configured by read-only memory (ROM), random access memory (RAM), electrically erasable programmable read-only memory (EPROM), flash memory, hard drive, registers, cache memory, computer-readable storage media, and / or combinations thereof. One or more memories 104 and 204 may be located internally and / or externally to one or more processors 102 and 202. One or more memories 104 and 204 may be connected to one or more processors 102 and 202 via various technologies such as wired or wireless connections.

[0070] One or more transceivers 106 and 206 may transmit user data, control information, and / or radio signals / channels mentioned in the methods and / or operational flow charts of the present disclosure to one or more other devices. One or more transceivers 106 and 206 may receive user data, control information, and / or radio signals / channels mentioned in the functions, processes, proposals, methods, and / or operational flow charts disclosed in the present disclosure from one or more other devices. For example, one or more transceivers 106 and 206 may be connected to one or more processors 102 and 202 and transmit and receive radio signals. For example, one or more processors 102 and 202 may execute control so that one or more transceivers 106 and 206 may transmit user data, control information, or radio signals to one or more other devices. One or more processors 102 and 202 may execute control so that one or more transceivers 106 and 206 may receive user data, control information, or radio signals from one or more other devices. One or more transceivers 106 and 206 may be connected to one or more antennas 108 and 208. One or more transceivers 106 and 206 may be configured to transmit and receive user data, control information, and / or radio signals / channels mentioned in the functional, process, proposal, method, and / or operational flow charts disclosed in the present disclosure through one or more antennas 108 and 208. In the present disclosure, one or more antennas may be multiple physical antennas or multiple logical antennas (e.g., antenna ports). One or more transceivers 106 and 206 may convert received radio signals / channels, etc. from RF band signals to baseband signals so that the received user data, control information, radio signals / channels, etc. may be processed using one or more processors 102 and 202. One or more transceivers 106 and 206 may convert user data, control information, radio signals / channels, etc. processed using one or more processors 102 and 202 from baseband signals to RF band signals. To this end, one or more of the transceivers 106 and 206 may include (analog) oscillators and / or filters.

[0071] Figure 3 Another example of a wireless device capable of executing implementations of the present disclosure is shown. Figure 3 , the wireless devices 100 and 200 may correspond to Figure 2 The wireless devices 100 and 200 may be configured by various elements, components, units / portions and / or modules. For example, each of the wireless devices 100 and 200 may include a communication unit 110, a control unit 120, a memory unit 130 and an additional component 140. The communication unit may include a communication circuit 112 and a transceiver 114. For example, the communication circuit 112 may include Figure 2The one or more processors 102 and 202 and / or the one or more memories 104 and 204 of the present invention may include: Figure 2 The control unit 120 is electrically connected to the communication unit 110, the memory 130, and the additional components 140, and controls the overall operation of the wireless device. For example, the control unit 120 may control the electrical / mechanical operation of the wireless device based on the program / code / command / information stored in the memory unit 130. The control unit 120 may transmit information stored in the memory unit 130 to the outside (e.g., other communication devices) via the communication unit 110 via a wireless / wired interface, or store information received from the outside (e.g., other communication devices) via the communication unit 110 via the wireless / wired interface in the memory unit 130.

[0072] The additional components 140 may be configured differently depending on the type of wireless device. For example, the additional components 140 may include at least one of a power supply unit / battery, an input / output (I / O) unit, a drive unit, and a computing unit. The wireless device may be configured in accordance with, but not limited to, a robot ( Figure 1 100a), vehicles ( Figure 1 100b-1 and 100b-2), XR devices ( Figure 1 100c), handheld device ( Figure 1 100d), household appliances ( Figure 1 100e), IoT devices ( Figure 1 100f), digital broadcast UE, holographic device, public safety device, MTC device, medical device, fintech device (or financial device), security device, climate / environmental device, AI server / device ( Figure 1 400), BS( Figure 1 The wireless device may be used in a mobile or fixed location depending on the usage / service.

[0073] exist Figure 3In the wireless devices 100 and 200, the various elements, components, units / portions, and / or modules may all be connected to each other via a wired interface, or at least a portion thereof may be wirelessly connected via the communication unit 110. For example, in each of the wireless devices 100 and 200, the control unit 120 and the communication unit 110 may be wired, and the control unit 120 and the first unit (e.g., 130 and 140) may be wirelessly connected via the communication unit 110. The various elements, components, units / portions, and / or modules within the wireless devices 100 and 200 may also include one or more elements. For example, the control unit 120 may be configured by a collection of one or more processors. As an example, the control unit 120 may be configured by a collection of a communication control processor, an application processor, an electronic control unit (ECU), a graphics processing unit, and a memory control processor. As another example, the memory 130 may be configured by a random access memory (RAM), a dynamic RAM (DRAM), a read-only memory (ROM), a flash memory, a volatile memory, a non-volatile memory, and / or a combination thereof.

[0074] In the present disclosure, at least one memory (e.g., 104 or 204) may store instructions or programs, and these instructions or programs, when executed, may cause at least one processor operatively connected to the at least one memory to perform operations according to some embodiments or implementations of the present disclosure.

[0075] In the present disclosure, a computer-readable (non-volatile) storage medium may store at least one instruction or program, and when the at least one instruction or program is executed by at least one processor, the at least one processor may cause the at least one processor to perform operations according to some embodiments or implementations of the present disclosure.

[0076] In the present disclosure, a processing device or apparatus may include at least one processor and at least one computer memory operatively connected to the at least one processor. The at least one computer memory may store instructions or programs, and when executed, these instructions or programs may cause the at least one processor operatively connected to the at least one memory to perform operations according to some embodiments or implementations of the present disclosure.

[0077] In the present disclosure, a computer program may include program code stored on at least one computer-readable (non-volatile) storage medium, and when executed, is configured to perform operations according to some implementations of the present disclosure or cause at least one processor to perform operations according to some implementations of the present disclosure. The computer program may be provided in the form of a computer program product. The computer program product may include at least one computer-readable (non-volatile) storage medium.

[0078] The communication device of the present disclosure includes: at least one processor; and at least one computer memory, which is operatively connected to the at least one processor and configured to store instructions that, when executed, cause the at least one processor to perform operations according to examples of the present disclosure described later.

[0079] Figure 4 An example of a frame structure used in a 3GPP-based wireless communication system is shown.

[0080] Figure 4 The frame structure is only exemplary, and the number of subframes, the number of time slots, and the number of symbols in a frame may vary. In the NR system, different OFDM parameter sets (e.g., subcarrier spacing (SCS)) may be configured for multiple cells aggregated for one UE. Therefore, the (absolute time) duration of time resources including the same number of symbols (e.g., subframes, time slots, or transmission time intervals (TTIs)) may be configured differently for the aggregated cells. Here, the symbol may include an OFDM symbol (or a cyclic prefix-OFDM (CP-OFDM) symbol) and an SC-FDMA symbol (or a discrete Fourier transform-spread-OFDM (DFT-s-OFDM) symbol). In the present disclosure, symbols, OFDM-based symbols, OFDM symbols, CP-OFDM symbols, and DFT-s-OFDM symbols may be used interchangeably.

[0081] Reference Figure 4 In the NR system, UL transmission and DL transmission are organized into frames. Each frame has T f =(△f max *N f / 100)*T c = 10 ms duration and is divided into two half frames of 5 ms each. The basic time unit of NR is T c =1 / (△f max *N f ), where △f max =480*10 3 Hz and N f =4096. For reference, the basic time unit of LTE is T s =1 / (△f ref *N f,ref ), where △f ref =15*10 3 Hz and N f,ref =2048. T c and T f With constant κ = T c / T f =64. Each half frame includes 5 subframes, and the duration of a single subframe is T sfThe subframe is further divided into slots, and the number of slots in a subframe depends on the subcarrier spacing. Each slot includes 14 or 12 OFDM symbols based on the cyclic prefix. In normal CP, each slot includes 14 OFDM symbols, and in extended CP, each slot includes 12 OFDM symbols. The parameter set depends on the exponentially scalable subcarrier spacing Δf=2 u *15kHz. The following table shows the number of OFDM symbols per time slot (N slot symb ), the number of time slots per frame (N frame，u slot ) and the number of time slots per subframe (N subframe，u slot ).

[0082] [Table 1]

[0083] u <![CDATA[N slot symb ]]> <![CDATA[N frame,u slot ]]> <![CDATA[N subframe,u slot ]]> 0 14 10 1 1 14 20 2 2 14 40 4 3 14 80 8 4 14 160 16

[0084] The following table shows the subcarrier spacing Δf = 2 u *15 kHz, number of OFDM symbols per slot, number of slots per frame, and number of slots per subframe.

[0085] [Table 2]

[0086] u <![CDATA[N slol symb ]]> <![CDATA[N frame,u slot ]]> <![CDATA[N subframe，u slot ]]> 2 12 40 4

[0087] For subcarrier spacing configuration u, the slots may be indexed in ascending order within a subframe as follows: n u s ∈{0,...,n subframe ,u slot -1}, and are indexed in ascending order within the frame as follows: n u s，f ∈{0,...,n frame,u slot -1}.

[0088] A slot includes multiple (e.g., 14 or 12) symbols in the time domain. For each parameter set (e.g., subcarrier spacing) and carrier, a common resource block (CRB) N is allocated from a common resource block (CRB) N indicated by higher layer signaling (e.g., RRC signaling). start,u grid Begins to define N size,u grid，x *N RB sc subcarriers and N subframe,u symb OFDM symbol resource grid, where N size,u grid，xis the number of resource blocks (RBs) in the resource grid, and the subscript x is DL for the downlink and UL for the uplink. N RB sc is the number of subcarriers per RB. In 3GPP-based wireless communication systems, N RB sc Typically 12. For a given antenna port p, subcarrier spacing configuration u, and transmission link (DL or UL), there is one resource grid. The carrier bandwidth N for subcarrier spacing configuration u is given to the UE through higher layer parameters (e.g., RRC parameters) size,u grid . Each element in the resource grid for antenna port p and subcarrier spacing configuration u is called a resource element (RE), and one complex symbol can be mapped to each RE. Each RE in the resource grid is uniquely identified by an index k in the frequency domain and an index l representing the symbol position relative to a reference point in the time domain. In the NR system, an RB is defined by 12 consecutive subcarriers in the frequency domain. In the NR system, RBs are classified into CRBs and physical resource blocks (PRBs). For subcarrier spacing configuration u, CRBs are numbered from 0 upwards in the frequency domain. The center of subcarrier 0 of CRB 0 of subcarrier spacing configuration u is equal to "point A" used as a common reference point for the RB grid. PRBs for subcarrier spacing configuration u are defined within a bandwidth part (BWP) and are numbered from 0 to N size,u BWP,i -1 numbering, where i is the number of BWPs. PRB n in BWP i PRB With CRBn u CRB The relationship between n u PRB =n u CRB +N size,u BWP,i Given, where N size BWP,i is the CRB where the BWP starts relative to CRB 0. A BWP consists of multiple consecutive RBs in the frequency domain. For example, a BWP may be a CRB with a given parameter set u in BWP i on a given carrier. i A carrier may include up to N (e.g., 5) BWPs. A UE may be configured with one or more BWPs on a given component carrier. Data communication is performed via enabled BWPs, and only a predetermined number of BWPs (e.g., one BWP) among those configured for the UE may be active on the component carrier.

[0089] Figure 5 Shows the process of processing a transport block (TB) on the transmitting side.

[0090] To enable the receiving side to correct errors experienced by radio signals in a radio channel, the transmitting side encodes information using a forward error correction code and then transmits the encoded information. The receiving side demodulates the received signal and decodes the error correction code, thereby recovering the information transmitted by the transmitting side. During this decoding process, errors in the received signal caused by the radio channel are corrected.

[0091] Data arrives at the coding block in each DL / UL cell in the form of at most two transport blocks per transmission time interval (TTI). The following coding steps can be applied to each transport block of the DL / UL cell:

[0092] - Cyclic redundancy check (CRC) attachment to the transport block;

[0093] - Code block segmentation and CRC attachment to the code blocks;

[0094] - Channel coding;

[0095] - Rate matching; and

[0096] - Code block concatenation.

[0097] In a practical communication system, a transport block of a predetermined size or larger is divided into multiple smaller data blocks and then encoded to facilitate the actual implementation of encoding. The smaller data blocks are called code blocks. Although code blocks usually have the same size, one of the code blocks may have a different size due to the limited size of the internal interleaver of the channel encoder. Error correction coding is performed on each code block of a predetermined interleaver size, and then interleaving is performed to reduce the impact of burst errors generated during transmission over the radio channel. The error-corrected and interleaved code blocks are transmitted by being mapped to actual radio resources. The amount of radio resources designated for actual transmission is specified. Therefore, the encoded code blocks are rate-matched to the amount of radio resources. Usually, rate matching is performed by puncturing or repetition. For example, if the amount of radio resources (i.e., the number of transmission bits that can be transmitted on the radio resources) is M, and if the encoded bit sequence (i.e., the number of output bits of the encoder) is N (where M is different from N), then rate matching is performed to make the length of the encoded bit sequence match M. If M > N, then all or part of the bits of the encoded bit sequence are repeated to make the length of the rate-matched sequence match M. If M < N, then a part of the bits of the encoded bit sequence is punctured to make the length of the rate-matched sequence match M, and the punctured bits are excluded from the transmission.

[0098] In a wireless communication system, the transmitting side encodes the data to be transmitted based on channel coding with a specific code rate. Then, the transmitting side adjusts the code rate of the data to be transmitted through a rate matching process involving puncturing and repetition.

[0099] There are various types of error correction codes, but the size of the information block that can achieve the best performance is determined according to the error correction code. Although many coding schemes can be used to achieve high-capacity information performance with long information block lengths, most coding schemes do not consistently show good performance over a wide range of information block lengths and code rates. . However, turbo codes, low-density parity-check (LPDC) codes, and polar codes show promising BLER performance over a wide range of code rates and code lengths, and are therefore being considered for use in NR systems. As requirements for various scenarios such as eMBB, massive IoT, and URLLC increase, there is a need for coding schemes that provide higher channel coding efficiency than turbo codes. In addition, there is a need to increase the maximum number of subscribers that can be accommodated by a channel, that is, to increase capacity. Polar codes, among various error-correcting codes, provide a new framework that can address the problems of traditional channel codes. Invented by E. Arikan of Bilkent University (see: "Channel Polarization: A Method for Constructing Capacity-Achieving Codes for Symmetric Binary-Input Memoryless Channels," IEEE Transactions on Information Theory, Vol. 55, No. 7, pp. 3051-3073, July 2009). Polar codes are mathematically proven to be the first capacity-achieving codes with low encoding and decoding complexity. Polar codes outperform turbo codes over large information block lengths without the presence of error streams. Hereinafter, channel coding using polar codes is referred to as polar coding.

[0100] Polar codes are said to be able to achieve the capacity of a given binary discrete memoryless channel. This is only possible when the block size is large enough. In other words, polar codes are codes that can achieve channel capacity even when the code size N increases infinitely. Polar codes have low encoding and decoding complexity and can be successfully decoded. Polar codes are a type of linear block error-correcting code. Multiple recursive connections are the basic building blocks of polar codes and the foundation of code construction. A physical conversion of the channel occurs, converting the physical channel into a virtual channel, and this conversion is based on multiple recursive cascades. If multiple channels are multiplied and accumulated, the majority of channels can become better or worse. The basic idea of polar codes is to use good channels. For example, data is sent at rate 1 through a good channel, and data is sent at rate 0 through a bad channel. That is, through channel polarization, the channel enters a polarized state from a normal state.

[0101] Figure 6 is an example of a block diagram of a polar code encoder.

[0102] Figure 6 (a) shows the basic module of polar code, specifically, the first level channel combination for polar coding. Figure 6 In (a), W2 represents the entire equivalent channel obtained by combining two binary-input discrete memoryless channels (B-DMCs) Ws. In this paper, u1 and u2 are the binary input source bits, and y1 and y2 are the output coded bits. Channel combining is the process of connecting B-DMCs in parallel.

[0103] Figure 6 (b) shows a basic matrix F of the basic module. The binary input source bits u1 and u2 input to the basic matrix F and the output coded bits x1 and x2 of the basic matrix F have the following relationship.

[0104] [Formula 1]

[0105]

[0106] Channel W2 can achieve the highest rate symmetric capacity I(W). In B-DMC W, symmetric capacity is an important parameter for measuring rate and is the highest rate at which reliable communication can occur on channel W. B-DMC can be defined as follows.

[0107] [Formula 2]

[0108]

[0109] A second set of N binary input channels can be synthesized or created from N independent copies of a given B-DMC W, and the channels have the property {W N (i) :1<=i<=N}. If N increases, there is a tendency for some channels to have a capacity close to 1 and the rest to have a capacity close to 0. This is called channel polarization. In other words, channel polarization is the process of creating a second set of N channels {W N (i) :1<=i<=N}. The influence of channel polarization means that when N increases, all symmetric capacity terms {I(W N (i))} tends to 0 or 1. In other words, the concept behind channel polarization in polar codes is to transform N copies (i.e., N transmissions) of a channel with a symmetric capacity of I(W) (e.g., an additive white Gaussian noise channel) into extreme channels with a capacity close to 1 or 0. Of the N channels, the I(W) portion will be perfect channels, while the 1-I(W) portion will be completely noisy channels. Then, only the information bits are sent through the good channels, while the bits input to the other channels are frozen at 1 or 0. The amount of channel polarization increases with the block length. Channel polarization consists of two stages: the channel combination stage and the channel splitting stage.

[0110] Figure 7 The concepts of channel combining and channel splitting for channel polarization are shown. Figure 7 As shown in , when N copies of the original channel W are appropriately combined to create the vector channel W vec , and then split into new polarized channels, if N is large enough, the new polarized channels are classified into channels with capacity C(W) = 1 and channels with capacity C(W) = 0. In this case, since bits passing through a channel with capacity C(W) = 1 are transmitted without error, it is preferable to transmit information bits through this channel. However, since bits passing through a channel with capacity C(W) = 0 cannot convey information, it is preferable to transmit frozen bits, which are meaningless bits, through this channel.

[0111] Reference Figure 7 (a), given a B-DMC W, copies are recursively combined to output the N →Y N Given the vector channel W vec , where N = 2 n Where n is an integer equal to or greater than 0. Recursion always starts at level 0 with W1 = W. If n is 1 (n=1), this represents the first level of recursion, in which two independent copies of W1 are combined. If these two copies are combined, a channel W2 is obtained: X2 → Y2. The transitional probability of this new channel W2 can be expressed as follows.

[0112] [Formula 3]

[0113]

[0114] If channel W2 is obtained, the two copies of channel W2 are combined to obtain a single copy of channel W4. This recursion can be represented by W4:X4→Y4 with the following transition probability.

[0115] [Formula 4]

[0116]

[0117] exist Figure 7 In (b), G N is a generator matrix of size N. Figure 7 (b) Input to G N u N 1 and G N The output x N The relationship between 1 can be expressed as x N 1=u N 1G N , where x N 1={x1,...,x N},u N 1={u1,...,u N When N B-DMCs are combined, each B-DMC can be expressed recursively. That is, G N It can be expressed by the following formula.

[0118] [Formula 5]

[0119]

[0120] In this article, represents the Kronecker product, N=2 n , n>=1, and B N is the permutation matrix known as the bit-reversal operation, and can be calculated recursively. I2 is the two-dimensional identity matrix, and the recursion is initialized to B2=I2. N is a bit-reversal interleaver and is used to convert the input s N 1={s1,...,s N} is mapped to the output x N 1={s1,s3,...,s N-1 ,s2,...,s N}. For example, G2 corresponds to Figure 6 The basis matrix F shown in (b) is G4, which can be expressed as the following matrix.

[0121] [Formula 6]

[0122]

[0123] The bit reversal interleaver may not be included in the transmitting side. Without considering the bit reversal interleaver, G N It can be expressed as (where N=2 n ). represents the nth Kronecker power of the matrix G2, where G2 is Figure 6 The same as the basis matrix F shown in (b).

[0124] The relationship in formula 5 is Figure 8 Shown in.

[0125] Figure 8 The Nth level channel combination for polar codes is shown.

[0126] The process of defining an equivalent channel for a specific input after combining N B-DMC Ws is called channel splitting. Channel splitting can be expressed as a channel transition probability indicated by the following equation.

[0127] [Formula 7]

[0128]

[0129] Signal polarization has the following characteristics:

[0130] > Conservative: C(W - )+C(W + )=2C(W),

[0131] >Polarization: C(W - )≤C(W)≤C(W + ).

[0132] When channel combining and channel splitting are performed, the following theorem can be obtained.

[0133] *Theorem: For any B-DMC W, the channel {W N (i)} is polarized in the following sense. For any fixed δ∈{0,1}, when N becomes infinite through powers of 2, the channel capacity I(W N (i) )∈(1-δ,1] becomes I(W) and is used for the channel capacity I(W N (i) )∈[0,δ) becomes 1-I(W). Therefore, if N→∞, the signal is completely noisy or polarized to be free of noise. These channels can be accurately identified by the transmitter. Therefore, bad channels are fixed, and non-fixed bits can be transmitted on good channels.

[0134] That is, if the polar code size N is infinite, the channel has a lot of noise or no noise relative to a specific input bit. This has the same meaning as if the capacity of the equivalent channel for a specific input bit is divided into 0 or I(W).

[0135] The input of the polar encoder is divided into bit channels to which information data is mapped and bit channels to which information data is not mapped. As previously mentioned, according to the polar code theorem, if the codeword of the polar code becomes infinite, the input bit channels can be classified as noiseless channels and noisy channels. Therefore, if information is assigned to the noiseless bit channels, channel capacity can be achieved. However, in reality, it is impossible to configure codewords of infinite length, calculate the reliability of the input bit channels, and assign data bits to the input bit channels in order of reliability. In this disclosure, bit channels to which data bits are assigned are referred to as good bit channels. Good bit channels can be input bit channels to which data bits are mapped. Bit channels to which data is not mapped are referred to as frozen bit channels. A known value (e.g., 0) is input to the frozen bit channel, and encoding is then performed. Any value known by the transmitting and receiving sides can be mapped to the frozen bit channel. Information about the good bit channels can be used when performing puncturing or repetition. For example, the positions of the codeword bits (i.e., output bits) corresponding to the positions of the input bits to which information bits are not assigned can be punctured.

[0136] The decoding scheme for polar codes is the successive cancellation (SC) decoding scheme. The SC decoding scheme obtains the channel transition probability and then uses it to calculate the log-likelihood ratio (LLR) of the input bit. In this case, if the channel combination and channel splitting processes use recursive features, the channel transition probability can be calculated recursively. Therefore, the final LLR value can also be calculated recursively. First, the input bit u can be obtained as follows i The channel transition probability W N (i) (y1 N ,u1 i-1 |u1). u1 i Splittable into odd index u 1,o i and even index u 1,e i The channel transition probability can be expressed as follows.

[0137] [Formula 8]

[0138]

[0139] in,

[0140] [Formula 9]

[0141]

[0142] The polar decoder uses the known values of the polar code (e.g., received bits, frozen bits, etc.) to retrieve the information and generate u N 1's estimate u^N 1. LLR is defined as follows.

[0143] [Equation 10]

[0144]

[0145] The LLR can be calculated recursively as follows.

[0146] [Equation 11]

[0147]

[0148] The recursive calculation of LLR is based on LLR L (1) 1(y i )=W(y i |0) / W(y i |1) Backtrack to code length 1. L (1) 1(y i ) is the soft information observed from the channel.

[0149] The complexity of the polar encoder and SC decoder scales with the length N of the polar code and is considered to be O(NlogN). Assuming K input bits are used for a polar code of length N, the coding rate becomes N / K. If the generator matrix of the polar encoder for a data payload size N is G N , then the coded bits can be expressed as x N 1=u N 1G N Assume u N The K bits in 1 correspond to the payload bits, then G N The row index corresponding to the payload bit is i, and G N The row index corresponding to (NK) bits is F. The minimum distance of the polar code can be assumed to be d min (C) = min i∈I 2 wt(i) , where wt(i) is the number of 1s in the binary expansion of i, and i=0, 1, ..., N-1.

[0150] SC list (SCL) decoding is an extension of the basic SC decoder. In this type of decoder, L decoding paths are considered simultaneously during each decoding phase. Here, L is an integer. In other words, in the case of polar codes, the List L decoding algorithm is used to track L paths simultaneously during the decoding process.

[0151] Figure 9The evolution of the decoding path during the decoding of a list L is illustrated. For ease of description, it is assumed that the number of bits to be determined is n and that all bits are not frozen. If the list size L is 4, each level includes a maximum of 4 nodes with paths that continue downward. Figure 9 The discontinuous paths are indicated by dashed lines. Figure 9 Describe the process of evolution of decoding paths in List L decoding. i) If List L decoding starts, the first unfrozen bit can be 0 or 1. ii) Continue with List L decoding. The second unfrozen bit can be 0 or 1. Since the number of paths is not greater than L=4, no pruning is needed yet. iii) Consideration of all options for the first bit (i.e., the bit of the first level), the second bit (i.e., the bit of the second level), and the third bit (i.e., the bit of the third level) results in 8 decoding paths, which are too many because L=4. iv) Prune the 8 decoding paths to L(=4) promising paths. v) Continue with the 4 valid paths by considering both options for the fourth unfrozen bit. In this case, the number of paths is doubled, i.e., 8 paths, which are too many because L=4. vi) Prune the 8 paths to L(=4) best paths. In Figure 7 In the example, four candidate codewords are obtained: 0100, 0110, 0111, and 1111. One of these codewords is determined to be the most similar to the original codeword. Similar to the normal decoding process, for example, during pruning or final codeword determination, the path with the largest sum of LLR absolute values can be selected as the surviving path. If a CRC is present, the surviving path can be selected based on the CRC.

[0152] In addition, CRC-assisted SCL decoding is SCL decoding using CRC, and improves the performance of polar codes. CRC is the most widely used error detection and correction technology in the fields of information theory and coding. For example, if the input block of the error correction encoder has K bits and the length of the information bits is k, and the length of the CRC sequence is m bits, then K=k+m. The CRC bits are part of the source bits of the error correction code. If the size of the channel code used for encoding is N, the code rate R is defined as R=K / N. CRC-assisted SCL decoding is used to detect error-free paths while the receiving device confirms the CRC code for each path. The SCL decoder outputs the candidate sequence to the CRC detector. The CRC detector feeds back the check result to help determine the codeword.

[0153] Although more complex than the SC algorithm, SCL decoding or CRC-assisted SCL decoding offers superior decoding performance. For more details on the List L decoding algorithm for polar codes, see I. Tal and A. Vardy, “List decoding of polar codes,” Proc. IEEE Int. Symp. Inf. Theory, July 2011, pp. 1–5.

[0154] Figure 10 The concept of selecting positions to which information bits are to be allocated in polar codes is illustrated.

[0155] exist Figure 10 In the example of , it is assumed that the size N of the mother code is 8 (that is, the size N of the polar code is 8) and the code rate is 1 / 2.

[0156] exist Figure 10 In, C(W i ) represents the channel W i The capacity of the channel corresponding to the input bit position of the polar code is as follows: Figure 10 When the reliability of the input bit position is shown as Figure 10 In order to transmit data at a code rate of 1 / 2, the transmitting device allocates the 4 bits constituting the data to the 4 input bit positions with high channel capacity among the 8 input bit positions (i.e., Figure 10 The input bit positions u1 to u8 are represented as input position bits u4, u6, u7 and u8) and the other input bit positions are frozen. Figure 10 The generator matrix G8 corresponding to the polar code is as follows. To obtain the generator matrix G8.

[0157] [Equation 12]

[0158]

[0159] Figure 10 The input bit positions represented as u1 to u8 correspond one-to-one to the rows from the highest row to the lowest row of G8. Figure 10, it can be understood that the input bit corresponding to u8 affects all output coded bits. On the other hand, it can be understood that the input bit corresponding to u1 only affects y1 among the output coded bits. Referring to Formula 12, when the binary input source bits u1 to u8 are multiplied by G8, the row in which the input bit appears at all output bits is the lowest row [1,1,1,1,1,1,1,1,1] among the rows of G8 in which all elements are 1. In addition, the row in which the binary input source bit appears at only one output bit is the row in which one element is 1 among the rows of G8, that is, the row [1,0,0,0,0,0,0,0] where the row weight is 1. Similarly, it can be understood that a row with a row weight of 2 reflects the input bit corresponding to the row into two output bits. Referring to Figure 10 As shown in Equation 12, u1 to u8 correspond to the rows of G8 one by one and bit indexes for distinguishing input positions of u1 to u8 (ie, bit indexes for distinguishing input positions) may be allocated to the rows of G8.

[0160] In the following, for polar codes, it can be assumed that the number of bits is calculated starting from the highest row with the smallest row weight relative to the N input bits toward G N Each row of is assigned a bit index from 0 to N-1. Figure 10 , bit index 0 is assigned to the input position of u1, i.e., the first row of G8, and bit index 7 is assigned to the input position of u8, i.e., the last row of G8. However, since the bit index is used to indicate the input position of the polar code, a scheme different from the above assignment scheme can be used. For example, bit indices 0 to N-1 can be assigned starting from the lowest row with the largest row weight.

[0161] In the case of output bit indexing, such as Figure 10 As shown in Equation 12, it may be assumed that bit indices of 0 to N-1 or bit indices of 1 to N are allocated to each column from the first column having the largest column weight to the last column having the smallest column weight among the columns of the GN.

[0162] In polar codes, the arrangement of information bits and frozen bits is one of the most important elements in the configuration and performance of polar codes. In other words, determining the rank of input bit positions can be a key factor in the performance and configuration of polar codes. For polar codes, bit indices can distinguish input positions or output bit positions of the polar code. In the present invention, a sequence obtained by enumerating the reliabilities of bit positions in ascending or descending order is called a bit index sequence or polar sequence. In other words, the bit index sequence represents the reliability of input bit positions or output bit positions of the polar code in ascending or descending order. A transmitting device inputs information bits to input bits with high reliability based on the input bit index sequence and performs encoding using the polar code. A receiving device can use the same or corresponding input bit index sequence to identify input positions to which information bits or frozen bits are assigned. In other words, the receiving device can use the same or corresponding input bit index sequence as that used by the transmitting device and perform polar decoding using the corresponding polar code. In the following description, it can be assumed that the input bit index sequence is predetermined so that information bits can be assigned to input bit positions with high reliability. In this disclosure, the input bit index sequence is also referred to as a polarization sequence.

[0163] Figure 11 The puncturing and information bit allocation of polar codes are illustrated. Figure 11 In the , F represents frozen bits, D represents information bits, and 0 represents skipped bits.

[0164] Among the coded bits, a situation in which an information bit is changed to a frozen bit may occur according to the index or position of the punctured bit. For example, if the output coded bits of the mother code of N=8 should be punctured in the order of Y8, Y7, Y6, Y4, Y5, Y3, Y2, and Y1 and the target code rate is 1 / 2, then Y8, Y7, Y6, and Y4 are punctured, and only U8, U7, U6, and U4 connected to Y8, Y7, Y6, and Y4 are frozen to 0, and these input bits are not transmitted, as shown in FIG. Figure 9 As shown. The input bits that are changed into frozen bits by puncturing the coded bits are called skipped bits or shortened bits, and the corresponding input positions are called skipped positions or shortened positions. Shortening is a rate matching method that inserts known bits into the input bit positions connected to the positions of the output bits to be transmitted while maintaining the size of the input information (i.e., the size of the information block). The matrix G can be generated from NThe input corresponding to the column with a column weight of 1 in the matrix is shortened starting from the input corresponding to the column with a column weight of 1 in the matrix, and the next shortening can be performed with respect to the input corresponding to the column with a column weight of 1 in the remaining matrix, in which the columns and rows with a column weight of 1 are removed. In order to prevent all information bits from being punctured, the information bits already assigned to the information bit positions can be reallocated in an order of high reliability within a set of frozen bit positions.

[0165] In the case of polar codes, decoding can generally be performed in the following order.

[0166] >1. Recover bits with low reliability first. Although the reliability varies depending on the structure of the decoder, since the input index in the encoder with a low value (hereinafter referred to as the encoder input bit index or bit index) usually has low reliability, decoding is usually performed starting from the low encoder input bit index.

[0167] >2. When there are known bits for recovery bits, the known bits are used together with the recovery bits or the processing of 1 is omitted and the known bits for the specific input bit positions are immediately used to recover the information bits that were unknown bits. The information bits can be source information bits (e.g., bits of a transport block) or CRC bits.

[0168] As described above, since the polar encoder has different reliabilities depending on the input position, the transmitting side can assign data blocks (i.e., information blocks before encoding) to bit channels in order of reliability based on the size of the corresponding data blocks, and perform encoding by setting all other items to frozen (e.g., a value of '0'). For example, if the mother code size of the polar encoder (i.e., the maximum size of a code block that the polar encoder can encode) is N, and if the size of the data block input to the polar encoder is K, polar encoding is performed by arranging the bits of the data block in order of reliability in K bit channels and setting 0 to NK bit channels.

[0169] The following shows the polarization sequence used in the NR (5G) system (see the polarization sequence defined in 3GPP TS 38.212 Rel-15).

[0170] <Polarization Sequence>

[0171]

[0172]

[0173]

[0174]

[0175] The table above shows the polarization sequence Q0Nmax-1 and the reliability of the polarization sequence W(Q i Nmax In the above table, W represents W(Q i Nmax ), I represents Q i Nmax That is, the table above gives the polarization sequence Q0 Nmax-1 ={Q0 Nmax ,Q1 Nmax ,...,Q Nmax-1 Nmax}, where 0 <= Q i Nmax <=Nmax-1 represents the bit index before polarization coding (ie, bit channel index), where i=0, 1, ..., Nmax-1. For 3GPP TS 38.212Rel-15, Nmax=1024. Polarization sequence Q0 Nmax-1 is the ascending order of reliability W(Q0 Nmax ) <W(Q1 Nmax )<... <W(Q Nmax-1 Nmax ), where W(Q i Nmax ) represents the bit index Q i Nmax For example, referring to the table above, bit index Q i Nmax =4 reliability W(Q i Nmax ) = 3 below the bit index Q i Nmax =3 reliability W(Q i Nmax ) = 7. That is, the above table lists the bit indices 0 to 1023 representing the 1024 input positions of the polar code with Nmax = 1024 in ascending order of reliability.

[0176] For any information block encoded as N bits, the same polarization sequence Q0 is used N-1 ={Q0 N ,Q1 N ,Q2 N ,...,Q N-1 N Polarization sequence Q0 N-1 is the polarization sequence Q0 Nmax-1 In ascending order of reliability W(Q0 N ) <W(Q1 N ) <W(Q2 N )<... <W(Q N-1 NAll elements Q with values less than N sorted i Nmax subset. For example, when N = 8, the polarization sequence Q0 7 includes among the elements of the polarization sequence Q0 Nmax-1 Q among i Nmax elements less than 8, and Q i Nmax elements less than 8 are sorted in ascending order of reliability W(0) < W(1) < W(2) < W(4) < W(3) < W(5) < W(6).

[0177] For example, Table 3 lists the input bit positions of the information blocks of size K(=10) of the polarization code input to the polarization sequence of N = 512.

[0178] [Table 3]

[0179] Polarization sequence 1 505 2 506 3 479 4 508 5 495 6 503 7 507 8 509 9 510 10 511

[0180] Table 3 shows 10 elements for K = 10 among the elements of the polarization sequence of N = 512 in ascending order of reliability. Referring to the above <polarization sequence> table, among the I(=Q i Nmax ) values less than N = 512, the 10 I values with reliability W(Q i Nmax ) are {479, 495, 503, 505, 506, 507, 508, 509, 510, 511}. If {479, 495, 503, 505, 506, 507, 508, 509, 510, 511} is arranged in ascending order of reliability W, then {505, 506, 479, 508, 495, 503, 507, 509, 510, 511} can be obtained, which is the set of bit indices for K = 10 in the polarization sequence of N = 512 shown in Table 3.

[0181] If the bit sequence input to the channel coding is represented by c0, c1, c2, c3, ···, c K-1 is represented, then the coded bits are represented by d0, d1, d2, d3, ···, d N-1 is represented, where K is the number of bits to be coded, where N = 2 n .

[0182] For any information block coded into N bits, the same polarization sequence Q0 N-1 ={Q0 N , Q1 N , Q2 N ,..., Q N-1 NPolarization sequence Q0 N-1 is the polarization sequence Q0 Nmax-1 A subset of Q where all elements Q are smaller than the value of N i Nmax In ascending order of reliability W(Q0 N ) <W(Q1 N ) <W(Q2 N )<... <W(Q N-1 N ) sort.

[0183] In some embodiments, if the input in polar encoding is u=[u0 u1 u2 ... u N-1 ], then the encoded output d=[d0 d1 d2 d3...d N-1 ]Through d=uG N is obtained, and the encoding can be performed in GF(2).

[0184] Because the list-based continuous cancellation decoder performs a sorting operation on each bit, the list-based continuous cancellation decoder used in conventional polar code decoding suffers from high implementation complexity and latency. To address this issue, a permutation decoding method has been proposed that performs decoding by shuffling the order of received signals.

[0185] If the vector π(c) formed by rearranging codewords c of code C using a permutation π belongs to the original code C, then the permutation π is called an automorphism. The permutation decoder generates candidate vectors by permuting the received signal based on several automorphisms and decodes these vectors in parallel. Unlike the successive cancellation list (SCL) decoder, the permutation decoder allows full parallelization and does not require a sorter, significantly improving implementation complexity and latency.

[0186] To apply permutation decoding, it is important to find automorphisms (permutations) that make the codewords valid even after the code order is shuffled. Algebraic codes such as Bose-Chaudhuri-Hocquenghem (BCH) codes can be easily shuffled using cyclic shifts. That is, when using cyclic shifts, simple permutations that satisfy the automorphism condition can be found. For example, in the case of BCH codes, different permutations can be generated by changing the shift parameter s based on the cyclic shift defined as follows: s ([c1,c2,c3,...,c N ])=[c N-s+1 ,c N-s+2 ,...,c N ,c1,c2,...,c N-s ].

[0187] Figure 12An example of a permutation decoder is shown. Figure 12 In some embodiments, a permutation decoder generates P decoding paths, performs decoding by applying different permutations to the P paths, and performs permutation decoding by selecting a codeword that is closest to the received signal. However, with permutation decoding, decoding performance may vary depending on the value of s, and an optimal value of s needs to be determined based on the received signal.

[0188] Figure 13 The decoding performance may vary depending on the permutation (shift) parameters. Figure 13 The figure shows the simulation results of permutation decoding for code size N=64, information size K=32 and SNR=3dB. Figure 13 In the decoding process, "Dec Success" indicates that the decoding is successful, and "Dec Fail" indicates that the decoding is unsuccessful. The success or failure of decoding is related to the minimum value of the log-likelihood ratio (LLR) of the message. Figure 13 ,It is observed that as the LLR of the message decreases, the probability of decoding failure increases.,If the shift parameter that increases the message LLR can be identified,,the decoding success rate can be improved.

[0189] Therefore, it is necessary to perform appropriate permutations to improve the performance of permutation decoding. If applying different permutations to the received vectors results in the same decoding result as SC decoding without permutation, the performance gain relative to complexity may be reduced. Therefore, it is important to find a permutation that guarantees different decoding results. In other words, the decoding performance of the permutation decoder depends on the shift parameter s, and selecting the shift parameter s based on the received vectors is crucial. Below, some embodiments of the present disclosure for determining the optimal shift parameter by analyzing the signal strength distribution from the received signal will be described.

[0190] Figure 14 The decoding process according to some embodiments of the present disclosure is illustrated.

[0191] The Continuous Cancellation (SC) or List-Continuous Cancellation (List-SC) decoder for polar codes extracts the LLRs of the channel: L0, L1, ..., L N-1 , calculate the LLR of the message in sequence: L (0) 、L (1) ,...,L (N-1)During this process, the distribution of message LLRs becomes polarized. As the code length N increases, some messages approach a confidence level close to 0, while other LLRs approach a confidence level close to ∞. In this case, the message with a confidence level of 0 is set to a value shared between the receiver and transmitter, and communication can be performed by embedding the actual message in the message with infinite confidence. In a finite-length code, because there are messages that experience partial polarization, there are messages that do not have infinite confidence. Such vulnerable messages with partial polarization are the cause of the degradation of decoding performance.

[0192] Some embodiments of the present disclosure relate to methods for improving the performance of a permutation decoder in a polar code decoder to reduce computational complexity. According to some embodiments of the present disclosure, given the LLR values of a received vector, the impact of the received LLR on a vulnerable message is analyzed, and the LLR value of the vulnerable message is directed to increase based on a shifted permutation. Some embodiments of the present disclosure may include shifting the received signal so that the highest LLR is transmitted to the most vulnerable message (e.g., the most vulnerable bit) within the information message (e.g., the bits of the received signal).

[0193] To this end, the decoder according to some embodiments of the present disclosure selects a vulnerable message (i.e., a vulnerable bit) from the received information message (i.e., the received bits) (S1401). For example, the decoder according to some embodiments of the present disclosure may select the vulnerable bit i based on SC error analysis, which analyzes the relationship between the bit with error and the LLR value. The vulnerable message LLR L is selected from the received signal of length N. N (i) The value of can be expressed as channel LLR L0, L1, ..., L N-1 Function: L N (i) =f(L0,L1,…,L N-1 ).

[0194] The index of the polar code bit channel can be partitioned (or grouped) based on the index i of the vulnerable message (S1402). The index of the vulnerable message (hereinafter referred to as the vulnerable index) is predefined. The vulnerable message index changes only depending on the channel and does not change according to the shift parameter.

[0195] Figure 15 is a diagram for explaining a process of partitioning channels according to some embodiments of the present disclosure.

[0196] In some embodiments of the present disclosure, the index [0:N-1] may be partitioned by the following process. For example, when N=64 and message 7 among N messages (messages 0 to N-1) is a vulnerable message, the index of the vulnerable message is i=7. According to the binary representation with the most significant bit (MSB) placed in the rightmost position (i.e., the binary representation for index i is defined as (b1, b2, ..., b n )), since N=64, i=7 can be represented by a 6-bit binary representation (111000). If you search for consecutive 0s starting from the MSB in the binary representation (111000), you get (111 000 ). If all possible binary vectors are substituted into consecutive 0 positions, the binary vectors that can be substituted into three consecutive 0 positions range from (---000) to (---111), resulting in 8 possible vectors. When (000) is substituted into the first three positions, the partition is created / determined using elements L0, L8, ..., L56. By converting the binary representation within the partition into decimal form, the channel index can be obtained. Therefore, each element in the partition can be matched with a channel index. Channel inputs belonging to the same partition participate in the boxplus operation during the decoding process.

[0197] In summary, the message LLR is generated by applying the following boxplus operation or summation operation to the channel LLR:

[0198] >boxplus operation:

[0199] >Summation operation: x+y

[0200] When the binary representation of index i for the vulnerable message is (b1,b2,...,b n ), if b j If b is 0, the boxplus operation is applied. j is 1, then the sum operation is applied. Based on the binary vector of the vulnerable message (b n ,b n-1 ,...,b1), the channel LLR index obtained by boxplus operation can be identified. For example, if the index of the vulnerable message is i=1=(100), the LLR of the message can be calculated according to the following formula, L (1) .

[0201] Formula 13

[0202]

[0203] When i=1=(100) is partitioned according to the above process, for (-00), (-01), (-10), (-11), if 0 is substituted into the "-" position, the following groups are obtained: one group (000)=0, (001)=4, (010)=2, (011)=6; and another group (100)=1, (101)=5, (110)=3, (111)=7. Because the elements of the index within each group are grouped using the boxplus operation, Equation 13 can be derived.

[0204] In Equation 13, it is assumed that the channel inputs (L0, L2, L4, L6) grouped together by the boxplus operation are group 1 (i.e., partition 1), and the other channel inputs (L1, L3, L5, L7) grouped together by the boxplus operation are group 2 (i.e., partition 2). The value of the boxplus operation for each group is determined by the minimum LLR value within the group. That is, the message L (7) The LLR can be approximated as follows.

[0205] Formula 14

[0206] L (1) =min(|L0|, |L2|,|L4|,|L6|)+min(|L1|, |L3|, |L5|, |L7|)

[0207] If even one low LLR value exists within a group, the overall calculation result may be reduced. Therefore, by applying a shift so that low received LLR values are not spread across multiple groups but are concentrated in a specific group, the reduction of the message LLR value can be prevented. With this in mind, in some embodiments of the present disclosure, the difference between the maximum LLR and the minimum LLR in each group is calculated, and the sum of these differences across all groups (minimum-maximum delta sum) is defined as a measure of the distribution state of the LLR.

[0208] When the decoder according to some embodiments of the present disclosure performs a shift on the received signal (S1403), the decoder will continue decoding (S1406) only when the minimum-maximum Δ sum value exceeds a certain threshold (yes in S1405). Each time the metric exceeds the threshold, decoding can be attempted. Decoding can be attempted up to P times, where P is the number of permutations. If the minimum-maximum Δ sum value for a certain shift parameter s is less than a specified threshold (no in S1405), the minimum-maximum Δ sum value for another shift parameter can be calculated to determine whether the other shift parameter is applied to the permutation. The specified threshold can be determined by iterative experimentation.

[0209] Reference again Figure 14For example, assume that group (0, 3) and group (1, 2) are generated by grouping. If the received signals L0, L1, L2, and L3 are not shifted, (L0, L3) are grouped together into one group, and (L1, L2) are grouped together into another group. Assuming that the signal obtained by shifting the received signal once is L3, L0, L1, L2, and (L3, L2) are grouped together into one group, and (L0, L1) are grouped together into another group. The decoding result depends on how the grouping is formed. As the minimum-maximum Δ and the shift increase, the success probability also increases. Therefore, when this value exceeds the threshold, decoding is attempted.

[0210] Figure 16 The figure shows the comparison of the sum of the minimum and maximum differences of the LLR groups based on the permutation (shift) parameters. Specifically, Figure 16 Simulation results of permutation decoding by assuming mother code size N=64, information size K=32, and noise strength (SNR)=3dB in additive white Gaussian noise (AWGN) channel are shown. Figure 16 Showing the min-max delta and can be used as a metric for embodiments of the present disclosure. This data can also be used to determine the specific thresholds mentioned above.

[0211] Figure 17 FIGURE 1 compares the performance of a decoding method for selecting a shift parameter according to some embodiments of the present disclosure with other decoding methods. Specifically, Figure 17 A comparison of word error rate (WER) performance among SCL decoding, SC decoding, permutation decoding in which a shift parameter is randomly selected and applied, and permutation decoding in which a shift parameter is selected and applied according to some embodiments of the present disclosure is shown when the number of paths is set to 8 (i.e., list size L=8) for an extended BCH code with a code length of 64 and a message length of 36 (i.e., mother code size N=64 and information size K=36 for the encoder). Figure 17 According to some embodiments of the present disclosure, permutation decoding applying shift parameters shows similar WER performance to that of SCL decoding, and permutation decoding shows better WER performance than existing permutation decoding applying random shift parameters.

[0212] Figure 18 The figure shows a comparison of the performance between the decoding method of selecting the shift parameter and the SCL decoding method according to some embodiments of the present disclosure. Specifically, Figure 18 A comparison of the WER performance between SCL decoding with a list size of L and decoding according to some embodiments of the present disclosure for an extended BCH code with a mother code size of N=64 and an information size of K=32 is shown. Figure 18, for different list sizes, decoding according to some embodiments of the present disclosure shows similar WER performance to SCL decoding. Unlike the SCL decoder, the decoder according to some embodiments of the present disclosure allows full parallelization and does not require a sequencer, thereby significantly improving implementation complexity and latency while still achieving similar WER performance to the SCL decoder.

[0213] Figure 19 Another example of a permutation decoder is shown. Figure 19 , SCL decoding with a list size of L is applied to each permutation path.

[0214] In some embodiments of the present disclosure, the decoding process is performed by arranging the received signals using different shift parameters to generate P received signals, and then decoding the P received signals using a list-SC decoder (i.e., SCL decoder) with a list size of L.

[0215] In this case, even if the total parallelism of L*P is the same, the total complexity and performance may vary depending on how each value is configured. For example, for a code with a mother code size N=64 and a total parallelism of L*P=32, if L=32 and P=1, it is equivalent to performing a single list-SC decoding, which provides excellent performance. However, because the list-SC decoder requires an additional sorting operation, the decoding complexity increases. On the other hand, for a total parallelism of L*P=32, if L=1 and P=32, each decoder can operate in parallel without a sorting operation, thereby reducing the complexity. However, the same result can be obtained from the parallel path. For example, when Figure 19 When decoding of the topmost path in

[15] fails, decoding of the paths below it, to which the shift is applied, is expected to succeed. However, there may be cases where decoding also fails. Therefore, for a total parallelism of L*P=32, decoding performance may degrade when L=1 and P=32. By combining the number of parallel paths associated with the permutation (i.e., the number of permutation paths) P and the list size L, an optimal balance between performance and complexity can be found.

[0216] Figure 20 The figure shows the performance of permutation-based SCL decoding depending on the number of permutation paths and the list size. Specifically, Figure 20 A comparison of WER performance depending on P and L without normalization is shown for an extended BCH code with a code length of N=64 and a message length of 36. Here, P represents the number of parallel permutation paths, and L represents the list size for SCL decoding. Figure 20 , it is observed that for the same total parallelism, the performance is optimal when L = 32 and P = 1. However, if L and P are reasonably combined, similar performance can be achieved when L = 8 and P = 4.

[0217] The following table shows the required sorter area based on the list size L (reference: Balatsoukas-Stimming, M. Bastani Parizi, and A. Burg, “On Metric Ordering for Decoding Consecutive Cancellation Lists for Polar Codes,” in 2015 IEEE International Symposium on Circuits and Systems (ISCAS), 2015, pp. 1993-1996).

[0218] Table 4

[0219]

[0220] At the same time, each time the list size increases by 1, the SCL decoder requires approximately 226,000 μm 2 (Reference: S.A. Hashemi, A. Balatsoukas-Stimming, P. Giard, C. Thibeault, and W.J. Gross, “Partitioned Continuous Cancellation List Decoding of Polar Codes,” in 2016 IEEE International Conference on Acoustics, Speech, and Signal Processing (ICASSP), 2016, pp. 957-960).

[0221] Figure 21 The figure shows the area required for the sequencer and SCL decoder depending on the list size. Figure 21 ,The area of the sorter increases exponentially with the list size, while the area of the SCL decoder increases linearly with the list size.

[0222] The following table shows the required delay (number of cycles) depending on the list size L, where a mother code size N = 512 and a code rate R = 0.5 are used (Reference: K. Lee and I. Park, "Big-small ordering for consecutive cancellation list decoding of polar codes," IEEE Access, vol. 8, pp. 96955-96962, 2020).

[0223] Table 5

[0224] L＝4 L＝8 L＝16 Sorting delay 1584 3168 5280 SCL Delay 5697 7281 9391

[0225] Referring to Table 5, as the list size L increases, the sorter overhead also increases.

[0226] The complexity of the SCL decoder and the sequencer can be compared by adding the operations of the adder and comparator to obtain the total computational load. The following table shows the comparison of the total computational load of the sequencer and the SCL decoder depending on the list size L for an extended BCH code with a mother code size N=64 and an information size K=36.

[0227] Table 6

[0228] L 8 16 32 64 128 Sequencer 864 2304 5760 13824 32256 SCL 3072 6144 12288 24576 49152

[0229] The following table shows the complexity of SCL decoding and permutation decoding for an extended BCH code with mother code size N = 64 and information size K = 36. Here, P represents the number of parallel permutation paths, and L represents the list size.

[0230] Table 7

[0231]

[0232] When normalization is performed by assuming that permutation decoding has the same operation as SCL decoding, parameters normalized with respect to P and L can be obtained as follows.

[0233] Table 8

[0234]

[0235] In this case, when the complexity of each combination is converted into the total number of operations for comparison, it can be expressed as follows.

[0236] Table 9

[0237]

[0238] Therefore, when the number of parallel paths P for permutation decoding is adjusted by assuming a fixed computational complexity (for example, the same as when P=1 and L=32, which is 18048), the following can be obtained: Figure 22 WER performance normalized to complexity shown in . Figure 22 FIGURES compare the normalized WER performance when the computational complexity is P=32 and L=1 for an extended BCH code with a mother code size of N=64 and an information size of K=36. Figure 22 , if the complexity is the same, it is observed that the permutation decoding with P = 5 and L = 8 has the best performance, and the permutation decoding has better performance than when SCL = 32. In some embodiments of the present disclosure, when the WER performance is compared for normalized combinations of P and L based on a fixed complexity, a (P, L) combination with lower WER performance can be determined.

[0239] In summary, in some embodiments of the present disclosure, a normalized combination of P and L can be obtained for a mother code size of N based on a fixed computational complexity. By comparing the permutation decoding performance for each combination of P and L, the combination of P and L with the best performance (e.g., the lowest WER) can be selected as the number of parallel paths P and the list size L for permutation decoding.

[0240] According to some embodiments of the present disclosure, a communication device or decoder may perform operations related to channel coding. The communication device or decoder may include: at least one transceiver; at least one processor; and at least one computer memory, the at least one computer memory being operably connected to the at least one processor and storing instructions that, when executed, cause the at least one processor to perform operations according to some embodiments of the present disclosure. A processing device for a communication device or decoder may include: at least one processor; and at least one computer memory, the at least one computer memory being operably connected to the at least one processor and storing instructions that, when executed, cause the at least one processor to perform operations according to some embodiments of the present disclosure. A computer-readable (non-volatile) storage medium may store at least one computer program including instructions that, when executed by at least one processor, cause the at least one processor to perform operations according to some embodiments of the present disclosure. A computer program or computer program product may include instructions that are recorded in at least one computer-readable (non-volatile) storage medium and, when executed, cause (at least one processor) to perform operations according to some embodiments of the present disclosure. For a communication device, a decoder, a computer-readable (non-volatile) storage medium and / or a computer program product, the operation may include: receiving a signal associated with a first bit sequence having a length of N from another communication device, where N is an integer greater than 1; determining P different shift parameters for the first bit sequence, where P is an integer greater than 1; generating P second bit sequences by cyclically shifting the first bit sequence based on the P different shift parameters; and determining a codeword by performing successive cancellation decoding with a list size of L on each of the P second bit sequences, where L is an integer greater than 0. Determining P different shift parameters for a first bit sequence includes: determining a shift parameter s that satisfies the following: determining an index i of a vulnerable bit among N bits of the first bit sequence; grouping operation values (e.g., channel LLR values) for the first bit sequence into a plurality of groups based on the index i of the vulnerable bit; and a predefined metric (e.g., a sum of differences between minimum and maximum values) for the plurality of groups exceeds a predetermined threshold, wherein the predefined metric is obtained by summing the difference between the maximum operation value and the minimum operation value for each of the plurality of groups. In some embodiments, grouping the operation values into the plurality of groups based on the index i of the vulnerable bit may include: determining, based on a binary representation (b_1, b_2, ..., b_(n-1)) of the index i of the vulnerable bit, applying a boxplus operation to b_j=0 and applying a summation operation to b_j=1; and determining the indices of the operation values combined by the boxplus operation as a group.

[0241] In some embodiments, the first bit sequence may include bits obtained by another communication device by encoding information bits having a length of K using an error correction code having a size of N.

[0242] In some embodiments, these operations may include: determining a combination of candidates for the number of permutation paths and a candidate list size based on a fixed complexity for a parallelism of N / 2; and determining a combination with the best performance based on the decoding performance of each of the combinations. The number of candidates for the permutation paths and the candidate list size of the combination with the best performance may be used as P and L, respectively.

[0243] In some embodiments of the present disclosure, a communication device or a decoder may establish a radio resource control (RRC) connection for communicating with another communication device (e.g., a UE or a BS), and receive a signal related to a first bit sequence of length N from the other communication device. Figure 23 and Figure 24 In the signal transmission / reception process described in

[15] , a communication device may receive a signal including part or all of a first bit sequence from another communication device through a physical channel (e.g., a physical uplink control channel (PUCCH), a physical uplink shared channel (PUSCH), a physical downlink control channel (PDCCH), a physical downlink shared channel (PDSCH), a physical broadcast channel (PBCH), a physical sidelink control channel (PSCCH), a physical sidelink shared channel (PSSCH), or a physical sidelink broadcast channel (PSBCH)). The physical channel may carry a rate-matched codeword obtained by performing rate matching on a codeword (e.g., a codeword of length N) of uplink control information (UCI), uplink shared channel (UL-SCH) data, a master information block (MIB) as part of minimum system information, downlink control information (DCI), downlink shared channel (DL-SCH) data, or sidelink shared channel (SL-SCH) data. The communication device or encoder that transmits the rate-matched codeword may perform rate matching on the codeword based on the radio resources allocated for transmission of the codeword and the target code rate of the corresponding physical channel. The communication device or encoder can send the rate-matched codeword as a transmission sequence to another communication device. The communication device or decoder receiving the rate-matched codeword can obtain a first bit sequence of length N from the received signal.

[0244] Figure 23 Physical channels used in a communication system based on 3GPP as an example of a wireless communication system and a signal transmission / reception process using the channels are illustrated.

[0245] When the UE is powered on or when the UE has been disconnected from the wireless communication system, the UE searches for a cell to reside in and performs an initial cell search (S11) involving synchronization with a base station (BS) in the cell. For the initial cell search, the UE receives a synchronization signal block (SSB) (also called an SSB / PBCH block) from the BS. The SSB includes a primary synchronization signal (PSS), a secondary synchronization signal (SSS), and a physical broadcast channel (PBCH). The UE establishes synchronization with the base station based on the PSS / SSS and obtains information such as a cell identity (ID). The UE can obtain broadcast information in the cell based on the PBCH. The UE can receive a DL reference signal (RS) to monitor the DL channel status during the initial cell search process.

[0246] After the initial cell search, the UE can camp on the cell. Subsequently, the UE can monitor the PDCCH in the cell and obtain more specific system information by receiving the PDSCH based on the DCI carried on the PDCCH (S12).

[0247] Subsequently, in order to complete the connection to the BS, the UE may perform a random access procedure (S13 to S16). In the random access procedure, for example, the UE may transmit a preamble on the PRACH (S13) and receive a PDCCH and a random access response (RAR) to the preamble on the PDSCH corresponding to the PDCCH (S14). When the UE fails to receive the RAR directed to the UE, the UE may attempt to retransmit the preamble. In the case of contention-based random access, the UE may transmit a PUSCH based on the UL resource assignment included in the RAR (S15) and perform a contention resolution procedure including receiving the PDCCH and the PDSCH corresponding to the PDCCH (S16).

[0248] After completing the above process, the UE can receive PDCCH / PDSCH from the BS (S17) and send PUSCH / PUCCH to the BS during normal UL / DL signal transmission (S19). The control information sent by the UE to the BS is generally referred to as uplink control information (UCI). UCI includes hybrid automatic repeat and request acknowledgment / negative acknowledgment (HARQ ACK / NACK), scheduling request (SR) and channel state information (CSI). CSI includes channel quality indicator (CQI), precoding matrix indicator (PMI) and / or rank indicator (RI). Typically, UCI is sent on PUCCH. However, when control information and data need to be sent simultaneously, control information can be sent on PUSCH. In addition, the UE can send UCI on PUSCH aperiodically after receiving a request / command from the network.

[0249] Figure 24 The figure shows a random access process applicable to the embodiment of the present disclosure. Specifically, Figure 24 (a) illustrates the 4-step random access process, and Figure 24 (b) illustrates the two-step random access process.

[0250] The random access procedure can be used for various purposes, including initial access, UL synchronization adjustment, resource allocation, handover, radio link reconfiguration after radio link failure, and positioning. Random access procedures are categorized as contention-based procedures and dedicated (i.e., non-contention-based) procedures. Contention-based random access procedures are generally used for initial access, while dedicated random access procedures are used for handover, UL synchronization reconfiguration when DL data arrives at the network, and positioning.

[0251] The PRACH preamble configuration to be used can be provided to the UE. Multiple RACH preamble formats (i.e., PRACH preamble formats) are defined by one or more RACH OFDM symbols and different cyclic prefixes (CPs) (and / or guard periods). The PRACH preamble configuration for a cell provides the PRACH preamble formats and RACH opportunities available on the cell to the UE. The RACH opportunity refers to the time-frequency resources that can be used to send / receive RA preambles. In some scenarios, a single RACH opportunity (RO) for all available RA preambles that can be sent on the cell is configured through an RRC message (e.g., SIB2 of the cell). In other scenarios, SSBs associated with different beams can be selected, and the association between the SSBs and RACH opportunities can be provided to the UE by the BS. SSBs associated with different DL beams of the cell can be identified by different SSB indices, where each SSB index can represent a different DL beam. The base station (BS) provides a set of available RACH opportunities for transmitting RA preambles through a PRACH configuration including the PRACH preamble configuration, and also provides the RACH opportunities associated with the SSBs. For example, the number of SSBs associated with a single RACH opportunity may be provided to the UE via a higher layer (e.g., RRC) parameter SSB-perRACH-Occasion. Based on the PRACH configuration for the cell, each SSB transmitted on the cell is associated with one or more RACH opportunities. The base station may provide the number of preambles for each SSB to the UE via the PRACH configuration. For example, the number of preambles for each SSB may be provided based on the value of the higher layer parameter cb-preamblePerSSB. Based on the values of SSB-perRACH-Occasion and cb-preamblePerSSB, the total number of preambles for each SSB in each RACH opportunity may be determined. The SSB index may be mapped to the RACH opportunity in the following order:

[0252] - First, in the order of increasing preamble index within a single RACH opportunity;

[0253] - Secondly, in the order of increasing frequency resource index for RACH opportunities for frequency reuse;

[0254] - Third, in order of increasing time resource index for time-multiplexed RACH opportunities within a RACH slot; or

[0255] - Fourth, in order of increasing RACH slots.

[0256] In some scenarios where SSBs are associated with different DL beams, the UE may detect one or more SSBs on the cell. The UE may select an SSB from the detected SSBs (randomly or based on reference signal received power (RSRP)). The UE may then determine the RACH timing associated with the selected SSB based on the PRACH configuration. The UE may send an RA preamble on the determined RACH timing. The BS may monitor the available RACH timings on the cell. Based on the RACH timing at which the RA preamble is received, the BS may determine which SSB the UE selects from the SSBs with different SSB indices sent on the cell when sending the RA preamble. Based on the SSB selected by the UE, the BS may determine an appropriate DL beam for the UE.

[0257] In a contention-based random access procedure, the UE selects a random access (RA) preamble. In a contention-based random access procedure, multiple UEs may simultaneously transmit the same RA preamble, requiring a subsequent contention resolution procedure. In a dedicated random access procedure, the UE uses an RA preamble uniquely assigned to the UE by the base station. This allows the UE to perform a random access procedure without colliding with other UEs.

[0258] refer to Figure 24 (a) The contention-based random access procedure includes the following four steps: The messages sent in steps 1 to 4 may be referred to as message 1 (Msg1) to message 4 (Msg4), respectively.

[0259] - Step 1: UE sends RA preamble on PRACH.

[0260] -Step 2: The UE receives the RAR from the BS on the PDSCH.

[0261] -Step 3: The UE sends UL data to the BS on the PUSCH. The UL data includes a Layer 2 (L2) / Layer 3 (L3) message.

[0262] -Step 4: The UE receives a contention resolution message from the BS on the PDSCH.

[0263] The UE may receive random access information in system information from the BS. When the UE requires random access, the UE sends Msg1 (e.g., preamble) to the BS on the PRACH. The BS may identify each RA preamble by the time / frequency resource carrying the RA preamble (hereinafter referred to as RA opportunity (RO)) and the random access preamble index (PI). Upon receiving the RA preamble from the UE, the BS sends a random access response (RAR) message to the UE on the PDSCH. In order to receive the RAR message, the UE monitors the L1 / L2 control channel (e.g., PDCCH) with a cyclic redundancy check (CRC) masked with the random access RNTI (RA-RNTI) within a preconfigured time window (e.g., ra-ResponseWindow), which includes scheduling information for the RAR message. When the scheduling information is received on the PDCCH masked with the RA-RNTI, the UE may receive the RAR message on the PDSCH indicated by the scheduling information. The UE then checks whether there is a RAR directed to the UE in the RAR message. The presence or absence of a RAR pointing to a UE can be determined by checking whether there is a random access preamble ID (RAPID) for the preamble sent by the UE. The index of the preamble sent by the UE can be the same as the RAPID. The RAR includes the index of the corresponding RA preamble, timing offset information for UL synchronization (e.g., timing advance command (TAC)), UL scheduling information for Msg3 transmission (e.g., UL grant), and UE temporary identification information (e.g., temporary C-RNTI (TC-RNTI)). Upon receiving the RAR, the UE sends Msg3 on the PUSCH based on the UL scheduling information and timing offset value in the RAR. Msg3 may include the UE's ID (or global ID). In addition, Msg3 may also include information related to the RRC connection request for initial access to the network (e.g., RRCSetupRequest message). After receiving Msg3, the base station (BS) sends a contention resolution message, i.e., Msg4, to the UE. When the UE receives the contention resolution message and successfully resolves the contention, the TC-RNTI is changed to the C-RNTI. Msg4 may include the UE's ID / RRC connection related information (e.g., RRCSetup message). When the information sent in Msg3 does not match the information received in Msg4, or when the UE has not received Msg4 within a predetermined time, the UE may determine that contention resolution has failed and retransmit Msg3. Upon successful contention resolution, the UE may transition to the RRC_CONNECTED state. When the UE is in the RRC_CONNECTED state, RRC messages may be exchanged between the UE's RRC layer and the BS's RRC layer. In other words, a UE in the RRC_CONNECTED state refers to a UE that has established an RRC connection with a base station.

[0264] The dedicated random access procedure includes the following three steps. The messages sent in steps 0 to 2 are respectively referred to as Msg0 to Msg2. The BS can trigger the dedicated random access procedure through the PDCCH serving the purpose of commanding RA preamble transmission (hereinafter referred to as PDCCH command).

[0265] -Step 0: The BS allocates an RA preamble to the UE through dedicated signaling.

[0266] - Step 1: UE sends RA preamble on PRACH.

[0267] -Step 2: The UE receives the RAR from the BS on the PDSCH.

[0268] Step 1 and step 2 of the dedicated random access procedure may be the same as step 1 and step 2 of the contention-based random access procedure.

[0269] The wireless communication system to be introduced may require lower latency than the legacy system. In particular, for latency-sensitive services such as URLLC, the 4-step random access procedure may not be preferred. A low-latency random access procedure may be required for various scenarios in wireless communication systems. When the embodiments of the present disclosure are implemented together with the random access procedure, the embodiments of the present disclosure can be performed together with the following 2-step random access procedure to reduce the latency involved in the random access procedure.

[0270] refer to Figure 24 (b) The two-step random access procedure can be performed in two steps: transmission of MsgA from the UE to the BS and transmission of MsgB from the BS to the UE. MsgA transmission may include transmission of an RA preamble on the PRACH and transmission of an UL payload on the PUSCH. In MsgA transmission, the PRACH and PUSCH may be transmitted in time division multiplexing (TDM). Alternatively, the PRACH and PUSCH may be transmitted in frequency division multiplexing (FDM) in the MsgA transmission.

[0271] Upon receiving MsgA, the BS may send MsgB to the UE. MsgB may include the RAR for the UE.

[0272] The payload of MsgA may include an RRC connection request-related message (e.g., an RRCSetupRequest message) requesting establishment of a connection between the RRC layer of the BS and the RRC layer of the UE. In this case, MsgB may be used to transmit RRC connection-related information (e.g., an RRCSetup message). In contrast, an RRC connection request-related message (e.g., an RRCSetupRequest message) may be transmitted on the PUSCH based on the UL grant in MsgB. In this case, the RRC connection-related information (e.g., an RRCSetup message) related to the RRC connection request may be transmitted on the PDSCH associated with the PUSCH transmission after the PUSCH transmission based on MsgB.

[0273] Upon successfully receiving MsgB associated with MsgA sent by the UE, the UE may transition to the RRC_CONNECTED state. When the UE is in the RRC_CONNECTED state, RRC messages may be exchanged between the UE's RRC layer and the BS's RRC layer. In other words, a UE in the RRC_CONNECTED state refers to a UE that has established an RRC connection with the BS.

[0274] A communication device on the receiving side may receive a radio frequency (RF) signal at a carrier frequency via at least one antenna. In some embodiments, the RF signal may include a signal associated with a first bit sequence having a length of N. For example, a method or operation of the communication device may include: receiving a radio signal from another communication device via a physical channel on a serving cell of the communication device; obtaining an orthogonal frequency division multiplexing (OFDM) baseband signal by performing down-conversion on the received radio signal; obtaining a complex-valued modulation symbol of the physical channel based on the OFDM baseband signal; and obtaining a first bit sequence having a length of N based on the complex-valued modulation symbol of the physical channel. The first bit sequence having a length of N may be decoded according to some embodiments of the present disclosure.

[0275] The examples of the present disclosure described above have been presented to enable those skilled in the art to implement and practice the present disclosure. Although the present disclosure has been described with reference to examples, those skilled in the art may make various modifications and variations in the examples of the present disclosure. Therefore, the present disclosure is not intended to be limited to the examples set forth herein, but rather to be accorded the widest scope consistent with the principles and features disclosed herein.

[0276] Implementations of the present disclosure may be used in a BS, a UE, or other devices in a wireless communication system.

Claims

1. A method for decoding a received signal by a communication device in a wireless communication system, the method comprising: Detecting synchronization signal blocks (SSBs) including the primary synchronization signal (PSS), secondary synchronization signal (SSS), and physical broadcast channel (PBCH); initiating a random access procedure including transmission of a physical random access channel (PRACH) based on the SSB; receiving a radio signal from another communication device via a physical channel on a serving cell of the communication device based on the random access procedure; obtaining an Orthogonal Frequency Division Multiplexing (OFDM) baseband signal by performing down-conversion on the received radio signal; Obtaining a complex-valued modulation symbol of the physical channel based on the OFDM baseband signal; Obtaining a first bit sequence having a length N based on the complex-valued modulation symbol of the physical channel, where N is an integer greater than 1; Determining P different shift parameters for the first bit sequence, where P is an integer greater than 1; generating P second bit sequences by cyclically shifting the first bit sequence based on the P different shift parameters; and The codeword is determined by performing successive cancellation decoding with a list size of L on each of the P second bit sequences, where L is an integer greater than 0, Determining the P different shift parameters for the first bit sequence includes determining a shift parameter s that satisfies the following: determining an index i of a vulnerable bit among the N bits of the first bit sequence; Grouping the operation values for the first bit sequence into a plurality of groups based on the index i of the vulnerable bit; and A predefined metric for the plurality of groups exceeds a predetermined threshold, wherein the predefined metric is obtained by summing a difference between a maximum operational value and a minimum operational value for each of the plurality of groups.

2. The method according to claim 1, wherein Grouping the operation values into the plurality of groups based on the index i of the vulnerable bit comprises: Based on the binary representation of the index i of the vulnerable bit (b_1, b_2, ..., b_(n-1)), determine to apply a boxplus operation to b_j=0 and a sum operation to b_j=1; and The indices of the operation values combined by the boxplus operation among the operation values are determined as one group.

3. The method according to claim 1, wherein The first bit sequence includes bits obtained by the other communication device by encoding information bits having a length of K using an error correction code having a size of N.

4. The method according to claim 1, further comprising: Determining a combination of candidates for the number of permutation paths and a candidate list size based on a predetermined complexity for a parallelism of N / 2; as well as determining a combination with the best performance based on the decoding performance of each of the combinations, Therein, the number of candidates for the permutation paths and the size of the candidate list for the combination with the best performance are used as P and L, respectively.

5. A communication device configured to decode a received signal in a wireless communication system, the communication device comprising: at least one transceiver; at least one processor; as well as at least one computer memory operatively connected to the at least one processor and storing instructions that, when executed, cause the at least one processor to perform operations comprising: Detecting synchronization signal blocks (SSBs) including the primary synchronization signal (PSS), secondary synchronization signal (SSS), and physical broadcast channel (PBCH); initiating a random access procedure including transmission of a physical random access channel (PRACH) based on the SSB; receiving a radio signal from another communication device via a physical channel on a serving cell of the communication device based on the random access procedure; obtaining an Orthogonal Frequency Division Multiplexing (OFDM) baseband signal by performing down-conversion on the received radio signal; Obtaining a complex-valued modulation symbol of the physical channel based on the OFDM baseband signal; Obtaining a first bit sequence having a length N based on the complex-valued modulation symbol of the physical channel, where N is an integer greater than 1; Determining P different shift parameters for the encoded bit sequence, where P is an integer greater than 1; generating P second bit sequences by cyclically shifting the encoded bit sequence based on the P different shift parameters; and The codeword is determined by performing successive cancellation decoding with a list size of L on each of the P second bit sequences, where L is an integer greater than 0, Determining the P different shift parameters for the first bit sequence includes determining a shift parameter s that satisfies the following: determining an index i of a vulnerable bit among the N bits of the first bit sequence; Grouping the operation values for the first bit sequence into a plurality of groups based on the index i of the vulnerable bit; and A predefined metric for the plurality of groups exceeds a predetermined threshold, wherein the predefined metric is obtained by summing a difference between a maximum operational value and a minimum operational value for each of the plurality of groups. The communication device according to claim 5 , wherein: Grouping the operation values into the plurality of groups based on the index i of the vulnerable bit comprises: Based on the binary representation of the index i of the vulnerable bit (b_1, b_2, ..., b_(n-1)), determine to apply a boxplus operation to b_j=0 and a sum operation to b_j=1; and The indices of the operation values combined by the boxplus operation among the operation values are determined as one group.

7. The communication device according to claim 5, wherein: The first bit sequence includes bits obtained by the other communication device by encoding information bits having a length of K using an error correction code having a size of N. The communication device according to claim 5 , wherein: The operations include: Determining a combination of candidates and candidate list size for the number of permutation paths based on a predetermined complexity for a parallelism of N / 2; and determining a combination with the best performance based on the decoding performance of each of the combinations, Therein, the number of candidates for the permutation paths and the size of the candidate list for the combination with the best performance are used as P and L, respectively.

9. A processing device in a wireless communication system, the processing device comprising: at least one processor; as well as at least one computer memory operatively connected to the at least one processor and storing instructions that, when executed, cause the at least one processor to perform operations comprising: Detecting synchronization signal blocks (SSBs) including the primary synchronization signal (PSS), secondary synchronization signal (SSS), and physical broadcast channel (PBCH); initiating a random access procedure including transmission of a physical random access channel (PRACH) based on the SSB; receiving a radio signal from another communication device via a physical channel on a serving cell of the communication device based on the random access procedure; obtaining an Orthogonal Frequency Division Multiplexing (OFDM) baseband signal by performing down-conversion on the received radio signal; Obtaining a complex-valued modulation symbol of the physical channel based on the OFDM baseband signal; Obtaining a first bit sequence having a length N based on the complex-valued modulation symbol of the physical channel, where N is an integer greater than 1; Determining P different shift parameters for the first bit sequence, where P is an integer greater than 1; generating P second bit sequences by cyclically shifting the first bit sequence based on the P different shift parameters; and The codeword is determined by performing successive cancellation decoding with a list size of L on each of the P second bit sequences, where L is an integer greater than 0, Determining the P different shift parameters for the first bit sequence includes determining a shift parameter s that satisfies the following: determining an index i of a vulnerable bit among the N bits of the first bit sequence; Grouping the operation values for the first bit sequence into a plurality of groups based on the index i of the vulnerable bit; and A predefined metric for the plurality of groups exceeds a predetermined threshold, wherein the predefined metric is obtained by summing a difference between a maximum operational value and a minimum operational value for each of the plurality of groups.

10. A computer-readable storage medium configured to store at least one program code comprising instructions that, when executed, cause at least one processor to perform operations comprising: A processing device in a wireless communication system, the processing device comprising: at least one processor; and at least one computer memory operatively connected to the at least one processor and storing instructions that, when executed, cause the at least one processor to perform operations comprising: Detecting synchronization signal blocks (SSBs) including the primary synchronization signal (PSS), secondary synchronization signal (SSS), and physical broadcast channel (PBCH); initiating a random access procedure including transmission of a physical random access channel (PRACH) based on the SSB; receiving a radio signal from another communication device via a physical channel on a serving cell of the communication device based on the random access procedure; obtaining an Orthogonal Frequency Division Multiplexing (OFDM) baseband signal by performing down-conversion on the received radio signal; Obtaining a complex-valued modulation symbol of the physical channel based on the OFDM baseband signal; Obtaining a first bit sequence having a length N based on the complex-valued modulation symbol of the physical channel, where N is an integer greater than 1; Determining P different shift parameters for the first bit sequence, where P is an integer greater than 1; generating P second bit sequences by cyclically shifting the first bit sequence based on the P different shift parameters; and The codeword is determined by performing successive cancellation decoding with a list size of L on each of the P second bit sequences, where L is an integer greater than 0, Determining the P different shift parameters for the first bit sequence includes determining a shift parameter s that satisfies the following: determining an index i of a vulnerable bit among the N bits of the first bit sequence; Grouping the operation values for the first bit sequence into a plurality of groups based on the index i of the vulnerable bit; and A predefined metric for the plurality of groups exceeds a predetermined threshold, wherein the predefined metric is obtained by summing a difference between a maximum operational value and a minimum operational value for each of the plurality of groups.

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