Method and system for designing broadband active EMI (Electro-Magnetic Interference) filter based on phasor analysis

Abstract

The invention belongs to the technical field of electromagnetic interference filter design, and discloses a broadband active EMI filter design method and system based on phasor analysis, and the method comprises the steps: carrying out the modeling of a transfer function and impedance of each link of an AEF in a target frequency band through employing an initial active parameter; calculating a phase difference theta delta between an interference current # imgabs0 # generated by a common-mode interference source and an AEF output compensation current # imgabs1 #, and calculating a phase difference theta c between a current # imgabs2 # and a current # imgabs3 # at a power grid side; and adjusting the initial active parameter, so as to reduce an absolute value delta theta delta of a difference value between theta delta and 180 degrees or reduce an absolute value delta theta c of a difference value between theta c and 90 degrees in a frequency band with an out-of-limit interference amplitude, measuring a common-mode EMI frequency spectrum after the AEF corresponding to the adjusted active parameter is inserted into the common-mode interference source, and when the common-mode EMI frequency spectrum meets an EMC standard within a preset margin, determining that the common-mode EMI frequency spectrum is not the same as the AEF. The corresponding active parameter is the target design parameter of the AEF. According to the invention, the interference suppression performance of the active EMI filter can be improved in a wide frequency band.

Description

Technical Field

[0001] The present invention belongs to the technical field of electromagnetic interference filter design, and more specifically, relates to a wideband active EMI filter design method and system based on phasor analysis. Background Art

[0002] Electromagnetic interference (EMI) is an unavoidable problem in the operation of power electronic devices. This is because pulse width modulation (PWM) methods dictate that power switches are in a high-speed on-off state, resulting in large voltage (dv / dt) and current (di / dt) transitions, forming an EMI source rich in high-frequency components. Common-mode (CM) EMI interference can cause excessive ground leakage current and exacerbate radiated EMI, ultimately leading to device operating abnormalities. Therefore, CM EMI is a key target for suppression during the design of power electronic devices. Traditional strategies for CM EMI suppression rely on inserting passive EMI filters, primarily composed of inductors and capacitors, into the interference propagation path to block or shunt the interference. However, the trend toward higher power density in power electronic equipment places even greater demands on the miniaturization and lightweighting of EMI filters.

[0003] To reduce the size and weight of EMI, active suppression methods, such as active EMI filters (AEFs), have attracted widespread attention from academia and industry. AEFs cancel EMI currents by generating a signal inversely proportional to the interference, eliminating the need for passive components. Their size and weight are significantly smaller than passive EMI filters composed of inductors and capacitors at the same power level. Furthermore, AEFs offer greater controllability, meaning their insertion loss can be precisely controlled across different frequency bands, reducing the interference spectrum to below electromagnetic compatibility (EMC) standards across a wide frequency range. In order to optimize the interference suppression performance of AEF, topology optimization, parameter refinement design and impedance matching are generally adopted. Among them: topology optimization refers to the use of a multi-stage cascade structure in the internal amplification circuit of AEF, thereby improving the effective bandwidth and loop gain of AEF; parameter refinement design refers to calculating the required insertion loss of AEF based on the interference spectrum and standard limits, so as to accurately design the parameters so that the actual insertion loss approaches the required value in the entire frequency band; impedance matching refers to adjusting the impedance on the interference source side and the grid side impedance to ensure that the AEF output current is diverted to the grid side to the maximum extent, thereby achieving effective cancellation of interference on the grid side.

[0004] AEFs designed based on existing methods can theoretically output a compensation signal that is in the opposite direction of the interference. However, the non-ideal characteristics of various links in the actual circuit can cause a phase difference between the compensation signal and the interference signal, which seriously weakens the interference compensation effect. There is an urgent need to improve AEF design methods that consider phase relationships. Summary of the Invention

[0005] In response to the above defects or improvement needs of the prior art, the present invention provides a wide-band active EMI filter design method and system based on phasor analysis, the purpose of which is to improve the interference suppression performance of the active EMI filter within a wide frequency band.

[0006] To achieve the above object, the present invention provides a method for designing a broadband active EMI filter based on phasor analysis, comprising:

[0007] Determine the initial active parameters of the active EMI filter AEF;

[0008] Using the initial active parameters, the transfer function of each link of AEF in the target frequency band and the common mode interference source impedance Z CM , grid side impedance modeling;

[0009] Based on the transfer function, the common mode interference source impedance Z CM and grid side impedance, calculate the interference current generated by the common mode interference source Compensation current with AEF output The phase difference θ Δ , and calculate the current on the grid side Compensation current with AEF output The phase difference θ c ;

[0010] Adjust the initial active parameters to reduce the phase difference θ in the frequency band where the interference amplitude exceeds the limit Δ The absolute value of the difference from 180° Δθ Δ , or reduce the phase difference θ c The absolute value of the difference from 90° Δθ c , measure the common-mode EMI spectrum after the AEF corresponding to the adjusted active parameters is inserted into the common-mode interference source. Within the target frequency band, when the common-mode EMI spectrum meets the EMC standard within a preset margin, the corresponding active parameters are the target design parameters of the AEF.

[0011] Furthermore, the AEF links include sampling link, signal amplification link and signal compensation link; the initial active parameters include: the primary-to-secondary turns ratio n of the current transformer CT used in the sampling link, the feedback resistor R of the inverting amplifier used in the signal amplification link f and the ratio of the input resistance R1f / R1, compensation resistor R in the signal compensation link comp With compensation capacitor C comp .

[0012] Furthermore, the initial active parameters are adjusted to reduce the phase difference θ in the frequency band where the interference amplitude exceeds the limit. Δ The absolute value of the difference from 180° Δθ Δ , or reduce the phase difference θ c The absolute value of the difference from 90° Δθ c ,include:

[0013] Calculate the low corner frequency f of the CT transfer function in the frequency domain L-CT and high corner frequency f H-CT ;Frequency f within the target frequency band <f L-CT When f>f H-CT When , reduce the CT primary-to-secondary turns ratio n to reduce Δθ Δ ;f L-CT ≤f≤f H-CT When f and θ c The turning frequency f of crossing 90° c-θc If the relationship between <f c-θc , reduce the CT primary-to-secondary turns ratio n to reduce Δθ c ; If f≥f c-θc , then increase the CT primary-to-secondary turns ratio n to reduce Δθ c ;

[0014] Calculate the corner frequency f of the closed-loop gain of the inverting amplifier c-op ; f≥f c-op When R f / R1, to reduce Δθ Δ ;f <f c-op When f and f c-θc If the relationship between <f c-θc , reduce R f / R1, to reduce Δθ c ; If f≥f c-θc , then increase R f / R1, to reduce Δθ c ;

[0015] Calculate the compensation resistance R comp With the compensation capacitor C comp The corner frequency f of the series equivalent impedance c-comp ;f <f c-comp When the compensation capacitor C comp , to reduce Δθ Δ ; f≥f c-comp When f and fc-θc If the relationship between <f c-θc , increase the compensation resistor R comp to reduce the value of Δθ c ; If f≥f c-θc , then reduce the compensation resistance R comp to reduce the value of Δθ c .

[0016] Furthermore, the low corner frequency f L-CT and high corner frequency f H-CT The calculation method is:

[0017]

[0018] Among them, L m1 is the equivalent inductance of CT, R c is the equivalent resistance that characterizes the loss, R sense is the sensing resistor; C pp is the parasitic capacitance of the primary side of CT;

[0019] The turning frequency f c-op With R f / R1 satisfies:

[0020]

[0021] Among them, G op is the open-loop gain of the operational amplifier in the inverting amplifier, and f1 and f2 are two open-loop corner frequencies of the operational amplifier respectively;

[0022] The turning frequency f c-comp The calculation method is:

[0023] Furthermore, the phase difference θ Δ and the phase difference θ c The calculation method is:

[0024]

[0025] Among them, A is Amplitude, Represents the compensation current output by AEF and interference current The quotient of Δ ; G CT is the transfer function of CT, G A is the transfer function of the inverting amplifier; || represents the parallel symbol; Z LISN is the impedance on the grid side, Z L is the equivalent impedance of an external magnetic ring, which is an empirical value; Zout It is the output impedance of the push-pull amplifier used in the signal amplification link.

[0026] Furthermore, the transfer function G of CT CT and the transfer function of the inverting amplifier G A The calculation method is:

[0027]

[0028] Among them, s is the Laplace operator, R c is the equivalent resistance that characterizes the loss, R sense is the sensing resistor; C pp is the parasitic capacitance of the primary side of CT, L m is the excitation inductance; Z o is the operational amplifier output impedance in the inverting amplifier, Z in It is the input impedance of the push-pull amplifier used in the signal amplification link.

[0029] The present invention also provides a broadband active EMI filter obtained by the broadband active EMI filter design method based on phasor analysis as described in any one of the above.

[0030] The present invention also provides a wideband active EMI filter design system based on phasor analysis, comprising a computer-readable storage medium and a processor;

[0031] The computer-readable storage medium is used to store executable instructions;

[0032] The processor is configured to read the executable instructions stored in the computer-readable storage medium to execute any one of the above-mentioned methods for designing a wide-band active EMI filter based on phasor analysis.

[0033] The present invention also provides a computer-readable storage medium having a computer program stored thereon, wherein when the program is executed by a processor, the method for designing a wideband active EMI filter based on phasor analysis as described above is implemented.

[0034] The present invention also provides a computer program product, comprising a computer program, which, when executed on a computer, enables the computer to execute any of the above-mentioned methods for designing a wideband active EMI filter based on phasor analysis.

[0035] In general, the above technical solutions conceived by the present invention can achieve the following beneficial effects:

[0036] (1) The present invention takes into account the non-ideal characteristics of the AEF internal circuit, the interference current Compensation current with AEF output Phase difference θΔ Usually deviates from 180°, at this time, AEF outputs compensation current Neutralizing interference current The component with a phase difference of 90° not only does not contribute to the interference compensation, but also amplifies the interference. Therefore, the compensation current output by AEF is After phase shift occurs, the interference compensation performance of AEF will be seriously affected. Through further analysis, it is found that the factors that determine the interference compensation effect are not only and The phase difference θ Δ , and AEF output compensation current The amplitude I c size; when θ Δ Fixed, I c When the current on the grid side changes and The phase difference θ c When the angle is 90°, the current amplitude on the grid side is I r Based on the above analysis, the optimization goal of the constructed AEF is to reduce and Phase difference θ Δ The absolute value of the difference from 180° Δθ Δ ,as well as and The phase difference θ c The absolute value of the difference from 90° Δθ c By adjusting the active parameters of AEF, Δθ is reduced in the frequency band where the interference amplitude exceeds the limit. Δ or Δθ c , so that the CM EMI spectrum of the AEF corresponding to the adjusted active parameters, after being inserted into the target power electronic device, meets EMC standards within the target frequency band. The resulting AEF parameters are the required active EMI filter parameters, and the active EMI filter derived from these AEF parameters also meets the requirements. This invention addresses the problem of traditional common-mode EMI filter design methods in power electronic devices ignoring the phase relationship between the compensation current and the interference current, thereby fully optimizing the filter's interference suppression performance across a wide frequency band.

[0037] (2) Furthermore, the present invention simplifies the complex broadband optimization problem into the calculation of several cutoff frequencies, establishing a mapping relationship between key AEF parameters and the interference suppression performance of each frequency band. This has important guiding significance for optimizing the broadband performance of the AEF. The method of the present invention is highly feasible in actual engineering practice. In addition, the proposed design method does not require additional software or hardware and is highly transferable.

[0038] In summary, the present invention takes into account the non-ideal characteristics of the internal links of the AEF, the compensation current and the interference current are not necessarily in an anti-phase relationship, but an additional phase offset is introduced, and this characteristic will change with frequency, resulting in the interference suppression performance of the AEF in a wide frequency band (10kHz to 10MHz range) being seriously affected. To solve this problem, the present invention proposes an AEF optimization design method based on the phase relationship between the compensation current and the interference current, breaking through the limitation of the traditional AEF design scheme that uses insertion loss as the single evaluation factor without considering phase information. Based on the cutoff frequency where the phase response of each link turns, the influence mechanism of each key parameter of the AEF on its attenuation is revealed, which can help the AEF to optimize the interference suppression performance in the entire frequency band. BRIEF DESCRIPTION OF THE DRAWINGS

[0039] Figure 1 Schematic diagram of the common mode path of the inverter in an embodiment of the present invention;

[0040] Figure 2 Schematic diagram of the principle of AEF in an embodiment of the present invention;

[0041] Figure 3 : This is an equivalent circuit diagram of the common mode path after AEF is adopted in an embodiment of the present invention;

[0042] Figure 4 This is a circuit diagram of the internal structure of the AEF in an embodiment of the present invention;

[0043] Figure 5 This is a comparison chart of the frequency response modeling results of key links in the embodiment of the present invention and the actual measurement results of the network analyzer; wherein, Figure 5 (a) is the frequency response of the current transformer CT; Figure 5 (b) is the frequency response of the signal amplification link; Figure 5 (c) in the figure is the output impedance of the operational amplifier and the push-pull amplifier;

[0044] Figure 6 1 is a simplified equivalent circuit diagram of a common-mode path including an AEF in an embodiment of the present invention;

[0045] Figure 7 is the phase relationship among the AEF compensation current, the common-mode interference current and the compensated current in the embodiment of the present invention; wherein, Figure 7 (a) in the figure means that no phase shift is introduced, i.e., θ Δ =180°; Figure 7 (b) in the figure introduces a phase shift, i.e., θ Δ ≠180°

[0046] Figure 8: is the time domain waveform of the AEF compensation current, common mode interference current and compensated current in the embodiment of the present invention; wherein, Figure 8 (a) in the equation is θ Δ =180°, the interference cancellation effect is significant; Figure 8 (b) in the equation is θ Δ When the angle is ≠180°, the interference cancellation effect is greatly weakened;

[0047] Figure 9 is θ in the embodiment of the present invention Δ ≠180° when the current phase relationship; Figure 9 (a) in the equation is θ Δ The situation when the compensation current amplitude changes; Figure 9 (b) in the figure shows that the compensation current amplitude remains unchanged, while θ Δ the situation when it changes;

[0048] Figure 10 is θ in the embodiment of the present invention Δ and θ c Calculated values in the frequency range of 10kHz to 10MHz;

[0049] Figure 11 Schematic diagram of the frequency response of the key links of AEF in an embodiment of the present invention; wherein, Figure 11 (a) is the frequency response of CT; Figure 11 (b) shows the frequency response of the open-loop gain and closed-loop gain of the inverting amplifier; Figure 11 (c) in the figure is the impedance of the compensation link;

[0050] Figure 12 The amplitude-frequency characteristics of the CT and the inverting amplifier and the impedance characteristics of the compensation link in the embodiment of the present invention are shown;

[0051] Figure 13 Flowchart of the AEF design method based on phasor analysis in an embodiment of the present invention;

[0052] Figure 14 Schematic diagram of an AEF design method based on phasor analysis in an embodiment of the present invention;

[0053] Figure 15 These are two measured EMI spectra that did not exceed the standard in the embodiment of the present invention;

[0054] Figure 16 The EMI spectrum measured after optimizing the AEF parameters in the embodiment of the present invention is compared with the spectrum of the working condition that does not meet the standard. DETAILED DESCRIPTION

[0055] In order to make the objectives, technical solutions and advantages of the present invention more clearly understood, the present invention is further described in detail below with reference to the accompanying drawings and embodiments. It should be understood that the specific embodiments described herein are only intended to illustrate the present invention and are not intended to limit the present invention. In addition, the technical features involved in the various embodiments of the present invention described below may be combined with each other as long as they do not conflict with each other.

[0056] Example 1

[0057] In the embodiment of the present invention, the power electronic device selects an inverter and analyzes the common mode path of the inverter.

[0058] The common mode interference path of a typical inverter is as follows: Figure 1 As shown in the figure, the high-speed on-off of the power switch causes a large voltage jump dv / dt at the midpoint of the bridge arm, and the parasitic capacitance C between the power module and the load to the heat sink is p The induced common mode current i CM , as shown in formula (1). Then, i CM The power then returns to the line impedance stabilization network (LISN) through the device's protective ground, and ultimately to the DC bus (grid). The LISN isolates interference introduced by the DC power supply and simulates grid-side impedance, providing a reference for measuring common-mode interference in the target device (power electronic device).

[0059]

[0060] In order to compensate for the common-mode EMI current, AEF needs to sample the EMI current in the interference loop, generate the corresponding compensation current through the signal processing circuit, and finally inject it back into the interference loop to achieve EMI current compensation. The principle is as follows Figure 2 As shown in the figure. For the sampling stage, current-sampling AEFs typically use a current transformer (CT) to collect common-mode current and convert it into a voltage signal. For the signal amplification stage, an inverting amplifier typically performs signal inversion and voltage amplification, while a push-pull amplifier performs power amplification. The compensation stage is typically an RC branch that isolates the AEF's internal circuit from the DC power loop and injects the amplified compensation signal (the output current of the signal amplification stage) into the interference loop, i.e., the power grid, to offset common-mode electromagnetic interference. Because the AEF constructs a loop from the converter to the PE, the compensation capacitor should be considered a Y capacitor, and its value should not exceed the Y capacitor capacitance limit set by the leakage current standard. Figure 3 The common-mode loop circuit after AEF insertion is shown.

[0061] Figure 4The equivalent circuit diagrams of CT, inverting amplifier and push-pull amplifier are shown. First, CT converts the common mode current i CM Converted into the input voltage V of the signal amplification link in , where C pp with C sp are the parasitic capacitances of the primary and secondary sides of CT, C sp =n 2 C pp , L m is the magnetizing inductance, R c To characterize the equivalent resistance of the loss, the transfer function of CT G CT It can be expressed by formula (2), I CM is the output current of the CT, which is also the input current of the signal amplifier, n is the primary-to-secondary turns ratio of the CT, s is the Laplace operator, R sense is the sensing resistor. Then, the inverting amplifier circuit uses an operational amplifier to convert the input voltage V in Invert and amplify to V out , after considering the non-ideal characteristics of the op amp, the transfer function G of the inverting amplifier circuit is A It can be expressed as formula (3). Where, V out is the output voltage of the inverting amplifier circuit, R f and R1 are feedback resistor and input resistor respectively; G op is the open-loop gain of the operational amplifier, which can be found in the technical manual, Z o is the operational amplifier output impedance, Z in is the input impedance of the push-pull amplifier. The main function of the push-pull amplifier is to keep the voltage constant (V AEF ≈V op ), constructing a smaller output impedance Z out , which is expressed as formula (5), thereby improving the AEF load capacity. E is the transistor emitter resistance, Z ZoCBC Represents Z o , base resistor R B and the base-collector capacitance C BC The parallel impedance, Z BE Represents the base-emitter resistance R BE With capacitor C BE The parallel impedance, r o is the transistor output impedance, and β is the transistor gain. Figure 5 The modeling results of the above three links are shown respectively and compared with the measured results. It can be seen that the model established by the embodiment of the present invention has a high degree of accuracy in the 10kHz-10MHz frequency band (target frequency band) of interest. At the same time, it can be seen that the output impedance Z out Less than the operational amplifier impedance Z o.

[0062]

[0063] In the embodiment of the present invention, the AEF compensation current and the interference current are analyzed in phasor. After the equivalent model of each link of AEF is established, Figure 3 Simplified to Figure 6 The circuit shown; where V CM is the common mode interference voltage source, Z CM is the common mode interference source impedance, both of which are obtained by measurement; Z comp is the compensation resistor R comp With compensation capacitor C comp The series equivalent impedance, which serves as the transfer function of the compensation link; Z out is the output impedance of the push-pull amplifier circuit, calculated using formula (4); V AEF is the output voltage of AEF; Z LISN is the equivalent impedance of LISN, which is the impedance of the grid side in actual application scenarios and is obtained by measurement; Z L The equivalent impedance of an external magnetic ring (empirical value) is used to provide a high impedance circuit at high frequencies to compensate for the shortcomings of AEF in high frequency performance. According to the principle of AEF, the interference current (That is, the common mode current i CM ) is sampled as the input of AEF, and then AEF outputs the corresponding compensation current Finally, the compensated current on the LISN side is achieved The amplitude is the smallest. Theoretically, and The phase difference θ Δ It is determined only by the inverting amplifier, that is, θ Δ =180°, so All components are used for The offset, such as Figure 7 (a) in the equation (5) is shown. However, due to the additional phase shift introduced by the non-ideal characteristics of the AEF internal circuit, θ Δ Usually deviates by 180°, such as Figure 7 (b) in the equation (6) is shown; at this time, the compensation current corresponding to the AEF output can be decomposed into two orthogonal components and It is the component that plays the role of interference compensation. Its current direction (phase) is the same as the interference current. On the contrary, it can play a role in interference compensation; Phase and interference current The 90° difference not only does not contribute to interference compensation, but also amplifies the interference. Therefore, the compensation current corresponding to the AEF output When phase offset occurs, the interference compensation performance of AEF will be seriously affected. Figure 8 The time domain waveform further shows the consequences of phase offset: Figure 8 (a) shows θ Δ When the compensation current is 180°, the compensation current and the interference current are nearly equal and opposite in phase, so the current after compensation (the current on the LISN side) ) The amplitude is very small and the AEF suppression performance is excellent; Figure 8 (b) shows θ Δ At a 180° deviation, the current amplitudes before and after compensation are similar, and the AEF barely compensates for the interference. Therefore, performing phasor analysis of the AEF compensation current and the interference current is crucial for optimizing the AEF's interference compensation capabilities.

[0064]

[0065] From formula (6), we can see that the interference compensation effect is determined by Amplitude I r Not only the size and The phase difference θ Δ , and the compensation current corresponding to the AEF output The amplitude I c . Figure 9 The influence of these two factors is analyzed separately. Δ Fixed, I c When changes occur (I c by I c1 Gradually increases to I c3 ),like Figure 9 As shown in (a); Obviously, when and The phase difference θ c When it is 90°, I r Get the minimum value. When I c Fixed, θ Δ When changes occur, such as Figure 9 As shown in (b), when θ Δ When it is equal to 180°, I r Get the minimum value. Figure 6 Based on Kirchhoff's circuit law, the circuit shown in Figure 7 can be used to calculate θ Δ ;in, Represents the compensation current output by AEF and interference current The quotient ofΔ ;|| represents the circuit parallel symbol. At the same time, based on the sine theorem, θ can be calculated c , as shown in formula (8). Figure 10 Shows θ Δ and θ c Calculated value in the 10kHz-10MHz frequency band; where f c-θc represents θ c Crossing the 90° corner frequency, which means that at f c-θc Before, due to θ c Less than 90°, I needs to be reduced c So that θ c Close to 90°; and at f c-θc Afterwards, due to θ c Less than 90°, I needs to be increased c So that θ c Close to 90°. To simplify the expression, Equation (9) defines Δθ Δ and Δθ c , are the absolute values of the differences between the two and 180° and 90° respectively; therefore, according to phasor analysis, the optimization goal of AEF is to minimize Δθ Δ or Δθ c (equivalent to the current on the LISN side The amplitude of the filter is the smallest), and the AEF parameters designed based on the optimization target are the required active EMI filter parameters. In the embodiment of the present invention, f c-θc It can be directly based on θ c The value of is read, for example, according to Figure 10 As shown, f c-θc represents θ c The frequency of the 90° transition can be read as f c-θc =10 6 Hz; or, according to formula (8), let θ c =90°, the calculated frequency is f c-θc .

[0066]

[0067] Furthermore, in the embodiment of the present invention, the design method of specific AEF parameters is described based on the analysis of the internal electrical characteristics of the AEF. Δ and Δθ c , it is necessary to conduct an in-depth analysis of the frequency characteristics of the internal links of AEF.

[0068] For the sampling link, based on formula (2), formula (10) uses two corner frequencies f L-CT With f H-CT The CT transfer function is divided into three parts in the frequency domain, where fL-CT and are the low corner frequency and high corner frequency of the CT transfer function in the frequency domain, respectively. m1 is the equivalent inductance of CT, like Figure 11 As shown in (a), and f L-CT With f H-CT It can be calculated by formula (11). When the frequency f in the target frequency band <f L-CT When θ is θ, CT shows inductive characteristics, so CT introduces an advanced phase shift, that is, Δ ,θ c Contributes, and the gain G at this time CT-lowfre (f <f L-CT The gain of CT increases with the increase of frequency; when f L-CT <f<f H-CT When CT shows resistive characteristics, the gain G CT-ideal It does not change with frequency and does not introduce phase shift. H-CT When CT shows capacitive characteristics, it will introduce a lagging phase shift, and the gain G CT-highfre Ideally, when designing AEF, we want the resistive region of CT to be as wide as possible, i.e., f L-CT As small as possible, while f H-CT As large as possible; however, combining equations (10) and (11), it can be seen that the width of the resistive region is contradictory to the gain. For example, reducing the CT primary-to-secondary turns ratio n can reduce f L-CT And increase f H-CT , that is, widening the width of the resistive region, which will lead to G CT-ideal In other words, the change of n in each frequency band has an impact on Δθ Δ and Δθ c The impact caused is different and needs to be balanced according to the actual spectrum distribution of the interference. Specifically, when CT introduces a phase offset, that is, the frequency f in the target frequency band <f L-CT When f>f H-CT When increasing the primary-to-secondary turns ratio n of CT, Δθ will increase Δ On the contrary, reducing the primary-to-secondary turns ratio n of the CT will reduce Δθ Δ When CT does not introduce phase shift, that is, f L-CT ≤f≤f H-CT When n increases, the gain will also increase. At this time, it is necessary to judge the relationship between f and θ c The turning frequency f of crossing 90° c-θc The relationship between <f c-θc When the gain increases, Δθ c Increase, that is, f <f c-θcWhen increasing the primary-to-secondary turns ratio n of CT, Δθ will increase c On the contrary, reducing the primary-to-secondary turns ratio n of the CT will reduce Δθ c ; f≥f c-θc When the gain increases, Δθ c Reduce, that is, f≥f c-θc When increasing the primary-to-secondary turns ratio n of CT, Δθ will be reduced. c On the contrary, reducing the primary-to-secondary turns ratio n of the CT will increase Δθ c CT primary-to-secondary turns ratio n versus Δθ Δ or Δθ c The specific impact is shown in Table 1 below.

[0069]

[0070] For an inverting amplifier, Equation (12) shows the expression of its gain, where G op , f1 and f2 are the open-loop gain and two open-loop corner frequencies of the operational amplifier, respectively. The open-loop gain and open-loop corner frequency can be obtained from the technical manual. c-op is the corner frequency corresponding to the closed-loop gain of the inverting amplifier. Therefore, the frequency response of the inverting amplifier can be expressed as Figure 11 (b) in the figure indicates that G op is the open-loop gain of the operational amplifier, G close is the closed-loop gain of the operational amplifier, and the feedback resistor R f and the ratio of the input resistance R1 is equal. <f c-op When the inverting amplifier has a fixed closed-loop gain R f / R1, and the inverting amplifier does not introduce additional phase shift; when f ≥ f c-op When the inverting amplifier gain is equal to the open-loop gain G op , at this time, the gain decreases with increasing frequency, and additional phase shift will be introduced. The intersection of the open-loop gain and the horizontal axis is called the gain bandwidth product (GBW), which is determined by the properties of the operational amplifier itself. Therefore, when designing AEF, it is hoped that the operational amplifier used has as high a GBW as possible. In summary, once the operational amplifier is determined, the parameter that determines the frequency response of the inverting amplifier is R f / R1:R f The larger the R1, the higher the closed-loop gain, but f c-op The smaller it is, the narrower the effective bandwidth is; R f The smaller R1 is, the lower the closed-loop gain is, but f c-op The higher it is, the wider the effective bandwidth is. In summary, R fThe effect of / R1 on the inverting amplifier is similar to the effect of n on CT. The adjustment of effective bandwidth and gain is contradictory and needs to be adjusted in combination with the actual interference characteristics. Specifically, when the inverting amplifier introduces a phase offset, that is, f≥f c-op When R f / R1, will increase Δθ Δ , on the contrary, reduce R f / R1, it will reduce Δθ Δ ; When the inverting amplifier does not introduce a phase shift, that is, f <f c-op When R f As R1 increases, the gain will also increase. At this time, it is necessary to judge the relationship between f and θ. c The turning frequency f of crossing 90° c-θc The relationship between <f c-θc When the gain increases, Δθ c Increase, that is, f <f c-θc When R f / R1, will increase Δθ c , on the contrary, reduce R f / R1, it will reduce Δθ c ; f≥f c-θc When the gain increases, Δθ c Reduce, that is, f≥f c-θc When R f / R1, will reduce Δθ c , on the contrary, reduce R f / R1, it will increase Δθ c . R f / R1 vs. Δθ Δ or Δθ c The specific impact is shown in Table 1 below.

[0071]

[0072] For the output compensation link, equation (13) expresses its turning frequency f c-comp ; Obviously, for an RC series branch, in the low frequency band, that is, f <f c-comp When the overall capacitance is capacitive, the compensation capacitor will introduce a 90° lag phase shift, and the impedance decreases with increasing frequency; in the high frequency band, that is, f≥f c-comp When , the capacitance impedance can be ignored and the whole is resistive, so no phase shift is introduced, and the impedance value is equal to the compensation resistor R comp ,like Figure 11 As shown in (c) in the figure. Ideally, we hope that f c-comp The smaller the better, that is, the compensation resistor R comp With compensation capacitor C compThe larger the product, the better; however, due to the leakage current limit, the compensation capacitor has a maximum value, and R comp The increase in the AEF resistance is equivalent to the increase in the internal resistance of the AEF, which will lead to the increase of f c-comp The compensation current decreases in the subsequent frequency band. In summary, R comp with C comp The effects on the gain and effective bandwidth of the compensation circuit are also contradictory. Specifically, when the compensation capacitor introduces a phase shift, that is, f <f c-comp When increasing the compensation capacitor C comp , which will reduce Δθ Δ , on the contrary, reduce the compensation capacitance C comp , it will increase Δθ Δ When the compensation capacitor does not introduce phase shift, that is, f≥f c-comp When the compensation resistor R comp The increase of f will lead to a decrease in gain. At this time, it is necessary to judge the relationship between f and θ c The turning frequency f of crossing 90° c-θc The relationship between <f c-θc When increasing the compensation resistor R comp The value of Δθ will reduce c , on the contrary, reduce the compensation resistance R comp The value of , will increase Δθ c ; f≥f c-θc When increasing the compensation resistor R comp The value of Δθ will increase c , on the contrary, reduce the compensation resistance R comp The value of , will reduce Δθ c . R comp with C comp For Δθ Δ or Δθ c The specific impact is shown in Table 1 below.

[0073]

[0074] Table 1 Key parameters for Δθ Δ or Δθ c The influence of

[0075]

[0076] In Table 1, ↑: positive correlation with parameter change; ↓: negative correlation with parameter change; --: no direct correlation.

[0077] Figure 12 The frequency response and impedance of the three links of sampling, amplification and compensation are combined. It can be seen that the phase shift inside the AEF is affected by the three links at the same time, and the influence can be characterized by each cutoff frequency. For example, when f c-op>f>f H-CT When , only CT introduces phase shift, while the inverting amplifier and compensation link are in the resistive working area and will not introduce phase shift. When the key parameters corresponding to each link change, the gain and phase shift θ of each frequency band Δ will change accordingly, which also needs to be based on θ c The value of crossing 90° is used to determine f c-θc , thus characterizing the effect of gain change on Δθ c In addition, these cutoff frequencies will also change with the changes in parameters, so they need to be recalculated after each parameter adjustment.

[0078] like Figure 13 As shown in the figure, the AEF design process based on phasor analysis in the embodiment of the present invention is shown in the figure. First, the initial active parameters of the AEF are determined based on empirical values, including: the primary-to-secondary turns ratio n of the CT in the sampling circuit, the compensation resistor R in the signal compensation circuit, and the initial active parameters of the AEF. comp With compensation capacitor C comp , the feedback resistor R of the inverting amplifier in the signal amplification circuit f and the ratio of the input resistance R1 f / R1;

[0079] Based on the determined initial active parameters, the transfer function of the AEF internal link in the target frequency band is modeled and the converter impedance Z is measured. CM , grid side impedance; related calculation methods refer to formula (2) to formula (4).

[0080] Then, the CM EMI spectrum after inserting the AEF into the target power electronic device is measured. If the EMC standard is met, the initial active parameters are directly used as the parameter results of the AEF; if the standard is not met, the CM EMI spectrum is calculated separately. and The phase difference θ Δ ,and and The phase difference θ c , the relevant calculation method is shown in formula (7) and formula (8); and calculate each cutoff frequency: f c-comp 、f L-CT 、f c-θc 、f H-CT 、f c-op .

[0081] According to each cutoff frequency, the target frequency band is divided into multiple sub-intervals; for the sub-interval frequency band where the interference amplitude exceeds the limit, refer to the rules in Table 1 and adjust the active parameters to reduce Δθ in the sub-interval frequency band where the interference amplitude exceeds the limit. Δ or Δθ c, then measure the interference spectrum again, and repeat this process until the CM EMI spectrum measured in the target frequency band under the current AEF active parameters meets the standard limits.

[0082] Figure 15 The two working conditions of interference exceeding the limit and the corresponding parameter influence rules are shown. Table 2 shows the various cutoff frequencies corresponding to the two working conditions. Figure 15 In (a), the interference exceeds the limit in the Ie band (i.e., it does not exceed the standard and does not meet the EMC standard); since there is no sufficient margin for the interference distance standard limit in the Id band (the empirical value of the margin for the interference distance standard limit is 6db), this band should also be optimized at the same time. According to the parameter influence law, it can be adopted to increase R f / R1 and reduce R comp Ways to reduce Δθ c Even though this method will result in Δθ in the three frequency bands Ia, Ib and Ic c However, there is sufficient margin for interference in these three frequency bands, so a certain degree of AEF performance degradation is acceptable. Figure 15 In (b), the interference exceeds the limit in the II-a frequency band. An effective method is to increase C comp To reduce the Δθ in this frequency band Δ ; In addition, R comp Reduce Δθ in the II-e band c , thereby improving the margin within this frequency band. Figure 16 The following table shows the operating conditions before and after parameter optimization using the proposed AEF design method. After optimization, the interference in both frequency bands that had previously exceeded the standard limit is now below the standard limit, and attenuation is improved by up to 10dB. Table 3 compares the key AEF parameters corresponding to these operating conditions.

[0083] Table 2 Cut-off frequencies of two working conditions

[0084]

[0085] Table 3 Key parameter values of AEF

[0086]

[0087] In summary, the AEF design method based on phasor analysis takes into account the phase relationship between the compensation current and the interference current, reveals the influence mechanism of key parameters in different frequency bands on the AEF interference suppression performance, and has a high reference value for parameter optimization design within a wide frequency band.

[0088] In an embodiment of the present invention, an AEF design method based on phasor analysis of compensation current and interference current is proposed to identify the dominant factors and influencing mechanisms of attenuation in each frequency band. First, by combining the construction of a transfer function model with actual measurement using a network analyzer, the frequency response and impedance of each link in the common-mode interference loop are obtained, including the impedance characteristics of the inverter and line impedance stabilization network and the transfer function of the internal links of the AEF. Subsequently, based on the equivalent circuit of the common-mode interference loop and the characteristics of the AEF control loop, the AEF output compensation current can be calculated. and interference current The optimization of the interference suppression performance of the AEF can ultimately be attributed to the adjustment of the phase difference between the currents. There is a mapping relationship between the phase angle and the active control parameters within the AEF. Therefore, based on this relationship, the phase angle between the currents can be adjusted by optimizing the AEF parameters, thereby achieving targeted improvements in the interference suppression performance of the AEF in different frequency bands.

[0089] Example 2

[0090] The embodiment of the present invention provides a broadband active EMI filter obtained by the broadband active EMI filter design method based on phasor analysis as described in the first embodiment above.

[0091] The relevant technical solutions are the same as above and will not be repeated here.

[0092] Example 3

[0093] An embodiment of the present invention provides a wideband active EMI filter design system based on phasor analysis, including a memory and a processor. The memory stores a computer program, and when the processor executes the computer program, the steps of the wideband active EMI filter design method based on phasor analysis in the above-mentioned embodiment 1 are implemented.

[0094] The relevant technical solutions are the same as above and will not be repeated here.

[0095] Example 4

[0096] An embodiment of the present invention provides a computer-readable storage medium having a computer program stored thereon. When the computer program is executed by a processor, the steps of the method for designing a wideband active EMI filter based on phasor analysis in the above-mentioned embodiment 1 are implemented.

[0097] Specifically, the memory may include a high-speed random access memory, and may also include a non-volatile memory, such as a hard disk, a memory, a plug-in hard disk, a smart memory card (Smart Media Card, SMC), a secure digital (Secure Digital, SD) card, a flash card (Flash Card), at least one disk storage device, a flash memory device, or other volatile solid-state storage device.

[0098] The relevant technical solutions are the same as above and will not be repeated here.

[0099] Example 5

[0100] An embodiment of the present application provides a computer program product, including a computer program. When the computer program is run on a computer, the computer executes the steps of the wideband active EMI filter design method based on phasor analysis in the above-mentioned embodiment 1.

[0101] The relevant technical solutions are the same as above and will not be repeated here.

[0102] It will be easily understood by those skilled in the art that the above description is merely a preferred embodiment of the present invention and is not intended to limit the present invention. Any modifications, equivalent substitutions, and improvements made within the spirit and principles of the present invention should be included in the scope of protection of the present invention.

Claims

1. A broadband active EMI filter design method based on phasor analysis, characterized in that: include: Determine the initial active parameters of the active EMI filter AEF; Using the initial active parameters, the transfer function of each link of AEF in the target frequency band and the common mode interference source impedance Z CM , grid side impedance modeling; Based on the transfer function, the common mode interference source impedance Z CM and grid side impedance, calculate the interference current generated by the common mode interference source Compensation current with AEF output The phase difference θ Δ , and calculate the current on the grid side Compensation current with AEF output The phase difference θ c ; Adjust the initial active parameters to reduce the phase difference θ in the frequency band where the interference amplitude exceeds the limit Δ The absolute value of the difference from 180° Δθ Δ , or reduce the phase difference θ c The absolute value of the difference from 90° Δθ c , measure the common-mode EMI spectrum after the AEF corresponding to the adjusted active parameters is inserted into the common-mode interference source. Within the target frequency band, when the common-mode EMI spectrum meets the EMC standard within a preset margin, the corresponding active parameters are the target design parameters of the AEF.

2. The method for designing a wideband active EMI filter based on phasor analysis according to claim 1, wherein: The AEF links include sampling link, signal amplification link and signal compensation link; the initial active parameters include: the primary and secondary turns ratio n of the current transformer CT used in the sampling link, the feedback resistor R of the inverting amplifier used in the signal amplification link f and the ratio of the input resistance R1 f / R1, compensation resistor R in the signal compensation link comp With compensation capacitor C comp .

3. The method for designing a wideband active EMI filter based on phasor analysis according to claim 2, wherein: Adjust the initial active parameters to reduce the phase difference θ in the frequency band where the interference amplitude exceeds the limit Δ The absolute value of the difference from 180° Δθ Δ , or reduce the phase difference θ c The absolute value of the difference from 90° Δθ c ,include: Calculate the low corner frequency f of the CT transfer function in the frequency domain L-CT and high corner frequency f H-CT ;Frequency f within the target frequency band <f L-CT When f>f H-CT When , reduce the CT primary-to-secondary turns ratio n to reduce Δθ Δ ;f L-CT ≤f≤f H-CT When f and θ c The turning frequency f of crossing 90° c-θc If the relationship between <f c-θc , reduce the CT primary-to-secondary turns ratio n to reduce Δθ c ; If f≥f c-θc , then increase the CT primary-to-secondary turns ratio n to reduce Δθ c ; Calculate the corner frequency f of the closed-loop gain of the inverting amplifier c-op ; f≥f c-op When R f / R1, to reduce Δθ Δ ;f <f c-op When f and f c-θc If the relationship between <f c-θc , reduce R f / R1, to reduce Δθ c ; If f≥f c-θc , then increase R f / R1, to reduce Δθ c ; Calculate the compensation resistance R comp With the compensation capacitor C comp The corner frequency f of the series equivalent impedance c-comp ;f <f c-comp When the compensation capacitor C comp , to reduce Δθ Δ ; f≥f c-comp When f and f c-θc If the relationship between <f c-θc , increase the compensation resistor R comp to reduce the value of Δθ c ; If f≥f c-θc , then reduce the compensation resistance R comp to reduce the value of Δθ c .

4. The method for designing a wideband active EMI filter based on phasor analysis according to claim 3, wherein: The low corner frequency f L-CT and high corner frequency f H-CT The calculation method is: Among them, L m1 is the equivalent inductance of CT, R c is the equivalent resistance that characterizes the loss, R sense is the sensing resistor; C pp is the parasitic capacitance of the primary side of CT; The turning frequency f c-op With R f / R1 satisfies: Among them, G op is the open-loop gain of the operational amplifier in the inverting amplifier, and f1 and f2 are two open-loop corner frequencies of the operational amplifier respectively; The turning frequency f c-comp The calculation method is:

5. The method for designing a wideband active EMI filter based on phasor analysis according to any one of claims 2 to 4, characterized in that: The phase difference θ Δ and the phase difference θ c The calculation method is: Among them, A is Amplitude, Represents the compensation current output by AEF and interference current The quotient of Δ ; G CT is the transfer function of CT, G A is the transfer function of the inverting amplifier; || represents the parallel symbol; Z LISN is the impedance on the grid side, Z L is the equivalent impedance of an external magnetic ring, which is an empirical value; Z out It is the output impedance of the push-pull amplifier used in the signal amplification link.

6. The method for designing a wideband active EMI filter based on phasor analysis according to claim 5, wherein: CT transfer function G CT and the transfer function of the inverting amplifier G A The calculation method is: Among them, s is the Laplace operator, R c is the equivalent resistance that characterizes the loss, R sense is the sensing resistor; C pp is the parasitic capacitance of the primary side of CT, L m is the excitation inductance; Z o is the operational amplifier output impedance in the inverting amplifier, Z in It is the input impedance of the push-pull amplifier used in the signal amplification link.

7. A broadband active EMI filter obtained by the broadband active EMI filter design method based on phasor analysis according to any one of claims 1 to 6.

8. A broadband active EMI filter design system based on phasor analysis, characterized in that: comprising a computer-readable storage medium and a processor; The computer-readable storage medium is used to store executable instructions; The processor is configured to read the executable instructions stored in the computer-readable storage medium to execute the wideband active EMI filter design method based on phasor analysis according to any one of claims 1 to 6.

9. A computer-readable storage medium having a computer program stored thereon, characterized in that: When the program is executed by a processor, the wideband active EMI filter design method based on phasor analysis as described in any one of claims 1 to 6 is implemented.

10. A computer program product, characterized in that The invention comprises a computer program, which, when running on a computer, enables the computer to execute the wide-band active EMI filter design method based on phasor analysis according to any one of claims 1 to 6.

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