Boost converter and control method thereof

Abstract

The invention discloses a boost converter and a control method thereof, and relates to the technical field of power electronic power conversion, and the boost converter comprises a DC input power supply, a first boost module, a second boost module and a load. The cathode of the DC input power supply is connected with the first input end, the anode of the DC input power supply is connected with the second input end of the first boost module, the first boost module is connected with the second boost module through the transformer, and the first output end of the second boost module is connected with one end of the load. The second output end of the second boost module is connected with the other end of the load. The technical problems that a boost converter of a traditional electric energy quality treatment device is mainly a Boost converter, but large input current ripples exist under high-power operation, and the conduction loss of the electric energy quality treatment device is increased are solved.

Description

Technical Field

[0001] The present invention relates to the technical field of power electronic power conversion, and in particular to a boost converter and a control method thereof. Background Art

[0002] With the large-scale integration of renewable energy sources (such as photovoltaic and wind power), the widespread use of power electronic loads, and the nonlinear application of industrial equipment, power quality issues in power grids are becoming increasingly prominent, primarily manifesting as voltage sags and swells. These issues not only affect the normal operation of power equipment but can also damage precision instruments, malfunction of relay protection, and reduce grid stability. Therefore, power quality management technologies have become a key component in the development of smart grids and new power systems. As the core power unit of power quality management devices, the performance of the boost converter directly affects the management effectiveness, dynamic response speed, and system energy efficiency.

[0003] The boost converter of traditional power quality management devices is mainly a Boost converter, but there is a large input current ripple under high power operation, which increases the conduction loss of the power quality management device. Summary of the Invention

[0004] The present invention provides a boost converter and a control method thereof, which solves the technical problem that the boost converter of a traditional power quality management device is mainly a Boost converter, but has large input current ripple under high power operation, which increases the conduction loss of the power quality management device.

[0005] A first aspect of the present invention provides a boost converter comprising a DC input power supply, a first boost module, a second boost module and a load;

[0006] The negative electrode of the DC input power supply is connected to the first input terminal of the first boost module, and the positive electrode of the DC input power supply is connected to the second input terminal of the first boost module;

[0007] The first boost module is connected to the second boost module via a transformer;

[0008] The first output end of the second boost module is connected to one end of the load, and the second output end of the second boost module is connected to the other end of the load.

[0009] Optionally, the first boost module includes a first inductor and a boost sub-module;

[0010] The other end of the first inductor is connected to the first end of the boost submodule, and the second end of the boost submodule serves as the first input end of the first boost module;

[0011] One end of the first inductor serves as a second input end of the first boost module.

[0012] Optionally, the boost submodule includes a first switch tube, a second switch tube, a third switch tube, a fourth switch tube, a first capacitor, a second capacitor, a leakage inductor and a coupling inductor;

[0013] The negative electrode of the first capacitor is connected to the non-identical end of the first winding and the identical end of the second winding of the coupled inductor respectively;

[0014] The source electrode of the first switching tube is connected to the source electrode of the third switching tube and the negative electrode of the second capacitor respectively;

[0015] One end of the leakage inductor is connected to the same-name end of the first winding of the coupled inductor, and the other end of the leakage inductor is connected to the drain of the first switching tube, the source of the second switching tube and the same-name end of the primary winding of the transformer respectively;

[0016] The non-identical ends of the second winding of the coupled inductor are respectively connected to the drain of the third switching tube, the source of the fourth switching tube and the non-identical ends of the primary winding of the transformer;

[0017] The positive electrode of the first capacitor is connected to the drain of the second switch tube, the drain of the fourth switch tube and the positive electrode of the second capacitor respectively.

[0018] Optionally, the negative electrode of the first capacitor serves as the first end of the boost submodule;

[0019] The source of the first switch tube serves as the second end of the boost submodule.

[0020] Optionally, the coupled inductor is a reverse coupled inductor.

[0021] Optionally, the second boost module includes a fifth switch tube, a sixth switch tube, a seventh switch tube, an eighth switch tube, a third capacitor and a fourth capacitor;

[0022] The like-named ends of the secondary winding of the transformer are respectively connected to the drain of the fifth switching tube, the source of the sixth switching tube, and the source of the seventh switching tube;

[0023] The non-identical ends of the secondary winding of the transformer are respectively connected to the source of the eighth switch tube, the positive electrode of the third capacitor, and the negative electrode of the fourth capacitor;

[0024] The drain of the seventh switching tube is connected to the drain of the eighth switching tube;

[0025] The drain of the sixth switch tube is connected to the positive electrode of the fourth capacitor, and the source of the fifth switch tube is connected to the negative electrode of the third capacitor.

[0026] Optionally, the positive electrode of the fourth capacitor serves as the first output end of the second boost module.

[0027] Optionally, the negative electrode of the third capacitor serves as the second output end of the second boost module.

[0028] A second aspect of the present invention provides a control method for a boost converter, comprising:

[0029] Obtaining an input voltage and an output voltage of a boost converter, and subjecting the turns ratio of the boost converter, the input voltage, and the output voltage to dynamic modulation processing to obtain a corresponding duty cycle;

[0030] generating a driving signal according to the duty cycle;

[0031] The driving signal is used to regulate the first boost module and the second boost module of the boost converter.

[0032] Optionally, the step of dynamically modulating the turns ratio of the boost converter, the input voltage, and the output voltage to obtain a corresponding duty cycle includes:

[0033] multiplying a preset modulation coefficient, a turns ratio of the boost converter, and the input voltage to obtain a first product value;

[0034] Performing difference processing on the output voltage and the first multiplication value to obtain the first difference;

[0035] The first difference is ratioed to the output voltage to obtain a corresponding duty cycle.

[0036] It can be seen from the above technical solutions that the present invention has the following advantages:

[0037] The boost converter of the present invention includes a DC input power supply, a first boost module, a second boost module and a load. The negative pole of the DC input power supply is connected to the first input terminal, the positive pole of the DC input power supply is connected to the second input terminal of the first boost module, the first boost module is connected to the second boost module through a transformer, the first output terminal of the second boost module is connected to one end of the load, and the second output terminal of the second boost module is connected to the other end of the load. The boost converter of the traditional power quality management device is mainly a Boost converter, but there is a large input current ripple under high power operation, which increases the technical problem of the conduction loss of the power quality management device. Compared with the traditional Boost converter, the present invention reduces the current ripple in the boost converter through the first boost module, thereby reducing the conduction loss of the boost converter. BRIEF DESCRIPTION OF THE DRAWINGS

[0038] In order to more clearly illustrate the embodiments of the present invention or the technical solutions in the prior art, the following briefly introduces the drawings required for use in the embodiments or the description of the prior art. Obviously, the drawings described below are only some embodiments of the present invention. For ordinary technicians in this field, other drawings can be obtained based on these drawings without paying any creative work.

[0039] Figure 1 A schematic diagram of a boost converter provided by an embodiment of the present invention;

[0040] Figure 2 A voltage and current waveform diagram of the main components in a boost converter provided by an embodiment of the present invention;

[0041] Figure 3 A first circuit modal diagram of a boost converter provided by an embodiment of the present invention;

[0042] Figure 4 A second circuit modal diagram of a boost converter provided by an embodiment of the present invention;

[0043] Figure 5 A third circuit modal diagram of a boost converter provided by an embodiment of the present invention;

[0044] Figure 6 A fourth circuit modal diagram of a boost converter provided by an embodiment of the present invention;

[0045] Figure 7 A coupled inductor voltage and current waveform diagram of a boost converter provided by an embodiment of the present invention when the duty cycle is less than 0.5;

[0046] Figure 8 A coupled inductor voltage and current waveform diagram of a boost converter provided by an embodiment of the present invention when the duty cycle is greater than 0.5;

[0047] Figure 9 The inductor voltage and current waveforms of a boost converter provided by an embodiment of the present invention when the coupling coefficient is 0;

[0048] Figure 10 The inductor voltage and current waveforms of a boost converter provided by an embodiment of the present invention when the coupling coefficient is less than 0 and the duty cycle is less than 0.5;

[0049] Figure 11 The inductor voltage and current waveforms of a boost converter provided by an embodiment of the present invention when the coupling coefficient is less than 0 and the duty cycle is greater than 0.5;

[0050] Figure 12A schematic diagram of a curve showing a change in the inductor current ratio with respect to the duty cycle at different coupling coefficients for a boost converter provided by an embodiment of the present invention;

[0051] Figure 13 This is a waveform diagram of a boost converter provided by an embodiment of the present invention at a circuit output of 3kW;

[0052] Figure 14 This is a waveform diagram of the traditional interleaved boost circuit under 3kW output provided by an embodiment of the present invention;

[0053] Figure 15 A comparison chart of the efficiency of a boost converter provided by an embodiment of the present invention and a traditional interleaved boost circuit;

[0054] Figure 16 A flowchart of a control method for a boost converter provided by an embodiment of the present invention. DETAILED DESCRIPTION

[0055] An embodiment of the present invention provides a boost converter and a control method thereof, which are used to solve the technical problem that the boost converter of a traditional power quality management device is mainly a Boost converter, but has large input current ripple under high power operation, which increases the conduction loss of the power quality management device.

[0056] In order to make the purpose, features, and advantages of the present invention more obvious and easy to understand, the technical solutions in the embodiments of the present invention will be clearly and completely described below in conjunction with the drawings in the embodiments of the present invention. Obviously, the embodiments described below are only part of the embodiments of the present invention, not all of the embodiments. Based on the embodiments of the present invention, all other embodiments obtained by ordinary technicians in this field without making creative work are within the scope of protection of the present invention.

[0057] For easier understanding, see Figure 1-16 The embodiment of the present invention provides a boost converter, including a DC input power supply V in , a first boost module, a second boost module and a load;

[0058] DC input power supply V in The negative electrode is connected to the first input terminal of the first boost module, and the DC input power supply V in The positive electrode is connected to the second input terminal of the first boost module;

[0059] The first step-up module is connected to the transformer T r connected to the second boost module;

[0060] The first output end of the second boost module is connected to one end of the load, and the second output end of the second boost module is connected to the other end of the load.

[0061] In an embodiment of the present invention, the boost converter includes a DC input power supply V in , first boost module, second boost module and load. DC input power supply V in The negative electrode is connected to the first input terminal, the DC input power supply V in The positive electrode is connected to the second input terminal of the first boost module, and the first boost module is connected to the second input terminal of the first boost module through the transformer T r The first boost module is connected to the second boost module. The first output terminal of the second boost module is connected to one end of the load, and the second output terminal of the second boost module is connected to the other end of the load. The first boost module is controlled by two sets of drive signals. The two sets of drive signals are phase-shifted by 180 degrees, and the duty cycle is dynamically adjusted according to the input and output voltages. The second boost conversion module performs phase shift control based on the drive signals.

[0062] See Figure 1 As shown, the first boost module includes a first inductor L in and boost submodule; first inductor L in The other end is connected to the first end of the boost submodule, and the second end of the boost submodule serves as the first input end of the first boost module; the first inductor L in One end serves as the second input end of the first boost module.

[0063] In the embodiment of the present invention, the first boost module includes a first inductor L in and the boost submodule, the first inductor L in The other end is connected to the first end of the boost submodule, and the second end of the boost submodule is connected to the DC input power supply V in The negative pole of the first inductor L is connected in One end is connected to the DC input power supply V in positive connection.

[0064] See Figure 1 As shown, the boost submodule includes a first switch tube S1, a second switch tube S2, a third switch tube S3, a fourth switch tube S4, a first capacitor C1, a second capacitor C2, a leakage inductor L k and coupled inductor L C The negative electrode of the first capacitor C1 is connected to the coupling inductor L C The non-identical end of the first winding is connected to the identical end of the second winding; the source of the first switch tube S1 is connected to the source of the third switch tube S3 and the negative electrode of the second capacitor C2 respectively; the leakage inductance L k One end and the coupled inductor L C The first winding of the same name end is connected, the leakage inductance L k The other end is connected to the drain of the first switch tube S1, the source of the second switch tube S2 and the transformer T r The primary winding of the same name is connected; the coupling inductor LC The non-identical ends of the second winding are connected to the drain of the third switch tube S3, the source of the fourth switch tube S4 and the transformer T r The positive electrode of the first capacitor C1 is respectively connected to the drain of the second switch tube S2, the drain of the fourth switch tube S4 and the positive electrode of the second capacitor C2.

[0065] In the embodiment of the present invention, the boost submodule includes a first switch tube S1, a second switch tube S2, a third switch tube S3, a fourth switch tube S4, a first capacitor C1, a second capacitor C2, a leakage inductor L k and coupled inductor L C The negative electrode of the first capacitor C1 is connected to the coupling inductor L C The non-identical terminals of the first winding are connected to the identical terminals of the second winding. The source of the first switch tube S1 is connected to the source of the third switch tube S3 and the negative electrode of the second capacitor C2 respectively. k One end and the coupled inductor L C The first winding of the same name end is connected, the leakage inductance L k (ie transformer T r The other end of the inductor is connected to the drain of the first switch tube S1, the source of the second switch tube S2 and the transformer T r The primary winding of the same name is connected. Coupled inductor L C The non-identical ends of the second winding are connected to the drain of the third switch tube S3, the source of the fourth switch tube S4 and the transformer T r The positive electrode of the first capacitor C1 is connected to the drain of the second switch tube S2, the drain of the fourth switch tube S4 and the positive electrode of the second capacitor C2 respectively.

[0066] See Figure 1 As shown, the negative electrode of the first capacitor C1 serves as the first end of the boost submodule; the source electrode of the first switch tube S1 serves as the second end of the boost submodule.

[0067] In the embodiment of the present invention, the negative electrode of the first capacitor C1 serves as the first end of the boost submodule, and the negative electrode of the first capacitor C1 is connected to the first inductor L in Connect the other end.

[0068] It should be noted that the coupled inductor L C is the reverse coupling inductor L C .

[0069] In the embodiment of the present invention, the coupled inductor L C is the reverse coupling inductor L C , reverse coupling inductor L C Compared with the uncoupled inductor L C It can improve the dynamic performance of the converter.

[0070] See Figure 1 As shown, the second boost module includes a fifth switch tube S5, a sixth switch tube S6, a seventh switch tube S7, an eighth switch tube S8, a third capacitor C3 and a fourth capacitor C4; a transformer T r The same-named terminals of the secondary winding of the transformer are connected to the drain of the fifth switch tube S5, the source of the sixth switch tube S6 and the source of the seventh switch tube S7 respectively; r The non-identical ends of the secondary winding are respectively connected to the source of the eighth switch tube S8, the positive electrode of the third capacitor C3, and the negative electrode of the fourth capacitor C4; the drain of the seventh switch tube S7 is connected to the drain of the eighth switch tube S8; the drain of the sixth switch tube S6 is connected to the positive electrode of the fourth capacitor C4, and the source of the fifth switch tube S5 is connected to the negative electrode of the third capacitor C3.

[0071] In the embodiment of the present invention, the second boost module includes a fifth switch tube S5, a sixth switch tube S6, a seventh switch tube S7, an eighth switch tube S8, a third capacitor C3 and a fourth capacitor C4. r The same-named terminals of the secondary winding of the transformer are connected to the drain of the fifth switch tube S5, the source of the sixth switch tube S6 and the source of the seventh switch tube S7 respectively. r The non-identical terminals of the secondary winding are connected to the source of the eighth switch S8, the positive terminal of the third capacitor C3, and the negative terminal of the fourth capacitor C4, respectively. The drain of the seventh switch S7 is connected to the drain of the eighth switch S8. The drain of the sixth switch S6 is connected to the positive terminal of the fourth capacitor C4, and the source of the fifth switch S5 is connected to the negative terminal of the third capacitor C3.

[0072] See Figure 1 As shown, the positive electrode of the fourth capacitor C4 serves as the first output end of the second boost module.

[0073] In the embodiment of the present invention, the positive electrode of the fourth capacitor C4 serves as the first output terminal of the second boost module and is connected to one end of the load.

[0074] See Figure 1 As shown, the negative electrode of the third capacitor C3 serves as the second output end of the second boost module.

[0075] In the embodiment of the present invention, the negative electrode of the third capacitor C3 serves as the second output terminal of the second boost module and is connected to the other end of the load.

[0076] It should be noted that, see Figure 2 As shown, V GS1 is the driving signal of the first switch tube S1, V GS2 is the driving signal of the second switch tube S2, i L1 is the current flowing through the first inductor L inCurrent, I L1,max is the current flowing through the first inductor L in Maximum current, I L1,min is the current flowing through the first inductor L in The minimum current, i L2 is the current flowing through the second inductor, I L2,max is the maximum current flowing through the second inductor, I L2,min is the minimum current flowing through the second inductor, i in is the input current.

[0077] See Figure 2-Figure 3 As shown, when the boost converter is in the first stage (t0 (first stage start time) - t1 (first stage end time)), the first switch tube S1, the fourth switch tube S4, the seventh switch tube S7 and the eighth switch tube S8 are turned on under the action of the driving signal, and the DC input power supply V in For the first inductor L in charging, the second inductor discharges, therefore, flows through the first inductor L in The current flowing through the second inductor increases linearly, and the current flowing through the second inductor decreases linearly (the input current ripple is offset at this time). C The first capacitor C1 is charged through the fourth switch tube S4, and the transformer T r The primary winding voltage of the transformer T is clamped by the first capacitor C1. r The secondary winding of the seventh switch tube S7 and the eighth switch tube S8 are short-circuited, and the leakage inductance L k The current is subjected to negative pressure and decreases linearly. The third capacitor C3 and the fourth capacitor C4 are connected in series to discharge the load.

[0078] See Figure 2 and Figure 4 As shown, when the boost converter is in the second stage (t1-t2 (end time of the second stage)), the first switch tube S1, the fourth switch tube S4 and the fifth switch tube S5 are turned on under the action of the driving signal, and the DC input power supply V in For the first inductor L in charging, the second inductor discharges, therefore, flows through the first inductor L in The current flowing through the second inductor increases linearly, and the current flowing through the second inductor decreases linearly (the input current ripple is offset at this time). C The first capacitor C1 is charged through the fourth switch tube S4, and the transformer T r The primary winding voltage of the transformer T is clamped by the first capacitor C1. r The secondary winding of is clamped by the third capacitor C3. Since the voltage turns ratio relationship is satisfied, the leakage inductance L k The current withstand voltage is basically zero and the current is almost constant. The third capacitor C3 and the fourth capacitor C4 are connected in series to discharge the load.

[0079] See Figure 2 and Figure 5 As shown, when the boost converter is in the third stage (t2-t3 (end time of the third stage)), the second switch tube S2, the fourth switch tube S4 and the fifth switch tube S5 are turned on under the action of the driving signal, and the first inductor L in and the second inductor discharges, therefore, flows through the first inductor L in The current flowing through the second inductor decreases linearly. C The first capacitor C1 is charged through the second switch tube S2 and the fourth switch tube S4. r The primary winding voltage of the transformer T is short-circuited by the second switch tube S2 and the fourth switch tube S4. r The secondary winding is clamped by the third capacitor C3. k When subjected to positive voltage, the current rises linearly. The third capacitor C3 and the fourth capacitor C4 are connected in series to discharge the load.

[0080] See Figure 2 and Figure 6 As shown, when the boost converter is in the fourth stage (t3-t4 (end time of the fourth stage)), the second switch tube S2, the fourth switch tube S4, the seventh switch tube S7 and the eighth switch tube S8 are turned on under the action of the driving signal, and the first inductor L in and the second inductor discharges. Therefore, the current flowing through the first inductor L in The current flowing through the second inductor decreases linearly. C The first capacitor C1 is charged through the second switch tube S2 and the fourth switch tube S4. r The primary winding voltage of the transformer T is short-circuited by the second switch tube S2 and the fourth switch tube S4. r The secondary winding of the seventh switch tube S7 and the eighth switch tube S8 are short-circuited, and the leakage inductance L k When the voltage is zero, the current remains unchanged. The third capacitor C3 and the fourth capacitor C4 are connected in series to discharge the load.

[0081] It should be noted that, assuming that the duty cycle ,in is the circuit's duty cycle, is the on-time of the first switch S1 (i.e., the time during which the driving signal of the first switch S1 is at a high level). All devices are ideal devices. The capacitance value of the capacitor is large enough, and the voltage across the capacitor remains constant during the switching cycle. C The two windings are highly symmetrical, with only a phase difference in the drive signals. Therefore, we only need to analyze the case of one phase when D < 0.5. When the first switch S1 is on and the third switch S3 is off, Kirchhoff's voltage law yields:

[0082] V L1(on) = V in ; (1)

[0083] Among them, V L1(on) When the first switch tube S1 is turned on and the third switch tube S3 is turned off, the coupling inductor L C The voltage across the first winding, V in is the input voltage.

[0084] When the first switch S1 is turned off, Kirchhoff's voltage law gives:

[0085] V L1(off) = V in - V O ; (2)

[0086] Among them, V L1(off) When the first switch tube S1 is turned off, the coupling inductor L C The voltage across the first winding, V O is the output voltage.

[0087] Under steady-state conditions, the first inductor L in Using the volt-second balance condition, we can get:

[0088] V L1(on) DT S + V L1(off) (1-D)T S = 0; (3)

[0089] Substituting equations (1) and (2) into equation (3), we can obtain:

[0090] V C2 = V in / (1-D); (4)

[0091] Among them, V C2 is the voltage across the second capacitor C2.

[0092] For the input power supply, the first inductor L in , the first capacitor C1 and the second capacitor C2, Kirchhoff's law gives:

[0093] -V in + V Lin - V C1 + V C2 = 0; (5)

[0094] Among them, V Lin is the first inductor L in The average voltage of the inductor is V. Linis 0.

[0095] Substituting (4) into (5) we can obtain:

[0096] V C1 = DV in / (1-D); (6)

[0097] Among them, V C1 is the voltage across the first capacitor C1.

[0098] The third capacitor C3 and the fourth capacitor C4 are connected in series on both sides of the output to evenly divide the output voltage:

[0099] V C3 = V C4 = V O / 2; (7)

[0100] Among them, V C3 is the voltage across the third capacitor C3, V C4 is the voltage across the fourth capacitor C4

[0101] It is worth mentioning that in order to simplify the analysis process of zero input current ripple characteristics, it is assumed that all devices are ideal devices; the capacitance value is large enough, and the voltage across the capacitor remains constant during the switching cycle.

[0102] See Figure 1 As shown, writing Kirchhoff's voltage equation yields:

[0103] V in = v Lin -v C1 +v C2 ; (8)

[0104] Among them, v Lin is the first inductor L in The voltage across the terminals, v C1 is the voltage across the first capacitor C1, v C2 is the voltage across the second capacitor C2.

[0105] v C1 = V C1 , v C2 =V C2 ; (9)

[0106] Among them, V C1 is the average voltage of the first capacitor C1, V C2 is the average voltage of the second capacitor C2.

[0107] Substituting formula (8) into formula (7) yields:

[0108] v Lin=V in +V C1 -V C2 ; (10)

[0109] See Figure 14 As shown, the first inductor L in The transient voltage is equal to the constant value, that is, V Lin .

[0110] When the first inductor L in In steady state, according to the volt-second balance, the first inductance L in Average voltage V Lin is 0, we can get:

[0111] v Lin =V Lin =0; (11)

[0112] Substituting equation (10) into the inductor induced voltage formula v = LΔi / Δt, we can obtain:

[0113] Δi Lin = v Lin * Δt / L = 0; (12)

[0114] Among them, Δi is the change of single-circuit inductor current caused by ΔD, ΔD is the change of duty cycle, Δt is the change of time, Δi Lin is the first inductor L in The change in the inductor current.

[0115] It is worth mentioning that in order to simplify the low winding current ripple coupled inductor L C In the analysis process, it is assumed that the coupling coefficient α = M / L, where M is the mutual inductance, L is the self-inductance, and -1<α<1; all devices are ideal devices; the capacitance value is large enough, and the voltage across the capacitor remains constant during the switching cycle; and the inductors of the two paths are exactly the same.

[0116] See Figure 1 As shown, the coupled inductor L C Relationship to write the voltage equation:

[0117] ; (13)

[0118] in, is the coupled inductor L C The voltage value of the first inductor, is the coupled inductor L C The voltage value of the second inductor, is the coupled inductor L C The inductance value of the first inductor, is the coupled inductor L CThe inductance of the second inductor, t is the time, is the coupled inductor L C The current value of the first inductor, is the coupled inductor L C The current value of the second inductor.

[0119] See Figure 7 As shown, the inductor voltage in each time period can be obtained, where V a =V in , V b = V in -V o Where, V a is the voltage across the two windings of the coupled inductor when the first switch S1 and the third switch S3 are turned on, V b It is the voltage borne by the two windings of the coupled inductor when the second switch tube S2 and the fourth switch tube S4 are turned on.

[0120] In the time period t0~t1, there are:

[0121] V L1 =V a , V L2 =V b ; (14)

[0122] Substituting formula (13) into formula (12) yields:

[0123] ; (15)

[0124] in, is the equivalent inductance during the time period t0~t1.

[0125] According to the above calculation ideas, the equivalent inductances of other time periods can be obtained, which are:

[0126] ; (16)

[0127] ; (17)

[0128] Among them, L eq2 is the equivalent inductance during the time period t1~t2, L eq3 is the equivalent inductance during the time period t2~t3, L eq4 is the equivalent inductance during the time period t3 to t4. The equivalent inductance formulas derived for each time period are consistent with the results for the case where the duty cycle D is less than 0.5, so they are not repeated here.

[0129] See Figure 7-Figure 8 As shown, when D<0.5, the magnitude of the steady-state inductor current ripple is given by When D>0.5, the size of the steady-state inductor current ripple is determined by L eq3 Decide.

[0130] Substituting α=0 into equations (14) and (16), we can obtain:

[0131] L=L eq1 =L eq3 ; (18)

[0132] Therefore, in the uncoupled case, the steady-state equivalent inductance is equal to the self-inductance and has nothing to do with the duty cycle.

[0133] It is worth mentioning that the dynamic response performance can be expressed as Δi / ΔD within a switching cycle, where ΔD is the change in duty cycle and Δi is the change in single-circuit inductor current caused by ΔD.

[0134] See Figure 9 As shown in the figure, in the non-coupling case, when the duty cycle changes by ΔD, we can get:

[0135] ; (19)

[0136] in, is the increase in the current of the first winding of the coupled inductor during the ΔD*T time period when α=0 and the duty cycle increases by ΔD. It is the reduction of the current of the first winding of the coupled inductor during the ΔD*T period when α=0 and the duty cycle does not change.

[0137] V a =V in , V b = V in –V o Substituting into formula (18) we can get:

[0138] ; (20)

[0139] See Figure 10 As shown in the figure, in the case of reverse coupling and duty cycle D<0.5, when the duty cycle changes by ΔD, we can get:

[0140] ;(twenty one)

[0141] in, The change in the current of the first winding of the coupled inductor during the time period t0~t1 when α<0 and the duty cycle increases by ΔD (D<0.5) is: The change in the current of the first winding of the coupled inductor during the time period t1 to t2 when α<0 and the duty cycle increases by ΔD (D<0.5) is: is the change in the current of the first winding of the coupled inductor during the time period t2 - t3 when α < 0 and the duty cycle increases by ΔD (D < 0.5). is the change in the current of the first winding of the coupled inductor during the time period t3 - t4 when α < 0 and the duty cycle increases by ΔD (D < 0.5). is the change in the current of the first winding of the coupled inductor during the time period t4 - t5 when α < 0 and the duty cycle increases by ΔD (D < 0.5).

[0142] According to Equation (20), it can be obtained that:

[0143] ; (22)

[0144] where k is the time period serial number.

[0145] Refer to Figure 7 As shown, in the steady state, using volt - second balance, it can be obtained that:

[0146] ; (23)

[0147] Substituting Equation (22) into Equation (21), it can be obtained that:

[0148] ; (24)

[0149] Refer to Figure 11 As shown, in the case of reverse coupling and duty cycle D > 0.5, when the duty cycle changes by ΔD, it can be obtained that:

[0150] (25) <00​​​​​​​​​​​​​​​​​​​​​​​​​​​​​​​

[0158] It is worth mentioning that if the uncoupled inductor L is designed C and reverse coupling inductor L C With equal dynamic equivalent inductance, the ratio of steady-state inductor current ripple is:

[0159] (29)

[0160] in, is the reverse coupling inductor L C The current ripple of the inductor in is the uncoupled inductor L C The current ripple of the inductor in one circuit is as follows. Figure 12 As shown, under appropriate parameter settings, the boost converter of the present application can reduce the current ripple by more than 50% compared with the uncoupled case, and accordingly can reduce the conduction loss of each winding.

[0161] It is worth mentioning that, from the modal analysis, it can be seen that the transformer T r The primary winding voltage is clamped by the second capacitor C2, and the transformer T r The voltage of the secondary winding third capacitor C3 is clamped, then the transformer T r Leakage inductance L k The electrical quantity characteristic expression is:

[0162] V Lk = L k *di Lk / dt (30)

[0163] V Lk = -V C2 + V C3 / n = V O / 2n - V in / (1-D)(31)

[0164] Among them, V Lk Transformer T r Leakage inductance L k Voltage, i Lk Transformer T r Leakage inductance L k current, and n is the turns ratio.

[0165] When the input and output voltages change, the control duty cycle satisfies formula (32), which can ensure that V Lk Constantly set to 0, the leakage inductance L k The current will not produce large current ripple.

[0166] D = (V O -2nVin ) / V O (32)

[0167] See Table 1 and Figure 13 As shown, the duty cycle D of the first switch tube S1 is set to 0.33. C The current waveforms of the first and second windings have segmented slopes, and the ripple current of each phase is 3.1A. Figure 15 It can be seen that the input current waveform of the boost converter is a constant value, its AC component is about 110mA, and the ripple rate of the input current is only 1.2%; the transformer T r Leakage inductance L k The current in this state satisfies formula (32), which can control the leakage inductance L k The current is in a small ripple range (flat-top wave).

[0168] Table 1

[0169]

[0170] See Figure 14 As shown in the figure, under the same parameter configuration, the input current ripple of the traditional boost converter is 2.23A, and the ripple rate increases to 25%, thus introducing more conduction loss. In addition, due to the use of coupled inductor L C The ripple current of the two-phase inductor branch is 4.41A, which is 42% higher than that of the boost converter of this application.

[0171] It should be noted that, see Figure 15 As shown, the boost converter of the present invention can improve the efficiency by 5% under light load conditions and by 2% under heavy load conditions compared to the traditional boost converter.

[0172] In an embodiment of the present invention, the boost converter includes a DC input power supply V in , first boost module, second boost module and load. DC input power supply V in The negative electrode is connected to the first input terminal, the DC input power supply V in The positive electrode is connected to the second input terminal of the first boost module, and the first boost module is connected to the second input terminal of the first boost module through the transformer T r The first output terminal of the second boost module is connected to one end of the load, and the second output terminal of the second boost module is connected to the other end of the load. This overcomes the technical problem that the boost converter of the traditional power quality management device is mainly a Boost converter, but there is a large input current ripple under high power operation, which increases the conduction loss of the power quality management device. Compared with the traditional Boost converter, the present invention reduces the current ripple in the boost converter through the first boost module, thereby reducing the conduction loss of the boost converter.

[0173] See also Figure 16 The present invention provides a control method for a boost converter, comprising:

[0174] Step 101: Obtain the input voltage and output voltage of the boost converter, and dynamically modulate the turns ratio, input voltage, and output voltage of the boost converter to obtain a corresponding duty cycle.

[0175] Step 102: Generate a driving signal according to the duty cycle;

[0176] Step 103: Use the driving signal to regulate the first boost module and the second boost module of the boost converter.

[0177] Furthermore, step 101 includes the following sub-steps:

[0178] S11, multiplying a preset modulation coefficient, a turns ratio of the boost converter, and an input voltage to obtain a first product value;

[0179] S12, performing difference processing on the output voltage and the first product value to obtain a first difference;

[0180] S13 , performing ratio processing on the first difference and the output voltage to obtain a corresponding duty cycle.

[0181] Those skilled in the art will clearly understand that, for the convenience and brevity of description, the specific working processes of the systems, devices and units described above can refer to the corresponding processes in the aforementioned method embodiments and will not be repeated here.

[0182] In the several embodiments provided in this application, it should be understood that the disclosed systems, devices and methods can be implemented in other ways. For example, the device embodiments described above are merely schematic. For example, the division of units is only a logical function division. In actual implementation, there may be other division methods, such as multiple units or components can be combined or integrated into another system, or some features can be ignored or not executed. Another point is that the mutual coupling or direct coupling or communication connection shown or discussed can be an indirect coupling or communication connection through some interface, device or unit, which can be electrical, mechanical or other forms.

[0183] The above embodiments are only used to illustrate the technical solutions of the present invention, rather than to limit the same. Although the present invention has been described in detail with reference to the aforementioned embodiments, those skilled in the art should understand that they can still modify the technical solutions described in the aforementioned embodiments, or make equivalent replacements for some of the technical features therein. However, these modifications or replacements do not deviate the essence of the corresponding technical solutions from the spirit and scope of the technical solutions of the various embodiments of the present invention.

Claims

1. A boost converter, characterized in that: It includes a DC input power supply, a first boost module, a second boost module and a load; The negative electrode of the DC input power supply is connected to the first input terminal of the first boost module, and the positive electrode of the DC input power supply is connected to the second input terminal of the first boost module; The first boost module is connected to the second boost module via a transformer; The first output end of the second boost module is connected to one end of the load, and the second output end of the second boost module is connected to the other end of the load.

2. The boost converter according to claim 1, wherein: The first boost module includes a first inductor and a boost submodule; The other end of the first inductor is connected to the first end of the boost submodule, and the second end of the boost submodule serves as the first input end of the first boost module; One end of the first inductor serves as a second input end of the first boost module.

3. The boost converter according to claim 2, wherein: The boost submodule includes a first switch tube, a second switch tube, a third switch tube, a fourth switch tube, a first capacitor, a second capacitor, a leakage inductor and a coupling inductor; The negative electrode of the first capacitor is connected to the non-identical end of the first winding and the identical end of the second winding of the coupled inductor respectively; The source electrode of the first switching tube is connected to the source electrode of the third switching tube and the negative electrode of the second capacitor respectively; One end of the leakage inductor is connected to the same-name end of the first winding of the coupled inductor, and the other end of the leakage inductor is connected to the drain of the first switching tube, the source of the second switching tube and the same-name end of the primary winding of the transformer respectively; The non-identical ends of the second winding of the coupled inductor are respectively connected to the drain of the third switching tube, the source of the fourth switching tube and the non-identical ends of the primary winding of the transformer; The positive electrode of the first capacitor is connected to the drain of the second switch tube, the drain of the fourth switch tube and the positive electrode of the second capacitor respectively.

4. The boost converter according to claim 3, wherein: The negative electrode of the first capacitor serves as the first end of the boost submodule; The source of the first switch tube serves as the second end of the boost submodule.

5. The boost converter according to claim 3, wherein: The coupled inductor is a reverse coupled inductor.

6. The boost converter according to claim 1, wherein: The second boost module includes a fifth switch tube, a sixth switch tube, a seventh switch tube, an eighth switch tube, a third capacitor and a fourth capacitor; The like-named ends of the secondary winding of the transformer are respectively connected to the drain of the fifth switching tube, the source of the sixth switching tube, and the source of the seventh switching tube; The non-identical ends of the secondary winding of the transformer are respectively connected to the source of the eighth switch tube, the positive electrode of the third capacitor, and the negative electrode of the fourth capacitor; The drain of the seventh switching tube is connected to the drain of the eighth switching tube; The drain of the sixth switch tube is connected to the positive electrode of the fourth capacitor, and the source of the fifth switch tube is connected to the negative electrode of the third capacitor.

7. The boost converter according to claim 6, wherein: The positive electrode of the fourth capacitor serves as the first output end of the second boost module.

8. The boost converter according to claim 6, wherein: The negative electrode of the third capacitor serves as the second output end of the second boost module.

9. A control method for a boost converter, characterized in that: include: Obtaining an input voltage and an output voltage of a boost converter, and subjecting the turns ratio of the boost converter, the input voltage, and the output voltage to dynamic modulation processing to obtain a corresponding duty cycle; generating a driving signal according to the duty cycle; The driving signal is used to regulate the first boost module and the second boost module of the boost converter.

10. The control method of the boost converter according to claim 9, characterized in that: The step of dynamically modulating the turns ratio of the boost converter, the input voltage, and the output voltage to obtain a corresponding duty cycle includes: multiplying a preset modulation coefficient, a turns ratio of the boost converter, and the input voltage to obtain a first product value; Performing difference processing on the output voltage and the first multiplication value to obtain the first difference; The first difference is ratioed to the output voltage to obtain a corresponding duty cycle.

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