Low-carrier-ratio permanent magnet motor sensorless control method based on disturbance suppression

Abstract

The invention discloses a low-carrier-ratio permanent magnet motor sensorless control method based on disturbance suppression, and belongs to the field of motor control. Sampling current and voltage of the permanent magnet motor under a low carrier wave ratio working condition, and listing a current equation of the permanent magnet motor under an alpha-beta coordinate system; calculating the difference between the actual value and the observed value of the motor stator current, and designing a sliding mode surface; a quasi-proportional resonance controller is adopted, discretization design is carried out, and the difference value of the current is input into the quasi-proportional resonance controller to obtain extended back electromotive force; performing calculation processing on the back electromotive force by using a phase-locked loop to obtain observation values of the rotating speed and the position of the motor; a nonlinear disturbance observer model is constructed, observation compensation is carried out on disturbance, and the robustness of the system is improved.

Description

Technical Field

[0001] The present invention relates to the field of motor drive control, and in particular to a sensorless control method for a low-carrier ratio permanent magnet motor based on disturbance suppression. Background Art

[0002] Permanent magnet synchronous motors (PMSMs) are widely favored in high-performance drive applications due to their high power density and reliability. The vector control strategy of a PMSM relies on precise speed and position information. Traditionally, mechanical sensors are mounted on the motor shaft to obtain these signals, but this increases system size and cost, as well as hardware wiring complexity. Consequently, sensorless control technologies that do not require additional position sensors have become a research hotspot in recent years.

[0003] Research methods for sensorless control technology, both domestically and internationally, fall into two main categories: one is applicable to low-speed applications, with high-frequency voltage injection being the mainstream method. The other is suitable for medium- and high-speed applications, typically using back-electromotive force to estimate motor speed and position. Common methods include sliding mode observer (SMO) control, neural network control, and model reference adaptation.

[0004] In motor drive systems, a reduction in the carrier ratio (i.e., the ratio of switching frequency to fundamental frequency) is typically caused by two typical operating conditions: in high-power operation, the inverter switching frequency needs to be reduced to minimize power loss; in high-speed control mode, the increase in fundamental frequency is limited by the switching losses and efficiency of power devices, preventing the inverter switching frequency from increasing accordingly, resulting in a decrease in the carrier ratio. Under low-carrier operating conditions, the chattering phenomenon of traditional sliding mode observers is exacerbated, and a large number of harmonics are present in the current and back EMF, significantly reducing the observation accuracy of the rotor speed and position. In addition, traditional sliding mode observers have poor anti-interference capabilities, resulting in large speed fluctuations when subjected to load disturbances. Summary of the Invention

[0005] To overcome the aforementioned issues with permanent magnet synchronous motors operating at low carrier ratios, this paper proposes a sensorless control method for low-carrier ratio permanent magnet motors based on disturbance suppression. A quasi-proportional resonant controller replaces the switching function and low-pass filter of the traditional sliding mode observer. This method effectively suppresses system chattering at low carrier ratios, reduces back-EMF harmonics, improves motor speed estimation accuracy, and mitigates phase lag. Furthermore, to enhance the system's dynamic stability, a nonlinear disturbance observer is proposed to mitigate the impact of sudden load changes on motor operation, reduce speed fluctuations, and improve system robustness.

[0006] In order to achieve the above objectives, the present invention uses the following technical solutions:

[0007] The present invention proposes a sensorless control method for a low-carrier ratio permanent magnet motor based on disturbance suppression, and the specific steps are as follows:

[0008] S1. Use a sampling circuit to sample the current and voltage of the permanent magnet motor under low-carrier ratio conditions. Write the current equation of the permanent magnet motor in the α-β coordinate system.

[0009] S2. The actual value of the motor stator current i α ,i β With the observed value Make the difference and design the sliding surface accordingly;

[0010] S3. Using a quasi-proportional resonant controller and performing a discrete design, the current error in step S2 is input to the quasi-proportional resonant controller to obtain an extended back EMF;

[0011] S4. Calculate and process the extended back EMF using a phase-locked loop to obtain observed values ​​of the motor speed and position;

[0012] S5. A nonlinear disturbance observer is adopted, including defining a disturbance observation error, deriving a state equation of the disturbance observer according to a mechanical motion characteristic equation, and constructing a disturbance observer model.

[0013] Preferably, the current equation of the permanent magnet motor in step S1 in the α-β coordinate system is expressed as:

[0014]

[0015] Where u α ,u β ,i α ,i β , e α , e β are the components of voltage, current and back electromotive force in the α-β coordinate system, p represents the differential operator, L represents inductance, and R represents resistance.

[0016] Preferably, the specific design of the sliding surface in step S2 is

[0017]

[0018] Where s αβ represents the designed sliding surface, represents the current observation value, i αβ Indicates the actual current value.

[0019] Preferably, the quasi-proportional resonant controller is adopted and discretized design is performed in step S3, and the process includes: writing the discrete domain equation of the permanent magnet motor mathematical model:

[0020]

[0021] Where, T s represents the sampling time, u αβ Indicates the voltage value, Respectively represent back electromotive force and current observation value, A, B, C represent coefficients, K represents discrete time coefficient, G QPR represents the transfer function of the quasi-proportional resonant controller;

[0022] Discretize the quasi-proportional resonant observer and the expression is:

[0023]

[0024] Wherein, b2, b1, b0, a2, a1, and a0 all represent coefficients.

[0025] Preferably, the structure of the phase-locked loop in step S4 is specifically expressed as follows:

[0026]

[0027] Where θ e , represents the actual angle value and its observed value, k represents the product of angular velocity and magnetic flux, Represents the back electromotive force observation value on two coordinate axes. Approaching 0,

[0028] The position information can be extracted using a phase-locked loop. The specific expression is as follows:

[0029]

[0030] Where K p ,K i are the parameters of the PI controller, and s represents the complex variable.

[0031] Preferably, the disturbance observation error is defined in step S5, and the specific expression is:

[0032]

[0033] Where d is the actual disturbance of the system, To observe the disturbance;

[0034] The nonlinear disturbance observer can be designed as:

[0035]

[0036] Where L represents the observer gain.

[0037] Preferably, the mechanical motion characteristic equation in step S5 is expressed as:

[0038]

[0039] Where J represents the moment of inertia, b is the friction coefficient of the motor, and k t is the torque constant, ω m is the mechanical angular velocity, T L is the load torque.

[0040] Preferably, the state equation of the nonlinear disturbance observer in step S5 is expressed as:

[0041]

[0042] In the formula, z represents a defined state variable, represents the derivative of the state variable, d is the actual disturbance of the system, To observe the disturbance, represents the derivative of the observed disturbance, i q Represents the q-axis component of the current.

[0043] Preferably, the position sensorless control method described in step S5 is used for low carrier ratio working conditions.

[0044] Compared with the prior art, the beneficial effects of the technical solution of the present invention are:

[0045] The present invention proposes a sensorless control method for a low-carrier ratio permanent magnet motor based on disturbance suppression. The method samples the current and voltage of the permanent magnet motor under low-carrier ratio conditions, and lists the current equation of the permanent magnet motor in the α-β coordinate system; the actual value of the motor stator current is subtracted from the observed value, and a sliding mode surface is designed based on this; the current difference is input into a discretized quasi-proportional resonant controller, and the quasi-proportional resonant controller is used to replace the traditional sliding mode observer switching function to obtain the extended back electromotive force, thereby weakening the vibration, reducing the harmonic content of the back electromotive force, and improving the observation accuracy of the motor position and speed under low carrier ratio; the back electromotive force is calculated and processed by a phase-locked loop to obtain the observed values ​​of the motor speed and position; a nonlinear disturbance observer model is constructed to observe and compensate for the system disturbance caused by external disturbances, reduce the motor speed fluctuation, and improve the system robustness. BRIEF DESCRIPTION OF THE DRAWINGS

[0046] Figure 1 A flow chart showing a sensorless control method for a low-carrier ratio permanent magnet motor based on disturbance suppression proposed by the present invention;

[0047] Figure 2 The schematic block diagram of the sensorless control system for a low-carrier ratio permanent magnet motor based on disturbance suppression proposed by the present invention is shown; DETAILED DESCRIPTION

[0048] In order to more clearly understand the above-mentioned objects, features and advantages of the present invention, the present invention is further described in detail below in conjunction with the accompanying drawings and specific embodiments. It should be noted that, in the absence of conflict, the embodiments of the present application and the features therein can be combined with each other.

[0049] In the following description, many specific details are set forth to facilitate a full understanding of the present invention. However, the present invention may also be implemented in other ways different from those described herein. Therefore, the scope of protection of the present invention is not limited to the specific embodiments disclosed below.

[0050] For ease of understanding, please refer to the flowchart of the low carrier ratio permanent magnet motor sensorless control method based on disturbance suppression proposed in the present invention. Figure 1 , please refer to the control system principle block diagram Figure 2 .

[0051] The present invention provides an embodiment of a sensorless control method for a low-carrier ratio permanent magnet motor based on disturbance suppression, comprising:

[0052] S1. Use a sampling circuit to sample the current and voltage of the permanent magnet motor under low-carrier ratio conditions. Write the current equation of the permanent magnet motor in the α-β coordinate system.

[0053] Specifically, the current equation is expressed as:

[0054]

[0055] Where u α 、u β 、i α 、i β 、e α 、e β where are the voltage, current, and back EMF components on the α and β axes, respectively. R and L are the stator resistance and inductance, respectively. p is the differential operator.

[0056] S2. The actual value of the motor stator current i α ,i β With the observed value Make the difference and design the sliding surface accordingly;

[0057] Specifically, the actual value of the stator current i α ,i β With the observed value Making a difference:

[0058]

[0059] And the sliding surface s is defined as:

[0060]

[0061] S3. Using a quasi-proportional resonant controller and performing a discrete design, the current error in step S2 is input to the quasi-proportional resonant controller to obtain an extended back EMF;

[0062] Specifically, based on the internal model principle, a proportional resonant controller (PR) can act on the AC current error signal to achieve steady-state error-free current regulation. Although the PR controller has characteristics such as high gain at resonance and excellent frequency selection performance, its narrow bandwidth is not conducive to system stability. This paper adopts quasi-proportional resonant (QPR) control. The current difference is input into the QPR. In other words, the QPR replaces the switching function and low-pass filter (LPF) of the traditional sliding mode observer and discretizes the QPR. The derivation process is as follows:

[0063] The QPR controller expression is:

[0064]

[0065] Where ω0 is the resonant frequency; k p is the proportional gain; k r is the resonant gain. c is the cutoff frequency of QPR, which adjusts the controller bandwidth.

[0066] Specifically, the QPR observer equation can be expressed as:

[0067]

[0068] Since the control system based on the QPR observer is a dual-input single-output system, according to the linear superposition principle, it can be regarded as i αβ and u αβ The transfer functions entered individually are then superimposed.

[0069] i αβ and u αβ The transfer function when input alone is as follows:

[0070]

[0071] According to the linear superposition principle, we can get:

[0072]

[0073] By discretizing the motor model, we can obtain the motor discrete domain state equation:

[0074]

[0075] The motor model is a surface-mount permanent magnet synchronous motor, where: A = e -RT / L , T represents the sampling time.

[0076] The QPR observer is discretized using the bilinear method, namely Tustin discrete transformation, and the transformation formula is:

[0077]

[0078] The discretized expression of the quasi-proportional resonant controller can be obtained:

[0079]

[0080] In the formula, the parameter design is:

[0081]

[0082] S4 inputs the expanded back EMF of S3 into the phase-locked loop to obtain the observed values ​​of the motor speed and position;

[0083] Specifically, the structure of the phase-locked loop is expressed as follows

[0084]

[0085] Where θ e , represents the actual value of the angle and its observed value, and k represents the product of angular velocity and magnetic flux. Represents the back electromotive force observation value on two coordinate axes. Approaching 0,

[0086] Furthermore, a phase-locked loop is used to calculate and process the back electromotive force to extract the rotor position information.

[0087]

[0088] Where K p ,K i are the parameters of the PI controller, and s is a complex variable.

[0089] S5. A nonlinear disturbance observer is proposed to observe and compensate for disturbances, including defining the disturbance observation error, deriving the state equation of the disturbance observer based on the mechanical motion characteristic equation, and constructing the disturbance observer model.

[0090] Specifically, this paper proposes a nonlinear disturbance observer (NDOB) designed as follows:

[0091] Definition of perturbation observation error for:

[0092]

[0093] Where: d is the actual disturbance of the system, To observe the disturbance, NDOB can be designed as

[0094]

[0095] Where L represents the observer gain. The mechanical motion characteristic equation can be obtained from the motor mathematical model:

[0096]

[0097] Where k t is the torque constant, ω m is the mechanical angular velocity, T L is the load torque, b is the motor friction coefficient, and J is the moment of inertia.

[0098] The load torque As a disturbance d, the above formula can be rearranged to obtain the disturbance equation:

[0099]

[0100] because Cannot be measured directly, define a state variable The state equation of the disturbance observer is obtained:

[0101]

[0102] In the formula, z represents a defined state variable, represents the derivative of the state variable, d is the actual disturbance of the system, To observe the disturbance, represents the derivative of the observed disturbance, i q Represents the q-axis component of the current.

[0103] The following proves the stability of the observer:

[0104] Derivative of the perturbation observation error:

[0105]

[0106] Since d changes slowly with time, its derivative is approximately zero

[0107]

[0108] Solving the differential equation, we get

[0109]

[0110] In the formula is the initial disturbance error. When the observer gain L>0, the observation error can be guaranteed to converge to zero.

[0111] In summary, the present invention proposes a sensorless control method for a low-carrier ratio permanent magnet motor based on disturbance suppression. It adopts a quasi-proportional resonant controller, a phase-locked loop, and a nonlinear disturbance observer, which effectively solves the problems of severe jitter, low observer accuracy, and poor anti-disturbance performance of traditional sliding mode observers under low carrier ratios, and provides a design basis for improving the observation accuracy of the sensorless control system of a low-carrier ratio permanent magnet motor.

Claims

1. A sensorless control method for a low-carrier ratio permanent magnet motor based on disturbance suppression, characterized in that: The specific steps are as follows: S1. Use a sampling circuit to sample the current and voltage of the permanent magnet motor under low-carrier ratio conditions. Write the current equation of the permanent magnet motor in the α-β coordinate system. S2. The actual value of the motor stator current i α ,i β With the observed value Make the difference and design the sliding surface accordingly; S3. Using a quasi-proportional resonant controller and performing a discrete design, the current error in step S2 is input to the quasi-proportional resonant controller to obtain an extended back EMF; S4. Calculate and process the extended back EMF using a phase-locked loop to obtain observed values ​​of the motor speed and position; S5. A nonlinear disturbance observer is used, including: defining a disturbance observation error, deriving a state equation of the disturbance observer according to a mechanical motion characteristic equation, and constructing a disturbance observer model.

2. A sensorless control method for a low-carrier ratio permanent magnet motor based on disturbance suppression according to claim 1, characterized in that: The current equation of the permanent magnet motor in the α-β coordinate system in step S1 is expressed as: Where u α ,u β ,i α ,i β , e α , e β are the components of voltage, current, and back electromotive force on the α-β coordinate axis, p represents the differential operator, L represents inductance, and R represents resistance.

3. The sensorless control method for a low-carrier ratio permanent magnet motor based on disturbance suppression according to claim 1 is characterized in that: The design of the sliding surface described in S2 is specifically as follows: Where s αβ represents the designed sliding surface, represents the current observation value, i αβ Indicates the actual current value.

4. The sensorless control method for a low-carrier ratio permanent magnet motor based on disturbance suppression according to claim 1, characterized in that: S3 describes the use of a quasi-proportional resonant controller and a discretization design, which includes writing the discrete domain equations of the permanent magnet motor mathematical model: Where, T s represents the sampling time, u αβ Indicates the voltage value, Respectively represent back electromotive force and current observation value, A, B, C represent coefficients, K represents discrete time coefficient, G QPR represents the transfer function of the quasi-proportional resonant controller; The quasi-proportional resonant observer is discretized and expressed as: Wherein, b2, b1, b0, a2, a1, and a0 all represent coefficients.

5. The sensorless control method for a low-carrier ratio permanent magnet motor based on disturbance suppression according to claim 1, characterized in that: The specific structure expression of the phase-locked loop in step S4 is: Where, represents the actual angle value and its observed value, k represents the product of angular velocity and magnetic flux, Represents the back electromotive force observation value on two coordinate axes. Approaching 0, The position information can be extracted using a phase-locked loop. The specific expression is as follows: Where K p ,K i are the parameters of the PI controller, and s is a complex variable.

6. The sensorless control method for a low-carrier ratio permanent magnet motor based on disturbance suppression according to claim 1, characterized in that: The disturbance observation error is defined in step S5, and the specific expression is: Where d is the actual disturbance of the system, To observe the disturbance; The nonlinear disturbance observer is designed as: Where L represents the observer gain.

7. A sensorless control method for a low-carrier ratio permanent magnet motor based on disturbance suppression according to claim 6, characterized in that: The mechanical motion characteristic equation in step S5 is expressed as: Where J represents the moment of inertia, b is the friction coefficient of the motor, and k t is the torque constant, ω m is the mechanical angular velocity, T L is the load torque.

8. The sensorless control method for a low-carrier ratio permanent magnet motor based on disturbance suppression according to claim 7, characterized in that: The state equation of the nonlinear disturbance observer in step S5 is: In the formula, z represents a defined state variable, represents the derivative of the state variable, d is the actual disturbance of the system, To observe the disturbance, represents the derivative of the observed disturbance, i q Represents the q-axis component of the current.

9. A sensorless control method for a low-carrier ratio permanent magnet motor based on disturbance suppression according to claim 8, characterized in that: The sensorless control method is used for low carrier ratio working conditions.

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