Wide-range efficiency optimization method for full-soft switching of isolated SR-DAB converter

Abstract

The invention relates to the technical field of converter modulation and artificial intelligence, in particular to a full-soft switching wide-range efficiency optimization method for an isolated SR-DAB converter, and the method comprises the steps: S1, building a general model of the SR-DAB converter based on the frequency domain analysis of a topological structure of the SR-DAB converter; s2, on the basis of ZVS characteristic analysis of the SR-DAB converter, constructing a ZVS condition model of the SR-DAB converter; s3, based on the general model of the SR-DAB converter and the ZVS condition model, constructing a multi-target constraint model including resonance current RMS optimization, ZVS condition constraint and efficiency maximization; s4, after given power serves as input of the multi-target constraint model, the multi-target constraint model is solved through a genetic algorithm, and an optimal control variable combination of the SR-DAB converter is obtained; and S5, taking the optimal control variable combination as the input of the SR-DAB converter to realize modulation. According to the invention, the full working range efficiency of the SR-DAB converter can be improved.

Description

Technical Field

[0001] The present invention relates to the field of converter modulation and artificial intelligence technology, and in particular to a full soft switching wide-range efficiency optimization method for an isolated SR-DAB converter. Background Art

[0002] With the rapid development of renewable energy systems, high-efficiency, high-power-density power electronic converters have become a research focus. Among them, the Series Resonant Dual Active Bridge (SR-DAB) converter is a commonly used converter.

[0003] The main application scenarios of the SR-DAB converter include: 1) Renewable energy generation and energy storage systems: achieving efficient bidirectional energy conversion (charging and discharging) between the DC bus voltage and the energy storage battery; using soft switching to reduce switching losses and adapt to a wide input / output voltage range (such as photovoltaic panel voltage fluctuations or battery charge and discharge characteristics); supporting multi-port integration to coordinate the energy flow of renewable energy, energy storage, and loads. 2) Electric vehicles and charging infrastructure: supporting bidirectional energy transmission and energy exchange between the vehicle battery and the grid (V2G / V2H); high-frequency resonance technology reduces the size of magnetic components and increases power density, suitable for the compact design of on-board equipment; efficiently matching a wide range of battery voltages (such as 400V / 800V platforms) with the voltage of the grid / charging station. 3) DC microgrids and data center power supply: achieving efficient isolation and energy allocation between DC buses of different voltage levels (such as the photovoltaic DC bus, energy storage battery, and load bus); using soft switching to improve system efficiency, reduce heat dissipation requirements, and enhance reliability; supporting the parallel connection of multiple modules to expand power capacity and meet the high-reliability power supply requirements of data centers.

[0004] For SR-DAB, the modulation method determines the efficiency of the converter. Currently, there are three main modulation methods: traditional variable frequency modulation strategy, modulation harmonic burst control strategy, and multi-degree-of-freedom modulation strategy.

[0005] 1) Traditional variable-frequency modulation strategies can change the impedance characteristics of the resonant tank by adjusting the switching frequency, thereby reducing circulating current and improving efficiency under variable load conditions. However, as the port voltage gain increases or the transmitted power decreases, the switching frequency increases nonlinearly. This not only increases control complexity and complicates the design of magnetic components under wide frequency variations, but also exacerbates losses caused by parasitic effects in the high-frequency range. Therefore, the operating range of SR-DAB using variable-frequency modulation is generally significantly limited. 2) Harmonic burst control strategies select different harmonic components for power transmission and combine burst control techniques to achieve soft switching (ZVS) over a wide load range while reducing pulse power. Experiments show that this method can significantly improve efficiency under light load conditions, but the efficiency optimization effect is limited when the voltage gain deviates from unity or in the mid-power range. 3) Multi-degree-of-freedom modulation strategies often sacrifice some ZVS capability in the pursuit of circulating current suppression and efficiency optimization, resulting in increased hard-switching losses in specific modes, which in turn weakens the overall efficiency improvement.

[0006] In summary, how to design a method to improve the efficiency of the SR-DAB converter in the full operating range is a technical problem that needs to be solved urgently. Summary of the Invention

[0007] In view of the deficiencies of the above-mentioned prior art, the technical problem to be solved by the present invention is: how to provide a full soft-switching wide-range efficiency optimization method for an isolated SR-DAB converter, by modeling a multi-objective constraint model that comprehensively considers the RMS optimization of the resonant current, the ZVS condition constraint and the efficiency maximization, and adjusting the control variables under a wide range of power input and output conditions to minimize the RMS of the resonant current, meet the ZVS condition and achieve efficiency maximization, so that the SR-DAB converter can achieve efficient and stable modulation in a wider power range, thereby improving the efficiency of the SR-DAB converter over the entire operating range.

[0008] In order to solve the above technical problems, the present invention adopts the following technical solutions:

[0009] A full soft-switching wide-range efficiency optimization method for an isolated SR-DAB converter, comprising:

[0010] S1: Based on the frequency domain analysis of the SR-DAB converter topology, a general model of the SR-DAB converter is constructed;

[0011] S2: Based on the ZVS characteristics analysis of the SR-DAB converter, a ZVS condition model of the SR-DAB converter is constructed;

[0012] S3: Based on the general model of the SR-DAB converter and the ZVS condition model, a multi-objective constraint model is constructed, which includes resonant current RMS optimization, ZVS condition constraint, and efficiency maximization.

[0013] S4: After taking the given power as the input of the multi-objective constraint model, the multi-objective constraint model is solved by genetic algorithm to obtain the optimal control variable combination of the SR-DAB converter;

[0014] S5: The optimal control variable combination is used as the input of the SR-DAB converter to achieve modulation.

[0015] Preferably, in step S1, the topology of the SR-DAB converter includes a main H-bridge arranged on the primary side, a secondary H-bridge arranged on the secondary side, and a transformer and a resonant circuit arranged between the main H-bridge and the secondary H-bridge;

[0016] The main H-bridge is an H-bridge composed of a first primary upper switch tube S1 and a first primary lower switch tube S2 connected in series, and a second primary upper switch tube S3 and a second primary lower switch tube S4 connected in parallel;

[0017] The secondary H-bridge is an H-full bridge consisting of a first secondary upper switch Q1 and a first secondary lower switch Q2 connected in series, a second secondary upper switch Q3 and a second secondary lower switch Q4 connected in parallel;

[0018] The resonant circuit is connected between the main H-bridge and the transformer and includes a series resonant inductor L r and resonant capacitor C r .

[0019] Preferably, in step S1, the formula of the general model of the SR-DAB converter is expressed as:

[0020]

[0021]

[0022] Where: I rms represents the RMS value of the primary-side inductor current of the SR-DAB converter; P represents the power of the SR-DAB converter in one switching cycle; V1 represents the input voltage of the SR-DAB converter; n represents the harmonic order of the SR-DAB converter; V2 represents the output voltage of the SR-DAB converter; ω represents the angular frequency; L r Represents the resonant inductance; C r represents the resonant capacitor; α1 represents the internal phase shift angle between the switches S1 and S4 of the main H-bridge of the SR-DAB converter; α2 represents the internal phase shift angle between the switches Q1 and Q4 of the auxiliary H-bridge of the SR-DAB converter; β represents the external phase shift angle between the main H-bridge and the auxiliary H-bridge of the SR-DAB converter.

[0023] Preferably, in step S1, the processing steps of constructing a general model of the SR-DAB converter include:

[0024] S101: Derivation of the primary side voltage v of the SR-DAB converter by Fourier transform h1 and secondary voltage v h2 ;

[0025] Formula 1 is expressed as:

[0026]

[0027] Where: ω s =2πf s , f s represents the switching frequency; α1 represents the internal phase shift angle between the switches S1 and S4 of the main H-bridge of the SR-DAB converter; α2 represents the internal phase shift angle between the switches Q1 and Q4 of the auxiliary H-bridge of the SR-DAB converter; β represents the external phase shift angle between the main H-bridge and the auxiliary H-bridge of the SR-DAB converter; V1 represents the input voltage of the SR-DAB converter; n represents the harmonic order of the SR-DAB converter; V2 represents the output voltage of the SR-DAB converter; t represents time;

[0028] S102: Calculate the phase of the primary and secondary voltage difference using formula 1

[0029] Formula 2 is expressed as:

[0030]

[0031] in:

[0032]

[0033] S103: Phase quantity based on primary and secondary voltage difference Calculate the phasors of the inductor current and capacitor voltage;

[0034] Formula 4 is expressed as:

[0035]

[0036] Where: ω represents the angular frequency; L r Represents the resonant inductance; C r represents the resonant capacitance;

[0037] Formula 5 is expressed as:

[0038]

[0039] Where: represents the phasor of the inductor current;

[0040] S104: Calculate the inductor current and the capacitor voltage based on Formula 4 and Formula 5;

[0041] Formula 6 is expressed as:

[0042]

[0043] Formula 7 is expressed as:

[0044]

[0045] Where:

[0046] S105: Derived the RMS value of the primary-side inductor current using Formula 6;

[0047] Formula 8 is expressed as:

[0048]

[0049] S106: Calculate the power of the SR-DAB converter in one switching cycle using Formula 6 and Formula 7. Formula 9 is expressed as:

[0050]

[0051] Preferably, in step S2, the formula of the ZVS condition model of the SR-DAB converter is expressed as:

[0052] 1) ZVS condition model for case 1

[0053]

[0054] Where: i Lx represents the resonant cavity current; k0, k1, k2, ω c Indicates the coefficient; X1 represents the switch tube S1 of the main H-bridge or the switch tube Q1 of the auxiliary H-bridge; L thx and C thx Indicates the equivalent inductance and current on the non-commutating side. When the primary side switch is in operation: C thx =C r ,L thx =L r ; When the secondary side switch tube is in action: C thx =C r *N 2 , L thx =L r / N 2 ; V thx Indicates the equivalent voltage on the non-commutating side, when the primary side switch is in operation: V thx =N*V h2 , when the secondary side switch tube is in action: V thx =V h1 / N;C ossRepresents the parasitic capacitance of the switching device; V Cx Indicates the capacitor voltage. When the primary side is in action: X is 1, when the secondary side is in action: X is 2; V x Indicates input or output voltage; I Lx represents the inductor current;

[0055] 2) ZVS condition model for case 2

[0056]

[0057] Where: X4 represents the switch tube S4 of the main H-bridge or the switch tube Q4 of the auxiliary H-bridge;

[0058] 3) ZVS condition model for case 3

[0059]

[0060] Preferably, in step S3, the formula of the multi-objective constraint model is expressed as:

[0061]

[0062] Preferably, in step S4, the optimal control variable combination of the SR-DAB converter includes, under given power conditions, the internal phase shift angle α1 between the switches S1 and S4 of the main H-bridge of the SR-DAB converter, the internal phase shift angle α2 between the switches Q1 and Q4 of the auxiliary H-bridge, the external phase shift angle β between the main H-bridge and the auxiliary H-bridge, and the switching frequency f s Four control variables.

[0063] Preferably, in the process of solving the multi-objective constraint model by the genetic algorithm, a control variable combination, ie, the value of the control variable, is found under given power conditions so that the inductor current reaches the minimum value and satisfies the ZVS condition.

[0064] Preferably, in step S4, the genetic algorithm used is the PSO algorithm; the PSO algorithm is used to solve the multi-objective constraint model with the control variables of the SR-DAB converter as input to obtain the optimal control variable combination

[0065] Compared with the prior art, the full soft-switching wide-range efficiency optimization method of the isolated SR-DAB converter in the present invention has the following beneficial effects:

[0066] The present invention constructs a universal model based on the frequency domain analysis of the topological structure of the SR-DAB converter. The frequency domain analysis can accurately capture the steady-state characteristics of the SR-DAB converter at different frequencies, thereby providing accurate basic data for the establishment of the subsequent multi-objective constraint model. In addition, the universal model can accurately reflect the parameter changes of the converter under different working conditions, so that the description of each target in the construction process of the multi-objective constraint model is more accurate, avoiding the deviation of the optimization results caused by inaccurate models, thereby improving the modeling accuracy of the multi-objective constraint model. At the same time, the universal model covers a variety of operating modes of the SR-DAB converter and has a wide range of applicability, so that various actual factors can be incorporated into the model when constructing the multi-objective constraint model, thereby reasonably setting the constraints and optimization targets in the multi-objective constraint model, ensuring that the model can be effectively optimized according to the actual operating conditions, and improving the effectiveness of the multi-objective constraint model in practical applications.

[0067] The present invention constructs a ZVS conditional model based on the ZVS characteristic analysis of the SR-DAB converter. The ZVS conditional model clarifies the specific conditions and boundaries for the SR-DAB converter to achieve zero voltage switching, provides key constraints for the multi-objective constraint model, and avoids sacrificing the ZVS characteristic due to blind pursuit of other objectives, thereby ensuring the feasibility and reliability of the optimization results of the multi-objective constraint model. At the same time, by introducing the ZVS conditional constraint into the multi-objective constraint model, the optimization algorithm can be guided to find the optimal modulation parameters while satisfying the ZVS condition, thereby further improving the modulation efficiency of the SR-DAB converter.

[0068] The present invention constructs a multi-objective constraint model based on the general model and ZVS condition model of the SR-DAB converter, which includes resonant current RMS optimization, ZVS condition constraints and efficiency maximization. The multi-objective constraint model integrates the general model and the ZVS condition model. The general model provides accurate converter performance data, and the ZVS condition model ensures the rationality of the optimization process, so that the multi-objective constraint model can accurately reflect the actual operation of the converter during the modeling process and reasonably set the weights and constraints of each objective. At the same time, the multi-objective constraint model of the modeling takes into account the resonant current RMS optimization and ZVS condition constraints to achieve the ultimate goal of maximizing efficiency. Under a wide range of power input and output conditions, the model can adjust the resonant current RMS according to actual conditions, meet the ZVS conditions and achieve efficiency maximization, so that the SR-DAB converter can achieve efficient and stable modulation within a wider power range, thereby expanding the operating range and efficiency of the SR-DAB converter and improving the efficiency of the SR-DAB converter over the entire operating range. In addition, solving the model by genetic algorithm can obtain more accurate, effective and reasonable optimization results, providing a reliable basis for the modulation control of the SR-DAB converter; the parameter mapping table generated by the PSO algorithm greatly reduces the real-time calculation complexity while maintaining global optimality.

[0069] The control variable combination generated by the multi-objective constraint model in the present invention includes the internal phase shift angle α1 between the switches S1 and S4 of the main H-bridge of the SR-DAB converter, the internal phase shift angle α2 between the switches Q1 and Q4 of the auxiliary H-bridge, the external phase shift angle β between the main H-bridge and the auxiliary H-bridge, and the switching frequency f s By introducing four degrees of freedom (DOF) coordinated control of the intra-bridge phase-shift angle, inter-bridge phase-shift angle, and switching frequency, the system constructs a multi-dimensional optimization space. This fully exploits the steady-state performance potential of the SR-DAB converter and achieves a dynamic balance between circulating current suppression and ZVS conditions by dynamically adjusting the coupling relationship between these parameters. Furthermore, a multi-DOF optimization approach combining TPS and adaptive VF regulation based on the phase-shift angle and frequency is used to modulate and optimize the resonant current waveform to reduce the RMS value and suppress reactive circulating current. Furthermore, variable frequency modulation is combined with the resonant cavity impedance characteristics to ensure soft switching over a wide voltage gain range. BRIEF DESCRIPTION OF THE DRAWINGS

[0070] In order to make the purpose, technical solutions and advantages of the invention more clear, the present invention will be further described in detail below with reference to the accompanying drawings, in which:

[0071] Figure 1 Flowchart of the full soft-switching wide-range efficiency optimization method for isolated SR-DAB converter.

[0072] Figure 2This is the topology diagram of the SR-DAB converter.

[0073] Figure 3 This is a typical waveform of the SR-DAB converter.

[0074] Figure 4 is the Thevenin equivalent model of the SR-DAB converter.

[0075] Figure 5 The transient commutation process of three cases: Figure 5 (a) Case 1: Only X1 is turned on; Figure 5 (b) Case 2: Only X4 is open; Figure 5 (c) Case 3: X1 and X4 are turned on at the same time.

[0076] Figure 6 The commutation process at different stages: Figure 6 (a) Phase 1; Figure 6 (b) Phase II; Figure 6 (c) Phase III.

[0077] Figure 7 This is the flow chart of the PSO algorithm.

[0078] Figure 8 The ZVS range of the four modulation schemes is: Figure 8 (a) Frequency conversion; Figure 8 (b) single phase shift; Figure 8 (c) three-phase shift; Figure 8 (d) Three-phase shift + frequency conversion.

[0079] Figure 9 The RMS current curves of the four modulation schemes are: Figure 9 (a) M = 0.4; Figure 9 (b)M=0.8.

[0080] Figure 10 The typical working waveform when P=300W: Figure 10 (a) VF; Figure 10 (b) SPS; Figure 10 (c)TPS; Figure 10 (d) The modulation method proposed by the present invention.

[0081] Figure 11 The typical working waveform when P=600W: Figure 11 (a) VF; Figure 11 (b) SPS; Figure 11 (c)TPS; Figure 11 (d) The modulation method proposed by the present invention.

[0082] Figure 12The switching waveform of the solution proposed in this invention when P=600W: Figure 12 (a) Waveforms of primary side switches S1 and S4; Figure 12 (b) Waveforms of secondary side switches Q1 and Q4.

[0083] Figure 13 The efficiency curves of the four schemes are shown in Figure 2. DETAILED DESCRIPTION

[0084] In order to make the purpose, technical solutions and advantages of the embodiments of the present invention clearer, the technical solutions in the embodiments of the present invention will be clearly and completely described below in conjunction with the drawings in the embodiments of the present invention. Obviously, the described embodiments are part of the embodiments of the present invention, not all of the embodiments. The components of the embodiments of the present invention generally described and shown in the drawings here can be arranged and designed in various different configurations. Therefore, the following detailed description of the embodiments of the present invention provided in the drawings is not intended to limit the scope of the invention claimed for protection, but only represents selected embodiments of the present invention. Based on the embodiments in the present invention, all other embodiments obtained by ordinary technicians in this field without making creative work are within the scope of protection of the present invention.

[0085] It should be noted that similar reference numerals and letters denote similar items in the following figures. Therefore, once an item is defined in one figure, it does not require further definition or explanation in subsequent figures. In the description of the present invention, it should be noted that the terms "center," "upper," "lower," "left," "right," "vertical," "horizontal," "inner," and "outer" indicate positions or relationships based on the positions or relationships shown in the figures, or the positions or relationships in which the inventive product is typically placed when in use. These terms are intended solely to facilitate the description of the present invention and simplify the description. They do not indicate or imply that the devices or components referred to must have a specific orientation, be constructed, or operate in a specific orientation, and are therefore not to be construed as limiting the present invention. Furthermore, the terms "first," "second," and "third," etc., are used solely to distinguish descriptions and are not to be construed as indicating or implying relative importance. Furthermore, terms such as "horizontal" and "vertical" do not imply that a component is absolutely horizontal or overhanging, but rather may be slightly tilted. For example, "horizontal" simply means that its direction is more horizontal than "vertical," and does not mean that the structure must be completely horizontal, but rather may be slightly tilted. In the description of the present invention, it should also be noted that, unless otherwise expressly specified or limited, the terms "disposed," "installed," "connected," and "connected" should be understood in a broad sense. For example, they may refer to fixed connections, detachable connections, or integral connections; they may refer to mechanical connections or electrical connections; they may refer to direct connections or indirect connections through an intermediate medium; and they may refer to internal communication between two components. Those skilled in the art will understand the specific meanings of the above terms in the present invention based on the specific circumstances.

[0086] The following is a further detailed description through specific implementation methods:

[0087] Example:

[0088] This embodiment discloses a full soft switching wide range efficiency optimization method for an isolated SR-DAB converter.

[0089] like Figure 1 As shown in Figure 1, the full soft-switching wide-range efficiency optimization method for the isolated SR-DAB converter includes:

[0090] S1: Based on the frequency domain analysis of the SR-DAB converter topology, a general model of the SR-DAB converter is constructed;

[0091] S2: Based on the ZVS characteristics analysis of the SR-DAB converter, a ZVS condition model of the SR-DAB converter is constructed;

[0092] S3: Based on the general model of the SR-DAB converter and the ZVS condition model, a multi-objective constraint model is constructed, which includes resonant current RMS (Root Mean Square) optimization, ZVS condition constraint, and efficiency maximization.

[0093] S4: After taking the given power as the input of the multi-objective constraint model, the multi-objective constraint model is solved by genetic algorithm to obtain the optimal control variable combination of the SR-DAB converter;

[0094] Among them, the optimal control variable combination includes the internal phase shift angle α1 between the switches S1 and S4 of the main H-bridge of the SR-DAB converter, the internal phase shift angle α2 between the switches Q1 and Q4 of the auxiliary H-bridge, the external phase shift angle β between the main H-bridge and the auxiliary H-bridge, and the switching frequency f under given power conditions. s Four control variables.

[0095] S5: The optimal control variable combination is used as the input of the SR-DAB converter to achieve modulation.

[0096] The present invention constructs a universal model based on the frequency domain analysis of the topological structure of the SR-DAB converter. The frequency domain analysis can accurately capture the steady-state characteristics of the SR-DAB converter at different frequencies, thereby providing accurate basic data for the establishment of the subsequent multi-objective constraint model. In addition, the universal model can accurately reflect the parameter changes of the converter under different working conditions, so that the description of each target in the construction process of the multi-objective constraint model is more accurate, avoiding the deviation of the optimization results caused by inaccurate models, thereby improving the modeling accuracy of the multi-objective constraint model. At the same time, the universal model covers a variety of operating modes of the SR-DAB converter and has a wide range of applicability, so that various actual factors can be incorporated into the model when constructing the multi-objective constraint model, thereby reasonably setting the constraints and optimization targets in the multi-objective constraint model, ensuring that the model can be effectively optimized according to the actual operating conditions, and improving the effectiveness of the multi-objective constraint model in practical applications.

[0097] The present invention constructs a ZVS conditional model based on the ZVS characteristic analysis of the SR-DAB converter. The ZVS conditional model clarifies the specific conditions and boundaries for the SR-DAB converter to achieve zero voltage switching, provides key constraints for the multi-objective constraint model, and avoids sacrificing the ZVS characteristic due to blind pursuit of other objectives, thereby ensuring the feasibility and reliability of the optimization results of the multi-objective constraint model. At the same time, by introducing the ZVS conditional constraint into the multi-objective constraint model, the optimization algorithm can be guided to find the optimal modulation parameters while satisfying the ZVS condition, thereby further improving the modulation efficiency of the SR-DAB converter.

[0098] The present invention constructs a multi-objective constraint model based on the general model and ZVS condition model of the SR-DAB converter, which includes resonant current RMS optimization, ZVS condition constraints and efficiency maximization. The multi-objective constraint model integrates the general model and the ZVS condition model. The general model provides accurate converter performance data, and the ZVS condition model ensures the rationality of the optimization process, so that the multi-objective constraint model can accurately reflect the actual operation of the converter during the modeling process and reasonably set the weights and constraints of each objective. At the same time, the multi-objective constraint model of the modeling takes into account the resonant current RMS optimization and ZVS condition constraints to achieve the ultimate goal of maximizing efficiency. Under a wide range of power input and output conditions, the model can adjust the resonant current RMS according to actual conditions, meet the ZVS conditions and achieve efficiency maximization, so that the SR-DAB converter can achieve efficient and stable modulation within a wider power range, thereby expanding the operating range and efficiency of the SR-DAB converter and improving the efficiency of the SR-DAB converter over the entire operating range. In addition, solving the model by genetic algorithm can obtain more accurate, effective and reasonable optimization results, providing a reliable basis for the modulation control of the SR-DAB converter; the parameter mapping table generated by the PSO algorithm greatly reduces the real-time calculation complexity while maintaining global optimality.

[0099] The control variable combination generated by the multi-objective constraint model in the present invention includes the internal phase shift angle α1 between the switches S1 and S4 of the main H-bridge of the SR-DAB converter, the internal phase shift angle α2 between the switches Q1 and Q4 of the auxiliary H-bridge, the external phase shift angle β between the main H-bridge and the auxiliary H-bridge, and the switching frequency f s By introducing four degrees of freedom (DOF) of coordinated control of the intra-bridge phase shift angle, inter-bridge phase shift angle, and switching frequency, the four control variables are used to construct a multi-dimensional optimization space. This fully exploits the steady-state performance potential of the SR-DAB converter and achieves a dynamic balance between circulating current suppression and ZVS conditions by dynamically adjusting the coupling relationship between these parameters. Furthermore, based on the phase shift angle and frequency, a multi-DOF optimization method combining TPS (Triangular Phase Shift Modulation) with adaptive VF regulation (Variable Frequency) is used to modulate and optimize the resonant current waveform to reduce the RMS value and suppress reactive circulating current. Furthermore, variable frequency regulation of the resonant cavity impedance characteristics is combined to ensure soft switching over a wide voltage gain range.

[0100] 1. SR-DAB Converter

[0101] like Figure 2 As shown, the topology of the SR-DAB converter includes a main H-bridge arranged on the primary side, a secondary H-bridge arranged on the secondary side, and a transformer and a resonant circuit arranged between the main H-bridge and the secondary H-bridge;

[0102] The main H-bridge is an H-bridge composed of a first primary upper switch tube S1 and a first primary lower switch tube S2 connected in series, and a second primary upper switch tube S3 and a second primary lower switch tube S4 connected in parallel;

[0103] The secondary H-bridge is an H-full bridge consisting of a first secondary upper switch Q1 and a first secondary lower switch Q2 connected in series, a second secondary upper switch Q3 and a second secondary lower switch Q4 connected in parallel;

[0104] The transformer's magnetizing inductance is L m ;

[0105] The resonant circuit is connected between the main H-bridge and the transformer and includes a series resonant inductor L r and resonant capacitor C r .

[0106] 2. General Model of SR-DAB Converter

[0107] Typical DC-DC waveforms of SR-DAB converters are as follows: Figure 3 As shown. α1 represents the internal phase shift angle between the switches S1 and S4 of the main H-bridge of the SR-DAB converter; α2 represents the internal phase shift angle between the switches Q1 and Q4 of the auxiliary H-bridge of the SR-DAB converter; β represents the external phase shift angle between the main H-bridge and the auxiliary H-bridge of the SR-DAB converter. Note that β not only represents v h1 and v h2 The phase shift angle between the center points also represents the phase shift angle between the corresponding sinusoidal components. α0 is the phase shift angle between the two bridges, and α0 = β + (α1 - α2) / 2.

[0108] In this embodiment, the processing steps for constructing a general model of the SR-DAB (DC-DC) converter include:

[0109] S101: Compared with the time domain analysis method, the frequency domain method can establish a unified mathematical model through Fourier transform, avoiding the complex modal classification problem in time domain analysis. It is more suitable for the system modeling and performance verification of the four-variable hybrid modulation scheme proposed in this paper. The primary side voltage v of the SR-DAB converter is derived (generally) through Fourier transform. h1 and secondary voltage v h2 ;

[0110] Formula 1 is expressed as:

[0111]

[0112] Where: ω s =2πf s , f srepresents the switching frequency; α1 represents the internal phase shift angle between the switches S1 and S4 of the main H-bridge of the SR-DAB converter; α2 represents the internal phase shift angle between the switches Q1 and Q4 of the auxiliary H-bridge of the SR-DAB converter; β represents the external phase shift angle between the main H-bridge and the auxiliary H-bridge of the SR-DAB converter; V1 represents the input voltage of the SR-DAB converter; n represents the harmonic order of the SR-DAB converter; V2 represents the output voltage of the SR-DAB converter; t represents time;

[0113] S102: Calculate the phase of the primary and secondary voltage difference using formula 1

[0114] Formula 2 is expressed as:

[0115]

[0116] in:

[0117]

[0118] S103: Phase quantity based on primary and secondary voltage difference Calculate the phasors of the inductor current and capacitor voltage;

[0119] Formula 4 is expressed as:

[0120]

[0121] Where: ω represents the angular frequency; L r Represents the resonant inductance; C r represents the resonant capacitance;

[0122] Formula 5 is expressed as:

[0123]

[0124] Where: Indicates the phasor of the inductor current; j represents a complex number, and j is used instead of i in the circuit to represent a complex number (because the letter for current is i);

[0125] S104: Calculate the inductor current and the capacitor voltage based on Formula 4 and Formula 5;

[0126] Formula 6 is expressed as:

[0127]

[0128] Formula 7 is expressed as:

[0129]

[0130] Where:

[0131] S105: Derived the RMS value of the primary-side inductor current using Formula 6;

[0132] Formula 8 is expressed as:

[0133]

[0134] S106: Calculate the power of the SR-DAB converter in one switching cycle using Formula 6 and Formula 7;

[0135] Formula 9 is expressed as:

[0136]

[0137] Finally, the general model of the SR-DAB converter is formulated as:

[0138]

[0139]

[0140] Where: I rms represents the RMS value of the primary-side inductor current of the SR-DAB converter; P represents the power of the SR-DAB converter in one switching cycle; V1 represents the input voltage of the SR-DAB converter; n represents the harmonic order of the SR-DAB converter; V2 represents the output voltage of the SR-DAB converter; ω represents the angular frequency; L r Represents the resonant inductance; C r represents the resonant capacitor; α1 represents the internal phase shift angle between the switches S1 and S4 of the main H-bridge of the SR-DAB converter; α2 represents the internal phase shift angle between the switches Q1 and Q4 of the auxiliary H-bridge of the SR-DAB converter; β represents the external phase shift angle between the main H-bridge and the auxiliary H-bridge of the SR-DAB converter.

[0141] 3. ZVS Condition Model of SR-DAB Converter

[0142] An SRC DAB converter can perform up to eight switching operations within a switching cycle. Analyzing and solving the transient commutation process for each switching operation within a switching cycle would be cumbersome. Therefore, to simplify the analysis of the switching device commutation process, the SRC DAB converter can be subjected to the Thevenin equivalent.

[0143] According to Thevenin's theorem, a linear two-port network consisting of an independent voltage source and a resistor can be represented by a voltage source in series with an impedance. Figure 4 The equivalent circuit of port x is given. Port x refers to the port where the switch tube is commutating. V x is the voltage source at the commutation end, V thx and i Lxare the AC square wave voltage and inductor current at the commutation end, L thx and V thx Represent the equivalent impedance and voltage at the non-commutation end respectively. For example, when the switch device at port 1 of the SRC DAB converter is commutating, x = 1, the switch devices X1-X4 are S1-S4 respectively, and V x =V1,v hx =v h1 ,i Lx =i L1 , C thx =C r , L thx =L r , V thx =N*V h2 When the switching device of port 2 of the SRC DAB converter is commutating, x=1, and the switching devices X1-X4 are Q1-Q4 respectively, V x =V2,v hx =v h2 ,i Lx =i L2 , C thx =N 2 *C r , L thx =L r / N 2 , V thx =V h1 / N.

[0144] Combine Figure 4 Based on the Thevenin equivalent circuit, this example analyzes and solves the commutation process and ZVS activation conditions for X1-X4. Because the upper and lower switching devices in the same bridge arm conduct complementary to each other, if one switching device achieves ZVS activation, the other switching device in the same bridge arm will also achieve ZVS activation. Therefore, the following analysis focuses only on the commutation process for X1 and X4 within half a switching cycle. Within half a switching cycle, the switching devices in HB can switch in three different ways. Figure 5 The transient commutation process of these three switching methods is demonstrated. Figure 5 (a) is the case where X1 is on, X2 is off, and X3 and X4 are inactive. Figure 5 (b) is the case where X4 is turned on, X3 is turned off, and X1 and X2 are not in action. Figure 5 (c) is the case where X1 and X4 are turned on at the same time, and X2 and X3 are turned off at the same time.

[0145] Below Figure 5 Taking the transient commutation process of X1 and X2 in (a) as an example, the critical current condition for the switching device to achieve ZVS turn-on is solved. The reference direction of the relevant electrical quantities is shown by the red arrow, and the orange dashed line shows the actual current flow direction. Figure 5 (a) gives the dead time t d Inside, the gate-source voltage of X1 and X2 is v gs_X1 and v gs_X2 , drain-source voltage v ds_X1 And the inductor current i Lx changes in the situation.

[0146] The turn-off time of X2 is defined as time 0, and X1 is turned on after the dead time. Figure 6 The main stages of the transient commutation process of X1 and X2 during the dead time are given accordingly:

[0147] (1) Phase 1 Figure 6 (a)]: Before t = 0, switches X2 and X3 are turned on, X1 and X4 are turned off, and the inductor current i Lx <0.

[0148] (2) Phase 2 Figure 6 (b)]:At t=0, the switch X2 is turned off. Inductor L thx and parasitic capacitance C oss1 、C oss2 Resonance occurs, current i Lx C oss1 Discharge, voltage v ds_X1 Decreases; at the same time, the current i Lx C oss2 Charging, voltage v ds_X2 When t=t1, the voltage vd s_X1 drops to 0 and v ds_X2 Rising to V x , the commutation process of phase 2 ends.

[0149] (3) Stage 3 Figure 6 (c)]: At t = t1, the body diode D1 of X1 is turned on, and the current i Lx The current continues through D1 and X3. At this time, the drain-source voltage of X1 is v ds = 0. ZVS can only be achieved if X1 is turned on during this phase.

[0150] Here, the loss caused by the parasitic capacitance charging and discharging process is ignored, and it is assumed that the total energy stored in the parasitic capacitance is constant before and after commutation. Therefore, the energy consumption of the switching device during the commutation process needs to be provided by the resonant cavity. Based on this, the realization of ZVS requires not only the correct direction of the resonant current (i.e., the direction of the resonant current). Lx <0); there must be enough energy in the resonant cavity to ensure that the drain-source voltage v ds_X1 It can drop to 0. The energy conditions for achieving ZVS are solved below.

[0151] Kirchhoff's current law, Figure 6 The current relationship of (a) can be expressed as:

[0152]

[0153] where v ds_X1 and v ds_X2 is the drain-source voltage of X1 and X2. Lx is the resonant cavity current, v Cx is the resonant capacitor voltage in the resonant cavity. Usually the switch device models in the same HB are the same, so the parasitic capacitance values ​​of X1 to X4 are uniformly expressed as C oss Due to the parasitic capacitance C of the switching device oss With voltage v ds The change is large and nonlinear. To improve the accuracy of the ZVS condition, based on the charge equivalence principle, the change of parasitic capacitance is linearized and C is derived. oss With v ds Taking the SCT3060 device as an example, combined with the device manual, its parasitic capacitance can be fitted as:

[0154]

[0155] It is known that at time t=0, the initial value of each state variable is i Lx (0)=I Lx (0), v Cx (0) = V Cx (0), v ds_x2 (0)=0, combined with the initial value conditions, solving the state equation can be obtained:

[0156]

[0157] Among them, the coefficients k0, k1, k2, ω c They are

[0158]

[0159] As mentioned above, if there is enough energy in the resonant cavity to make vds_X1 drop to 0, then the switch device X1 can be soft-turned on. Therefore, based on the above formula, the energy condition for achieving ZVS is:

[0160]

[0161] Based on the same idea, the ZVS current constraints for the action of switch device X4 and the simultaneous action of switch devices X1 and X4 can be derived. The ZVS condition models for the three switching actions of the SRC DAB converter are summarized as follows:

[0162] 1) ZVS condition model for case 1

[0163]

[0164] Where: i Lx represents the resonant cavity current; k0, k1, k2, ω c represents the coefficient; X1 represents the switch tube S1 (primary side switch tube 1) of the main H-bridge or the switch tube Q1 (secondary side switch tube 1) of the secondary H-bridge. Since the Thevenin equivalent is used in the derivation of the ZVS model, that is, when the primary side switch tube is in operation, the secondary side Thevenin equivalent is used. In this case, X1 is S1; the same applies to the secondary side; L thx and C thx Indicates the equivalent inductance and current on the non-commutating side. When the primary side switch is in operation: C thx =C r , L thx =L r ; When the secondary side switch tube is in action: C thx =C r *N 2 , L thx =L r / N 2 ; V thx Indicates the equivalent voltage on the non-commutating side, when the primary side switch is in operation: V thx =N*V h2 , when the secondary side switch tube is in action: V thx =V h1 / N;C oss Represents the parasitic capacitance of the switching device; V Cx Represents the capacitor voltage. When the primary side is in action: x is 1, when the secondary side is in action: x is 2; V x Indicates input or output voltage; I Lx represents the inductor current;

[0165] 2) ZVS condition model for case 2

[0166]

[0167] Where: X4 represents the switch tube S4 (primary side switch tube 4) of the main H-bridge or the switch tube Q4 (secondary side switch tube 1) of the secondary H-bridge;

[0168] 3) ZVS condition model for case 3

[0169]

[0170] Building on traditional energy condition analysis, this paper comprehensively considers the impact of non-circulating voltage, parasitic capacitance nonlinearity, and capacitor energy storage variations during switching, proposing a precise ZVS constraint model based on the Thevenin equivalent circuit. This comprehensive consideration of non-circulating voltage, parasitic capacitance nonlinearity, and capacitor energy storage variations during switching makes the resulting ZVS constraint more accurate than previous models.

[0171] 4. Multi-objective constraint model

[0172] In order to minimize both switching loss and conduction loss, this embodiment proposes a comprehensive optimization strategy for the SR-DAB converter. The strategy formulates a comprehensive optimization strategy based on four control variables (α1, α2, β and f s ) multi-constraint optimization framework. The mathematical formulation of the optimization problem is as follows. It is worth noting that the analytical expressions for transmission power and effective value contain many harmonic components. Although the more superimposed harmonics, the smaller the calculation error, the inclusion of higher-order harmonics significantly increases the runtime of the PSO algorithm. Therefore, to balance optimization accuracy and computational efficiency, this embodiment selects the sum of the 11th harmonic components (1st, 3rd, 5th, 7th, 9th, and 11th) as the effective value optimization target.

[0173] Specifically, the formula of the multi-objective constraint model is expressed as:

[0174]

[0175] 5. Genetic Algorithm

[0176] In this embodiment, in the process of solving the multi-objective constraint model by the genetic algorithm, a control variable combination that minimizes the inductor current and satisfies the ZVS condition is found under given power conditions, that is, the value of the control variables.

[0177] Specifically, such as Figure 7 As shown, the genetic algorithm used is the PSO (Particle Swarm Optimization) algorithm; the PSO algorithm is used to solve the multi-objective constraint model with the control variables of the SR-DAB converter as input to obtain the optimal control variable combination. However, the present invention does not improve the solution logic of the PSO algorithm.

[0178] 6. Comparison with Typical Modulation Methods

[0179] 1. ZVS range

[0180] The ZVS range distribution reveals the soft switching capability of the SR-DAB circuit under different power and voltage gain conditions. Figure 8 As shown, different colors represent different ZVS implementation situations: pink, green, blue, and orange respectively indicate that 4, 3, 2, and 1 bridge arms can achieve ZVS.

[0181] Experimental results show that traditional SPS and VF modulation strategies have obvious limitations in zero voltage switching capabilities due to their single degree of freedom control mechanism. Specifically, only two pairs of switches in the primary-side full bridge achieve ZVS operation, while the secondary-side switches are always in a hard-switching state, resulting in large switching losses. The uncovered area in the VF reveals the technical defect that the VF modulation cannot work due to the limitation of the gain range. TPS usually only maintains soft switching of 2-3 bridge arms, resulting in negligible efficiency improvement. In contrast, the proposed hybrid TPS combined with frequency scaling achieves a breakthrough in ZVS performance, maintaining stable ZVS operation in the entire working domain of all four bridge arms, such as Figure 8 The triple phase shift + frequency conversion dominates as shown in the pink area.

[0182] 2. Current RMS optimization

[0183] Figure 9 Simulation results reveal the dynamic relationship between the effective current (RMS) and transmission power in DAB circuits for different modulation strategies. Overall, under the two typical operating conditions of M = 0.6 and M = 0.8, the RMS current of all strategies increases approximately linearly with increasing power, consistent with the power transfer characteristics of resonant converters. Comparing the two voltage gain conditions, it can be observed that when M = 0.8, the RMS current of all strategies decreases by approximately 40%. This is due to the change in the resonant cavity impedance characteristics at high voltage gain, which reduces the current amplitude required for the same power transmission.

[0184] 7. Experimental Description

[0185] To test the performance of the SR-DAB converter, a 1 kW laboratory prototype was developed. The system parameters are shown in Table 1.

[0186] Table 1

[0187]

[0188] Figure 10 and Figure 11The steady-state operating waveforms of the four modulation strategies at 300W and 600W are shown. Key indicators including RMS current value and ZVS operation (red highlight indicates ZVS is on, blue highlight indicates non-ZVS is on) are directly marked on the waveform. The comparison of experimental waveforms shows that the multi-objective optimization strategy proposed in the present invention has significant advantages in suppressing RMS current. Under 300W operating conditions, the RMS current of the strategy is 3.34A, which is 0.23A lower than the VF modulation strategy, and 5.56A and 0.38A lower than the fixed-frequency SPS and TPS modulation strategies, respectively. Under 600W conditions, the RMS current of the proposed strategy is 6.51A, which is 5% lower than the VF strategy (6.82A), and 49% (9.72A) and 6.5% (6.93A) lower than the fixed-frequency SPS and TPS modulation strategies, respectively.

[0189] In terms of ZVS implementation, the strategy proposed in this paper ensures that all eight switching devices can complete ZVS operation under 300W and 600W conditions through optimized switching sequence design, effectively reducing switching losses. Comparative analysis shows that: (1) the variable frequency (VF) modulation strategy adopts a single-source modulation method, which causes two secondary-side devices to generate hard switching due to turning on at the current zero point; (2) the fixed-frequency SPS modulation strategy cannot achieve zero voltage switching (ZVS) of the secondary-side devices because its turning-on moment coincides with the negative current polarity; (3) the fixed-frequency TPS modulation strategy causes hard switching in the primary-side lagging bridge arm under 300W conditions, while at 600W, only the two devices in the primary-side leading bridge arm can achieve zero voltage switching (ZVS), and the remaining six devices (four secondary-side devices due to insufficient energy and two primary-side devices due to polarity mismatch) still operate in a hard switching state.

[0190] In order to further verify the zero voltage switch-on (ZVS) capability of the proposed strategy, Figure 12 The measured waveforms of the switching device under the working conditions of M = 0.5 and P = 600W are shown. The inductor current i Lx , gate-source voltage v gs and drain-source voltage v ds The changes in the switching time, where the dead time is set to 150ns. Figure 12 As shown, the gate drive signals of all switching devices are at the corresponding v ds The precise triggering when the voltage drops to zero confirms the successful realization of ZVS activation.

[0191] The results show that through the coordinated control of phase shift and frequency, all switches achieve soft switching, further verifying the effectiveness of the strategy proposed in this invention in maintaining ZVS operation over the full range.

[0192] like Figure 13As shown in Figure 2, experimental results show that the maximum efficiency of the proposed strategy is improved by 24% compared to the SPS strategy, while it is approximately 1.5% and 1.2% higher than the traditional VF and optimized TPS strategies, respectively. These results confirm the effectiveness of the proposed coordinated optimization strategy in expanding the ZVS implementation range, suppressing circulating currents, and improving overall power conversion efficiency.

[0193] Finally, it should be noted that the above embodiments are only used to illustrate the technical solutions of the present invention rather than to limit the technical solutions. Those skilled in the art should understand that modifications or equivalent replacements of the technical solutions of the present invention that do not depart from the purpose and scope of the technical solutions of the present invention should be included in the scope of the claims of the present invention.

Claims

1. A full soft switching wide range efficiency optimization method for an isolated SR-DAB converter, characterized in that: include: S1: Based on the frequency domain analysis of the SR-DAB converter topology, a general model of the SR-DAB converter is constructed; S2: Based on the ZVS characteristics analysis of the SR-DAB converter, a ZVS condition model of the SR-DAB converter is constructed; S3: Based on the general model of the SR-DAB converter and the ZVS condition model, a multi-objective constraint model is constructed, which includes resonant current RMS optimization, ZVS condition constraint, and efficiency maximization. S4: After taking the given power as the input of the multi-objective constraint model, the multi-objective constraint model is solved by genetic algorithm to obtain the optimal control variable combination of the SR-DAB converter; S5: The optimal control variable combination is used as the input of the SR-DAB converter to achieve modulation.

2. The method for optimizing the efficiency of an isolated SR-DAB converter over a wide range using full soft switching according to claim 1, wherein: In step S1, the topology of the SR-DAB converter includes a main H-bridge arranged on the primary side, a secondary H-bridge arranged on the secondary side, and a transformer and a resonant circuit arranged between the main H-bridge and the secondary H-bridge; The main H-bridge is an H-bridge composed of a first primary upper switch tube S1 and a first primary lower switch tube S2 connected in series, and a second primary upper switch tube S3 and a second primary lower switch tube S4 connected in parallel; The secondary H-bridge is an H-full bridge consisting of a first secondary upper switch Q1 and a first secondary lower switch Q2 connected in series, a second secondary upper switch Q3 and a second secondary lower switch Q4 connected in parallel; The resonant circuit is connected between the main H-bridge and the transformer and includes a series resonant inductor L r and resonant capacitor C r .

3. The method for optimizing the efficiency of an isolated SR-DAB converter with full soft switching over a wide range according to claim 1, wherein: In step S1, the formula of the general model of the SR-DAB converter is expressed as: Where: I rms represents the RMS value of the primary-side inductor current of the SR-DAB converter; P represents the power of the SR-DAB converter in one switching cycle; V1 represents the input voltage of the SR-DAB converter; n represents the harmonic order of the SR-DAB converter; V2 represents the output voltage of the SR-DAB converter; ω represents the angular frequency; L r Represents the resonant inductance; C r represents the resonant capacitor; α1 represents the internal phase shift angle between the switches S1 and S4 of the main H-bridge of the SR-DAB converter; α2 represents the internal phase shift angle between the switches Q1 and Q4 of the auxiliary H-bridge of the SR-DAB converter; β represents the external phase shift angle between the main H-bridge and the auxiliary H-bridge of the SR-DAB converter.

4. The method for optimizing the efficiency of an isolated SR-DAB converter with full soft switching over a wide range according to claim 3, wherein: In step S1, the processing steps for constructing a general model of the SR-DAB converter include: S101: Derivation of the primary side voltage v of the SR-DAB converter by Fourier transform h1 and secondary voltage v h2 ; Formula 1 is expressed as: Where: ω s =2πf s , f s represents the switching frequency; α1 represents the internal phase shift angle between the switches S1 and S4 of the main H-bridge of the SR-DAB converter; α2 represents the internal phase shift angle between the switches Q1 and Q4 of the auxiliary H-bridge of the SR-DAB converter; β represents the external phase shift angle between the main H-bridge and the auxiliary H-bridge of the SR-DAB converter; V1 represents the input voltage of the SR-DAB converter; n represents the harmonic order of the SR-DAB converter; V2 represents the output voltage of the SR-DAB converter; t represents time; S102: Calculate the phase of the primary and secondary voltage difference using formula 1 Formula 2 is expressed as: in: S103: Phase quantity based on primary and secondary voltage difference Calculate the phasors of the inductor current and capacitor voltage; Formula 4 is expressed as: Where: ω represents the angular frequency; L r Represents the resonant inductance; C r represents the resonant capacitance; Formula 5 is expressed as: Where: represents the phasor of the inductor current; S104: Calculate the inductor current and the capacitor voltage based on Formula 4 and Formula 5; Formula 6 is expressed as: Formula 7 is expressed as: Where: S105: Derived the RMS value of the primary-side inductor current using Formula 6; Formula 8 is expressed as: S106: Calculate the power of the SR-DAB converter in one switching cycle using Formula 6 and Formula 7; Formula 9 is expressed as:

5. The method for optimizing the efficiency of an isolated SR-DAB converter with full soft switching over a wide range according to claim 3, wherein: In step S2, the formula of the ZVS condition model of the SR-DAB converter is expressed as: 1) ZVS condition model for case 1 Where: i Lx represents the resonant cavity current; k0, k1, k2, ω c Indicates the coefficient; X1 represents the switch tube S1 of the main H-bridge or the switch tube Q1 of the auxiliary H-bridge; L thx and C thx Indicates the equivalent inductance and current on the non-commutating side. When the primary side switch is in operation: C thx =C r ,L thx =L r ; When the secondary side switch tube is in action: C thx =C r *N 2 , L thx =L r / N 2 ; V thx Indicates the equivalent voltage on the non-commutating side, when the primary side switch is in operation: V thx =N*V h2 , when the secondary side switch tube is in action: V thx =V h1 / N;C oss Represents the parasitic capacitance of the switching device; V Cx Indicates the capacitor voltage. When the primary side is in action: X is 1, when the secondary side is in action: X is 2; V x Indicates input or output voltage; I Lx represents the inductor current; 2) ZVS condition model for case 2 Where: X4 represents the switch tube S4 of the main H-bridge or the switch tube Q4 of the auxiliary H-bridge; 3) ZVS condition model for case 3 6. The method for optimizing the efficiency of an isolated SR-DAB converter with full soft switching over a wide range according to claim 5, wherein: In step S3, the formula of the multi-objective constraint model is expressed as: 0≤α1≤π；0≤α2≤π； and Lx (X1)<0;and Lx (X4)<0;and Lx (X1)<0; 7. The method for optimizing the efficiency of an isolated SR-DAB converter with full soft switching over a wide range according to claim 1, wherein: In step S4, the optimal control variable combination of the SR-DAB converter includes the internal phase shift angle α1 between the switches S1 and S4 of the main H-bridge of the SR-DAB converter, the internal phase shift angle α2 between the switches Q1 and Q4 of the auxiliary H-bridge, the external phase shift angle β between the main H-bridge and the auxiliary H-bridge, and the switching frequency f under given power conditions. s Four control variables.

8. The method for optimizing the efficiency of an isolated SR-DAB converter with full soft switching over a wide range according to claim 7, wherein: In the process of solving the multi-objective constraint model through genetic algorithm, the control variable combination that makes the inductor current reach the minimum value and meets the ZVS condition is found under given power conditions.

9. The method for optimizing the efficiency of an isolated SR-DAB converter with full soft switching over a wide range according to claim 1, wherein: In step S4, the genetic algorithm used is the PSO algorithm; the PSO algorithm is used to solve the multi-objective constraint model with the control variables of the SR-DAB converter as input to obtain the optimal control variable combination.

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