Magnetic switch zero-current connection auxiliary circuit for reducing switching loss of pulse generator

Abstract

The invention discloses a magnetic switch zero-current connection auxiliary circuit for reducing switching loss of a pulse generator. The magnetic switch zero-current connection auxiliary circuit comprises a magnetic switch MS, a load RL, an energy storage capacitor Cdc, a direct-current input voltage source Vdc and a driving switch. The single saturable magnetic core is used for achieving current phase regulation and control, a complex topological structure is replaced, the system architecture is simplified, the waveform rising edge is steeped, and waveform distortion is small. According to the invention, the slope of a voltage falling edge is directly optimized through driving resistance adjustment, and zero voltage switching (ZVS) is realized under the condition that an auxiliary resonance circuit or a high-frequency driving chip is not needed. A matching mechanism of the magnetic core saturation threshold value and the load voltage is established, and the soft switching adaptability in a wide load range is ensured. The magnetic switch zero-current connection auxiliary circuit provided by the invention has strong load adaptability, and shows excellent switching loss reduction capability in a range from light load to heavy load.

Description

Technical Field

[0001] The present invention relates to the technical field of pulse power, and in particular to a magnetic switch zero-current turn-on auxiliary circuit for reducing switching loss of a pulse generator. Background Art

[0002] As a non-thermal treatment method, pulsed electric fields (PEF) have been widely used in fields such as biomedicine, plasma science, materials modification, and food science. In the field of anti-icing conductor research, pulsed electric fields can be used as a plasma excitation source, spraying the generated plasma onto the surface of transmission lines to improve the conductor's hydrophobicity and enhance its anti-fouling, anti-corrosion, and self-cleaning capabilities in harsh environments. During plasma treatment, the conductor surface interacts with the high-energy-density plasma, causing changes in the surface chemical bond structure, such as the rearrangement of surface atomic bonds and the formation of new chemical bonds. These changes make the conductor surface more hydrophobic. Furthermore, the conductor surface may be affected by charged particles in the plasma, introducing surface charges. These charges can affect the wetting properties of liquids on the surface. For example, charged particles can attract charged particles in the liquid, causing the liquid to form a more stable film on the surface, thereby enhancing hydrophobicity. Therefore, pulsed electric field-excited plasma has garnered widespread attention in the field of improving the anti-icing properties of conductors.

[0003] As the core device for generating pulsed electric fields and plasma, the parameters and performance of high-voltage pulse power supplies directly affect the content and performance of plasma products. Generally speaking, higher pulsed electric field intensity and frequency produce more stable plasma. However, as the intensity and frequency of the electric field increase, switching losses also increase, exacerbating thermal effects, affecting switching performance and even causing equipment damage.

[0004] In recent years, numerous researchers have conducted extensive research on optimizing switching losses in power electronic devices and systems, focusing on four key areas: control strategy optimization, topology innovation, novel device design, and improved drive technology. However, these approaches rely heavily on multi-switch coordinated control, complex circuit topologies, optimized switching device structures, and complex drive algorithms. This significantly increases system complexity, cost, and debugging difficulties, limiting their application. Summary of the Invention

[0005] The purpose of the present invention is to provide a magnetic switch zero current connection auxiliary circuit for reducing the switching loss of a pulse generator, comprising: a magnetic switch MS, a load R L , energy storage capacitor C dc , DC input voltage source V dc , drive switch.

[0006] The DC input voltage source V dcThe negative terminal is grounded.

[0007] The DC input voltage source V dc One end of the series energy storage capacitor C dc Then, connect to the DC input voltage source V dc the other end.

[0008] The DC input voltage source V dc The positive end of the drive switch, magnetic switch MS, load R are connected in series in sequence. L Then, connect to the DC input voltage source V dc The negative terminal of the

[0009] The magnetic switch zero-current connection auxiliary circuit realizes the steepening of the current rising edge, the sharpening of the voltage falling edge and the current phase regulation through the coordinated regulation of the magnetic switch MS and the drive switch.

[0010] Furthermore, the driving switch includes a single-stage switch and a multi-stage series switch.

[0011] Furthermore, the single-stage switch includes a driving resistor R g , switch S, external power supply V IN .

[0012] The driving resistor R g One end is connected to the external power supply V IN One end is connected to the positive pole of , and the other end is connected to the gate of switch S.

[0013] The external power supply V IN The negative terminal is grounded.

[0014] The switch S is an IGBT switch.

[0015] The collector of the switch S is connected to the DC input voltage source V dc The positive pole of the transistor is located at one end, and the emitter is connected to the magnetic switch MS.

[0016] Furthermore, the multi-stage series switch includes n driving resistors R gi , n switches S i , n connection resistors R bi , where i=1,2,...,n, and n is a positive integer.

[0017] The driving resistor R gi One end is connected to the external power supply, and the other end is connected to the switch S i of the gate.

[0018] The switch S i All use IGBT switches.

[0019] The switch Sk The emitter of the switch S k+1 The collector of , k = 1, 2, ..., n-1.

[0020] The collector of the switch S1 is connected to the DC input voltage source V dc The positive terminal of the switch S n The emitter of is connected to the magnetic switch MS.

[0021] The connection resistance R bi One end of the switch S i The emitter of the switch S i collector.

[0022] Furthermore, the resistance of the driving resistor in the driving switch is adjustable.

[0023] The driving resistor R in the single-stage switch g The smaller the resistance value, the shorter the voltage falling edge time in the switch S turn-on overshoot.

[0024] The driving resistor R of the multi-stage series switch gi The smaller the resistance of the switch S i The shorter the voltage falling edge time in the turn-on overshoot.

[0025] Furthermore, the working state of the magnetic switch MS includes an unsaturated state and a saturated state.

[0026] Furthermore, when the magnetic switch MS works in an unsaturated state, the magnetic switch MS is equivalent to an open circuit, and the DC input voltage source V dc The voltage is transferred from the drive switch to the magnetic switch MS, causing the switch current phase to lag behind the switch voltage phase. The load R L Zero voltage turn-on.

[0027] Furthermore, when the magnetic switch MS works in saturation state, the core volt-second product of the magnetic switch MS reaches the core saturation threshold, and the magnetic switch MS is equivalent to a short circuit. L The load current i L and load voltage V L .

[0028] Furthermore, the calculation formula for the core volt-second product to reach the core saturation threshold is as follows:

[0029]

[0030] Where t is time. MS (t) is the voltage across the magnetic switch MS. N is the number of turns of the magnetic switch MS, A e is the effective cross-sectional area of ​​the core, and ΔB is the change in magnetic induction intensity.

[0031] Furthermore, when the core volt-second product does not reach the core saturation threshold, as the number of turns of the magnetic switch MS increases, the turn-on loss of the magnetic switch MS decreases.

[0032] When the core volt-second product reaches the core saturation threshold, the turn-on loss of the magnetic switch MS no longer decreases with the increase in the number of turns of the magnetic switch MS.

[0033] The turn-on loss of the magnetic switch MS is as follows:

[0034]

[0035] Where t is time, E on is the turn-on loss of the magnetic switch MS. DS 、V DS They are the waveform data of the current and voltage at both ends of the magnetic switch MS changing with time.

[0036] When the turn-on loss of the magnetic switch MS takes the minimum value, the number of turns of the magnetic switch MS is as follows:

[0037]

[0038] Where K is the margin coefficient, T s is the falling edge time of the switching voltage, U is the DC input voltage source V dc The voltage, N is the number of turns of the magnetic switch MS, A e is the effective cross-sectional area of ​​the core, and ΔB is the change in magnetic induction intensity.

[0039] The technical benefits of this invention are undeniable. The present invention proposes a soft-switching method for a load-side saturable magnetic core as a magnetic switch. This method leverages the nonlinear saturation characteristics of the core to achieve current phase lag and steepen the rising edge of the waveform. By varying the drive resistance to control the steepness of the voltage falling edge, the method significantly reduces switching losses and avoids complex topologies and algorithmic dependencies. Compared to existing technologies, the proposed method offers significant advantages in topological complexity, control algorithm complexity, and hardware cost. It is particularly suitable for scenarios with high power density and wide load variations, providing a new paradigm for the design of efficient power electronics systems.

[0040] The present invention utilizes a single saturable magnetic core to achieve current phase regulation, replaces complex topological structures, simplifies system architecture, and steepens the rising edge of the waveform, resulting in less waveform distortion.

[0041] The present invention directly optimizes the voltage falling edge slope by adjusting the driving resistance, and realizes zero voltage switching (ZVS) without the need for an auxiliary resonant circuit or a high-frequency driving chip.

[0042] The present invention establishes a matching mechanism between the core saturation threshold and the load voltage to ensure soft switching adaptability within a wide load range.

[0043] The magnetic switch zero-current connection auxiliary circuit proposed in the present invention can achieve flexible control of the current phase by adjusting the size of the saturable magnetic core and the number of turns of the magnetic core.

[0044] The magnetic switch zero-current turn-on auxiliary circuit proposed in the present invention has strong load adaptability and exhibits excellent ability to reduce switching losses in a range from light load to heavy load.

[0045] Compared with ordinary inductance control, the magnetic switch zero-current connection auxiliary circuit proposed in the present invention has significantly reduced the impact on the load waveform, low waveform distortion, and steep waveform rising edge; compared with traditional measures to reduce switching losses, the load measurement adopts a series saturable magnetic core combined with an adjustable drive resistor, with a small number of components and a simple structure, greatly improving system reliability and significantly reducing costs. BRIEF DESCRIPTION OF THE DRAWINGS

[0046] Figure 1 This is a topology diagram of the auxiliary circuit for zero current connection of the magnetic switch;

[0047] Figure 2 This is a schematic diagram of the unsaturated state of the magnetic core;

[0048] Figure 3 Schematic diagram of the core saturation state;

[0049] Figure 4 The simulation waveforms for different loads and different current phase adjustment strategies; Figure 4 (a) is the simulation waveform of different load phase adjustment strategies; Figure 4 (b) is a comparison of simulation waveforms of no phase adjustment strategy, MSZCT strategy and different inductance adjustment strategies;

[0050] Figure 5 Schematic diagram of the test platform;

[0051] Figure 6 This is the schematic diagram of the top PCB board of the drive circuit;

[0052] Figure 7 The experimental waveforms for different numbers of turns are shown in Figure 2. Figure 7 (a) is the voltage waveform on the switch S with different turns; Figure 7 (b) is the current waveform of the switch S with different numbers of turns; Figure 7 (c) is the voltage waveform on the magnetic switch with different numbers of turns;

[0053] Figure 8 Schematic diagram of the effect of the number of core turns on the switching loss of the magnetic switch;

[0054] Figure 9 The experimental waveforms of different driving resistances are shown; Figure 9 (a) is the waveform of the switch gate drive voltage under different drive resistances; Figure 9 (b) is the waveform of the switch gate drive current under different drive resistances; Figure 9 (c) is a schematic diagram of the rising edge of the switching current waveform with different driving resistances after smoothing; Figure 9 (d) is the voltage waveform of the switch under different driving resistances;

[0055] Figure 10 Schematic diagram of the effect of gate drive resistance on switching loss;

[0056] Figure 11 This is a schematic diagram of the topology of the 6kV experimental circuit;

[0057] Figure 12 This is the schematic diagram of the main circuit PCB board of the experiment;

[0058] Figure 13 This is the 6kV test waveform diagram; Figure 13 (a) is the 6kV test waveform without any measures to reduce switching loss; Figure 13 (b) is the 6kV test waveform when measures are taken to reduce switching losses. DETAILED DESCRIPTION

[0059] The present invention will be further described below with reference to the following examples, but it should not be understood that the scope of the present invention is limited to the following examples. Without departing from the above technical ideas of the present invention, various substitutions and modifications can be made according to common technical knowledge and customary means in the art, and all should be included in the scope of protection of the present invention.

[0060] Example 1:

[0061] See also Figures 1 to 13 A magnetic switch zero current turn-on auxiliary circuit for reducing the switching loss of a pulse generator, comprising: a magnetic switch MS, a load R L , energy storage capacitor C dc , DC input voltage source V dc , drive switch.

[0062] The DC input voltage source V dc The negative terminal is grounded.

[0063] The DC input voltage source V dc One end of the series energy storage capacitor C dc Then, connect to the DC input voltage source V dc the other end.

[0064] The DC input voltage source V dcThe positive end of the drive switch, magnetic switch MS, load R are connected in series in sequence. L Then, connect to the DC input voltage source V dc The negative terminal of the

[0065] The magnetic switch zero-current connection auxiliary circuit realizes the steepening of the current rising edge, the sharpening of the voltage falling edge and the current phase regulation through the coordinated regulation of the magnetic switch MS and the drive switch.

[0066] Example 2:

[0067] A magnetic switch zero-current turn-on auxiliary circuit for reducing the switching loss of a pulse generator. The main technical content is shown in Example 1. Furthermore, the driving switch includes a single-stage switch and a multi-stage series switch.

[0068] Example 3:

[0069] A magnetic switch zero current turn-on auxiliary circuit for reducing the switching loss of a pulse generator, the main technical content of which is shown in any one of embodiments 1 to 2, further, the single-stage switch includes a driving resistor R g , switch S, external power supply V IN .

[0070] The driving resistor R g One end is connected to the external power supply V IN One end is connected to the positive pole of , and the other end is connected to the gate of switch S.

[0071] The external power supply V IN The negative terminal is grounded.

[0072] The switch S is an IGBT switch.

[0073] The collector of the switch S is connected to the DC input voltage source V dc The positive pole of the transistor is located at one end, and the emitter is connected to the magnetic switch MS.

[0074] Example 4:

[0075] A magnetic switch zero current turn-on auxiliary circuit for reducing the switching loss of a pulse generator, the main technical content of which is shown in any one of embodiments 1 to 3. Further, the multi-stage series switch includes n driving resistors R gi , n switches S i , n connection resistors R bi , where i=1,2,...,n, and n is a positive integer.

[0076] The driving resistor R gi One end is connected to the external power supply, and the other end is connected to the switch S i of the gate.

[0077] The switch S i All use IGBT switches.

[0078] The switch S k The emitter of the switch S k+1 The collector of , k = 1, 2, ..., n-1.

[0079] The collector of the switch S1 is connected to the DC input voltage source V dc The positive terminal of the switch S n The emitter of is connected to the magnetic switch MS.

[0080] The connection resistance R bi One end of the switch S i The emitter of the switch S i collector.

[0081] Example 5:

[0082] A magnetic switch zero-current turn-on auxiliary circuit for reducing the switching loss of a pulse generator. The main technical content is shown in any one of Examples 1 to 4. Furthermore, the resistance value of the driving resistor in the driving switch is adjustable.

[0083] The driving resistor R in the single-stage switch g The smaller the resistance value, the shorter the voltage falling edge time in the switch S turn-on overshoot.

[0084] The driving resistor R of the multi-stage series switch gi The smaller the resistance of the switch S i The shorter the voltage falling edge time in the turn-on overshoot.

[0085] Example 6:

[0086] A magnetic switch zero-current connection auxiliary circuit for reducing the switching loss of a pulse generator. The main technical content is shown in any one of Examples 1 to 5. Furthermore, the working state of the magnetic switch MS includes an unsaturated state and a saturated state.

[0087] Example 7:

[0088] A magnetic switch zero current connection auxiliary circuit for reducing the switching loss of a pulse generator, the main technical content of which is shown in any one of embodiments 1 to 6. Further, when the magnetic switch MS works in an unsaturated state, the magnetic switch MS is equivalent to an open circuit, and the DC input voltage source V dc The voltage is transferred from the drive switch to the magnetic switch MS, causing the switch current phase to lag behind the switch voltage phase. The load R L Zero voltage turn-on.

[0089] Example 8:

[0090] A magnetic switch zero current connection auxiliary circuit for reducing the switching loss of a pulse generator, the main technical content of which is shown in any one of embodiments 1 to 7. Further, when the magnetic switch MS works in a saturated state, the core volt-second product of the magnetic switch MS reaches the core saturation threshold, and the magnetic switch MS is equivalent to a short circuit. L The load current i L and load voltage V L .

[0091] Example 9:

[0092] A magnetic switch zero-current turn-on auxiliary circuit for reducing switching losses of a pulse generator, the main technical content of which is shown in any one of Examples 1 to 8. Furthermore, the calculation formula for the magnetic core volt-second product reaching the magnetic core saturation threshold is as follows:

[0093]

[0094] Where t is time. MS (t) is the voltage across the magnetic switch MS. N is the number of turns of the magnetic switch MS, A e is the effective cross-sectional area of ​​the core, and ΔB is the change in magnetic induction intensity.

[0095] Example 10:

[0096] A magnetic switch zero-current turn-on auxiliary circuit for reducing the switching loss of a pulse generator. The main technical content is shown in any one of Examples 1 to 9. Furthermore, when the core volt-second product does not reach the core saturation threshold, the turn-on loss of the magnetic switch MS decreases as the number of turns of the magnetic switch MS increases.

[0097] When the core volt-second product reaches the core saturation threshold, the turn-on loss of the magnetic switch MS no longer decreases with the increase in the number of turns of the magnetic switch MS.

[0098] The turn-on loss of the magnetic switch MS is as follows:

[0099]

[0100] Where t is time, E on is the turn-on loss of the magnetic switch MS. DS 、V DS They are the waveform data of the current and voltage at both ends of the magnetic switch MS changing with time.

[0101] When the turn-on loss of the magnetic switch MS takes the minimum value, the number of turns of the magnetic switch MS is as follows:

[0102]

[0103] Where K is the margin coefficient, T sis the falling edge time of the switching voltage, U is the DC input voltage source V dc The voltage, N is the number of turns of the magnetic switch MS, A e is the effective cross-sectional area of ​​the core, and ΔB is the change in magnetic induction intensity.

[0104] Example 11:

[0105] See also Figures 1 to 13 A magnetic switch zero current turn-on auxiliary circuit for reducing the switching loss of a pulse generator, comprising: a magnetic switch MS, a load R L , energy storage capacitor C dc , DC input voltage source V dc , drive switch.

[0106] The DC input voltage source V dc The negative terminal is grounded.

[0107] The DC input voltage source V dc One end of the series energy storage capacitor C dc Then, connect to the DC input voltage source V dc the other end.

[0108] The DC input voltage source V dc The positive end of the drive switch, magnetic switch MS, load R are connected in series in sequence. L Then, connect to the DC input voltage source V dc The negative terminal of the

[0109] The magnetic switch zero-current connection auxiliary circuit realizes the steepening of the current rising edge, the sharpening of the voltage falling edge and the current phase regulation through the coordinated regulation of the magnetic switch MS and the drive switch.

[0110] By collaboratively designing the core parameters and drive resistance values, the matching relationship between the falling edge of the switch voltage and the core saturation time can be dynamically adjusted to achieve an ideal staggered switching voltage-current waveform, ultimately achieving zero-current turn-on of the switch and reducing turn-on losses.

[0111] Example 12:

[0112] A magnetic switch zero-current turn-on auxiliary circuit for reducing the switching loss of a pulse generator. The main technical content is shown in Example 11. Furthermore, the driving switch includes a single-stage switch and a multi-stage series switch.

[0113] Example 13:

[0114] A magnetic switch zero current turn-on auxiliary circuit for reducing the switching loss of a pulse generator, the main technical content of which is shown in any one of embodiments 11 to 12, further, the single-stage switch includes a driving resistor R g , switch S, external power supply VIN .

[0115] The driving resistor R g One end is connected to the external power supply V IN One end is connected to the positive pole of , and the other end is connected to the gate of switch S.

[0116] The external power supply V IN The negative terminal is grounded.

[0117] The switch S is an IGBT switch.

[0118] The collector of the switch S is connected to the DC input voltage source V dc The positive pole of the transistor is located at one end, and the emitter is connected to the magnetic switch MS.

[0119] Example 14:

[0120] A magnetic switch zero current turn-on auxiliary circuit for reducing the switching loss of a pulse generator, the main technical content of which is shown in any one of embodiments 11 to 13. Further, the multi-stage series switch includes n driving resistors R gi , n switches S i , n connection resistors R bi , where i=1,2,...,n, and n is a positive integer.

[0121] The driving resistor R gi One end is connected to the external power supply, and the other end is connected to the switch S i of the gate.

[0122] The switch S i All use IGBT switches.

[0123] The switch S k The emitter of the switch S k+1 The collector of , k = 1, 2, ..., n-1.

[0124] The collector of the switch S1 is connected to the DC input voltage source V dc The positive terminal of the switch S n The emitter of is connected to the magnetic switch MS.

[0125] The connection resistance R bi One end of the switch S i The emitter of the switch S i collector.

[0126] Example 15:

[0127] A magnetic switch zero-current turn-on auxiliary circuit for reducing the switching loss of a pulse generator. The main technical content is shown in any one of Examples 11 to 14. Furthermore, the resistance value of the driving resistor in the driving switch is adjustable.

[0128] The driving resistor in the driving switch is adjusted by directly replacing the driving resistor on the driving circuit board.

[0129] Generally, the drive resistance is within 33Ω. However, in order to make the switch have a steeper voltage falling edge waveform during the turn-on process and reduce turn-on loss, a smaller drive resistance is used. At the same time, to balance current overshoot and gate reliability constraints, the recommended adjustment range is 2-10Ω.

[0130] The driving resistor R in the single-stage switch g The smaller the resistance value, the shorter the voltage falling edge time in the switch S turn-on overshoot.

[0131] The driving resistor R of the multi-stage series switch gi The smaller the resistance of the switch S i The shorter the voltage falling edge time in the turn-on overshoot.

[0132] Example 16:

[0133] A magnetic switch zero-current turn-on auxiliary circuit for reducing the switching loss of a pulse generator. The main technical content is shown in any one of Examples 11 to 15. Furthermore, the working state of the magnetic switch MS includes an unsaturated state and a saturated state.

[0134] Example 17:

[0135] A magnetic switch zero current connection auxiliary circuit for reducing the switching loss of a pulse generator, the main technical content of which is shown in any one of embodiments 11 to 16. Furthermore, when the magnetic switch MS operates in an unsaturated state, the magnetic switch MS exhibits a high impedance, which is equivalent to an open circuit, and the DC input voltage source V dc The voltage is transferred from the drive switch to the magnetic switch MS, causing the switch current phase to lag behind the switch voltage phase. The load R L Zero voltage turn-on.

[0136] Example 18:

[0137] A magnetic switch zero-current connection auxiliary circuit for reducing the switching loss of a pulse generator, the main technical content of which is shown in any one of embodiments 11 to 17. Further, when the magnetic switch MS operates in a saturated state, the core volt-second product of the magnetic switch MS reaches the core saturation threshold, its magnetic permeability drops sharply, and the inductance value of MS decays sharply. After the core is fully saturated, the magnetic switch MS is equivalent to a short circuit. L The load current i L and load voltage VL .

[0138] Example 19:

[0139] A magnetic switch zero-current turn-on auxiliary circuit for reducing switching losses of a pulse generator, the main technical content of which is shown in any one of Examples 11 to 18. Furthermore, the calculation formula for the magnetic core volt-second product reaching the magnetic core saturation threshold is as follows:

[0140]

[0141] Where t is time. MS (t) is the voltage across the magnetic switch MS. N is the number of turns of the magnetic switch MS, A e is the effective cross-sectional area of ​​the core, and ΔB is the change in magnetic induction intensity.

[0142] Example 20:

[0143] A magnetic switch zero-current turn-on auxiliary circuit for reducing the switching loss of a pulse generator. The main technical content is shown in any one of Examples 11 to 19. Furthermore, when the core volt-second product does not reach the core saturation threshold, the turn-on loss of the magnetic switch MS decreases as the number of turns of the magnetic switch MS increases.

[0144] When the core volt-second product reaches the core saturation threshold, the turn-on loss of the magnetic switch MS no longer decreases with the increase in the number of turns of the magnetic switch MS.

[0145] The turn-on loss of the magnetic switch MS is as follows:

[0146]

[0147] Where t is time, E on is the turn-on loss of the magnetic switch MS. DS 、V DS They are the waveform data of the current and voltage at both ends of the magnetic switch MS changing with time.

[0148] When the turn-on loss of the magnetic switch MS is minimized, the switching voltage and current phases need to be completely staggered, and the magnetic switch saturation time T MS Greater than the falling edge of the switching voltage T s , and considering that after the switch is turned on, oscillation will occur due to the mutual coupling between the line parasitic parameters and the switch parasitic capacitance, a certain margin needs to be reserved. The number of turns of the magnetic switch MS is as follows:

[0149]

[0150] Where K is the margin coefficient, T s is the falling edge time of the switching voltage, U is the DC input voltage source V dcThe voltage, N is the number of turns of the magnetic switch MS, A e is the effective cross-sectional area of ​​the core, and ΔB is the change in magnetic induction intensity.

[0151] Example 21:

[0152] See also Figures 1 to 13 , a magnetic switch zero-current turn-on auxiliary circuit that reduces pulse generator switching losses. The main technical contents include:

[0153] The topology of the magnetic switch zero current turn-on auxiliary circuit of this embodiment is as follows: Figure 1 shown.

[0154] The zero turn-on loss auxiliary circuit proposed in this embodiment innovatively combines the nonlinear saturation mechanism of the saturable magnetic core with the gate drive resistor R g The coordinated regulation of the current rising edge, the voltage falling edge and the current phase are controlled by synchronous optimization. This topology can be equivalent to a capacitor discharge circuit in engineering applications ( Figure 1 In the unsaturated state ( Figure 2 ), the magnetic switch MS shows high impedance and can be approximated as an open circuit. At this time, the power supply voltage U is transferred from the switch S to the magnetic switch MS, forcing the switching current phase to lag behind the switching voltage, creating the necessary conditions for zero-voltage switching (ZVS); when the core volt-second product reaches the core saturation threshold ( Figure 3 ), its magnetic permeability drops sharply, the MS inductance value decreases sharply, and after the core is fully saturated, the MS can be approximated as a short circuit. The load current i L , load voltage V L By collaboratively designing the core parameters and drive resistance values, the matching relationship between the falling edge of the switch voltage and the core saturation time can be dynamically adjusted to achieve an ideal staggered switching voltage-current waveform, ultimately achieving zero-current turn-on of the switch and reducing turn-on losses.

[0155] According to the coil volt-second product formula, we can get

[0156]

[0157] Among them, u MS (t) is the voltage across the magnetic switch, N is the number of turns of the magnetic switch, A e is the effective cross-sectional area of ​​the core, and ΔB is the change in magnetic induction intensity.

[0158] According to the KVL law, we can get

[0159]

[0160] Where U is the power supply voltage, u S (t) is the voltage across the switch.

[0161] Voltage across the switch u S During the decline of (t), it can be approximately regarded as a straight line with a slope of dv / dt, so formula (2) can be simplified to

[0162]

[0163] If the minimum turn-on loss is to be achieved, the switching voltage and current phases need to be completely staggered, and the magnetic switch saturation time T MS Greater than the falling edge of the switching voltage T s , and considering that after the switch is turned on, the line parasitic parameters and the switch parasitic capacitance will produce oscillations due to the mutual coupling, a certain margin needs to be reserved.

[0164]

[0165] Among them, K is the margin coefficient, which can be adjusted according to the oscillation time.

[0166] Example 22:

[0167] A magnetic switch zero-current turn-on auxiliary circuit for reducing the switching loss of a pulse generator. The main technical content is shown in Example 21. Furthermore, in order to systematically verify the feasibility of this embodiment in suppressing turn-on loss and achieving zero-current switching under different load conditions, and to compare and analyze the control characteristics of the magnetic switch and conventional inductors on the current waveform, this study built a PSpice simulation platform based on the equivalent model shown in Fig. 1. The main simulation parameters are shown in Table 1.

[0168] In the simulation, the ability of zero-current turn-on (ZCT) to achieve zero current switching under light load and heavy load was explored respectively, and the control characteristics of the current waveform by magnetic switch and conventional inductor were compared and analyzed. The results are as follows: Figure 4 As shown in the figure, the simulation circuit is configured with 1kV DC input and 5μs pulse width parameters. It can be seen that under various load conditions, ZCT makes the switching current phase significantly lag behind the voltage phase, achieving zero current switching.

[0169] Table 1 Simulation parameters

[0170]

[0171] Example 23:

[0172] A magnetic switch zero current connection auxiliary circuit for reducing the switching loss of the pulse generator. The main technical content is shown in any one of Examples 21 to 22. Further, in order to verify the feasibility of this embodiment, a corresponding experimental prototype was developed and a Figure 5 The test platform shown in the figure has a top PCB board for the switch driver circuit. Figure 6 The main experimental parameters are shown in Table 2.

[0173] Table 2 Main experimental parameters

[0174]

[0175] To further quantify the mechanism by which design parameters regulate turn-on loss, this experiment uses an iron-based nanocrystalline saturable magnetic core as the core. Based on the circuit in Fig. (a), a dynamic magnetic switch test platform was constructed (Tab. 2 summarizes the main experimental parameters). The following key questions were systematically investigated:

[0176] The effect of the number of turns of the magnetic switch winding on the current phase lag and turn-on loss suppression during the conduction process;

[0177] The influence of switch drive resistance on switch voltage, current waveform and turn-on loss during turn-on.

[0178] On this basis, a magnetic switch test platform with three switches in series was built to verify the performance of the magnetic switch zero-current turn-on auxiliary circuit (MSZCT) at a higher voltage level. Figure 4 As shown, the three-stage switches are connected in series and only use a 5MΩ voltage-sharing resistor for static voltage balancing.

[0179] Figure 7 The voltage and current waveforms on the switch S, as well as the voltage waveform on the magnetic switch, are shown for magnetic switches with different numbers of core turns. It can be found that magnetic switches with different numbers of core turns have almost no effect on the switch voltage during the switch-on process, and each group has an almost synchronized voltage falling edge, which is the same as the simulation. As the number of turns of the magnetic switch increases, the phase lag of the switch current becomes more significant, and the regulation effect on the waveform only occurs at the front end of the rising edge, generating a gentle pre-pulse, while in the latter part of the rising edge, each group has an approximately steep rising edge, reflecting the low waveform distortion characteristics of the MSZCT. Further verifying the principle of MSZCT regulating the phase of the current waveform, at the moment the switch is turned on, the voltage across the switch drops rapidly, and the power supply voltage V dc Transfer from both ends of the switch to the magnetic switch and load. At this time, the magnetic core is not saturated, the magnetic switch is in a high impedance state and is much larger than the load impedance. The power supply voltage V dc Mainly applied to both ends of the magnetic switch. According to formula (1), the volt-second product The driving magnetic flux grows linearly. In this process, as the volt-second product increases, the magnetic core gradually enters the nonlinear region. The magnetic domain rotation is blocked, resulting in a significant decrease in relative permeability, a decrease in magnetic switch impedance, and a gradual increase in the line current. A gentle pre-pulse appears on the switch current, which plays a role in delaying the current phase. When the volt-second product reaches the critical value NAe ΔB, the core is fully saturated. At this point, the magnetic permeability approaches the vacuum permeability μ0, the impedance drops sharply, and the current increases rapidly. This ultimately lags the current phase and reduces turn-on losses.

[0180] Using the voltage and current waveforms at both ends of the switch measured experimentally, according to the formula Calculate the opening loss E on , turn-on loss E of different magnetic switch turns on like Figure 8 shown.

[0181] It can be found that as the number of magnetic switch turns increases, the turn-on loss decreases significantly, reaches saturation at 6 turns, and the turn-on loss no longer decreases significantly with the increase in the number of switch turns, reducing the switching loss by 73.42%.

[0182] Select the commonly used driving resistor value for experiment. Figure 9 The gate drive waveform, voltage on the switch, and current waveforms are shown below for different drive resistances. It can be seen that reducing the drive resistance can accelerate the voltage drop during the switch turn-on overshoot, while having little effect on the current waveform. Based on the above data, the turn-on loss is calculated and the switching loss waveform is fitted. The results are shown in Figure 1. Figure 10 shown.

[0183] In order to better compare the advantages of the MSZCT strategy in waveform control, a three-stage series magnetic switch test platform was further built. The three-stage series magnetic switch test platform was implemented under the conditions of 6kV input voltage and 1kΩ load. The circuit topology used in the experiment is as follows: Figure 11 As shown, the main circuit PCB board of the experiment is as follows Figure 12 In order to accommodate the high voltage stress on the magnetic switch and prevent premature saturation, a larger iron-based nanocrystalline core (size: 80×50×20mm, effective cross-sectional area A e =2.34cm 2 ). According to equation (4), the minimum winding requirement is calculated to be N>9.19, and the actual test is N=10. Figure 13 These are the experimental test results of a 6kV prototype. As can be seen in the figure, when there are no measures to reduce switching losses, the voltage-current overlap area is large during the turn-on transient. However, after applying the switching loss reduction measures, the significant phase separation between voltage and current limits the overlap to a smaller pre-pulse interval.

[0184] In summary, this invention reduces the voltage-current waveform overlap during the switch-on period by connecting a saturable magnetic core in series on the load side and steepening the falling edge of the voltage waveform by adjusting the switch drive resistor. This reduces system switching losses. This invention achieves flexible current phase adjustment with only one additional component, offering low cost and ease of debugging. It also offers the advantages of a wide range of applications.

Claims

1. A magnetic switch zero current turn-on auxiliary circuit for reducing pulse generator switching loss, characterized in that: include: Magnetic switch MS, load R L , energy storage capacitor C dc , DC input voltage source V dc , drive switch; The DC input voltage source V dc The negative end of the device is grounded; The DC input voltage source V dc One end of the series energy storage capacitor C dc Then, connect to the DC input voltage source V dc the other end; The DC input voltage source V dc The positive end of the drive switch, magnetic switch MS, load R are connected in series in sequence. L Then, connect to the DC input voltage source V dc The negative terminal of the The magnetic switch zero-current connection auxiliary circuit realizes the steepening of the current rising edge, the sharpening of the voltage falling edge and the current phase regulation through the coordinated regulation of the magnetic switch MS and the drive switch.

2. A magnetic switch zero current turn-on auxiliary circuit for reducing pulse generator switching loss according to claim 1, characterized in that: The driving switch includes a single-stage switch and a multi-stage series switch.

3. A magnetic switch zero current turn-on auxiliary circuit for reducing pulse generator switching loss according to claim 2, characterized in that: The single-stage switch includes a driving resistor R g , switch S, external power supply V IN ; The driving resistor R g One end is connected to the external power supply V IN The positive electrode of is located at one end, and the other end is connected to the gate of switch S; The external power supply V IN The negative end of the device is grounded; The switch S is an IGBT switch; The collector of the switch S is connected to the DC input voltage source V dc The positive pole of the transistor is located at one end, and the emitter is connected to the magnetic switch MS.

4. A magnetic switch zero current turn-on auxiliary circuit for reducing pulse generator switching loss according to claim 2, characterized in that: The multi-stage series switch includes n driving resistors R gi , n switches S i , n connection resistors R bi , where i = 1, 2, ..., n, and n is a positive integer; The driving resistor R gi One end is connected to the external power supply, and the other end is connected to the switch S i The gate; The switch S i All use IGBT switches; The switch S k The emitter of the switch S k+1 The collector of , k = 1, 2, ..., n-1; The collector of the switch S1 is connected to the DC input voltage source V dc The positive terminal of the switch S n The emitter is connected to the magnetic switch MS; The connection resistance R bi One end of the switch S i The emitter of the switch S i collector.

5. A magnetic switch zero-current turn-on auxiliary circuit for reducing switching loss of a pulse generator according to any one of claims 3 or 4, characterized in that: The resistance of the driving resistor in the driving switch is adjustable; The driving resistor R in the single-stage switch g The smaller the resistance value, the shorter the voltage falling edge time during the turn-on overshoot of switch S. The driving resistor R of the multi-stage series switch gi The smaller the resistance of the switch S i The shorter the voltage falling edge time in the turn-on overshoot.

6. The magnetic switch zero-current turn-on auxiliary circuit for reducing pulse generator switching loss according to claim 1, characterized in that: The working states of the magnetic switch MS include an unsaturated state and a saturated state.

7. A magnetic switch zero-current turn-on auxiliary circuit for reducing pulse generator switching loss according to claim 6, characterized in that: When the magnetic switch MS works in the unsaturated state, the magnetic switch MS is equivalent to an open circuit, and the DC input voltage source V dc The voltage is transferred from the drive switch to the magnetic switch MS, causing the switch current phase to lag behind the switch voltage phase. The load R L Zero voltage turn-on.

8. The magnetic switch zero-current turn-on auxiliary circuit for reducing switching loss of a pulse generator according to claim 6, characterized in that: When the magnetic switch MS works in saturation state, the core volt-second product of the magnetic switch MS reaches the core saturation threshold, and the magnetic switch MS is equivalent to a short circuit. L The load current i L and load voltage V L .

9. A magnetic switch zero-current turn-on auxiliary circuit for reducing switching loss of a pulse generator according to claim 8, characterized in that: The calculation formula for the core volt-second product to reach the core saturation threshold is as follows: Where t is time; u MS (t) is the voltage across the magnetic switch MS; N is the number of turns of the magnetic switch MS, A e is the effective cross-sectional area of ​​the core, and ΔB is the change in magnetic induction intensity.

10. A magnetic switch zero-current turn-on auxiliary circuit for reducing pulse generator switching loss according to claim 9, characterized in that: When the core volt-second product does not reach the core saturation threshold, the turn-on loss of the magnetic switch MS decreases as the number of turns of the magnetic switch MS increases; When the core volt-second product reaches the core saturation threshold, the turn-on loss of the magnetic switch MS no longer decreases with the increase of the number of turns of the magnetic switch MS; The turn-on loss of the magnetic switch MS is as follows: Where t is time, E on is the turn-on loss of the magnetic switch MS; I DS 、V DS They are the waveform data of the current and voltage at both ends of the magnetic switch MS changing with time. When the turn-on loss of the magnetic switch MS takes the minimum value, the number of turns of the magnetic switch MS is as follows: Where K is the margin coefficient, T s is the falling edge time of the switching voltage, U is the DC input voltage source V dc The voltage, N is the number of turns of the magnetic switch MS, A e is the effective cross-sectional area of ​​the core, and ΔB is the change in magnetic induction intensity.

Images