Transmitter and method for reducing local oscillation leakage in transmitter

Abstract

The invention provides a transmitter and a method for reducing local oscillation leakage in the transmitter. The transmitter includes an amplifier, a mixer, a self-mixer, a first signal source, a second signal source, and a correction logic circuit. The amplifier generates an amplified fundamental frequency signal, the mixer uploads the amplified fundamental frequency signal to generate a radio frequency signal, and the self-mixer performs self-mixing according to the radio frequency signal to generate a feedback signal. In the first stage, the correction logic circuit controls a first signal output by the first signal source to the amplifier so as to minimize a direct current signal in the amplified fundamental frequency signal. In a second stage, the correction logic circuit controls a second signal output by the second signal source to the mixer so as to minimize a feedback fundamental frequency signal in the feedback signal.

Description

Technical Field

[0001] This invention relates to the design of wireless communication circuits, and more particularly to a transmitter and a method for reducing local oscillation leakage in the transmitter. Background Technology

[0002] In the field of wireless communication, analog circuits on the signal output path typically exhibit DC bias. This DC bias, after being amplified by a mixer, can cause signal components not belonging to the transmitted signal to appear within the signal band. This phenomenon is known as local oscillation leakage (LO leakage). To address LO leakage, various correction mechanisms have been proposed. However, these mechanisms have certain drawbacks. For example, some techniques evaluate the DC bias correction status of the target circuit by detecting the power at a specific frequency. However, when the target circuit outputs a high-power signal due to the requirements of the correction mechanism, the signal components at that specific frequency can become interference sources due to signal coupling, making it difficult to properly evaluate the DC bias of the target circuit. Furthermore, the target circuit can have different DC biases under different gain settings. However, in actual operation, the target circuit needs to operate under different gain settings, making it difficult to meet the requirements of a large signal dynamic range by correcting the DC bias for only a single gain setting.

[0003] Therefore, a novel framework and related correction methods are needed to solve the problems of the relevant technologies with little or no side effects. Summary of the Invention

[0004] The object of the present invention is to provide a transmitter and a method for reducing local oscillation leakage in the transmitter in order to properly correct the DC bias of one or more analog circuits in the transmitter.

[0005] At least one embodiment of the present invention provides a transmitter. The transmitter may include an analog amplifier, a mixer, a self-mixer, a first correction signal source, a second correction signal source, and correction logic circuitry. The first correction signal source is coupled to the analog amplifier, the second correction signal source is coupled to the mixer, and the correction logic circuitry is coupled to both the first and second correction signal sources. The analog amplifier amplifies a baseband signal to generate an amplified baseband signal, the mixer up-converts the amplified baseband signal to generate a radio frequency (RF) signal, and the self-mixer performs self-mixing based on the RF signal to generate a feedback signal. The first correction signal source outputs a first correction signal to the analog amplifier, and the second correction signal source, coupled to the mixer, outputs a second correction signal to the mixer. In a first correction stage, the correction logic circuitry controls the first correction signal based on a direct current (DC) signal in the amplified baseband signal to minimize the DC signal. In the second correction stage following the first correction stage, the correction logic circuit controls the second correction signal based on the feedback base frequency signal in the feedback signal so that the feedback base frequency signal is minimized.

[0006] At least one embodiment of the present invention provides a method for reducing local oscillation leakage in a transmitter. The method may include: amplifying a baseband signal using an analog amplifier of the transmitter to generate an amplified baseband signal; up-converting the amplified baseband signal using a mixer of the transmitter to generate a radio frequency (RF) signal; performing self-mixing based on the RF signal using a self-mixer of the transmitter to generate a feedback signal; in a first correction phase, controlling a first correction signal source in the transmitter to output a first correction signal to the analog amplifier based on a direct current (DC) signal in the amplified baseband signal using a correction logic circuit of the transmitter, thereby minimizing the DC signal; and in a second correction phase following the first correction phase, controlling a second correction signal source in the transmitter to output a second correction signal to the mixer based on a feedback baseband signal in the feedback signal using the correction logic circuit, thereby minimizing the feedback baseband signal.

[0007] The transmitter and method provided in the embodiments of the present invention can correct the DC bias of the analog amplifier based on the DC component in the signal output by the analog amplifier. Compared with detecting the fundamental frequency signal generated by upsampling and then downsampling the signal output by the analog amplifier, the present invention can obtain a more accurate correction result. Furthermore, the embodiments of the present invention do not significantly increase additional costs; therefore, the present invention can solve the problems of related technologies without or with minimal side effects. Attached Figure Description

[0008] Figure 1This is a schematic diagram of a transmitter according to an embodiment of the present invention.

[0009] Figure 2 According to the embodiments of the present invention, Figure 1 The diagram shows a correction of the DC bias of the analog amplifier in the transmitter.

[0010] Figure 3 According to the embodiments of the present invention, Figure 1 A schematic diagram of the DC bias correction of the mixer in the transmitter shown.

[0011] Figure 4 This is a schematic diagram illustrating the workflow of a method for reducing local oscillation leakage in a transmitter according to an embodiment of the present invention. Detailed Implementation

[0012] Figure 1 This is a schematic diagram of a transmitter 10 according to an embodiment of the present invention. Figure 1 As shown, transmitter 10 may include a digital-to-analog converter (DAC) 101, a transimpedance amplifier (TIA) 102 coupled to the DAC 101, an analog amplifier such as a transmitter baseband (TXBB) amplifier 103 (denoted as "TXBB" in the figure for simplicity) coupled to the transimpedance amplifier 102, a mixer 104 coupled to the transmitter baseband amplifier 103, a power amplifier driver 105 (denoted as "PAD" in the figure for simplicity) coupled to the mixer 104, and a power amplifier coupled to the power amplifier driver 105. amplifier (PA) 106 (labeled "PA" in the figure for simplicity), self mixer 107 coupled to power amplifier driver 105, first correction signal source such as amplifier correction signal source 110 coupled to transmitter baseband amplifier 103, second correction signal source such as mixer correction signal source 120 coupled to mixer 104, and correction logic circuit 130 coupled to amplifier correction signal source 110 and mixer correction signal source 120.

[0013] In this embodiment, the analog-to-digital converter 101 can process the digital test signal D. CAL Perform digital-to-analog conversion to output analog test signal A0 CAL The transimpedance amplifier 102 can process the analog test signal A0. CAL (For example, a current test signal) undergoes current-to-voltage conversion to output a baseband signal A1. CAL(For example, a voltage test signal). The transmitter baseband amplifier 103 is used to amplify the baseband signal A1. CAL To generate the amplified fundamental frequency signal A2 CAL The mixer 104 is used to amplify the baseband signal A2. CAL Upsampling (e.g., based on a frequency of ω) LO (Up-converting the local oscillator signal ALO) to generate the radio frequency signal A0. RF Additionally, the power amplifier driver 105 can operate based on the radio frequency signal A0. RF Generate radio frequency signal A1 RF This drives the power amplifier 106. The power amplifier 106 can then output a radio frequency signal A2. RF The transmitter 10 is connected to the antenna for wireless transmission. When the transmitter 10 is operating in calibration mode, the self-mixer 107 is used to transmit according to the radio frequency signal A0. RF Perform self-mixing to generate feedback signal A0 FB In this embodiment, the self-mixer 107 can receive the radio frequency signal A1 output by the power amplifier driver 105. RF (It is based on radio frequency signal A0) RF (Generate), and for radio frequency signal A1 RF Perform self-mixing to generate feedback signal A0 FB In some embodiments, the self-mixer 107 may receive the radio frequency signal A0 output by the mixer 104. RF And for radio frequency signal A0 RF Perform self-mixing to generate feedback signal A0 FB In some embodiments, the self-mixer 107 may receive the radio frequency signal A2 output from the power amplifier 106. RF And for radio frequency signal A2 RF Perform self-mixing to generate feedback signal A0 FB .

[0014] In transmitter 10, factors affecting local oscillation leakage (LO leakage) include the DC bias V of transmitter baseband amplifier 103. DC,TXBB and the DC bias V of mixer 104 DC,MIXER For example, the local oscillation leakage of transmitter 10 can be combined with (G TXBB ×V DC,TXBB +V DC,MIXER ) positively correlated, where G TXBB This can represent the gain of the transmitter's baseband amplifier 103. In this embodiment, the amplifier correction signal source 110 is used to output the amplifier correction signal I1. CAL To the transmitter baseband amplifier 103 to correct the DC bias V of the transmitter baseband amplifier 103DC,TXBB The mixer correction signal source 120 is used to output the mixer correction signal I2. CAL To mixer 104 to correct the DC bias V of mixer 104 DC,MIXER During the amplifier calibration stage, the calibration logic circuit 130 can adjust the calibration logic based on the amplified base frequency signal A2. CAL The direct current (DC) signal in the amplifier controls the correction signal I1. CAL To minimize the DC signal (e.g., through control signal D1) CTRL Control amplifier calibration signal source 110 to adjust amplifier calibration signal I1 CAL The value of I1 is used to find the amplifier correction signal I1 that minimizes the DC signal. CAL (value). In the mixer calibration stage following the amplifier calibration stage, the calibration logic circuit 130 can adjust the value based on the feedback signal A0. FB The feedback fundamental frequency signal (e.g., feedback signal A0) in the middle. FB The specific frequency signal component in the signal controls the mixer correction signal I2. CAL To minimize the feedback baseband signal (e.g., through control signal D2) CTRL Control mixer calibration signal source 120 to adjust mixer calibration signal I2 CAL The value of the mixer correction signal I2 is used to find the value that minimizes the feedback baseband signal. CAL (value).

[0015] In this embodiment, the digital test signal D CAL The frequency is ω0, therefore the simulated test signal A0 CAL and baseband signal A1 CAL The frequencies are all ω0, where the DC bias V of the transmitter fundamental frequency amplifier 103 is... DC,TXBB It will then be loaded onto the amplified baseband signal A2 CAL At the DC frequency, the amplified base frequency signal A2 CAL It contains a signal component with a frequency of ω0 (corresponding to the digital test signal D). CAL ) and signal components with a frequency of 0, such as the DC signal (corresponding to the DC bias V of the transmitter baseband amplifier 103). DC,TXBB In particular, the magnitude of this DC signal can represent the DC bias V of the transmitter's baseband amplifier 103. DC,TXBB The magnitude of the DC bias V of the transmitter baseband amplifier 103 is thus minimized by the correction logic circuit 130. DC,TXBB The DC bias V of the fundamental frequency amplifier 103 of the correction transmitter is... DC,TXBBAfter being minimized (e.g., eliminated), mixer 104 can amplify the baseband signal A2 with frequency ω0. CAL (The signal component with a frequency of 0 has been minimized and therefore ignored) is up-frequencyed to make the radio frequency signal A0 RF (or radio frequency signal A1) RF A2 RF The included frequency is (ω) LO The signal component with frequency (ω + ω0) (assuming its signal amplitude is A) and frequency (ω) LO The signal component of -ω0 (assuming its signal amplitude is C), where the DC bias V of mixer 104 is... DC,MIXER It will also be up-frequency, causing the radio frequency signal A0 to be... RF (or radio frequency signal A1) RF A2 RF It also contains a signal component with a frequency of ωLO (assuming its amplitude is B). Therefore, the radio frequency signal A0 RF (or radio frequency signal A1) RF A2 RF The frequency in ) is ω LO The amplitude B of the signal component can represent the DC bias V of mixer 104. DC,MIXER Size.

[0016] For ease of understanding, radio frequency signal A0 RF (or radio frequency signal A1) RF A2 RF The frequency of (ω) LO +ω0), and the signal component with amplitude A can be represented by A(ω). LO +ω0) is used to represent the radio frequency signal A0. RF (or radio frequency signal A1) RF A2 RF The frequency of (ω) LO -ω0), and the signal component with amplitude C can be represented by C(ω). LO -ω0) is used to represent, and radio frequency signal A0 RF (or radio frequency signal A1) RF A2 RF The frequency in ) is ω LO The signal component with amplitude B can be represented by B(ω). LO The following description uses the self-mixer 107 to represent the radio frequency signal A1. For simplicity, the following description uses the self-mixer 107 to represent the radio frequency signal A1. RF The self-mixing architecture is explained, and the self-mixer 107 handles the radio frequency signal A0. RF Or A2 RF The architecture for self-mixing can be deduced similarly. When the self-mixer 107 mixes the radio frequency signal A1... RFDuring self-mixing, the radio frequency signal A1 received at the first input terminal of the self-mixer 107 (e.g., the input terminal on the left side of the self-mixer 107 in the figure) RF The signal component A(ω) LO +ω0) can be respectively compared with the radio frequency signal A1 received by the second input terminal of the self-mixer 107 (e.g., the input terminal above the self-mixer 107 in the figure). RF The signal component A(ω) LO +ω0), B(ω) LO ) and C(ω LO -ω0) mixing to generate feedback signal A0 FB The signal component with a frequency of DC (amplitude of G) SELFMIXER ×A×A), signal components with frequency ω0 (amplitude (G) SELFMIXER ×B×A)) and signal components with frequency (2×ω0) and amplitude (G SELFMIXER ×C×A)), where G SELFMIXER This can represent the conversion gain of the self-mixer 107. The radio frequency signal A1 received at the first input terminal of the self-mixer 107 (e.g., the input terminal on the left side of the self-mixer 107 in the figure) is... RF The signal component B(ω) LO The radio frequency signal A1 can be received by the second input terminal of the self-mixer 107 (e.g., the input terminal above the self-mixer 107 in the figure). RF The signal component A(ω) LO +ω0), B(ω) LO ) and C(ω LO -ω0) mixing to generate feedback signal A0 FB The signal component with a frequency of DC (amplitude of G) SELFMIXER ×B×B)) and signal components with frequency ω0 (amplitude (G) SELFMIXER ×A×B) and (G SELFMIXER ×C×B)). The radio frequency signal A1 received from the first input terminal of the mixer 107 (e.g., the input terminal on the left side of the mixer 107 in the figure). RF The signal component C(ω) LO -ω0) can be respectively received by the radio frequency signal A1 from the second input terminal of the self-mixer 107 (e.g., the input terminal above the self-mixer 107 in the figure). RF The signal component A(ω) LO +ω0), B(ω) LO ) and C(ω LO -ω0) mixing to generate feedback signal A0 FB The signal component with a frequency of DC (amplitude of G) SELFMIXER×C×C), signal components with frequency ω0 (amplitude (G) SELFMIXER ×B×C)) and signal components with frequency (2×ω0) and amplitude (G SELFMIXER ×A×C)). Therefore, the feedback signal A0 FB The magnitude of the signal component with a DC frequency can be determined based on ((G) SELFMIXER ×A×A)+(G SELFMIXER ×B×B)+(G SELFMIXER The feedback signal A0 is determined by ×C×C)). FB The magnitude of the signal component with a mid-frequency ω0 can be determined according to ((G SELFMIXER ×B×A)+(G SELFMIXER ×A×B)+(G SELFMIXER ×C×B)+(G SELFMIXER The decision is made using ×B×C), while the feedback signal A0... FB The magnitude of the signal component with a mid-frequency (2×ω0) can be determined according to ((G SELFMIXER ×C×A)+(G SELFMIXER The decision is made using ×A×C). As shown above, the feedback signal A0... FB Each signal component with a mid-frequency of ω0 is related to the DC bias V of mixer 104. DC,MIXER The magnitude is related (e.g., including amplitude B), while the feedback signal A0 FB The signal component with a mid-frequency of DC frequency or (2×ω0) includes at least a portion of the DC bias V of mixer 104. DC,MIXER The magnitude is independent (e.g., excluding the portion containing amplitude B). Based on the above reasons, the correction logic circuit 130 is optimally based on the feedback signal A0. FB The signal component with a mid-frequency of ω0 controls the mixer correction signal I2. CAL So that the feedback signal A0 FB The signal component with a mid-frequency of ω0 is minimized, that is, when the fundamental frequency signal A1 CAL When the frequency is ω0, the feedback signal A0 FB The feedback base frequency signal in this context is feedback signal A0. FB The signal component has a mid-frequency of ω0. Therefore, the correction logic circuit 130 can correct the DC bias V of the mixer 104 by minimizing the feedback base frequency signal. DC,MIXER .

[0017] Additionally, transmitter 10 may further include an analog-to-digital converter 140 and a power spectral density (PSD) circuit 150 (denoted as "PSD circuit" in the figure for simplicity), wherein the power spectral density circuit 150 is coupled to the analog-to-digital converter 140 and the correction logic circuit 130. In this embodiment, the analog-to-digital converter 140 is used to adjust the amplified baseband signal A2 during the amplifier correction stage. CAL Perform analog-to-digital conversion to generate a first digital signal (e.g., the digital signal D obtained during the amplifier calibration phase). FB ), and in the mixer calibration phase, based on the feedback signal A0 FB Perform analog-to-digital conversion to generate a second digital signal (e.g., the digital signal D obtained during the mixer calibration stage). FB The power spectral density circuit 150 is used to calculate the power of the signal component with a frequency of 0 in the first digital signal to obtain a first calculation result (e.g., the calculation result D obtained during the amplifier calibration phase). PSD And calculate the power of the signal component with frequency ω0 in the second digital signal to obtain a second calculation result (e.g., the calculation result D obtained in the mixer correction stage). PSD The first calculation result and the second calculation result respectively represent the power of the DC signal (corresponding to the DC bias V of the transmitter baseband amplifier 103). DC,TXBB The power of the feedback baseband signal (corresponding to the DC bias V of mixer 104) and the power of the feedback baseband signal. DC,MIXER In particular, the correction logic circuit 130 can control the amplifier correction signal I1 based on the first calculation result. CAL And based on the second calculation result, control the mixer correction signal I2. CAL .

[0018] In this embodiment, the transmitter 10 may further include an attenuator 160, wherein the attenuator 160 is coupled between the transmitter baseband amplifier 103 and the analog-to-digital converter 140. In some cases, the amplified baseband signal A2 CAL The signal range may exceed the input dynamic range of the analog-to-digital converter 140, causing the output of the analog-to-digital converter 140 to saturate. To avoid saturation of the digital converter 140, the attenuator 160 can reduce the amplified fundamental frequency signal A2 during the amplifier correction stage. CAL The amplitude is used to generate the attenuated fundamental frequency signal A0. ATT Therefore, the analog-to-digital converter 140 can adjust the attenuated baseband signal A0 during the amplifier correction stage. ATT (It is based on the amplified fundamental frequency signal A2) CAL The generated signal is converted from analog to digital to produce a digital signal D.FB Additionally, transmitter 10 may further include a programmable-gain amplifier (PGA) 170 (denoted as "PGA" in the figure for simplicity), wherein the programmable-gain amplifier 170 is coupled between mixer 107 and analog-to-digital converter 140. The programmable-gain amplifier 170 is used to adjust the feedback signal A0 during the mixer calibration phase. FB The amplitude is used to generate the adjusted feedback signal A1 FB To ensure the adjusted feedback signal A1 FB The amplitude meets the input dynamic range requirements of the analog-to-digital converter 140, which can adjust the feedback signal A1 during the mixer calibration phase. FB (It is based on feedback signal A0) FB The generated signal is converted from analog to digital to produce a digital signal D. FB .

[0019] Figure 2 According to the embodiments of the present invention, Figure 1 The DC bias V of the correction transmitter fundamental frequency amplifier 103 in the transmitter 10 shown is... DC,TXBB The diagram illustrates the relevant test signal paths, as shown by the dashed arrows. Specifically, the analog-to-digital converter 140 can perform attenuation of the baseband signal A0 during this amplifier calibration phase. ATT Perform analog-to-digital conversion to output a digital signal D FB1 (This can be considered as the digital signal D obtained during the amplifier calibration phase mentioned above) FB (For example), while the power spectral density circuit 150 can calculate the digital signal D. FB1 The power of the signal component with a frequency of 0 is used to obtain the calculation result D. PSD1 (This can be considered as the calculation result D obtained during the amplifier calibration phase mentioned above) PSD (Example).

[0020] Figure 3 According to the embodiments of the present invention, Figure 1 The DC bias V of the corrected mixer 104 in the transmitter 10 shown DC,MIXER The diagram illustrates the relevant test signal paths, as shown by the dashed arrows. Specifically, the analog-to-digital converter 140 can adjust the feedback signal A1 during the mixer calibration phase. FB Perform analog-to-digital conversion to output a digital signal D FB2 (This can be considered as the digital signal D obtained during the mixer calibration stage mentioned above) FB (For example), while the power spectral density circuit 150 can calculate the digital signal D. FB2 The power of the signal component with a mid-frequency ω0 is used to obtain the calculation result D.PSD2 (This can be considered as the calculation result D obtained during the mixer calibration stage mentioned above) PSD (Example).

[0021] It should be noted that the transmitter baseband amplifier 103 may have multiple candidate amplification gains; however, when the amplification gain of the transmitter baseband amplifier 103 changes, the DC bias V of the transmitter baseband amplifier 103 will change. DC,TXBB This will also change. If only the multiple candidate amplification gains are corrected and the correction results are applied to all candidate amplification gains, the local oscillation leakage problem of transmitter 10 can reappear when the amplification gain of transmitter baseband amplifier 103 changes. Therefore, amplifier correction signal source 110 may include transmitter baseband amplifier correction table 112 (labeled "TXBB table" in the figure) and current-mode digital-to-analog converter 111 corresponding to transmitter baseband amplifier 103 (labeled "TXBB IDAC" in the figure for simplicity), wherein current-mode digital-to-analog converter 111 is coupled to transmitter baseband amplifier correction table 112. For example, current-mode digital-to-analog converter 111 is used to adjust the DC bias value of transmitter baseband amplifier 103. The transmitter baseband amplifier calibration table 112 is used to record multiple digital amplifier calibration values ​​(e.g., calibration values ​​obtained under the settings of the multiple candidate amplification gains) corresponding to the multiple candidate amplification gains of the transmitter baseband amplifier 103. The transmitter baseband amplifier calibration table 112 can output the corresponding digital amplifier calibration value (e.g., digital amplifier calibration value D1) among the multiple digital amplifier calibration values ​​when the amplification gain of the transmitter baseband amplifier 103 is set to a specific amplification gain of the multiple candidate amplification gains. CAL The current-mode digital-to-analog converter 111 is used to adjust the corresponding digital amplifier correction value (e.g., digital amplifier correction value D1). CAL Output amplifier correction signal I1 CAL .

[0022] Similarly, mixer 104 may have multiple candidate conversion gains, and when the conversion gain of mixer 104 changes, the DC bias V of mixer 104... DC,MIXERThis will also change. Therefore, the mixer calibration signal source 120 may include a mixer calibration table 122 (labeled "Mixer Table" in the figure) and a current-type digital-to-analog converter 121 (labeled "Mixer IDAC" in the figure for simplicity) corresponding to the mixer 104, wherein the current-type digital-to-analog converter 121 is coupled to the mixer calibration table 122. For example, the current-type digital-to-analog converter 121 is used to adjust the DC bias value of the mixer 104. The mixer calibration table 122 is used to record a plurality of digital mixer calibration values ​​(e.g., calibration values ​​obtained under the settings of the plurality of candidate conversion gains) corresponding to the plurality of candidate conversion gains of the mixer 104, wherein the mixer calibration table 122 may output the corresponding digital mixer calibration value (e.g., digital mixer calibration value D2) among the plurality of digital mixer calibration values ​​when the conversion gain of the mixer 104 is set to a specific conversion gain of the plurality of candidate conversion gains. CAL The current-mode digital-to-analog converter 121 is used to adjust the corresponding digital amplifier correction value (e.g., digital mixer correction value D2). CAL Output mixer correction signal I2 CAL .

[0023] Figure 4 According to an embodiment of the present invention, this invention is used in a transmitter (e.g., Figure 1 Reduce local oscillation leakage in the transmitter 10) (e.g., by correcting, reducing, or eliminating the DC bias V of the transmitter's baseband amplifier 103). DC,TXBB and the DC bias V of mixer 104 DC,MIXER A schematic diagram of the workflow of a method for reducing local oscillation leakage is provided, wherein steps S410 to S430 belong to the first correction stage (e.g., the amplifier correction stage described above), and steps S440 to S470 belong to the second correction stage following the first correction stage (e.g., the mixer correction stage described above). It should be noted that... Figure 4 The illustrated workflow is for illustrative purposes only and is not intended to limit the scope of the invention. For example, one or more steps may be performed... Figure 4 The workflow shown has been added, deleted, or modified. Furthermore, these steps do not necessarily need to be followed exactly to achieve the same result. Figure 4 Execute in the order shown.

[0024] In step S410, the transmitter may shut down its internal mixer (e.g., Figure 1 Mixer 104 shown.

[0025] In step S420, the transmitter may utilize its internal analog amplifier (e.g., Figure 1 The transmitter baseband amplifier 103 shown amplifies the baseband signal to generate an amplified baseband signal.

[0026] In step S430, the transmitter may utilize its internal correction logic circuitry (e.g., Figure 1 The correction logic circuit 130 shown controls the first correction signal source in the transmitter to output a first correction signal to the analog amplifier based on the direct current (DC) signal in the amplified base frequency signal, so that the DC signal is minimized.

[0027] In step S440, the transmitter can turn on the mixer.

[0028] In step S450, the transmitter can use the mixer to up-convert the amplified baseband signal to generate a radio frequency signal.

[0029] In step S460, the transmitter may utilize its internal self-mixer (e.g., Figure 1 The self-mixer 107 shown performs self-mixing based on the radio frequency signal to generate a feedback signal.

[0030] In step S470, the transmitter can use the correction logic circuit to control the second correction signal source in the transmitter to output a second correction signal to the mixer according to the feedback base frequency signal in the feedback signal, so as to minimize the feedback base frequency signal.

[0031] In summary, this invention detects the amplified baseband signal A2 without frequency upscaling or downscaling. CAL (or attenuated base frequency signal A0) ATT Under these conditions, the DC bias V of the transmitter's baseband amplifier 103... DC,TXBB The information is loaded at the DC frequency, not ω0. Therefore, when the amplification gain of the transmitter's baseband amplifier 103 increases, the amplified baseband signal A2... CAL (or attenuated base frequency signal A0) ATT When the power at frequency ω0 in the equation increases, the DC bias V... DC,TXBB The detection of information will not be interfered with. Furthermore, the amplification gain of the transmitter baseband amplifier 103 can be adjusted to correct the DC bias V of the mixer 104. DC,MIXER When the time is minimized, and at this time the DC bias V of the transmitter baseband amplifier 103 is minimized. DC,TXBB The correction has been completed (e.g., minimized). Therefore, by detecting the feedback signal A0 FB (or adjusted feedback signal A1) FB The DC bias V of mixer 104 can be determined by the power at frequency ω0 in the equation. DC,MIXER Furthermore, by establishing a calibration table to record the calibration values ​​required for different gain values, the present invention can properly correct the DC bias V of the transmitter fundamental frequency amplifier 103 under various gain settings. DC,TXBBand / or the DC bias V of mixer 104 DC,MIXER Therefore, the present invention can effectively solve the problems of related technologies.

[0032] The above description is only the preferred embodiment of the present invention. All equivalent changes and modifications made within the scope of the claims of the present invention shall be covered by the present invention.

[0033] [Symbol Explanation]

[0034] 10: Teleporter

[0035] 101: Digital-to-Analog Converter

[0036] 102: Transimpedance Amplifier

[0037] 103: Transmitter baseband amplifier

[0038] 104: Mixer

[0039] 105: Power Amplifier Driver

[0040] 106: Power Amplifier

[0041] 107: Self-mixer

[0042] 110: Amplifier calibration signal source

[0043] 111: Current-mode digital-to-analog converter

[0044] 112: Transmitter Baseband Amplifier Calibration Table

[0045] 120: Mixer calibration signal source

[0046] 121: Current-mode digital-to-analog converter

[0047] 122: Mixer Calibration Table

[0048] 130: Correction Logic Circuit

[0049] 140: Analog-to-digital converter

[0050] 150: Power Spectrum Density Circuit

[0051] 160: Attenuator

[0052] 170: Programmable gain amplifier

[0053] D CAL Digital test signal

[0054] A LO Local oscillation signal

[0055] A0 CAL Simulated test signal

[0056] A1 CAL :Fundamental frequency signal

[0057] A2 CAL Amplified base frequency signal

[0058] A0 RF A1 RF A2 RF Radio frequency signal

[0059] A0 FB Feedback signal

[0060] A1 FB Feedback signal after adjustment

[0061] A0 ATT Attenuated base frequency signal

[0062] I1 CAL Amplifier correction signal

[0063] I2 CAL Mixer correction signal

[0064] D FB D FB1 D FB2 Digital signal

[0065] D PSD D PSD1 D PSD2 Calculation results

[0066] D1 CTRL D2 CTRL Control signals

[0067] ω0,ω LO :frequency

[0068] S410~S450: Steps

Claims

1. A transmitter, comprising: An analog amplifier is used to amplify a fundamental frequency signal to produce an amplified fundamental frequency signal. A mixer is used to up-convert the amplified baseband signal to generate a radio frequency signal. A self-mixer is used to self-mix the radio frequency signal to generate a feedback signal. A first correction signal source is coupled to the analog amplifier to output a first correction signal to the analog amplifier; A second correction signal source is coupled to the mixer to output a second correction signal to the mixer; as well as A correction logic circuit is coupled to the first correction signal source and the second correction signal source, wherein: In the first correction stage, the correction logic circuit controls the first correction signal based on the DC signal in the amplified baseband signal to minimize the DC signal; and In the second correction stage following the first correction stage, the correction logic circuit controls the second correction signal based on the feedback base frequency signal in the feedback signal so that the feedback base frequency signal is minimized.

2. The transmitter according to claim 1, wherein the frequency of the baseband signal is ω0, and the feedback baseband signal is the signal component with frequency ω0 in the feedback signal.

3. The transmitter according to claim 2, further comprising: An analog-to-digital converter is used to perform analog-to-digital conversion based on the amplified baseband signal in the first correction stage to generate a first digital signal, and to perform analog-to-digital conversion based on the feedback signal in the second correction stage to generate a second digital signal; and A power spectral density circuit, coupled to the analog-to-digital converter and the correction logic circuit, is used to calculate the power of the signal component with frequency 0 in the first digital signal to obtain a first calculation result, and to calculate the power of the signal component with frequency ω0 in the second digital signal to obtain a second calculation result, wherein the first calculation result and the second calculation result represent the power of the DC signal and the power of the feedback baseband signal, respectively. The correction logic circuit controls the first correction signal based on the first calculation result and controls the second correction signal based on the second calculation result.

4. The transmitter according to claim 3, further comprising: An attenuator, coupled between the analog amplifier and the analog-to-digital converter, is used to reduce the amplitude of the amplified fundamental frequency signal in the first correction stage to generate an attenuated fundamental frequency signal. In the first correction stage, the analog-to-digital converter performs analog-to-digital conversion on the attenuated baseband signal to generate the first digital signal.

5. The transmitter according to claim 3, further comprising: A programmable gain amplifier, coupled between the self-mixer and the analog-to-digital converter, is used to adjust the amplitude of the feedback signal in the second correction stage to generate an adjusted feedback signal. In the second correction stage, the analog-to-digital converter performs analog-to-digital conversion on the adjusted feedback signal to generate the second digital signal.

6. The transmitter of claim 1, wherein the correction logic circuit corrects a first DC bias of the analog amplifier by minimizing the DC signal, and the correction logic circuit corrects a second DC bias of the mixer by minimizing the feedback baseband signal.

7. The transmitter of claim 1, wherein the first correction signal source comprises: A first calibration table is used to record multiple first digital calibration values ​​corresponding to multiple first candidate gains of the analog amplifier, wherein the first calibration table outputs the corresponding first digital calibration value among the multiple first digital calibration values ​​when the gain of the analog amplifier is set to a first specific gain among the multiple first candidate gains. A first digital-to-analog converter is coupled to the first calibration table and is used to output the first calibration signal according to the corresponding first digital calibration value.

8. The transmitter of claim 1, wherein the second correction signal source comprises: The second calibration table is used to record multiple second digital calibration values ​​corresponding to multiple second candidate gains of the mixer, wherein the second calibration table outputs the corresponding second digital calibration value among the multiple second digital calibration values ​​when the gain of the mixer is set to the second specific gain among the multiple second candidate gains; A second digital-to-analog converter is coupled to the second calibration table to output the second calibration signal based on the corresponding second digital calibration value.

9. A method for reducing local oscillation leakage in a transmitter, comprising: In the first calibration stage, the analog amplifier of the transmitter is used to amplify the fundamental frequency signal to generate an amplified fundamental frequency signal; In the first correction stage, the correction logic circuit of the transmitter controls the first correction signal source in the transmitter to output a first correction signal to the analog amplifier based on the DC signal in the amplified base frequency signal, so that the DC signal is minimized. In the second correction stage following the first correction stage, the amplified baseband signal is up-frequencyed using the mixer of the transmitter to generate a radio frequency signal. In the second calibration stage, the transmitter's self-mixer performs self-mixing based on the radio frequency signal to generate a feedback signal; and In the second correction stage, the correction logic circuit uses the feedback base frequency signal in the feedback signal to control the second correction signal source in the transmitter to output a second correction signal to the mixer, so that the feedback base frequency signal is minimized.

10. The method according to claim 9, wherein, The baseband signal has a frequency of ω0, and the feedback baseband signal is the signal component with frequency ω0 in the feedback signal.

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