Current source type PWM rectifier control method under unbalanced working condition of power grid

Abstract

The invention discloses a current source type PWM rectifier control method under a power grid imbalance working condition, and the method comprises the steps: firstly designing a current source type PWM rectifier and a corresponding control system, carrying out the small-signal linearization modeling of the circuit topology of the control system, and building an open-loop transfer function of the control system; secondly, providing a direct current side direct control strategy which does not need a complex phase-locked loop and positive and negative sequence calculation; finally, a modulation strategy for the current source type PWM rectifier is designed, and the modulation performance of the control system is further improved. The novel control method is provided for the current source type PWM rectifier under the unbalanced power grid, compared with a traditional double-closed-loop control strategy, the THD value of the control method is obviously reduced, and the control method has the advantages of being high in power factor, low in input current harmonic wave and the like.

Description

Technical Field

[0001] This invention relates to a control method for a current-source PWM rectifier under unbalanced power grid conditions, belonging to the field of rectifier control technology. Background Technology

[0002] With the development of power electronics technology, PWM rectifiers have become increasingly diverse. Based on the different energy storage components, common PWM rectifiers are divided into two structures: voltage source PWM rectifiers (VSRs) and current source PWM rectifiers (CSRs). Voltage source PWM rectifiers are also known as boost rectifiers, while current source PWM rectifiers are also known as buck rectifiers.

[0003] Currently, research on three-phase CSRs mainly focuses on grid voltage balance conditions. However, voltage imbalance is prevalent in actual operating conditions, leading to low-frequency pulsations on the DC side of the CSR and a large number of harmonics in the grid-side current, thus degrading rectifier performance. Under three-phase unbalanced grid voltage conditions, the traditional control strategies commonly used in CSRs require extracting the positive and negative sequence components of voltage and current, which is a complex and computationally intensive process. Summary of the Invention

[0004] Purpose of the invention: In order to overcome the shortcomings of the existing technology, the present invention provides a control method for a current source type PWM rectifier under unbalanced grid conditions. The control method adopts a DC-side dual closed-loop control strategy and does not require positive and negative sequence components. By using a combination of notch filter and regulator, the second harmonic pulsation of DC-side current is suppressed, so that the power quality of the system is significantly improved under unbalanced conditions.

[0005] Technical solution: To achieve the above objectives, the technical solution adopted by this invention is as follows:

[0006] A control method for a current-source PWM rectifier under unbalanced grid conditions is presented. This method improves the input three-phase bus current waveform by adding a second-harmonic trap filter and a dedicated regulator for the second harmonic to the voltage and current dual-loop control system. The method also achieves rapid tracking of the output voltage and current through a designed voltage and current dual-loop controller. Specifically, in a control system based on a current-source PWM rectifier, a direct DC-side control strategy that eliminates the need for complex phase-locked loops (PLLs) and positive / negative sequence calculations is employed. The control parameters for the voltage and current dual-loop control are determined based on the open-loop transfer function of the control system. Simultaneously, a trap filter is used. Eliminate the DC current reference value i of the outer voltage loop output ref The second harmonic in, and, through the regulator The second harmonic component of the DC current i is extracted and canceled at the output side of the inner current loop, thereby eliminating the second harmonic in the DC bus; where s is the negative frequency, ω0 is the grid frequency, K is the controller coefficient, and k r This is the gain coefficient.

[0007] Specifically, the DC-side direct control strategy includes the following steps:

[0008] Step 1: The outer voltage loop uses a PI controller to convert the DC voltage u... o With DC voltage reference value u o,ref After comparing the PI controller connected to the outer voltage loop, observe the Bode plot of the open-loop transfer function after adding the PI controller, and select the proportional coefficient and integral coefficient of the PI controller according to the gain margin and phase margin.

[0009] Step 2: Under unbalanced grid conditions, due to the influence of the negative sequence component of the grid, the DC voltage u o The DC current i will contain a second harmonic. A notch filter F(s) is used to eliminate the DC current reference value i output by the outer voltage loop. ref The second harmonic in the middle is used to obtain the corrected DC current reference value i. ref A notch filter F(s) can rapidly attenuate the input signal at a specified frequency point to achieve a filtering effect that blocks the passage of signals at a specified frequency.

[0010] Step 3: The inner current loop uses a PI controller to compare the DC current i with the corrected DC current reference value i. ref 'Compare the PI controller connected to the inner current loop, observe the Bode plot of the open-loop transfer function after adding the PI controller, and select the proportional coefficient and integral coefficient of the PI controller according to the gain margin and phase margin.'

[0011] Step 4: The regulator D(s) can extract the component at a specified frequency point. The second harmonic component in the DC current i is extracted by the regulator D(s), and this second harmonic component is subtracted on the output side of the current inner loop to eliminate the second harmonic in the DC bus, thereby obtaining the modulation ratio m with reduced second harmonic content. The modulation strategy is executed based on the modulation ratio m.

[0012] Specifically, the current-source PWM rectifier includes a three-phase bridge arm consisting of six bridge arm switches S, with each bridge arm switch S connected in series with a diode D to block reverse current; in the control system, the AC input voltage, after being filtered by an LC filter to remove high-frequency current harmonics, is connected to the current-source PWM rectifier, and a freewheeling diode D is connected at the rear end of the current-source PWM rectifier. F The load resistor R and the DC-side filter capacitor C are connected through the DC-side filter inductor L.

[0013] Specifically, two freewheeling diodes D are connected to the back end of the current source PWM rectifier. F These are the freewheeling diodes D connected in parallel with the bridge arm switch S on the positive side of the three-phase bridge arm, respectively. FP The negative freewheeling diode D is connected in parallel with the bridge arm switch S on the negative side of the three-phase bridge arm. FN When the bridge arm switch S on the positive side of the three-phase bridge arm is turned off, the DC side filter inductor L... P Because the current cannot change abruptly to generate a reverse induced electromotive force, the forward freewheeling diode D... FP Turn on, turning on the DC-side filter inductor L P The stored energy is fed back to the DC positive bus, while suppressing voltage spikes when the bridge arm switch S is turned off; when the bridge arm switch S on the negative side of the three-phase bridge arm is turned off, the DC side filter inductor L... N Because the current cannot change abruptly to generate a reverse induced electromotive force, the negative freewheeling diode D... FN Turn on, turning on the DC-side filter inductor L N The stored energy is fed into the DC negative bus and connected to the positive freewheeling diode D. FP A bidirectional current-sequence loop is formed to ensure continuous current.

[0014] Specifically, the DC-side filter capacitor C includes a DC positive bus terminal capacitor C. P and DC negative bus terminal capacitor C N The DC positive bus terminal capacitor C P Used to filter out high-frequency ripple on the DC positive bus, and works in conjunction with the LC filter to reduce voltage disturbances caused by the operation of the bridge arm switch S; the DC negative bus terminal capacitor C N Used to stabilize the DC negative bus potential, in conjunction with freewheeling diode D F This enables energy feedback pathways while suppressing second harmonics introduced by grid imbalances.

[0015] Specifically, the DC-side filter inductor L includes a DC positive bus series inductor L. P Inductor L connected in series with DC negative bus N ;

[0016] The DC positive bus series inductor L P With the positive freewheeling diode D FP In coordination, when the bridge arm switch S on the positive side of the three-phase bridge arm is turned off, the DC positive bus series inductor L... P Storing inductor energy to form a DC positive bus series inductor L P →Front-side freewheeling diode D FP →Load resistor R→This serves as a freewheeling path for the DC negative bus, preventing voltage spikes caused by sudden current changes when the bridge arm switch S is turned off; simultaneously, the DC positive bus is connected in series with the inductor L. P Capacitor C at the DC positive bus terminal PIn combination, they form an LC filter branch to reduce the high-frequency current ripple generated by the operation of the bridge arm switch S.

[0017] The DC negative bus series inductor L N With negative side freewheeling diode D FN In coordination, when the bridge arm switch S on the negative side of the three-phase bridge arm is turned off, the DC negative bus series inductor L... N Storing inductor energy to form a DC negative bus series inductor L N →Negative-side freewheeling diode D FN →Load resistor R→This serves as a freewheeling path for the DC positive bus, preventing voltage spikes caused by sudden current changes when the bridge arm switch S is turned off; simultaneously, the DC negative bus is connected in series with inductor L. N Capacitor C at the DC negative bus terminal N In combination, they form an LC filter branch to reduce the high-frequency current ripple generated by the operation of the bridge arm switch S.

[0018] Specifically, the inner current loop takes the DC-side output current deviation as input, and outputs a modulation ratio m after being adjusted by a PI controller; in the modulation strategy of the current source type PWM rectifier, the AC-side input voltage period is first divided into 12 sectors, then the sector where the target vector is located is determined, and then the target vector is synthesized by combining the modulation ratio m and the sector basic vector, thereby generating the trigger pulse of each bridge arm switch to adjust the DC-side output.

[0019] Specifically, the open-loop transfer function of the control system is constructed through the following process:

[0020] S1. Based on the state-space averaging method, small-signal linearization modeling is performed. Considering the characteristic that the switching frequency of the control system is slower than that of PWM, the bridge arm switch S and diode D are converted into controlled voltage sources and controlled current sources, respectively. The circuit topology of the control system is linearized. Line voltage sampling is performed at the front end of the LC filter and the phase voltage is calculated. An AC disturbance signal is introduced in steady state to solve the transfer function of the DC-DC converter.

[0021] S2. Write out the differential formulas for the input and output inductor voltage and capacitor current, perform small-signal linearization modeling, and obtain the small-signal linearized equivalent model:

[0022]

[0023] in: The perturbation component represented by ·. The first derivative is represented by , where s is the negative frequency and M is the steady-state quantity of the modulation ratio. U is the disturbance component of the modulation ratio; N,eq i is the equivalent value of the AC power supply voltage. LF L is the current flowing through the AC side filter inductor.F,eq C is the equivalent value of the AC side filter inductance. F,eq u is the equivalent value of the AC side filter capacitor. LF The voltage across the AC-side filter inductor, u CF For AC measurement, the voltage across the filter capacitor is: L is the DC-side filter inductor, C is the DC-side filter capacitor, R is the load resistor, i is the current flowing through the DC-side filter inductor L (i.e., DC current), i0 is the current flowing through the load resistor R, and u0 is the voltage across the load resistor R (i.e., DC voltage).

[0024] S3. Derive the open-loop transfer function based on the equivalent model of signal linearization:

[0025]

[0026] Where: m is the modulation ratio.

[0027] Specifically, when implementing the modulation strategy of the current-source PWM rectifier, the AC input voltage period is first divided into 12 sectors, then the sector in which the target vector is located is determined, and then the target vector is synthesized using the sector basic vectors. At the same time, the 0-vector mode in the traditional circuit topology is optimized to the mode where all bridge arm switches S are fully off, and the duty cycle δ of the three-phase bridge arm of the current-source PWM rectifier is calculated. i :

[0028]

[0029] Where: the subscript i represents the three phases of a three-phase electrical system, i = R, S, T; u o,ref The set output voltage of the current source PWM rectifier is given by m, where m is the modulation ratio, ranging from 0 to 1, and u is the modulation ratio. CF,i To measure the voltage across the filter capacitor of phase i AC, V m This represents the amplitude of the AC input voltage.

[0030] Beneficial Effects: The current-source PWM rectifier control method under unbalanced grid conditions provided by this invention has the following advantages compared with the prior art: 1. By constructing a modulation strategy suitable for the circuit topology, this invention can reduce the impact of the modulation strategy on the total harmonic distortion of the input three-phase AC bus and reduce the THD value; 2. By adding a second harmonic notch filter and a dedicated regulator for the second harmonic on the basis of voltage and current dual closed loop, this invention can suppress the DC bus current, thereby improving the waveform of the input three-phase bus current; 3. Through the voltage and current dual closed loop controller design, this invention can achieve rapid tracking of the output voltage and current. Attached Figure Description

[0031] Figure 1 This is the main circuit of the method of the present invention;

[0032] Figure 2 This is a control block diagram of the method of the present invention;

[0033] Figure 3 This is the small-signal linearization equivalent model of the current-source PWM rectifier of the present invention;

[0034] Figure 4 Open-loop Bode plots are shown for the cases where the inner current loop has no current controller and the case has a current controller.

[0035] Figure 5 Open-loop Bode plots are shown for the cases with and without a voltage controller in the outer voltage loop.

[0036] Figure 6 This refers to the switching mode in sector one;

[0037] Figure 7 The simulated waveform of the three-phase input current under full load at rated voltage under traditional dual closed-loop control;

[0038] Figure 8 THD analysis of three-phase input current under traditional dual closed-loop control;

[0039] Figure 9 This is a simulation waveform of the three-phase input current under the rated voltage full load condition under the control strategy of this invention;

[0040] Figure 10 This is a simulated waveform of the DC output voltage under full load conditions at rated voltage under the control strategy of this invention;

[0041] Figure 11 THD analysis of three-phase input current under the control strategy of this invention;

[0042] Figure 12 This refers to the power factor under the rated voltage full load condition under the control strategy of this invention. Detailed Implementation

[0043] The present invention will now be described in detail with reference to the accompanying drawings and specific embodiments.

[0044] like Figure 1 , 2 The diagram shows the main circuit and control block diagram of a current source type PWM rectifier control method under power grid imbalance conditions.

[0045] like Figure 1As shown, the current-source PWM rectifier in this case includes a three-phase bridge consisting of six bridge arm switches S. Each bridge arm switch S is connected in series with a diode D to block reverse current. In the control system, the AC input voltage is filtered by an LC filter to remove high-frequency current harmonics before being connected to the current-source PWM rectifier. Two freewheeling diodes D are first connected to the downstream end of the current-source PWM rectifier. F Two DC-side filter inductors L are used to maintain the continuity and stability of the DC current, and a freewheeling diode D is used. F The DC-side filter inductor L is the core component that constitutes the "current source" characteristic; the load resistor R and two DC-side filter capacitors C are connected after the DC-side filter inductor L to filter out the ripple of the DC voltage.

[0046] like Figure 1 As shown, two freewheeling diodes D F These are the freewheeling diodes D connected in parallel with the bridge arm switch S on the positive side of the three-phase bridge arm, respectively. FP The negative freewheeling diode D is connected in parallel with the bridge arm switch S on the negative side of the three-phase bridge arm. FN The two DC-side filter inductors L are respectively the series inductors L1 and L2 of the DC positive bus. P Inductor L connected in series with DC negative bus N The two DC-side filter capacitors C are the DC positive bus terminal capacitors C0 and C1 respectively. P and DC negative bus terminal capacitor C N .

[0047] As can be seen, the circuit topology of the control system based on the current source PWM rectifier in this case is simple in structure, can achieve single-stage step-down rectification with a wide range of output voltage regulation capability, is easy to connect in parallel, and has high reliability under DC short circuit. Compared with the traditional three-phase six-switch rectifier topology, this case can use switching devices with lower rated voltages, which typically have lower on-resistance, better switching characteristics, and lower cost. Therefore, the circuit topology of this case is more advantageous in terms of cost and efficiency.

[0048] The circuit topology of the control system in this case is modeled using small-signal linearization, and the open-loop transfer function of the control system is established, including the following steps:

[0049] S11. Based on the state-space averaging method, small-signal linearization modeling is performed. Taking into account the characteristic that the switching frequency of the control system is slower than that of PWM, the bridge arm switch S and diode D are converted into controlled voltage source and controlled current source.

[0050] To obtain the transfer function of the DC-DC converter, the circuit topology of the control system needs to be linearized, i.e., an AC disturbance signal needs to be introduced near the steady operating point. In this case, line voltage sampling needs to be performed after the input three-phase voltage to obtain the three-phase voltage u. RS u ST u TR Then, the phase voltage is calculated to obtain the three-phase voltage u. R u S u T , used for sector identification in SLO modulation strategy.

[0051] S12. Write the differential formulas for the input and output inductor voltage and capacitor current, perform small-signal linearization modeling, and obtain the following... Figure 3 The small-signal linearization equivalent model shown is as follows:

[0052]

[0053] in: The perturbation component represented by ·. The first derivative is represented by , where s is the negative frequency and M is the steady-state quantity of the modulation ratio. U is the disturbance component of the modulation ratio; N,eq i is the equivalent value of the AC power supply voltage. LF L is the current flowing through the AC side filter inductor. F,eq C is the equivalent value of the AC side filter inductance. F,eq u is the equivalent value of the AC side filter capacitor. LF The voltage across the AC-side filter inductor, u CF The AC filter capacitor is used to measure the voltage across its terminals; L is the DC filter inductor, C is the DC filter capacitor, R is the load resistor, i is the current flowing through the DC filter inductor L (i.e., DC current), i0 is the current flowing through the load resistor R, and u0 is the voltage across the load resistor R (i.e., DC voltage).

[0054] S13. Derive the open-loop transfer function based on the equivalent model of signal linearization:

[0055]

[0056] Where: m is the modulation ratio.

[0057] Based on the open-loop transfer function of the control system, this paper constructs a direct DC-side control strategy that does not require complex phase-locked loops and positive / negative sequence calculations. The outer voltage loop takes the DC-side output voltage deviation as input, and after adjustment by a PI controller, generates a DC current reference value. A notch filter is used to filter out the second harmonic introduced by grid imbalance, ensuring the purity of the input to the inner current loop. The inner current loop takes the DC-side output current deviation as input, and after adjustment by a PI controller, outputs a modulation ratio to drive a 12-sector space vector modulation module to generate a switching transistor trigger pulse, achieving coordinated control of fast current tracking and grid-side harmonic suppression. The direct DC-side control strategy specifically includes the following steps:

[0058] S21. The inner current loop uses a PI controller to construct an internal control loop, comparing the DC current i with the corrected DC current reference value i. ref 'Compare the PI controller connected to the inner current loop, observe the Bode plot of the open-loop transfer function after adding the PI controller, and select the proportional and integral coefficients of the PI controller based on the gain margin and phase margin; in this case, the open-loop Bode plots with and without a current controller in the inner current loop are as follows.' Figure 4 As shown, the horizontal axis represents frequency, the vertical axis in the upper figure represents amplitude, and the vertical axis in the lower figure represents phase.

[0059] S22. The outer voltage loop uses a PI controller to convert the DC voltage u o With DC voltage reference value u o,ref After comparison, the PI controller connected to the voltage outer loop is similar to that of the current inner loop. By observing the Bode plot of the open-loop transfer function after adding the PI controller, the proportional and integral coefficients of the PI controller are selected based on the gain margin and phase margin. In this case, the open-loop Bode plots with and without a voltage controller in the voltage outer loop are as follows: Figure 5 As shown, the horizontal axis represents frequency, the vertical axis in the upper figure represents amplitude, and the vertical axis in the lower figure represents phase.

[0060] S23. Under unbalanced grid conditions, due to the influence of the negative sequence component of the grid, the DC voltage u o Second harmonics will appear in the DC current i; since the notch filter F(s) can rapidly attenuate the input signal at a specified frequency point to achieve the filtering effect of blocking the passage of the specified frequency signal, the notch filter F(s) can be used to eliminate the DC current reference value i of the voltage outer loop output. ref The second harmonic in the middle is used to obtain the corrected DC current reference value i. ref The expression for the notch filter F(s) is:

[0061]

[0062] Where: s is the negative frequency, ω0 is the grid frequency, and K is the controller coefficient.

[0063] S24, In addition to filtering out the DC current reference value i ref To eliminate the second harmonic in the DC bus, this case also employs a regulator D(s). The expression for regulator D(s) is as follows:

[0064]

[0065] Where: k r This is the gain coefficient.

[0066] The regulator D(s) can extract the component at a specified frequency point. The second harmonic component in the DC current i is extracted by the regulator D(s), and this second harmonic component is subtracted on the output side of the current inner loop. The second harmonic in the DC bus can be eliminated, and the modulation ratio m with reduced second harmonic content is obtained. The subsequent modulation strategy is executed based on the modulation ratio m.

[0067] The modulation strategy of the current-source PWM rectifier constructed in this case first divides the AC input voltage period into 12 sectors, then determines the sector where the target vector is located, and then combines the modulation ratio and the sector's basic vector to synthesize the target vector, thereby generating trigger pulses for each bridge arm switch to regulate the DC output. The modulation strategy specifically includes the following steps:

[0068] S31. Divide the AC side input voltage cycle into 12 sectors, then determine the sector where the target vector is located, and then use the sector basic vectors to synthesize the target vector.

[0069] S32. Optimize the 0-vector mode (e.g., mode 010) in the traditional topology to a mode where all bridge arm switches S are fully off (e.g., mode 000) to adapt to the circuit topology of this case; calculate the three-phase bridge arm duty cycle δ of the current source PWM rectifier. i :

[0070]

[0071] Where: the subscript i represents the three phases of a three-phase electrical system, i = R, S, T; u o,ref The set output voltage of the current source PWM rectifier is given by m, where m is the modulation ratio, ranging from 0 to 1, and u is the modulation ratio. CF,i To measure the voltage across the filter capacitor of phase i AC, V m This represents the amplitude of the AC input voltage.

[0072] like Figure 6 As shown, this is the switching mode in sector one. The three-phase duty cycle of each sector is shown in Table 1.

[0073] Table 1 shows the duty cycle applied by the input voltage sector.

[0074] sector <![CDATA[δ eff,R ]]> <![CDATA[δ eff,S ]]> <![CDATA[δ eff,T ]]> 1,7 <![CDATA[δ R ]]> <![CDATA[1-δ T +d d ]]> <![CDATA[δ T ]]> 2,8 <![CDATA[δ R ]]> <![CDATA[1-δ R +d d ]]> <![CDATA[δ T ]]> 3,9 <![CDATA[1-δ S +d d ]]> <![CDATA[δ S ]]> <![CDATA[δ T ]]> 4,10 <![CDATA[1-δ T +d d ]]> <![CDATA[δ S ]]> <![CDATA[δ T ]]> 5,11 <![CDATA[δ R ]]> <![CDATA[δ S ]]> <![CDATA[1-δ R +d d ]]> 6,12 <![CDATA[δ R ]]> <![CDATA[δ S ]]> <![CDATA[1-δ S +d d ]]>

[0075] To verify the superiority of the control method provided by this invention, simulation analysis was performed in MATLAB / Simulink. The system parameters are shown in Table 2, and the simulation results are as follows. Figure 7-12 As shown.

[0076] Table 2 System Parameter Table

[0077]

[0078]

[0079] Under the traditional dual-closed-loop control strategy, the waveform of the three-phase input current under rated input voltage and full load condition after 0.3s is as follows: Figure 7 As shown, the THD of the AC side current is 4.59% (e.g. Figure 8 As shown, the horizontal axis represents frequency, and the vertical axis represents the content of each harmonic and its magnitude relative to the fundamental frequency. Under the control strategy of this case, the waveform of the three-phase input current under rated input voltage and full load condition after 0.3s is as follows. Figure 9 As shown, the DC voltage on the output side is stabilized at the rated value of 230V (e.g., Figure 10 As shown, the horizontal axis represents time, and the vertical axis represents voltage. The THD of the AC side current is 4.11% (e.g., Figure 11 As shown, the horizontal axis represents frequency, and the vertical axis represents the content of each harmonic and its magnitude relative to the fundamental frequency. This represents a significant improvement over the traditional dual-closed-loop design, and the harmonic distortion is less than 5%, meeting design requirements. At this point, the power factor can reach 1 (e.g., ...). Figure 12 As shown, the horizontal axis represents time, and the vertical axis represents the power factor value.

[0080] As can be seen from the simulation results, the current source PWM rectifier provided in this case has excellent characteristics such as high power factor and low input current harmonics.

[0081] The foregoing has shown and described the basic principles, main features, and advantages of the present invention. Those skilled in the art should understand that the above embodiments do not limit the present invention in any way, and all technical solutions obtained by equivalent substitution or equivalent transformation fall within the protection scope of the present invention.

Claims

1. A control method for a current-source PWM rectifier under unbalanced power grid conditions, characterized in that: In the control system based on a current-source PWM rectifier, a direct DC-side control strategy is adopted, and the control parameters for the voltage and current dual closed loops are determined based on the open-loop transfer function of the control system; simultaneously, a notch filter is used. Eliminate the DC current reference value i of the outer voltage loop output ref The second harmonic in, and, through the regulator The second harmonic component of the DC current i is extracted and canceled at the output side of the inner current loop, thereby eliminating the second harmonic in the DC bus; where s is the negative frequency, ω0 is the grid frequency, K is the controller coefficient, and k r This is the gain coefficient.

2. The control method for a current-source PWM rectifier under unbalanced grid conditions according to claim 1, characterized in that: The DC-side direct control strategy includes the following steps: Step 1: The outer voltage loop uses a PI controller to convert the DC voltage u... o With DC voltage reference value u o,ref After comparing the PI controller connected to the outer voltage loop, observe the Bode plot of the open-loop transfer function after adding the PI controller, and select the proportional coefficient and integral coefficient of the PI controller according to the gain margin and phase margin. Step 2: Under unbalanced grid conditions, a notch filter F(s) is used to eliminate the DC current reference value i output by the outer voltage loop. ref The second harmonic in the middle is used to obtain the corrected DC current reference value i. ref '; Step 3: The inner current loop uses a PI controller to compare the DC current i with the corrected DC current reference value i. ref 'Compare the PI controller connected to the inner current loop, observe the Bode plot of the open-loop transfer function after adding the PI controller, and select the proportional coefficient and integral coefficient of the PI controller according to the gain margin and phase margin.' Step 4: Extract the second harmonic component in the DC current i through the regulator D(s), and subtract this second harmonic component on the output side of the current inner loop to eliminate the second harmonic in the DC bus, and obtain the modulation ratio m with reduced second harmonic content. Execute the modulation strategy based on the modulation ratio m.

3. The control method for a current-source PWM rectifier under unbalanced grid conditions according to claim 1, characterized in that: The current-source PWM rectifier includes a three-phase bridge arm consisting of six bridge arm switches S, with each bridge arm switch S connected in series with a diode D to block reverse current. In the control system, the AC input voltage, after being filtered by an LC filter to remove high-frequency current harmonics, is connected to the current-source PWM rectifier, and a freewheeling diode D is connected to the downstream end of the current-source PWM rectifier. F The load resistor R and the DC-side filter capacitor C are connected through the DC-side filter inductor L.

4. The control method for a current-source PWM rectifier under unbalanced grid conditions according to claim 3, characterized in that: Two freewheeling diodes D are connected to the back end of the current source PWM rectifier. F These are the freewheeling diodes D connected in parallel with the bridge arm switch S on the positive side of the three-phase bridge arm, respectively. FP The negative freewheeling diode D is connected in parallel with the bridge arm switch S on the negative side of the three-phase bridge arm. FN .

5. The control method for a current-source PWM rectifier under unbalanced grid conditions according to claim 3, characterized in that: The DC-side filter capacitor C includes a DC positive bus terminal capacitor C. P and DC negative bus terminal capacitor C N The DC positive bus terminal capacitor C P Used to filter out high-frequency ripple on the DC positive bus, and works in conjunction with the LC filter to reduce voltage disturbances caused by the operation of the bridge arm switch S; the DC negative bus terminal capacitor C N Used to stabilize the DC negative bus potential, in conjunction with freewheeling diode D F This enables energy feedback pathways while suppressing second harmonics introduced by grid imbalances.

6. The control method for a current-source PWM rectifier under unbalanced grid conditions according to claim 3, characterized in that: The DC-side filter inductor L includes a DC positive bus series inductor L. P Inductor L connected in series with DC negative bus N .

7. The control method for a current-source PWM rectifier under unbalanced grid conditions according to claim 1, characterized in that: The inner current loop takes the DC-side output current deviation as input, and outputs the modulation ratio m after being adjusted by the PI controller. In the modulation strategy of the current source type PWM rectifier, the AC-side input voltage period is first divided into 12 sectors, then the sector where the target vector is located is determined, and then the target vector is synthesized by combining the modulation ratio m and the sector basic vector, thereby generating the trigger pulse of each bridge arm switch to adjust the DC-side output.

8. The control method for a current-source PWM rectifier under unbalanced grid conditions according to claim 1, characterized in that: The open-loop transfer function of the control system is constructed through the following process: S1. Based on the state-space averaging method, small-signal linearization modeling is performed. Considering the characteristic that the switching frequency of the control system is slower than that of PWM, the bridge arm switch S and diode D are converted into controlled voltage sources and controlled current sources, respectively. The circuit topology of the control system is linearized. Line voltage sampling is performed at the front end of the LC filter and the phase voltage is calculated. An AC disturbance signal is introduced in steady state to solve the transfer function of the DC-DC converter. S2. Write out the differential formulas for the input and output inductor voltage and capacitor current, perform small-signal linearization modeling, and obtain the small-signal linearized equivalent model: in: The perturbation component represented by ·. surface The first derivative is given, where s is the negative frequency and M is the steady-state quantity of the modulation ratio. U is the disturbance component of the modulation ratio; N,eq i is the equivalent value of the AC power supply voltage. LF L is the current flowing through the AC side filter inductor. F,eq C is the equivalent value of the AC side filter inductance. F,eq u is the equivalent value of the AC side filter capacitor. LF The voltage across the AC-side filter inductor, u CF For AC measurement, the voltage across the filter capacitor is: L is the DC-side filter inductor, C is the DC-side filter capacitor, R is the load resistor, i is the current flowing through the DC-side filter inductor L, i0 is the current flowing through the load resistor R, and u0 is the voltage across the load resistor R. S3. Derive the open-loop transfer function based on the equivalent model of signal linearization: Where: m is the modulation ratio.

9. The control method for a current-source PWM rectifier under unbalanced grid conditions according to claim 1, characterized in that: When implementing the modulation strategy of the current-source PWM rectifier, the AC input voltage period is first divided into 12 sectors, then the sector in which the target vector is located is determined, and then the target vector is synthesized using the sector basic vectors. At the same time, the 0-vector mode is optimized to the mode where all bridge arm switches S are fully turned off, and the duty cycle δ of the three-phase bridge arm of the current-source PWM rectifier is calculated. i : Where: the subscript i represents the three phases of a three-phase electrical system, i = R, S, T; u o,ref The set output voltage of the current source PWM rectifier is given by m, where m is the modulation ratio, ranging from 0 to 1, and u is the modulation ratio. CF,i To measure the voltage across the filter capacitor of phase i AC, V m This represents the amplitude of the AC input voltage.

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