A soft switching control method for a high voltage generator

Abstract

A soft switching control method for high-voltage generator, by measuring or calculating, the resonant period of LC resonant cavity composed of transformer leakage inductance and equivalent capacitor of voltage doubling rectifier circuit is determined. By setting the single conduction time of switch tube in full-bridge inverter circuit to half of the resonant period, the current flowing through the switch tube presents a sine wave shape due to the resonance effect in the conduction time, and the current naturally drops to near zero at the end of the conduction time, and the switch tube is turned off at this time to realize zero-current turn-off. And the voltage error signal obtained through the feedback loop is used to adjust the dead time between two adjacent conduction pulses. The application utilizes the inherent parasitic parameters (leakage inductance and equivalent capacitance) in the main power circuit to form a resonant cavity, without the need for additional resonant components; and can effectively avoid real-time detection of the switch tube current, and realize zero-current turn-off by matching the resonant half period through the preset conduction time.

Description

Technical Field

[0001] This invention relates to the field of switching power supply technology, and in particular to a soft-switching control method for a high-voltage generator. Background Technology

[0002] High-voltage generators are key components in many industrial and medical devices, typically comprising a DC bus capacitor, a full-bridge inverter circuit, a high-ratio step-up transformer, and a voltage multiplier rectifier circuit. Their operating principle (e.g.) Figure 1 As shown): The full-bridge inverter circuit converts DC voltage into a high-frequency AC square wave, which is then stepped up by a high-ratio step-up transformer, rectified and further stepped up by a voltage doubler rectifier circuit, and finally provides a stable high-voltage DC power to the load.

[0003] Because high-ratio step-up transformers have a large number of turns in their windings, their coupling coefficient is typically lower than that of conventional high-frequency transformers, resulting in significant leakage inductance. Simultaneously, the voltage doubler rectifier circuit exhibits equivalent capacitance characteristics at its input. Therefore, in the main power circuit, the transformer's leakage inductance and the equivalent capacitance of the voltage doubler rectifier circuit naturally form an LC resonant cavity (e.g., ...). Figure 2 , Figure 3 (As shown).

[0004] Currently, the main control technologies for high-voltage generators include the following:

[0005] 1. Fixed-frequency pulse width modulation (PWM) hard-switching control: This method stabilizes the output voltage by adjusting the duty cycle of the switching transistor. Due to the presence of the LC resonant cavity, the switching transistor can typically achieve zero-current turn-on (ZCS). However, the turn-off timing is not matched with the resonant characteristics, and the switching transistor is often forced to turn off when carrying a large current, resulting in huge switching losses and voltage stress. This not only reduces system efficiency but also affects the lifespan and reliability of power devices.

[0006] 2. LLC / LCC Resonant Control: This method achieves soft switching by adding an extra resonant inductor and capacitor to the main circuit to form an LLC or LCC resonant converter. However, this method has the following drawbacks: First, full soft switching can only be achieved under specific load and frequency conditions; second, the added resonant inductor and capacitor, especially in high-power applications, result in a larger size and weight, which is not conducive to the miniaturization and weight reduction of the equipment; finally, its compensation circuit design is relatively complex.

[0007] 3. Soft-switching control based on current sensing: Some existing technologies propose to perform a turn-off operation when the current crosses zero by detecting the current of the switching transistor. Although this method can ensure zero-current turn-off, it requires adding a current sensing circuit for each switching transistor, which significantly increases the hardware complexity, cost, and control difficulty of the system.

[0008] Therefore, how to effectively reduce the switching loss, improve the efficiency and power density of the high-voltage generator under the premise of simplifying the system topology and control strategy is a technical problem to be solved in the field SUMMARY

[0009] The application aims to provide a soft switching control method for a high-voltage generator, and in particular to provide a high-voltage generator control technology with simple structure, convenient control and high-efficiency soft switching.

[0010] To achieve the above-mentioned purpose, the application adopts the following technical scheme: a soft switching control method for a high-voltage generator, the high-voltage generator comprising a full-bridge inverter circuit, a high-ratio step-up transformer and a voltage doubler rectifier circuit, the leakage inductance of the high-ratio step-up transformer and the input end equivalent capacitance of the voltage doubler rectifier circuit constituting an LC resonant cavity,

[0011] and the following control method is adopted:

[0012] (1) setting the single conduction time of the switching tube in the full-bridge inverter circuit as a fixed conduction time, the fixed conduction time being equal to one-half of the resonant period of the LC resonant cavity, so as to realize zero-current turn-off of the switching tube;

[0013] (2) collecting the output voltage of the high-voltage generator, and dynamically adjusting the dead time between adjacent two conduction pulses according to the difference between the output voltage and a preset reference voltage, so as to stabilize the output voltage.

[0014] Preferably, the step of adjusting the dead time is specifically:

[0015] when the output voltage is lower than the reference voltage, the dead time is reduced to increase the switching frequency; when the output voltage is higher than or equal to the reference voltage, the dead time is increased to reduce the switching frequency.

[0016] Preferably, the resonant period of the LC resonant cavity is determined in advance by the following steps:

[0017] frequency-impedance characteristic scanning is performed on the combination of the high-ratio step-up transformer and the voltage doubler rectifier circuit, the resonant frequency corresponding to the resonant point is found, and the reciprocal of the resonant frequency is calculated to obtain the resonant period.

[0018] Preferably, the control method is realized through an analog control logic, wherein the step of setting the fixed conduction time comprises:

[0019] (11) generating a periodic triangular wave signal;

[0020] (12) generating a constant DC reference level, the level being higher than the peak value of the triangular wave signal;

[0021] (13) comparing the triangular wave signal with the DC reference level, and generating the fixed conduction time pulse during the period when the triangular wave signal is lower than the DC reference level.

[0022] Preferably, the step of adjusting the dead time comprises:

[0023] (21) generating a control signal according to the difference between the output voltage and the preset reference voltage;

[0024] (22) adjusting the falling edge slope of the triangular wave signal by using the control signal, wherein the falling edge slope is increased to shorten the dead time when the output voltage is low, and the falling edge slope is decreased to lengthen the dead time when the output voltage is high.

[0025] Preferably, during the generation of the fixed conduction time pulse, the adjustment of the falling edge slope of the triangular wave signal is suspended to avoid the influence of the fixed conduction time on the dead time adjustment.

[0026] Preferably, the method is implemented by a digital control logic,

[0027] wherein the fixed conduction time is set by setting the pulse width parameter of the PWM module in the digital controller to a value equal to one half of the resonance period of the LC resonant cavity.

[0028] Preferably, the step of adjusting the dead time comprises:

[0029] (21) collecting the output voltage by an analog-to-digital conversion module of the digital controller;

[0030] (22) calculating the target dead time according to the difference between the collected output voltage and the digitized reference value by a control algorithm inside the digital controller;

[0031] (23) writing the dead time value into the dead time register of the PWM module.

[0032] Preferably, the control algorithm is a PID control algorithm.

[0033] The present application has the following advantages:

[0034] 1. The present application makes full use of inherent parasitic parameters such as transformer leakage inductance and equivalent capacitance of voltage doubler rectifier circuit, without the need for additional resonant inductance and capacitance, thus simplifying the main power circuit topology, reducing hardware cost, volume and weight, and being conducive to the miniaturization of equipment.

[0035] 2、The application adopts fixed conduction time control, without real-time detection of switch tube current and complex zero-crossing judgment, greatly simplifying the design difficulty of control circuit and algorithm.

[0036] 3、The application sets the conduction time of the switch tube as a half cycle of resonance, ensures that the switch tube is turned off when the current is close to zero, realizes soft turn-off (ZCS), significantly reduces switching loss and voltage stress, improves the overall efficiency of the converter, and prolongs the service life of the power device.

[0037] 4、Compared with the traditional PWM control and frequency conversion control, the small signal model of the fixed conduction time control mode is simpler, which effectively reduces the design difficulty of the feedback compensation circuit. BRIEF DESCRIPTION OF DRAWINGS

[0038] In order to more clearly illustrate the technical solutions in the embodiments of the present application or the prior art, the drawings needed to be used in the embodiment or prior art description will be briefly introduced as follows.

[0039] Figure 1 is a schematic diagram of the main power circuit of the high-voltage generator.

[0040] Figure 2 is a schematic diagram of the actual main power circuit of the high-voltage generator considering the leakage inductance of the transformer.

[0041] Figure 3 is an equivalent circuit diagram of the high-voltage generator after simplifying and equivalent of the voltage doubler rectifier circuit and output load.

[0042] Figure 4 is a schematic diagram of the main power circuit of the high-voltage generator based on the LLC resonant converter.

[0043] Figure 5 is a schematic diagram of the main power circuit of the high-voltage generator based on the LCC resonant converter.

[0044] Figure 6 is an internal structure and signal flow block diagram of the high-voltage generator device in the embodiment of the application.

[0045] Figure 7 is a control circuit schematic diagram of one specific embodiment of the application.

[0046] Figure 8 is an example waveform diagram for realizing zero-current turn-off by using the application.

[0047] Figure 9 is a switch tube single turn-on time measurement diagram when the output voltage is low and the output voltage is high.

[0048] Figure 10 is the switch tube dead time measurement graph when the output voltage is low.

[0049] Figure 11 is the switch tube single turn-on time measurement graph when the output voltage is high.

[0050] Figure 12 is the switch tube dead time measurement graph when the output voltage is high. DETAILED DESCRIPTION

[0051] As shown in Figure 3 , the leakage inductance of the high-voltage-ratio step-up transformer and the equivalent input capacitance of the voltage doubler rectifier circuit form an LC resonant cavity, and the simplified equivalent circuit is consistent with the topology of an LC parallel resonant converter. In existing high-voltage generators, a hard switching control method with fixed-frequency PWM (pulse width modulation) control or a variable-frequency resonant control method of LLC / LCC is usually used.

[0052] In the hard switching control method with fixed-frequency PWM control, the output voltage is compared with a reference voltage to control the proportion of the conduction time of the switch tube of the full-bridge circuit in a single period, so as to adjust the output voltage.

[0053] In the variable-frequency resonant control method of LLC / LCC, a resonant inductor and a resonant capacitor are added to the full-bridge circuit to form a resonant cavity with the high-voltage-ratio step-up transformer (see Figure 4 , Figure 5 ). When the switch tube is turned on, although the input voltage of the resonant cavity is a square wave, the current starts to rise from 0, realizing zero-current soft switching. Due to the resonance, the current in the resonant cavity approaches a sine wave, and the switch tube can be turned off close to zero current.

[0054] Referring to Figure 3 , since the leakage inductance of the high-voltage-ratio step-up transformer and the input equivalent capacitance of the voltage doubler rectifier circuit automatically form an LC parallel resonant resonant cavity, when a square wave is input to the primary side of the transformer, the current starts to gradually rise from 0, and then gradually decreases in a sinusoidal manner. Near half of the resonant period of the LC resonant cavity, the current on the switch tube decreases to near 0, at which time the switch tube is turned off, realizing zero-current turn-off.

[0055] Therefore, the present application provides a soft switching control method for a high-voltage generator, as shown in Figure 6As shown, the high-voltage generator device mainly consists of a control system, an inverter circuit, a step-up transformer, a voltage doubling rectifier circuit and a feedback sampling circuit. The control system sends a fixed time width driving signal to the inverter circuit, the inverter circuit converts the direct current on the energy storage capacitor into an alternating square wave, the voltage is raised to the kV level through the step-up transformer, and then the alternating current is rectified to direct current through the voltage doubling rectifier circuit, and the direct current is provided to the load. The voltage feedback signal is collected from the output end of the voltage doubling rectifier circuit and fed back to the control system, and compared with the reference voltage signal, and a signal is output after compensation to control the time interval between the two driving signals. By adjusting the time interval of the driving signal, the duty cycle of the driving signal can be equivalent controlled, and the stability of the output voltage is ensured.

[0056] Further, the control method can be realized by an analog power supply control chip or a digital power supply control chip,

[0057] Embodiment 1

[0058] This embodiment introduces a method for realizing the control technology described in the present application using a conventional PWM type power supply control chip, and the circuit schematic diagram is as shown in Figure 7 .

[0059] In conventional PWM type power supply control chips, most of them adopt the form of programmable oscillation time and dead time. Usually, a capacitor (CT) is used as the charging and discharging of the oscillator, and a resistor (RT) controls the size of the charging current. The charging time of the CT capacitor is determined by the capacitor and RT, and the maximum width of a single pulse of a PWM signal is also determined. There is usually another resistor (RD) connected to the charging and discharging capacitor of the oscillator or the discharge triode, which is used to control the discharge speed of the charging and discharging capacitor of the oscillator. The discharge time of the oscillator capacitor usually corresponds to the dead time between the two PWM signals. During the dead time, the PWM signal output end continuously maintains low level. Due to the charging and discharging of the oscillator capacitor, the voltage on the oscillator capacitor usually presents a sawtooth wave. In order to facilitate the parallel use of the power supply, the oscillator of the power supply control chip usually has a synchronization signal output end, which alternately outputs high and low levels during the charging and discharging process of the oscillator charging and discharging capacitor.

[0060] In the PWM type control, a compensation amplifier is usually configured, which has a non-inverting input terminal, an inverting input terminal and an output terminal. In the conventional application, the non-inverting input terminal is usually connected to a reference voltage, the inverting input terminal is usually connected to an output feedback voltage, and a feedback and compensation network is connected between the output terminal and the inverting input terminal. In the normal working process, the output terminal of the compensation amplifier is compared with a triangular wave of an oscillator, when the output voltage of the compensation amplifier is greater than the instantaneous voltage of the triangular wave, the PWM signal output terminal outputs a high level; when the output voltage of the compensation amplifier is less than the instantaneous voltage of the triangular wave, the PWM signal output terminal outputs a low level. When the output voltage of the compensation amplifier is continuously higher than the peak value of the triangular wave, the PWM signal output terminal continuously outputs a high level in the charging process of the oscillator capacitor, and the signal width of each period is consistent.

[0061] The control method of the application needs to determine the resonance period of the resonance cavity formed by the transformer leakage inductance and the equivalent input capacitor of the voltage doubler rectifier circuit.

[0062] Referring to Figure 7 , the non-inverting input terminal of the compensation amplifier is connected to the reference voltage VRF of the controller, the inverting input terminal of the compensation amplifier is connected to the output terminal CMP pin of the compensation amplifier, and the compensation amplifier is connected as a voltage follower, so that the output terminal voltage of the compensation amplifier is kept at the value of the reference voltage, that is, always greater than the peak value of the triangular wave of the CT pin, so that the PWM signal output terminal continuously outputs a high level signal in the CT capacitor charging stage, and the width of each driving signal is kept consistent. Specifically, the connection mode of the enable pin and the soft start pin remains unchanged, and the control function of the external enable signal and the soft start function are still maintained.

[0063] When the parameters of the high-ratio step-up transformer and the voltage doubler rectifier circuit in the main power circuit are determined, the transformer leakage inductance can be measured by using a transformer comprehensive tester or an LCR bridge, the equivalent input capacitor of the voltage doubler rectifier circuit can be measured by using an LCR bridge, and then the resonance frequency and the resonance period of the resonance cavity formed by the transformer leakage inductance and the equivalent input capacitor of the voltage doubler rectifier circuit can be calculated.

[0064] The application provides a specific measurement method of leakage inductance (reference literature from Marian K. Kazimierczuk's High-Frequency Magnetic Components (Second Edition)), which is as follows:

[0065] 1. Basic parameter measurement: define the measurement value of the primary winding of the transformer when the secondary winding is open circuit as , then ​

[0066]

[0067] in, Defined as the leakage inductance of the primary winding of a transformer. Defined as the magnetizing inductance of the primary winding of a transformer.

[0068] When the primary winding of a transformer is open-circuited, the measured value of the secondary winding is defined as follows: Then there is

[0069]

[0070] in, Defined as the leakage inductance of the secondary winding of a transformer. Defined as the magnetizing inductance of the primary winding of a transformer.

[0071] The ratio of the primary winding to the secondary winding is defined as: Then, when the secondary winding of the transformer is short-circuited, the leakage inductance measured on the primary side is defined as follows: Then there is

[0072]

[0073] in, This is the leakage inductance of the secondary winding.

[0074] 2. Mutual Inductance Calculation: Defines the transformer primary winding and secondary winding as connected in series in phase when the transformer primary winding terminals are... and secondary winding terminals The measured values ​​between Then there is

[0075]

[0076] in, This refers to the leakage inductance of the primary and secondary windings.

[0077] When the primary and secondary windings of a transformer are connected in reverse series, the terminals of the primary winding of the transformer are defined as follows: and secondary winding terminals The measured values ​​between Then there is

[0078]

[0079] in, This represents the mutual inductance between the primary and secondary windings of the transformer. Therefore, the mutual inductance can be calculated as follows:

[0080]

[0081] The magnetizing inductance of the primary winding of the transformer can then be calculated as follows:

[0082]

[0083] The magnetizing inductance of the secondary winding of the transformer can be calculated as follows:

[0084]

[0085] Therefore, the leakage inductance of the primary and secondary windings can be calculated as follows:

[0086]

[0087]

[0088] 3. Leakage inductance calculation: When calculating the leakage inductance to the primary winding of the transformer,

[0089]

[0090] When the leakage inductance is factored into the secondary winding of the transformer, then

[0091]

[0092] The equivalent input capacitance of the voltage doubler rectifier circuit is determined by connecting the circuit according to the actual operating circuit structure, connecting the input terminal to the measurement terminal of the LCR bridge, and using the capacitance measurement function to obtain the equivalent input capacitance of the voltage doubler rectifier circuit. .

[0093] This invention provides another measurement method: connect the transformer output terminal and the voltage doubler rectifier circuit, use an impedance analyzer to scan the frequency-impedance characteristic curve of the connected circuit, obtain the frequency-impedance characteristic curve of the two, and observe the resonant point in the curve to obtain the transformer leakage inductance. (Since the transformer's secondary winding is connected to the voltage doubler rectifier circuit, the leakage inductance referred to the secondary winding needs to be used for calculation) and the equivalent input capacitance of the voltage doubler rectifier circuit. The resonant frequency of the formed resonant cavity and resonant period Then there is

[0094]

[0095]

[0096] The charging time of the CT capacitor in the oscillator of the control chip is set to half the resonant period of the resonant cavity, i.e. This ensures that the switching transistor turns off when the current approaches 0A.

[0097] This embodiment uses a conventional PWM control chip. Such chips typically include an oscillator defined by an external resistor (RT) and capacitor (CT). The charging time of the CT capacitor determines the maximum width of the PWM pulse.

[0098] To achieve a fixed on-time, the non-inverting input of the internal compensation amplifier is connected to the reference voltage (VRF), while its inverting input is shorted to the output (CMP), forming a voltage follower. This ensures the output voltage of the compensation amplifier remains constant at the reference voltage, which is always set higher than the peak value of the triangular wave on the oscillator CT pin. Consequently, during each charging cycle of the CT capacitor, the output of the PWM comparator remains high, generating a fixed-width drive pulse.

[0099] The conduction time of the switching transistor is the charging time of the CT capacitor. By selecting appropriate RT and CT values, this charging time is made equal to the resonant half-cycle determined in the previous step. .

[0100] The dead time corresponds to the discharge time of the CT capacitor. In this embodiment, an external circuit is used to control the discharge rate of the CT capacitor, thereby adjusting the dead time.

[0101] See Figure 7 The feedback voltage signal (Vfb) acquired from the output of the high-voltage generator is compared and compensated with the reference voltage (Vref) by an external operational amplifier (the internal amplifier ensures a fixed on-time, while the external amplifier implements voltage feedback regulation). Its output signal controls the base current of an NPN transistor. The collector of this transistor is connected to a current-collector (CT) capacitor for discharging the CT capacitor. Specifically, the reference voltage signal is connected to the non-inverting input of the operational amplifier, the feedback signal of the output voltage is connected to the inverting input, a feedback compensation circuit is connected between the inverting input and the output pin, and the output pin is connected to the base of the NPN transistor via a diode and a resistor to control the current in the transistor. The collector of the transistor is connected to the CT pin via a resistor to control the discharge rate of the CT capacitor. During the stable output phase, the output voltage of the operational amplifier is maintained at a certain value; it is necessary to avoid keeping the transistor on during the CT capacitor charging process, thus preventing continuous discharge of the CT capacitor.

[0102] See Figures 9-11When the output voltage is low, the difference between the feedback voltage Vfb and the reference voltage Vref is large, resulting in a higher operational amplifier output voltage. This increases the on-state current of the NPN transistor, accelerates the discharge speed of the CT capacitor, and shortens the dead time, thus allowing for a faster entry into the next conduction cycle and increasing power output. Conversely, when the output voltage is close to or higher than the set value, the operational amplifier output voltage decreases, the on-state current of the NPN transistor decreases, the discharge speed of the CT capacitor slows down, and the dead time lengthens, thereby reducing power output and stabilizing the output voltage at the set value.

[0103] To prevent the discharge transistor from remaining continuously on during CT capacitor charging, the SYNC pin of the PWM chip can be used. This pin alternates between high and low levels during the charge / discharge cycle. By using an additional PNP (or NPN) transistor, the NPN discharge transistor can be reliably turned off during the charging phase, ensuring that the charging and discharging processes do not interfere with each other.

[0104] like Figure 8 As shown, the yellow waveform represents the gate drive voltage of the switching transistor, and the purple-red waveform represents the current waveform of the switching transistor. When the conduction time is set to half a resonant cycle, the current of the switching transistor (purple-red waveform) exhibits a sinusoidal trend during the conduction phase. Since the current is the load current plus the excitation current of the transformer primary winding, the result is a sinusoidal wave rather than a standard sine wave. When the switching transistor is turned on, the current of the switching transistor gradually increases, reaches its peak, and then gradually decreases. When the turn-on time is equal to half of the resonant cycle of the main power circuit, the current of the switching transistor drops to close to 0A. At this point, turning off the switching transistor can achieve near-zero current turn-off, resulting in very small switching losses.

[0105] This invention fully utilizes the parasitic parameters of the step-up transformer and the equivalent capacitance of the voltage doubler rectifier circuit, achieving complete soft turn-off of the switching transistor in the simplest main power circuit topology without the need for additional components. It eliminates the need for current and voltage detection circuits for the switching transistor, making control simpler and easier to implement. Using a fixed on-time control mode also effectively reduces the design difficulty of the compensation circuit and the control section.

[0106] Example 2

[0107] The control technology of this invention can also be implemented by a digital controller (such as a DSP or MCU).

[0108] Similarly, the resonance period must first be determined by measurement or calculation. ).

[0109] Next, in the digital controller program, the pulse width of the PWM module is fixed to half the resonant period of the LC resonant cavity. .

[0110] Then, the controller's ADC (analog-to-digital converter) module is used to acquire the output voltage feedback signal, which is compared with the digitized reference voltage value inside the program. The required dead time is then calculated using PID (proportional-integral-derivative) or other control algorithms.

[0111] Finally, the calculated dead time value is written into the dead time register of the PWM module to dynamically adjust the switching frequency, thereby stabilizing the output voltage.

[0112] This method eliminates the need for complex external analog circuits, as the control logic is implemented in software, offering greater flexibility and scalability.

[0113] Of course, the above are only preferred embodiments of the present invention and are not intended to limit the scope of application of the present invention. Therefore, any equivalent changes made to the principle of the present invention should be included within the protection scope of the present invention.

Claims

1. A soft-switching control method for a high-voltage generator, characterized in that, The high-voltage generator includes a full-bridge inverter circuit, a high-ratio step-up transformer, and a voltage doubler rectifier circuit. The leakage inductance of the high-ratio step-up transformer and the equivalent capacitance at the input of the voltage doubler rectifier circuit form an LC resonant cavity. The following control method is adopted, which is implemented through analog control logic: (1) Set the single conduction time of the switching transistor in the full-bridge inverter circuit to a fixed conduction time, which is equal to half of the resonance period of the LC resonant cavity, so as to achieve zero current turn-off of the switching transistor. The step of setting the fixed conduction time includes: (11) Generates periodic triangular wave signals; (12) Generate a constant DC reference level that is higher than the peak value of the triangular wave signal; (13) Compare the triangular wave signal with the DC reference level, and generate the pulse with the fixed on time during the period when the triangular wave signal is lower than the DC reference level; (2) Acquire the output voltage of the high voltage generator, and dynamically adjust the dead time between two adjacent conduction pulses according to the difference between the output voltage and the preset reference voltage, so as to stabilize the output voltage; The steps for adjusting the dead time include: (21) First, generate a control signal based on the difference between the output voltage and the preset reference voltage; (22) The falling edge slope of the triangular wave signal is adjusted using the control signal, wherein when the output voltage is low, the falling edge slope is increased to shorten the dead time; when the output voltage is high, the falling edge slope is decreased to extend the dead time.

2. The soft-switching control method for a high-voltage generator as described in claim 1, characterized in that, The specific steps for adjusting the dead time are as follows: When the output voltage is lower than the reference voltage, the dead time is reduced to increase the switching frequency; when the output voltage is higher than or equal to the reference voltage, the dead time is increased to reduce the switching frequency.

3. A soft-switching control method for a high-voltage generator as described in claim 1 or 2, characterized in that, The resonant period of the LC resonant cavity is predetermined through the following steps: A frequency-impedance characteristic scan is performed on the combination of the high-ratio step-up transformer and the voltage doubler rectifier circuit to find the resonant frequency corresponding to the resonant point, and its reciprocal is calculated to obtain the resonant period.

4. The soft-switching control method for a high-voltage generator as described in claim 1, characterized in that, During the generation of the fixed on-time pulse, the adjustment of the falling edge slope of the triangular wave signal is paused to avoid the fixed on-time being affected by the dead-time adjustment.

5. The soft-switching control method for a high-voltage generator as described in claim 1, characterized in that, The method is implemented through a digital control logic. The fixed conduction time is defined as follows: the pulse width parameter of the PWM module in the digital controller is set to a value equal to half the resonant period of the LC resonant cavity.

6. The soft-switching control method for a high-voltage generator as described in claim 5, characterized in that, The steps for adjusting the dead time include: (21) The output voltage is acquired through the analog-to-digital conversion module of the digital controller; (22) Inside the digital controller, a control algorithm is used to calculate the dead time of the target based on the difference between the collected output voltage and the digital reference value; (23) Write the dead time value into the dead time register of the PWM module.

7. The soft-switching control method for a high-voltage generator as described in claim 6, characterized in that, The control algorithm is a PID control algorithm.

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