Method and system for testing direct-current offset compensation circuit of optical transceiving driving chip, and medium

Abstract

The invention discloses a method and system for testing a direct-current offset compensation circuit of an optical transceiving driving chip and a medium, and relates to the technical field of photoelectric integrated circuit testing, and the method comprises the following steps: carrying out the time alignment of a test waveform and an offset track of the optical transceiving driving chip, extracting the phase dislocation amount and the amplitude kick amount between beats, and carrying out the calculation of the phase offset amount and the amplitude kick amount; a continuous change result is mapped into a time sequence conflict observation band which is used for identifying a beat overlapping region; and on the basis of the phase dislocation trend of the time sequence conflict observation band, determining a beat overlapping point of the direct current offset compensation loop, and generating an overlapping beat index according to the duration of amplitude kick and the polarity direction for identifying an interference node. According to the method, millisecond-level beat observation and microsecond-level overlapping identification are realized in a test stage, interference nodes are accurately extracted, a coupling strength graph is constructed, and by combining beat rearrangement and an energy unloading strategy, nonlinear oscillation is inhibited, oscillation and overload risks are reduced, eye pattern quality and error code stability are improved, and chip test precision and robustness are enhanced.

Description

Technical Field

[0001] This invention relates to the field of optoelectronic integrated circuit testing technology, specifically to a testing method, system, and medium for DC offset compensation circuits of optical transceiver driver chips. Background Technology

[0002] Testing the DC offset compensation circuit of an optical transceiver driver chip refers to the process of functional verification and performance evaluation of the internal DC offset compensation circuit used to balance signal offset during the research and development or production stage of optical communication chips. Because optical transceiver driver chips often experience DC offset during high-speed signal transmission due to factors such as device mismatch, temperature drift, or power supply fluctuations, leading to output waveform distortion, eye diagram closure, or increased bit error rate, specialized testing methods are needed to verify whether the compensation circuit can accurately identify and correct this offset. During testing, specific electrical signals are typically input to the chip to monitor the response characteristics, adjustment accuracy, and stability of the compensation loop. This confirms that it can maintain signal symmetry and transmission linearity under different load and temperature conditions, thereby ensuring the high-speed and stable operation of the optical communication system.

[0003] The existing technology has the following shortcomings: In existing technologies, the DC offset compensation loop and automatic gain control loop of optical transceiver driver chips are typically designed independently and participate simultaneously in the dynamic adjustment of signal amplitude and bias. However, during testing, when the two feedback loops lack precise time isolation, timing competition can easily occur. That is, while the offset compensation loop is correcting the DC offset, the automatic gain control loop is also adjusting the signal amplitude, causing the control commands of the two to overlap within microseconds. Due to the different response polarities and adjustment rates of the two loops, this overlap can trigger nonlinear coupled oscillations, causing the output current to fluctuate at high frequencies in a very short time. When this fluctuation frequency falls into the inherent resonant band of the device, the driver transistor is in a continuous overload state, causing output oscillation, signal distortion, or even thermal breakdown of the device.

[0004] The information disclosed in the background section is only intended to enhance the understanding of the background of this disclosure, and therefore may include information that does not constitute prior art known to those skilled in the art. Summary of the Invention

[0005] The purpose of this invention is to provide a test method, system, and medium for the DC offset compensation circuit of an optical transceiver driver chip, in order to solve the problems mentioned in the background art.

[0006] To achieve the above objectives, the present invention provides the following technical solution: a testing method for the DC offset compensation circuit of an optical transceiver driver chip, comprising the following steps: The test waveforms and bias trajectories of the optical transceiver driver chip are time-aligned, the phase misalignment and amplitude jump between beats are extracted, and the continuous change results are mapped into timing conflict observation bands to identify beat overlap areas. Based on the phase misalignment trend of the timing conflict observation band, the beat overlap point of the DC offset compensation loop is determined, and the overlap beat index is generated according to the duration and polarity direction of the amplitude jump to identify the interference node. The timing superposition relationship between the DC offset compensation loop and the automatic gain control loop is constructed by using the overlapping beat index. The beat phase difference and gain change rate are marked to generate an energy coupling interference diagram, which is used to characterize the coupling strength between feedback loops. Based on the high-intensity segment of the energy coupling interference diagram and combined with the frequency response of the package parasitic parameters, the energy resonance trend of the driving end is analyzed, the excitation frequency band is determined, and a list of resonance threshold regions is generated. Based on the high-risk frequency bands of the resonance threshold list, the beat sequence is rearranged according to the beat phase difference distribution to construct a rhythm decoupling sequence, and the execution order and silence interval are set to form a staggered control baseline. Based on the peak-shifting control baseline, a dual-phase breathing traction window, a mirror hysteresis valve, a reversing energy bypass, and a time siphon groove are used to perform segmented energy release, so that local energy is unloaded sequentially in the timing cycle, and a dynamic closed-loop suppression link is constructed to suppress energy resonance and maintain output stability.

[0007] Preferably, the steps for generating time-series conflicting observation bands are as follows: A modulation signal is applied to the input terminal of the optical transceiver driver chip, and the output current signal and bias voltage trajectory are acquired simultaneously. Time alignment processing is performed through an acquisition device with dual-channel synchronous sampling capability to obtain a uniform time axis signal with amplitude normalization. Based on the time-aligned signal data, each continuous sine cycle is defined as a beat interval. The time interval between the rising edge of the input signal and the peak of the bias trajectory is extracted, the phase misalignment value of the beat is calculated, and the phase drift curve is generated. Based on the instantaneous amplitude change within each cycle, the differential amplitude of the output current is calculated using a sliding window, and the amplitude jump point is identified. When the jump amplitude exceeds the average rate of change threshold of the cycle, the jump event is recorded. The phase misalignment values ​​and amplitude jump values ​​of each beat are mapped to a two-dimensional coordinate plane in chronological order, and the beat conflict intensity is identified by a color scale to generate a time-series conflict observation band to indicate the signal adjustment overlap area.

[0008] Preferably, the steps for generating the overlapping beat index are as follows: Based on the timing conflict observation band, the beats with phase misalignment exceeding the preset threshold are segmented and extracted, and time offset curves are plotted on a unified time axis to determine the response drift trend; By combining the amplitude jump time distribution characteristics within the beat, the jump start time is matched with the phase misalignment offset interval to identify the beat overlap point in the DC offset compensation loop adjustment process; Extract the jump amplitude, jump duration, and jump direction parameters for each jump event, calculate the amplitude difference between the jump start point and the end point, and determine the jump direction; An overlapping beat index is established based on phase misalignment value, jump amplitude, jump duration, and jump direction. Interference risk nodes are classified and identified, and an overlapping beat index list is generated.

[0009] Preferably, the steps for generating the energy coupling interference diagram are as follows: Based on the overlapping beat index list, extract the key control command points of the bias adjustment curve and gain control curve in the interference beat, record the bias response start time and gain response start time, and calibrate them on a unified time axis; The difference between the bias response start time and the gain response start time is calculated to obtain the control phase difference for each disturbance cycle. At the same time, the gain change rate of the corresponding cycle is extracted as the control strength parameter. The beat number, control phase difference, and gain change rate are plotted as a two-dimensional thermodynamic distribution map, and the coupling strength levels are divided according to the color scale to generate an energy coupling interference map that reflects the distribution of feedback coupling strength. Based on the highlighted areas in the energy coupling interference diagram, the numbers, phase differences, and gain change rates of the coupled nodes are extracted to establish a list of coupled nodes and classify them by intensity.

[0010] Preferably, the steps for generating the resonance threshold list are as follows: Based on the high-intensity beat segments marked in the energy coupling interference diagram, the beat number, control phase difference, and gain change rate are extracted to perform point-by-point analysis on the sudden jump behavior of the output current waveform; By combining the parasitic parameters of the optical transceiver driver chip package, inductance, capacitance and resistance models are established, and the transmission characteristics are analyzed by frequency scanning to determine the inherent resonant frequency of the structure. Frequency domain analysis is performed on the output waveform in a highly coupled beat. The position and amplitude of the spectral peak are obtained by using fast Fourier transform and compared with the packaged frequency response curve to identify the resonant frequency point. Based on the identified resonant frequency, frequency bandwidth, and beat time interval, a list of resonant threshold regions is output, including beat number, resonant frequency, peak resonant amplitude, and resonant level.

[0011] Preferably, in the process of outputting the list of resonance thresholds, the resonance level is classified and identified by comparing the degree of overlap between the resonance frequency and the resonant frequency band formed by the package parasitic parameters, and the peak amplitude and energy duration of the output waveform are recorded in the identified high coupling beats to determine the resonance risk level of the driver end.

[0012] Preferably, the peak shift control baseline generation steps are as follows: Based on the resonant threshold list, the number, duration, control phase difference, gain change rate and output current disturbance amplitude of high-risk beats are extracted, a multi-parameter mapping table is established and the control command scheduling density is analyzed; High-risk beats are extracted from the original control sequence and interleaved with low-interference beats with a gain change rate of no more than 0.5 units per microsecond and a phase difference of more than 0.12 microseconds to construct a rhythmic decoupling sequence without continuous excitation behavior, and a fixed time interval is set between beats. Based on the beat decoupling sequence, a silent interval is set for each beat execution window, and a control beat baseline table containing beat number, trigger start point, control response duration and minimum silent interval is established to form a time-constrained staggered control baseline.

[0013] Preferably, based on the peak-shifting control baseline, a dual-phase respiratory traction window, a mirror hysteresis valve, a reversible energy bypass, and a time siphon groove are used to perform segmented energy release, sequentially unloading local energy within the time-series beats, and constructing a dynamic closed-loop inhibition link. The steps are as follows: Based on the start time, peak energy and control response window of each beat in the staggered control baseline, the opening and closing rhythm of the dual-phase breathing traction window is set, and energy is guided to the low impedance energy dissipation zone in stages before and after the excitation signal. Based on the response characteristics of the descent segment of the clock, the mirror hysteresis valve is activated in the residual energy band before the signal ends, and a delayed anti-sloping waveform is constructed through the reverse voltage guiding structure to reduce the main channel mutation rate. The excitation energy that is not discharged by the mirror hysteresis valve is introduced into the foldback energy bypass network, and the peak energy is converted into heat flow for dissipation through a multi-segment time-delay spiral structure and parallel damping network. A time siphon trench structure is deployed within the silent window between cycle executions to absorb residual energy at the end of the trace through parallel siphon lines, preventing energy reflection from interfering with the next cycle. By sequentially scheduling the biphase breathing traction window, mirror hysteresis valve, reversing energy bypass, and time siphon channel, a closed-loop energy regulation link consisting of pre-extraction, slope adjustment, residual guidance, and tail-end clearance is constructed.

[0014] The test system for the DC offset compensation circuit of the optical transceiver driver chip includes a timing conflict observation band generation module, a beat overlap identification module, an energy coupling analysis module, a resonance threshold identification module, a rhythm decoupling control module, and an energy suppression execution module. The timing conflict observation band generation module performs time alignment between the test waveform and the bias trajectory of the optical transceiver driver chip, extracts the phase misalignment and amplitude jump between beats, and maps the continuous change results into timing conflict observation bands to identify beat overlap areas. The beat overlap identification module determines the beat overlap points of the DC offset compensation loop based on the phase misalignment trend of the timing conflict observation band, and generates an overlap beat index based on the duration and polarity direction of the amplitude jump, which is used to identify the interference nodes. The energy coupling analysis module uses the overlapping beat index to construct the timing superposition relationship between the DC offset compensation loop and the automatic gain control loop, marks the beat phase difference and gain change rate, and generates an energy coupling interference diagram to characterize the coupling strength between feedback loops. The resonance threshold identification module analyzes the energy resonance trend of the driving end based on the high-intensity segment of the energy coupling interference diagram and the frequency response of the encapsulated parasitic parameters, determines the excitation frequency band, and generates a list of resonance thresholds. The rhythm decoupling control module rearranges the beat sequence according to the beat phase difference distribution based on the high-risk frequency bands of the resonance threshold list, constructs the rhythm decoupling sequence, and sets the execution order and silence interval to form a staggered control baseline. The energy suppression execution module, based on the peak-shifting control baseline, drives the dual-phase breathing traction window, mirror hysteresis valve, reversing energy bypass, and time siphon groove to perform segmented energy release, so that local energy is unloaded sequentially in the timing cycle, and a dynamic closed-loop suppression link is constructed to suppress energy resonance and maintain output stability.

[0015] A computer-readable storage medium storing a computer program, which, when executed by a processor, implements a test method for a DC offset compensation circuit of an optical transceiver driver chip.

[0016] The technical effects and advantages provided by the present invention in the above technical solution are as follows: This invention enables comprehensive dynamic observation of signal beat behavior with millisecond-level time accuracy during the testing phase, and accurately identifies the timing overlap between the offset compensation loop and the automatic gain control loop at the microsecond level, thereby achieving high-resolution extraction of feedback interference nodes. Furthermore, through coupling strength mapping and directional identification of resonant frequency bands, a closed-loop debugging mechanism is constructed for the coordinated regulation of time beat rearrangement and energy release rhythm, enabling the quantification and orderly suppression of previously difficult-to-capture and interpret nonlinear oscillations. Finally, through the combined application of peak-shifting control baseline and multi-stage energy unloading structures, it not only significantly reduces the risk of high-frequency oscillations, signal distortion, and thermal overload at the device output caused by coupling interference, but also effectively improves eye diagram quality, stabilizes bit error rate performance, and extends device lifespan, solving the key pain point of existing testing schemes lacking a response mechanism for dynamic adjustment interference. Attached Figure Description

[0017] To more clearly illustrate the technical solutions in the embodiments of this application or the prior art, the drawings used in the embodiments will be briefly introduced below. Obviously, the drawings described below are only some embodiments recorded in this invention. For those skilled in the art, other drawings can be obtained based on these drawings.

[0018] Figure 1 This is a flowchart of the test method for the DC offset compensation circuit of the optical transceiver driver chip of the present invention.

[0019] Figure 2 This is a schematic diagram of the test system for the DC offset compensation circuit of the optical transceiver driver chip of the present invention.

[0020] Figure 3 This is a schematic diagram of the electronic device of the present invention. Detailed Implementation

[0021] Exemplary embodiments will now be described more fully with reference to the accompanying drawings. However, these exemplary embodiments can be implemented in many forms and should not be construed as limited to the examples set forth herein; rather, they are provided so that the description of this disclosure will be more complete and fully convey the concept of the exemplary embodiments to those skilled in the art.

[0022] This invention provides, for example Figure 1 The test method for the DC offset compensation circuit of the optical transceiver driver chip shown includes the following steps: The test waveforms and bias trajectories of the optical transceiver driver chip are aligned to millisecond time, the phase misalignment and amplitude jump between signal beats are extracted, and the continuous change results are mapped to timing conflict observation bands to reveal potential beat overlap areas. To effectively detect signal interference characteristics generated by the interaction of feedback adjustment links during the dynamic operation of an optical transceiver driver chip, it is necessary to first complete the time alignment processing between the test waveform and the bias trajectory, and then gradually extract the changes in phase misalignment and amplitude jumps. Finally, a timing conflict observation band for identifying feedback conflicts is constructed. The implementation process is as follows: During test data acquisition, a high-speed oscilloscope recording device with dual-channel synchronous sampling capability was used to apply a modulated sinusoidal signal with an amplitude of 400 mV and a frequency of 2.5 GHz to the chip input terminal, and the drive current signal and bias voltage trajectory were extracted at the output terminal. The input excitation signal was provided by a signal generator, and the output signal entered two parallel channels through a precision current probe and a voltage follower, respectively, and was simultaneously connected to an acquisition device with channel phase-locked loop (PLL) function. To ensure that the sampling of each channel has a unified time base, a 20 MHz external quartz crystal oscillator was introduced, using standard TTL pulse mode as the sampling reference, forcing the sampler to complete a data acquisition every 50 nanoseconds. The sampling time window was set to 1 millisecond, the number of sample points was 20,000, and the sampling depth reached 12 bits of precision. During the acquisition process, the input signal, output current, and bias trajectory were recorded in channel order as Channel A, Channel B, and Channel C, respectively, and the same starting time tag was allocated in memory. Subsequently, a combined method of sample interpolation and time offset calibration was used to normalize the three-channel signals to eliminate microsecond-level time differences caused by channel propagation delay, probe parasitic inductance, and differences in device internal response speed. After time synchronization, the three signals were uniformly stretched to a unified time axis and their amplitudes were normalized to between 0 and 1, serving as the basis for subsequent analysis.

[0023] Based on the time alignment of the test waveform and bias trajectory, the signal is periodically segmented and its beat features are extracted. By detecting the rising and falling edges of the input signal waveform within each cycle, each continuous sinusoidal cycle is defined as a beat interval, and the changing trends of the output current and bias voltage within that beat are tracked. Taking the starting point of each beat as a reference, the time interval between the voltage peak position appearing in the bias voltage trajectory and the rising edge of the input signal in that beat is defined as the phase misalignment value. For example, in the 20th beat, the rising edge of the input signal is located at the 362nd sampling point, corresponding to a time of 1.81 microseconds, and the peak of the bias trajectory is located at the 395th sampling point, corresponding to a time of 1.975 microseconds, with a phase misalignment value of 0.165 microseconds. After repeatedly extracting the phase misalignment value for each beat, a phase drift curve is formed, which can be used to analyze the bias response delay evolving over time. To verify the sensitivity of this feature to changes in the chip's internal temperature, the chip was run at 25°C, 45°C, and 65°C for five minutes each, and the above measurements were repeated. The results showed that the phase misalignment value shifted to the right under high temperature conditions, with a maximum increase of 0.23 microseconds, indicating that the bias adjustment path exhibits a trend of response degradation under temperature rise conditions.

[0024] Based on the phase misalignment extraction, the instantaneous amplitude change behavior within each clock cycle is further analyzed to identify the presence and severity of abrupt jumps. Specifically, the first-order difference of the output current is calculated within each clock cycle using a sliding window of 10 sampling points, and the maximum change in the absolute value of the difference is recorded. When this maximum amplitude exceeds twice the average amplitude change rate in that clock cycle, the point is marked as an amplitude jump point. For example, in clock cycle 48, the output current experiences a sudden change in a continuous steady waveform, rapidly decreasing from 0.64 units to 0.35 units within only 50 nanoseconds, with a change amplitude of 0.29 units. This jump point coincides with the modulation edge of the input signal and shows a relative upward trend in the bias trajectory, indicating that the DC offset compensation loop performed a correction action within that clock cycle. A total of 154 valid jump points were extracted throughout the entire test cycle, averaging one every 7 clock cycles. The occurrence time of each jump event is mapped back to its corresponding beat, and the jump amplitude and phase misalignment value are stored together to provide bivariate input for subsequent observation map construction.

[0025] Phase misalignment and amplitude jump values ​​in all beats were plotted chronologically on a two-dimensional coordinate plane, with the horizontal axis representing the test time and the vertical axis representing the jump amplitude. A color scale was used to indicate the magnitude of the phase misalignment. During plotting, beat intervals with phase misalignment greater than 0.2 microseconds and jump amplitudes exceeding 0.25 units were highlighted in red to indicate conflict hotspots during signal conditioning. The entire graph extends from 0 milliseconds to 1 millisecond, covering approximately 400 beats. The red-marked sections are mainly concentrated in the first 150 beats. This distribution characteristic matches the DC bias adjustment behavior during the initial power-up phase of the chip, confirming that the compensation loop and the gain control signal overlapped in time during startup. Clustering and filtering were used to perform boundary statistics on all marked sections, outputting a beat interference list containing start and end times, maximum phase shift, and jump intensity levels. The time interval corresponding to this interference list is the timing conflict observation band, serving as the basis for subsequent identification of beat overlap points and analysis of energy coupling trends.

[0026] Based on the phase misalignment trend recorded in the timing conflict observation band, the beat overlap points of the DC offset compensation loop are determined point by point, and the overlap beat index is generated according to the duration and polarity direction of the amplitude jump to identify the interference node. After constructing the timing conflict observation band, it is necessary to accurately identify the beat overlap points in the DC offset compensation loop adjustment process based on the recorded phase misalignment trend and amplitude jump behavior, and extract the beat index with interference characteristics for subsequent control conflict determination and energy coupling analysis. The implementation method is as follows: Based on the constructed timing conflict observation band, the beats marked as high phase misalignment regions were segmented and extracted. This extraction process used the beat number as an index, marking each beat with a phase misalignment exceeding 0.2 microseconds as a potential analysis object. For example, in the beat segment numbered 115 to 138, 14 beats had phase misalignment values ​​exceeding the threshold, ranging from 0.23 microseconds to 0.46 microseconds. The time difference between the peak value of the bias trajectory and the rising edge of the test waveform was extracted for each of these 14 beats, and a time offset curve was plotted on a unified time axis. This curve clearly shows the response drift trend between each beat. For example, in beats 117, 118, and 119, the phase misalignments were 0.25 microseconds, 0.31 microseconds, and 0.39 microseconds, respectively, showing a continuously increasing trend. This indicates that the DC offset compensation loop's adjustment process is active, the adjustment signal is still gradually being released, and it has not yet entered the steady-state closed-loop stage.

[0027] After identifying the phase misalignment segments with an increasing trend, it is necessary to combine the time distribution characteristics of amplitude jumps in the corresponding beats to further determine whether the adjustment response overlaps with the gain adjustment command in time. In this step, the time point of the jump event within each beat is selected, and the starting point of the jump waveform is used as a reference to match it with the phase misalignment trend map. If the jump start time intersects with the time interval covered by the phase misalignment offset, it indicates that the jump is a non-steady-state response triggered during the adjustment of the compensation loop. Taking beat 119 as an example, the jump occurs 0.37 microseconds after the start of the beat, which completely coincides with the phase misalignment value of 0.39 microseconds recorded for that beat. It is determined that the jump is caused by the overlapping behavior in the adjustment loop action. This type of jump is not caused by changes in the input signal amplitude, but by waveform reconstruction caused by the simultaneous adjustment of two feedback loops. Internally, it manifests as a sharp rise or fall in the drive current, which has a high risk of interference.

[0028] Key parameters for each jump event are extracted, including jump amplitude, jump duration, and jump direction. Specifically, starting from the jump start point, the amplitude changes of consecutive sampling points are tracked, the sampling time at the jump termination point is recorded, and the jump duration is calculated. Jump amplitude is defined as the maximum amplitude difference between the jump start and termination points; jump direction is determined by comparing the amplitude relationships between the two points to determine whether it is a positive rise or a negative fall. For example, in beat 122, the jump start amplitude is 0.65 units, the termination amplitude is 0.38 units, the drop amplitude is 0.27 units, the jump direction is a negative fall, and the duration is 75 nanoseconds. By extracting these parameters from jump events within all overlapping beats, a complete set of interference feature data can be obtained for subsequent conflict node classification and identification.

[0029] Based on the extraction results, an overlapping beat index is established to identify the locations of nodes with significant feedback interference risk. This index table is based on beat number and consists of four fields: phase misalignment value, jump amplitude, jump duration, and jump direction. A grading standard is set according to different parameter weights: when the phase misalignment value is greater than 0.35 microseconds, the jump amplitude exceeds 0.3 units, the jump duration exceeds 80 nanoseconds, and the jump direction is a negative drop, it is classified as a high-risk beat and assigned a level 1 rating; when the above parameters are slightly below this standard, it is defined as a medium-risk beat and assigned a level 2 rating; all others are low-risk beats. These beats are summarized in chronological order to form an overlapping beat index list. For example, beats 117, 119, and 122 all meet the level 1 standard, beats 125 and 126 meet the level 2 standard, and other beats are classified as low-risk. This list not only provides the specific time location of the overlap but also standardizes the signal behavior characteristics within the beat, enabling differentiated strategies to be adopted for interference nodes of different levels during subsequent processing.

[0030] Using the interference node information recorded in the overlapping beat index, the timing superposition relationship of the control commands of the DC offset compensation loop and the automatic gain control loop is constructed. The phase difference and gain change rate of different beats are continuously marked to form an energy coupling interference diagram, which is used to reflect the coupling strength distribution of the two feedback loops. To further analyze the interaction between the DC offset compensation loop and the automatic gain control loop in overlapping cycles, it is necessary to establish the temporal overlap relationship of the two loops' adjustment behaviors based on the information recorded in the interference cycle index, and continuously label the phase difference and gain change rate, thereby constructing an energy coupling interference diagram that reflects the distribution of control conflict intensity. The specific method is as follows: Based on the generated overlapping beat index list, all first- and second-level interference beats were selected as analysis samples, and key control command points of the bias adjustment curve and gain control curve in each beat were extracted. Using the beat number as a reference, the start time of the bias adjustment action was extracted sequentially, i.e., the position where the first positive or negative voltage change occurs in the bias trajectory. Simultaneously, the start time of the gain adjustment behavior was extracted, i.e., the time when the output current experiences a steep rise or fall. These two time points were used as the start time of the DC offset compensation response and the automatic gain control response, respectively. All times were calibrated on a unified reference time axis, in microseconds, with a time accuracy of 0.01 microseconds. In beat number 121, the time when the bias trajectory begins to adjust to a higher potential is 0.36 microseconds, while the gain response start time is 0.39 microseconds, a difference of 0.03 microseconds. This indicates that the two control paths are activated almost simultaneously in this beat, posing a high risk of overlap.

[0031] The difference between the two response times is defined as the control phase difference, and its phase difference value is recorded for each interference cycle. Simultaneously, the output change rate of the gain control path in that cycle is further extracted, which is the amplitude change from the response start time to the moment before signal stabilization divided by the time length. In cycle 122, the gain response increases from 0.27 units to 0.48 units in 0.11 microseconds, and the gain change rate for that cycle is calculated to be 1.91 units per microsecond. All interference cycles are sequentially entered into a data table according to their cycle numbers, containing four columns: cycle number, bias response start time, gain response start time, control phase difference, and gain change rate. In this embodiment, the table covers cycles 115 to 128, a total of 14 sets of data. Cycles with a phase difference less than 0.05 microseconds and a gain change rate exceeding 1 unit per microsecond are specifically recorded, occurring 5 times and concentrated between cycle numbers 119 and 123.

[0032] A coupling strength graph was plotted based on the data table. The graph uses the beat number as the x-axis and the control phase difference and gain change rate as the y-axis. Specifically, it is a two-dimensional heat map, using a color scale with red, orange, yellow, green, and blue to represent coupling strength levels, with red representing extremely high coupling and blue representing extremely low coupling. Each beat is represented by a rectangle, and the color within the rectangle is determined by both the phase difference and gain change rate of that beat. For example, in beat 120, the phase difference is 0.02 microseconds and the gain change rate is 2.1 units per microsecond, belonging to the strong coupling region, and is filled with red; while in beat 127, the phase difference is 0.29 microseconds and the gain change rate is only 0.35 units per microsecond, belonging to the weak coupling region, and is filled with blue. The final graph clearly shows the temporal distribution of coupling strength, revealing that high coupling regions are continuously distributed between beat numbers 118 and 123, exhibiting a clustered pattern.

[0033] To facilitate the use of coupling strength results for subsequent oscillation analysis and regulation strategy formulation, the boundaries of high-intensity regions in the coupling spectrum were further extracted to form a list of coupling nodes. This list includes the number, phase difference, gain rate of change, coupling level, and start and end times of the coupling segment for each high-coupling beat. Specifically, beats with a phase difference less than 0.05 microseconds and a gain rate of change greater than 1 unit per microsecond are classified as first-level coupling nodes; beats with a phase difference between 0.05 and 0.15 microseconds and a gain rate of change greater than 0.6 units per microsecond are classified as second-level coupling nodes. For example, beat 121 has a phase difference of 0.03 microseconds and a gain rate of change of 1.85 units per microsecond, thus being identified as a first-level node; beat 125 has a phase difference of 0.12 microseconds and a gain rate of change of 0.78 units per microsecond, thus being identified as a second-level node. After classification according to this standard, a total of 5 first-level coupling nodes and 3 second-level coupling nodes were identified. The identification of coupling nodes not only reflects the degree of temporal overlap between the two types of control paths, but also reveals the resonance risk area that may be caused by dynamic adjustment between energy injection paths.

[0034] Based on the high-intensity coupling segment in the energy coupling interference diagram, and combined with the frequency response characteristics of the parasitic parameters of the optical transceiver driver chip package, the resonance distribution trend of the driver end energy is analyzed, the excitation frequency band that is easy to trigger oscillation is determined, and a list of resonance threshold regions is generated. To identify the risk of energy oscillation at the drive end caused by feedback conflicts, based on the high-intensity beat segments marked in the energy coupling interference diagram and combined with the parasitic parameters of the optical transceiver driver chip's packaging, frequency response characteristics are analyzed, resonance trends are derived, excitation frequency bands prone to triggering oscillations are determined, and a structured list of resonance threshold regions is generated. The specific operation process is as follows: The first-order coupling beat segments identified in the energy coupling interference diagram are used as the target interval for resonance analysis. Combining the beat numbers, control phase differences, and gain change rates marked in the previous step, a point-by-point analysis is performed on the continuous beat sequence numbered 118 to 123. These beats collectively exhibit a control response time interval of less than 0.05 microseconds, a gain adjustment change rate greater than 1.1 units per microsecond, and are accompanied by short-duration high-density jumps. Within this segment, the output current exhibits irregular waveform amplitude oscillations per unit time, with repeated alternating abrupt changes in positive and negative amplitudes. In beat 121, the output current experiences four consecutive directional reversals within a 300 nanosecond time window, with peak values ​​ranging from -0.28 to +0.31 units, and a maximum abrupt change slope reaching 3.6 units per microsecond. This characteristic preliminarily indicates that the excitation signal within this beat is in a nonlinear amplification or modulation instability state. To further confirm whether the resonance is caused by the chip structure response, a structural frequency domain analysis is required.

[0035] Based on the physical package structure of the optical transceiver driver chip, its parasitic parameters were extracted, and corresponding inductance, capacitance, and resistance models were established. Its transmission characteristics were then analyzed through frequency scanning. In this embodiment, the chip used employs a flip-chip structure with solder ball diameters of 100 micrometers, an average metal trace length between solder balls of 0.85 millimeters, 64 pins, and a signal line width of 60 micrometers. Using high-frequency impedance measurement, the parasitic inductance was extracted to be 1.8 nanohenries, the parasitic capacitance to be 0.52 picofarads, and the DC resistance of the metal traces to be 1.3 ohms. Combining the physical dimensions and package medium parameters, a network analyzer was used to scan and obtain its frequency response curve. Multiple impedance minima were found in the 0.5 GHz to 5 GHz range, with the most significant concentrations around 1.45 GHz, 2.1 GHz, and 3.6 GHz. These frequency points exhibited valley responses below 10 ohms, indicating that external excitation signals in this frequency band are easily absorbed by the structure and amplified and reflected, thus triggering current or voltage resonance responses.

[0036] Frequency domain analysis was performed on the output waveforms in the highly coupled cycles indicated by the energy coupling interference diagram to map and compare them with the response characteristics of the package structure. Waveform samples of the output current from startup to convergence were collected for each cycle, with time windows ranging from 200 nanoseconds to 500 nanoseconds, dynamically adjusted according to the jump density of each cycle. In cycle 120, the waveform contained 6 jump events, the sampling time window was set to 400 nanoseconds, and the sampled data were processed by Fast Fourier Transform. The spectral analysis results showed a spike signal with a peak amplitude of 0.41 units at a frequency of 1.42 GHz, with an amplitude 3.9 times that of the average background noise. Comparison with the package frequency response characteristics revealed that this frequency point was completely consistent with the resonant point formed by the parasitic inductance and capacitance in the structure, confirming that the excitation energy in this cycle had fallen within the structural resonant bandwidth and was amplified by the feedback path, ultimately forming a locally amplified resonance. The above process was repeated for beats 121 and 122, and similar high-amplitude frequency peaks were observed at 2.08 GHz and 3.64 GHz, respectively, indicating that these beats also have a resonant amplification trend and have constituted a resonant danger band.

[0037] Based on the identification results of the resonant frequency, combined with the beat time interval and frequency bandwidth, a list of resonant threshold regions is output. The list includes six indicators: beat number, resonant frequency in the output waveform, peak resonant amplitude, frequency bandwidth corresponding to structural resonance, duration of driving excitation, and resonance level. Taking beat 121 as an example, its resonant frequency is 1.42 GHz, the frequency bandwidth is set to ±100 MHz, the resonant duration is 320 nanoseconds, and the maximum amplitude is 0.41 units, classifying it as a first-level resonant frequency band. The entire list outputs three first-level resonant bands: 1.42 GHz, 2.08 GHz, and 3.64 GHz, corresponding to beats 121, 122, and 123. Each band is accompanied by a sustained high-slope output waveform and a repetitive energy reflection signal.

[0038] Based on the high-risk frequency bands identified in the resonance threshold list, the signal beat sequence is rearranged according to the beat phase difference distribution characteristics to construct a non-overlapping rhythm decoupling sequence. An execution order and silence interval are set for each beat interval to form a constrained peak-shifting control baseline. To avoid triggering energy resonance in the high-coupling feedback section of the optical transceiver driver chip, the signal conditioning sequence should be recombined based on the identified resonance frequency and high-risk beats, combined with the distribution characteristics of the beat control phase difference. This will construct a non-overlapping rhythmic decoupling sequence, and a strict scheduling order and silence interval should be set for each beat execution unit, thus forming a peak-shifting control baseline with clear time constraints. The implementation process is as follows: Based on the resonant threshold list constructed in the previous step, the beat numbers, durations, and spectral overlap regions corresponding to all first-order resonant frequency bands are extracted. Furthermore, based on the beat numbers, their respective control response times, gain change rates, and output current disturbance amplitudes are mapped to establish a multi-parameter mapping table. In this embodiment, beats 121, 122, and 123 correspond to excitation frequency bands of 1.42 GHz, 2.08 GHz, and 3.64 GHz, respectively. All exhibit gain control rates exceeding 1.2 units per microsecond, output current undergoing more than three directional flips within 200 nanoseconds, single jump amplitudes exceeding 0.25 units, and peak signals falling within the corresponding parasitic resonant bandwidth. Further extraction of the control phase difference values ​​for these beats reveals a difference of 0.038 microseconds between beats 121 and 122, and 0.035 microseconds between beats 122 and 123, both less than 0.05 microseconds, indicating excessively high control command scheduling density. To break the coupling relationship between timing overlap and energy amplification, the temporal order of the beats must be re-planned.

[0039] High-risk beats are extracted from the original control sequence and interleaved with low-coupling auxiliary beats to construct a rhythmic decoupling sequence without continuous excitation behavior. In this example, three low-interference beats, numbered 118, 119, and 124, are selected. Their gain change rate is less than 0.5 units per microsecond, their phase difference is greater than 0.12 microseconds, and they do not exhibit energy coupling amplification tendencies. The high-risk beat 122 is placed first, followed by beats 118 and 119 for adjustment buffering, then beat 121 is executed, and finally beats 124 and 123 are inserted to form a complete signal beat scheduling chain. The new sequence is: 122→118→119→121→124→123. The entire reorganization process ensures that control commands between any two first-level resonant beats are isolated by low-risk beats, avoiding the continuous superposition of adjustment behaviors. Meanwhile, a start execution time is set for all beats. Based on the principle of minimizing phase offset, the interval between beats 122 and 118 is set to 0.6 microseconds, between 118 and 119 to 0.4 microseconds, and between 121 and 124 to 0.7 microseconds. This ensures that the output current release of each segment is fully unloaded within the response range of the on-chip inductors and capacitors, avoiding regulation backlash caused by residual energy at the tail end.

[0040] After completing the decoupling and sequencing of the clock cycles, a clear silent interval is added to each cycle execution window, and a corresponding control scheduling list is constructed. The silent interval is the non-drive time window between the output signal of the previous cycle and the execution of the control signal of the next cycle after the output signal of the previous cycle has completely stabilized. Its duration is determined by the energy release rate and oscillation duration of the previous cycle. For example, in cycle 122, the output current drops from 0.82 units to 0.33 units within 350 nanoseconds after the excitation signal is issued, with a slope of 1.4 units per microsecond. There are still two short-cycle residual oscillations before the waveform stabilizes, so the silent interval is set to 300 nanoseconds. In contrast, cycle 119 takes only 190 nanoseconds from signal issuance to output stabilization, and the waveform residual amplitude is less than 0.05 units, so the silent period is set to 180 nanoseconds. The execution time, response time, silent interval, and scheduling order of all cycles are marked as a fixed template, called the control cycle baseline table. Each beat is identified by its number, trigger start point, control response duration, minimum silence interval, and corresponding frequency window. The entire table is used to ensure strict execution of the time interval logic at the hardware scheduling level. Under this baseline, simulation data shows that the energy spectral density of the output current decreases by 47% in the range of 1.4 GHz to 3.7 GHz, and the average settling time of the regulation behavior is shortened from 560 nanoseconds to 390 nanoseconds, indicating that peak-shifting scheduling effectively suppresses the closed-loop energy reconfiguration behavior of the resonant link.

[0041] Based on the beat sequence of the peak-shifting control baseline, the dual-phase breathing traction window, mirror hysteresis valve, reversing energy bypass and time siphon groove are driven to perform segmented energy release, so that local energy is unloaded sequentially in the time-series beat, and a dynamic closed-loop suppression link is constructed to suppress energy resonance and maintain output stability. To ensure that the staggered control timing sequence, after timing decoupling and silent configuration, can effectively release the energy gradually accumulated in the drive path, a multi-level segmented energy guiding and unloading mechanism needs to be constructed, forming a complete closed loop within the control timing, ultimately achieving energy resonance suppression and output waveform stability. The implementation process is as follows: Based on the start time, peak energy, and control response window corresponding to each cycle in the peak-shifting control baseline, the opening and closing rhythm of the dual-phase breathing traction window is set. 80 nanoseconds before the execution of each high-energy cycle, the first phase window is opened, and the pre-excitation energy is guided to the transition guide path through a low-capacitance coupling network on the conductive chip, allowing 10%-20% of the energy to be released first. After the excitation signal reaches its peak, the second phase window reverses its conduction for a 40 nanosecond delay, switching the energy release direction and transmitting the remaining waveform to the low-impedance energy dissipation region through parallel leads. For example, in cycle 122, the output peak is 0.92 units, the first traction window releases 0.16 units, the second traction window outputs 0.24 units, and the remaining signal enters the main path, reducing the load on the main drive path.

[0042] Based on the response characteristics of the falling segment of each beat, the activation window of the mirror hysteresis valve is set in the residual energy band before the signal ends. This structure constructs an anti-slope waveform with equal amplitude and phase delay at the point where the control waveform's falling slope is maximum by setting a reverse voltage guiding structure. Taking beat 121 as an example, the maximum slope of the main signal during its falling phase is -4.1 units / microsecond. The mirror hysteresis valve is activated at 420 nanoseconds of the signal. By delaying the input of an inverted control pulse by 120 nanoseconds, a compensating negative slope wave is formed in the output channel, effectively reducing the main channel's abrupt change rate to -2.3 units / microsecond, smoothing the control signal edges, and effectively eliminating waveform tearing.

[0043] The remaining unextracted excitation energy after the mirror hysteresis valve is configured to flow into the foldback energy bypass network, and guided by the subsequent dissipation path to the bias network far from the main drive channel. In this step, the foldback path adopts a three-segment delay spiral structure, set to 1.8 mm, 2.5 mm, and 3.1 mm respectively, and by configuring a parallel damping mesh at the end of each segment, the peak energy is dissipated as heat flow. Of the residual 0.33 units of energy generated by cycle 123, approximately 0.24 units are absorbed by this path, with the maximum heat rise not exceeding 4.1 degrees Celsius, far below the chip thermal damage threshold. This ensures that the current energy that cannot be immediately extracted from all main paths can be safely diverted.

[0044] To completely eliminate resonant link interference between cycles, a time siphon trench structure is deployed within the quiet window between cycle executions. Its function is to absorb the reverse waveform energy remaining in the encapsulation network or trace ends after the previous cycle is completed, bringing it to the ground plane and preventing it from interfering with the next cycle as excitation background. During the 220 nanosecond quiet interval between cycles 119 and 121, the time siphon trench responds with nanosecond-level scheduling. A parallel siphon line is pre-connected at the end of the signal transmission path, and through precise delay control, it starts for 12 nanoseconds in each quiet window, absorbing approximately 0.07 unit energy bands to prevent energy accumulation and reflected excitation.

[0045] The above physical structures are uniformly scheduled according to their cycle numbering and execution timing, forming a closed-loop energy control link consisting of four levels: pre-output, slope adjustment, residual guidance, and tail-end clearing. Each cycle is configured with a pre-execution window, main execution area, post-execution response, and empty window absorption section on the time axis, and the above control units are activated by level triggering. After running the entire sequence, the output signal is continuously monitored for 4000 cycles, revealing a 38% increase in eye diagram aperture ratio, a 62% decrease in peak reflectivity, and current disturbances at all feedback nodes within ±0.06 units, indicating a significant improvement in output stability.

[0046] The following example, using a real experimental process and a production test line, explains how the technical solution of this invention discovers and suppresses the timing conflict between the DC offset compensation loop and the automatic gain control loop, how it identifies high-risk resonant frequency bands, how it resolves the problem at its root through clock reordering and segmented energy release, and compares the improvement of existing technologies with that of this solution using specific data.

[0047] Example Background and Chip Operating Conditions: The device under test is a driver chip for a four-channel optical transceiver, with a single-channel rated data rate of 25Gb / s, a target differential output current of 8mA, a flip-chip ball grid package, a typical parasitic inductance of 1.8 nanohenries, and a parasitic capacitance of 0.52 picofarads. The upper-level excitation uses a 2.5 GHz sinusoidal superimposed pseudo-random binary sequence as input, with a temperature of 45 degrees Celsius, a power supply of 3.3 volts, and a peak-to-peak power supply ripple of 20 millivolts. The existing testing methods on the production line only perform steady-state eye diagram and bit error rate detection, without clock-level timing analysis; under these conditions, sporadic output oscillations and eye diagram closures frequently occur, resulting in a return rate of approximately 5.8%.

[0048] First, following the concept outlined in the first part of this invention, the input test waveform, output current trajectory, and bias voltage trajectory are aligned at the millisecond level under the same time base, with a sampling resolution of 0.1 microseconds and a time window of 1 millisecond. Each sine cycle is considered a beat unit, and the time difference between the rising edge of the input and the peak bias is extracted beat by beat as the phase misalignment value. Simultaneously, within each beat, a sliding window of 10 sampling points is used to search for amplitude jumps. When the temperature is 45 degrees Celsius and the input amplitude is 200 millivolts, the statistical results of 400 consecutive beats show that the median phase misalignment is 0.24 microseconds, and the 95th percentile is 0.41 microseconds; there are a total of 154 amplitude jump events, occurring on average once every 7 beats. After mapping the two types of features into a time-series conflict observation band in chronological order, a continuous red highlighted segment appears between beats 118 and 123, with a phase misalignment greater than 0.2 microseconds and a jump amplitude exceeding 0.25 units, indicating that the two control loops are overriding each other during this period.

[0049] Secondly, based on the six bright beats in the observation band, the overlapping positions of the beats were determined point by point, and an overlapping beat index was generated according to the duration and polarity direction of the jumps. Taking beat 121 as an example, the bias peak lags the input rising edge by 0.39 microseconds, and the output current experiences a negative jump of 0.31 units at the beginning of this beat, lasting for 90 nanoseconds. After organizing the index using five fields—beat number, phase misalignment, jump amplitude, duration, and direction—it can be seen that beats 119, 121, 122, and 123 meet the high-risk characteristics, with two jumps in opposite directions, indicating that there is a counter-superposition of compensation and gain adjustment.

[0050] Third, using the interference node information in the index, a timing superposition relationship between the two closed-loop control commands is constructed. For each interference cycle, the start time of the DC offset compensation response and the start time of the automatic gain control response are recorded, and the difference between the two is defined as the control phase difference. Simultaneously, the rate of change of the output current during the response phase is calculated and defined as the gain change rate. In cycle 120, the control phase difference is only 0.02 microseconds, and the gain change rate is 2.1 units per microsecond; in cycle 121, the control phase difference is 0.03 microseconds, and the gain change rate is 1.85 units per microsecond. Plotting the phase difference and change rate of consecutive cycles as an energy coupling interference diagram using a color heatmap shows that cycles 118 to 123 are represented by a continuous red band, indicating that the strong coupling section spans six cycles.

[0051] Fourth, frequency domain cross-validation was performed on the strongly coupled section using package parasitic parameters. First, parasitic inductance of 1.8 nanohenries, parasitic capacitance of 0.52 picofarads, and wiring resistance of 1.3 ohms were obtained from measured geometric and material parameters. Then, a network analyzer was used to scan the package impedance curve in the 0.5 to 5 GHz range, identifying three inherent frequency points with impedance valleys below 10 ohms at 1.45 GHz, 2.1 GHz, and 3.6 GHz. Spectral analysis of the output waveform from the 120th to the 123rd beat revealed significant energy peaks at 1.42 GHz, 2.08 GHz, and 3.64 GHz, with peak values ​​of 0.41 units, 0.36 units, and 0.29 units respectively, all exceeding three times the background noise. The two sets of frequency points corresponded one-to-one, proving that the energy within the strongly coupled beats indeed falls within the package resonant bandwidth. This generates a list of resonance thresholds: with the center frequency ±100 MHz as the effective bandwidth, 1.42, 2.08, and 3.64 GHz are marked as first-order resonance bands, and the corresponding beats are mainly concentrated between 121 and 123.

[0052] Fifth, based on the resonance threshold list, these high-risk beats are extracted from the original sequence and rearranged according to the beat phase difference distribution to construct a non-overlapping rhythmic decoupling sequence. Simultaneously, an execution order and silence interval are set for each beat segment. The actual sequence used is 122→118→119→121→124→123. Low-risk beats 118, 119, and 124 are inserted between high-risk beats as buffers to ensure that at least one non-interfering beat is isolated between any two first-order resonance beats. The silence interval is set according to the energy release rate: 300 nanoseconds after beat 122, 180 nanoseconds after beat 121, and 260 nanoseconds after beat 123. After rearrangement and silence configuration, in time-domain simulations, the energy spectral density from 1.4 GHz to 3.7 GHz decreases by approximately 47%, and the average settling time decreases from 560 nanoseconds to 390 nanoseconds.

[0053] Sixth, the segmented energy release phase begins. The four types of time-domain energy channels are driven according to the rearranged beat order, causing local energy to be unloaded sequentially, forming a closed-loop suppression link. Taking beat 122 as an example: 80 nanoseconds before the peak, the first phase traction window is opened, drawing out 0.16 units of leader energy; 40 nanoseconds after the peak, the second phase traction window is opened, unloading another 0.24 units; 120 nanoseconds after the maximum slope of the falling edge, the mirror hysteresis valve is triggered, slowing the falling slope from -4.1 units per microsecond to -2.3 units per microsecond; then, the residual energy is introduced into a three-stage delayed spiral bypass, where parallel damping at the ends of the three stages converts 0.24 units of energy into heat; finally, within the silent window before the next beat, a time siphon is opened to absorb 0.07 units of tail energy, preventing residual energy from reflecting in the encapsulation network. Similar operations are applied to beats 121 and 123, ensuring that the energy of all strongly coupled beats is diverted, delayed, dissipated, and absorbed within their respective time windows. After 2000 consecutive cycles of testing, the output current no longer exhibited continuous oscillation, and the highest temperature rise on the chip was reduced by 3.5 degrees Celsius compared to the original method. This indicates that the energy closed-loop suppression not only stabilized the waveform but also reduced thermal stress.

[0054] To make the results readily apparent, the following comparison of key performance indicators for the same chip under the same temperature, power supply, and excitation conditions, using the original production line method and the technical solution of this invention, is presented. All tests were conducted at 45 degrees Celsius, 3.3 volts, an input amplitude of 200 millivolts, and a time window of 1 millisecond; each indicator is calculated as the average of 10 repeated measurements, with the standard deviation in parentheses.

[0055] As can be seen from the table, the improvements in the critical link are systematic: the three inherent resonant peaks in the frequency domain are significantly reduced, ringing and overshoot in the time domain converge synchronously, eye diagram opening and bit error rate are optimized by orders of magnitude, and settling time and temperature rise also decrease synchronously. These data come from the same chip, the same test fixture, and the same environmental conditions; the only difference lies in whether the entire process proposed in this invention—observation band construction, overlapping beat identification, crosstalk diagram generation, resonant threshold delineation, beat staggering rearrangement, and segmented energy release—was adopted. This is sufficient to demonstrate that the technical solution of this invention has significant beneficial effects and engineering value compared to existing technologies.

[0056] This invention enables comprehensive dynamic observation of signal beat behavior with millisecond-level time accuracy during the testing phase, and accurately identifies the timing overlap between the offset compensation loop and the automatic gain control loop at the microsecond level, thereby achieving high-resolution extraction of feedback interference nodes. Furthermore, through coupling strength mapping and directional identification of resonant frequency bands, a closed-loop debugging mechanism is constructed for the coordinated regulation of time beat rearrangement and energy release rhythm, enabling the quantification and orderly suppression of previously difficult-to-capture and interpret nonlinear oscillations. Finally, through the combined application of peak-shifting control baseline and multi-stage energy unloading structures, it not only significantly reduces the risk of high-frequency oscillations, signal distortion, and thermal overload at the device output caused by coupling interference, but also effectively improves eye diagram quality, stabilizes bit error rate performance, and extends device lifespan, solving the key pain point of existing testing schemes lacking a response mechanism for dynamic adjustment interference.

[0057] This invention provides, for example Figure 2 The test system for the DC offset compensation circuit of the optical transceiver driver chip shown includes a timing conflict observation band generation module, a beat overlap identification module, an energy coupling analysis module, a resonance threshold identification module, a rhythm decoupling control module, and an energy suppression execution module. The timing conflict observation band generation module performs time alignment between the test waveform and the bias trajectory of the optical transceiver driver chip, extracts the phase misalignment and amplitude jump between beats, and maps the continuous change results into timing conflict observation bands to identify beat overlap areas. The beat overlap identification module determines the beat overlap points of the DC offset compensation loop based on the phase misalignment trend of the timing conflict observation band, and generates an overlap beat index based on the duration and polarity direction of the amplitude jump, which is used to identify the interference nodes. The energy coupling analysis module uses the overlapping beat index to construct the timing superposition relationship between the DC offset compensation loop and the automatic gain control loop, marks the beat phase difference and gain change rate, and generates an energy coupling interference diagram to characterize the coupling strength between feedback loops. The resonance threshold identification module analyzes the energy resonance trend of the driving end based on the high-intensity segment of the energy coupling interference diagram and the frequency response of the encapsulated parasitic parameters, determines the excitation frequency band, and generates a list of resonance thresholds. The rhythm decoupling control module rearranges the beat sequence according to the beat phase difference distribution based on the high-risk frequency bands of the resonance threshold list, constructs the rhythm decoupling sequence, and sets the execution order and silence interval to form a staggered control baseline. The energy suppression execution module, based on the peak-shifting control baseline, drives the dual-phase breathing traction window, mirror hysteresis valve, reversing energy bypass, and time siphon groove to perform segmented energy release, so that local energy is unloaded sequentially in the timing cycle, and a dynamic closed-loop suppression link is constructed to suppress energy resonance and maintain output stability.

[0058] The test method for the DC offset compensation circuit of the optical transceiver driver chip provided in this embodiment of the invention is implemented by the test system for the DC offset compensation circuit of the optical transceiver driver chip described above. For details of the specific methods and procedures of the test system for the DC offset compensation circuit of the optical transceiver driver chip described above, please refer to the embodiment of the test method for the DC offset compensation circuit of the optical transceiver driver chip described above, which will not be repeated here.

[0059] Please see Figure 3 , Figure 3 This is a schematic diagram illustrating an embodiment of a computer-readable storage medium provided by an embodiment of the present invention. For example... Figure 3 As shown, this embodiment provides a computer-readable storage medium 600, on which a computer program 611 is stored. When the computer program 611 is executed by a processor, it implements the test method of the DC offset compensation circuit of the optical transceiver driver chip described above.

[0060] The foregoing has only described certain exemplary embodiments of the present invention by way of illustration. Undoubtedly, those skilled in the art can modify the described embodiments in various ways without departing from the spirit and scope of the present invention. Therefore, the foregoing drawings and descriptions are illustrative in nature and should not be construed as limiting the scope of protection of the claims of the present invention.

Claims

1. A test method for the DC offset compensation circuit of an optical transceiver driver chip, characterized in that, Includes the following steps: The test waveforms and bias trajectories of the optical transceiver driver chip are time-aligned, the phase misalignment and amplitude jump between beats are extracted, and the continuous change results are mapped into timing conflict observation bands to identify beat overlap areas. Based on the phase misalignment trend of the timing conflict observation band, the beat overlap point of the DC offset compensation loop is determined, and the overlap beat index is generated according to the duration and polarity direction of the amplitude jump to identify the interference node. The timing superposition relationship between the DC offset compensation loop and the automatic gain control loop is constructed by using the overlapping beat index. The beat phase difference and gain change rate are marked to generate an energy coupling interference diagram, which is used to characterize the coupling strength between feedback loops. Based on the high-intensity segment of the energy coupling interference diagram and combined with the frequency response of the package parasitic parameters, the energy resonance trend of the driving end is analyzed, the excitation frequency band is determined, and a list of resonance threshold regions is generated. Based on the high-risk frequency bands of the resonance threshold list, the beat sequence is rearranged according to the beat phase difference distribution to construct a rhythm decoupling sequence, and the execution order and silence interval are set to form a staggered control baseline. Based on the peak-shifting control baseline, a dual-phase breathing traction window, a mirror hysteresis valve, a reversing energy bypass, and a time siphon groove are used to perform segmented energy release, so that local energy is unloaded sequentially in the timing cycle, and a dynamic closed-loop suppression link is constructed to suppress energy resonance and maintain output stability.

2. The test method for the DC offset compensation circuit of the optical transceiver driver chip according to claim 1, characterized in that, The steps for generating time-series conflicting observation bands are as follows: A modulation signal is applied to the input terminal of the optical transceiver driver chip, and the output current signal and bias voltage trajectory are acquired simultaneously. The time alignment is performed by an acquisition device with dual-channel synchronous sampling capability to obtain a uniform time axis signal with amplitude normalization. Based on the time-aligned signal data, each continuous sine cycle is defined as a beat interval. The time interval between the rising edge of the input signal and the peak of the bias trajectory is extracted, the phase misalignment value of the beat is calculated, and the phase drift curve is generated. Based on the instantaneous amplitude change within each beat, the differential amplitude of the output current is calculated using a sliding window, and the amplitude jump point is identified. When the jump amplitude exceeds the average rate of change threshold of the beat, the jump event is recorded. The phase misalignment values ​​and amplitude jump values ​​of each beat are mapped to a two-dimensional coordinate plane in chronological order, and the beat conflict intensity is identified by a color scale to generate a time-series conflict observation band to indicate the signal adjustment overlap area.

3. The test method for the DC offset compensation circuit of the optical transceiver driver chip according to claim 2, characterized in that, The steps for generating the overlapping beat index are as follows: Based on the timing conflict observation band, the beats with phase misalignment exceeding the preset threshold are segmented and extracted, and time offset curves are plotted on a unified time axis to determine the response drift trend; By combining the amplitude jump time distribution characteristics within the beat, the jump start time is matched with the phase misalignment offset interval to identify the beat overlap point in the DC offset compensation loop adjustment process; Extract the jump amplitude, jump duration, and jump direction parameters for each jump event, calculate the amplitude difference between the jump start point and the end point, and determine the jump direction; An overlapping beat index is established based on phase misalignment value, jump amplitude, jump duration, and jump direction. Interference risk nodes are classified and identified, and an overlapping beat index list is generated.

4. The test method for the DC offset compensation circuit of the optical transceiver driver chip according to claim 3, characterized in that, The steps for generating the energy coupling interference diagram are as follows: Based on the overlapping beat index list, extract the key control command points of the bias adjustment curve and gain control curve in the interference beat, record the bias response start time and gain response start time, and calibrate them on a unified time axis; The difference between the bias response start time and the gain response start time is calculated to obtain the control phase difference for each disturbance cycle. At the same time, the gain change rate of the corresponding cycle is extracted as the control strength parameter. The beat number, control phase difference, and gain change rate are plotted as a two-dimensional thermodynamic distribution map, and the coupling strength levels are divided according to the color scale to generate an energy coupling interference map that reflects the distribution of feedback coupling strength. Based on the highlighted areas in the energy coupling interference diagram, the numbers, phase differences, and gain change rates of the coupled nodes are extracted to establish a list of coupled nodes and classify them by intensity.

5. The test method for the DC offset compensation circuit of the optical transceiver driver chip according to claim 4, characterized in that, The steps for generating the resonance threshold list are as follows: Based on the high-intensity beat segments marked in the energy coupling interference diagram, the beat number, control phase difference, and gain change rate are extracted to perform point-by-point analysis on the sudden jump behavior of the output current waveform; By combining the parasitic parameters of the optical transceiver driver chip package, inductance, capacitance and resistance models are established, and the transmission characteristics are analyzed by frequency scanning to determine the inherent resonant frequency of the structure. Frequency domain analysis is performed on the output waveform in a highly coupled beat. The position and amplitude of the spectral peak are obtained by using fast Fourier transform and compared with the packaged frequency response curve to identify the resonant frequency point. Based on the identified resonant frequency, frequency bandwidth, and beat time interval, a list of resonant threshold regions is output.

6. The test method for the DC offset compensation circuit of the optical transceiver driver chip according to claim 5, characterized in that, During the process of outputting the list of resonance thresholds, the resonance level is classified and identified by comparing the degree of overlap between the resonance frequency and the resonant frequency band formed by the package parasitic parameters. The peak amplitude and energy duration of the output waveform are recorded in the identified high-coupling beats to determine the resonance risk level of the driver end.

7. The test method for the DC offset compensation circuit of the optical transceiver driver chip according to claim 5, characterized in that, The steps for generating the peak shift control baseline are as follows: Based on the resonant threshold list, the number, duration, control phase difference, gain change rate and output current disturbance amplitude of high-risk beats are extracted, a multi-parameter mapping table is established and the control command scheduling density is analyzed; High-risk beats are extracted from the original control sequence and interleaved with low-interference beats with a gain change rate of no more than 0.5 units per microsecond and a phase difference of more than 0.12 microseconds to construct a rhythmic decoupling sequence without continuous excitation behavior, and a fixed time interval is set between beats. Based on the beat decoupling sequence, a silent interval is set for each beat execution window, and a control beat baseline table containing beat number, trigger start point, control response duration and minimum silent interval is established to form a time-constrained staggered control baseline.

8. The test method for the DC offset compensation circuit of the optical transceiver driver chip according to claim 7, characterized in that, Based on the peak-shifting control baseline, a dual-phase breathing traction window, a mirror hysteresis valve, a reversible energy bypass, and a time siphon groove are used to perform segmented energy release. Local energy is unloaded sequentially within the time-series beats, and the dynamic closed-loop suppression link is constructed in the following steps: Based on the start time, peak energy and control response window of each beat in the staggered control baseline, the opening and closing rhythm of the dual-phase breathing traction window is set, and energy is guided to the low impedance energy dissipation zone in stages before and after the excitation signal. Based on the response characteristics of the descent segment of the clock, the mirror hysteresis valve is activated in the residual energy band before the signal ends, and a delayed anti-sloping waveform is constructed through the reverse voltage guiding structure to reduce the main channel mutation rate. The excitation energy that is not discharged by the mirror hysteresis valve is introduced into the foldback energy bypass network, and the peak energy is converted into heat flow for dissipation through a multi-segment time-delay spiral structure and parallel damping network. A time siphon trench structure is deployed within the silent window between cycle executions to absorb residual energy at the end of the trace through parallel siphon lines, preventing energy reflection from interfering with the next cycle. By sequentially scheduling the biphase breathing traction window, mirror hysteresis valve, reversing energy bypass, and time siphon channel, a closed-loop energy regulation link consisting of pre-extraction, slope adjustment, residual guidance, and tail-end clearance is constructed.

9. A test system for a DC offset compensation circuit of an optical transceiver driver chip, used to implement the test method for the DC offset compensation circuit of an optical transceiver driver chip as described in any one of claims 1-8, characterized in that, It includes a timing conflict observation band generation module, a beat overlap identification module, an energy coupling analysis module, a resonance threshold identification module, a rhythm decoupling control module, and an energy suppression execution module: The timing conflict observation band generation module performs time alignment between the test waveform and the bias trajectory of the optical transceiver driver chip, extracts the phase misalignment and amplitude jump between beats, and maps the continuous change results into timing conflict observation bands to identify beat overlap areas. The beat overlap identification module determines the beat overlap points of the DC offset compensation loop based on the phase misalignment trend of the timing conflict observation band, and generates an overlap beat index based on the duration and polarity direction of the amplitude jump, which is used to identify the interference nodes. The energy coupling analysis module uses the overlapping beat index to construct the timing superposition relationship between the DC offset compensation loop and the automatic gain control loop, marks the beat phase difference and gain change rate, and generates an energy coupling interference diagram to characterize the coupling strength between feedback loops. The resonance threshold identification module analyzes the energy resonance trend of the driving end based on the high-intensity segment of the energy coupling interference diagram and the frequency response of the encapsulated parasitic parameters, determines the excitation frequency band, and generates a list of resonance thresholds. The rhythm decoupling control module rearranges the beat sequence according to the beat phase difference distribution based on the high-risk frequency bands of the resonance threshold list, constructs the rhythm decoupling sequence, and sets the execution order and silence interval to form a staggered control baseline. The energy suppression execution module, based on the peak-shifting control baseline, drives the dual-phase breathing traction window, mirror hysteresis valve, reversing energy bypass, and time siphon groove to perform segmented energy release, so that local energy is unloaded sequentially in the timing cycle, and a dynamic closed-loop suppression link is constructed to suppress energy resonance and maintain output stability.

10. A computer-readable storage medium storing a computer program, characterized in that, When the computer program is executed by the processor, it implements the test method for the DC offset compensation circuit of the optical transceiver driver chip as described in any one of claims 1 to 8.

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