Discrete peak current control method and device for power factor correction converter

Abstract

The invention belongs to the technical field of power electronic converter control, and particularly relates to a discrete peak current control method and device of a power factor correction converter. The control of the discrete peak current on the output voltage can be realized without an error amplifier, the power factor correction of the input side of the converter is realized, the design of error amplifier parameters is not needed, and the realization is simple; an integral element in the error amplifier is not needed, so that rapid transient response can be realized; in addition, the converter switching tube can realize zero-current soft switching, the switching loss is low, and the efficiency is improved.

Description

TECHNICAL FIELD

[0001] The present application belongs to the technical field of power electronic converter control, and particularly relates to a discrete peak current control method and device for a power factor correction converter. BACKGROUND

[0002] Single-phase Boost power factor correction (PFC) converters have the advantages of simple structure and control, high power density, and the ability to achieve power factor correction and grid current harmonic suppression. Single-phase Boost PFC converters have become an indispensable part of electronic devices.

[0003] When a single-phase Boost PFC converter operates in critical conduction mode (CRM), the inductor current of the Boost PFC converter is exactly zero at the end of each switching period, which enables zero-current turn-on of the power switch, thereby reducing switching loss and improving efficiency. In addition, the CRM single-phase Boost PFC converter can avoid the problem of diode reverse recovery, thus simplifying circuit design and reducing electromagnetic interference. As a result, the CRM single-phase Boost PFC converter provides a "green" path that balances efficiency and cost for small and medium power supply devices, and has important engineering and academic value.

[0004] When the input voltage and load of a single-phase Boost PFC converter change suddenly, there are problems of large overshoot (significant overshoot and undershoot) and slow recovery of the output voltage: this not only affects the stable operation of the downstream load (such as precision chips or processors), but also may cause PFC system oscillation or even damage (such as overshoot voltage exceeding the stress tolerance of components, causing overheating and damage, shortening the service life of the entire power supply system); in addition, the output current oscillation caused by output voltage transients may also cause electromagnetic interference and other problems. Therefore, it is crucial to study the transient response characteristics of single-phase Boost PFC converters and design faster and more robust control strategies to improve the reliability, stability, and dynamic performance of single-phase PFC power supplies under complex operating conditions.

[0005] The existing transient response optimization methods for CRM single-phase Boost PFC converters are as follows:

[0006] ① Load current feedforward control: directly detect the load current variation of single-phase Boost PFC converter, and feed this signal to the controller. When the load changes, the controller can immediately adjust the switch-on behavior of the switch (such as temporarily increasing or decreasing the duty cycle) without waiting for the output voltage to change, and inject or reduce energy in advance. This method is one of the most direct and effective methods to improve transient response, which can significantly reduce the output voltage overshoot and recovery time; however, the control delay caused by the integral element in the error amplifier cannot be avoided.

[0007] ② Variable bandwidth voltage loop control: the traditional CRM single-phase Boost PFC converter has a narrow voltage control loop bandwidth (usually 10-20 Hz) to suppress the output voltage double-frequency ripple. This method allows the bandwidth of the voltage control loop to vary dynamically: in steady state, keep low bandwidth to filter out double-frequency ripple; when a transient process is detected, temporarily increase the bandwidth to improve the transient response speed. This method can achieve a trade-off between steady-state accuracy and dynamic speed; however, this method requires multiple bandwidths to be set, increasing the difficulty of controller implementation; in addition, when the controller switches between multiple bandwidths, oscillation phenomenon is easy to occur.

[0008] ③ Boundary adaptive control: temporarily let the converter out of CRM mode and enter continuous conduction mode (CCM) or discontinuous conduction mode (DCM) during the transient period. In this way, the single-phase Boost PFC converter can break through the upper limit of the switching frequency in CRM mode, thereby providing or absorbing more energy in a short time, quickly restoring the oscillating output voltage, and improving the transient response performance.

[0009] ④ Optimizing the output capacitor of the converter: appropriately reduce the output capacitor value of the converter under the premise of meeting the steady-state ripple requirement of the output voltage; a smaller capacitor will produce a larger voltage change rate under the same power disturbance, which will provide a stronger feedback signal to the voltage control loop to trigger faster correction action of the control loop. However, this method requires careful consideration of the output capacitor value to avoid excessive steady-state ripple.

[0010] ⑤ Increase parallel auxiliary circuit: based on the single-phase boost PFC converter, add a small power converter (such as a Buck or Boost converter) triggered by a transient signal. When the load changes, the above-mentioned parallel auxiliary circuit can quickly inject or absorb current to the output of the PFC converter, balance the energy, and alleviate the output voltage overshoot. This method can significantly improve the transient response performance of the single-phase boost PFC converter, but the auxiliary current will increase the complexity and hardware and software cost of the system.

[0011] In summary, the research of a CRM single-phase Boost PFC converter control method capable of achieving fast transient response performance is an academic and engineering problem that needs to be solved by the technical personnel in this research field, and provides key support for the improvement of PFC converter control performance and industrial implementation. SUMMARY

[0012] The purpose of the present application is to provide a fast transient high-efficiency PFC discrete peak current control method and device: the single-phase Boost PFC converter in the present application works in the inductor current critical conduction mode, and when the converter output load is suddenly changed, the converter can achieve faster output voltage response speed and lower output voltage overshoot, thereby protecting the output load; the control method provided by the present application does not require an error amplifier and its compensation network (proportional-integral controller) when generating the inductor peak current given value; in addition, in the present application, the inductor peak current given value is generated by the inductor current peak reference calculation module, the high-low inductor current coefficient calculation module, the discrete peak current calculation module and the average output voltage control loop, therefore, the double-frequency ripple in the PFC converter output voltage will not affect the inductor current, reducing the distortion of the converter input current; in addition, since it works in the inductor current critical conduction mode, the switch tube of the PFC converter can achieve zero-current soft switching, reducing the switching loss, and having certain academic and engineering application value.

[0013] The present application provides a fast transient high-efficiency PFC discrete peak current control method, and the specific technical solutions are as follows:

[0014] A fast transient high-efficiency PFC discrete peak current control method, comprising the following steps:

[0015] Step one: at the start of each control period, sample the input voltage u in , diode rectifier bridge output voltage u m , inductor current i m , output current I oref and output voltage U o of the single-phase Boost power factor correction converter; in rec L o o

[0016] Step two: determine the input voltage zero-crossing time and input voltage amplitude U in according to the input voltage u m ;

[0017] Step three: according to the input voltage amplitude U m , output voltage given value U oref , output current I o , rectifier bridge output voltage u rec ,, the peak inductance current reference i Lm : ;

[0018] Step four: according to the average value of the output voltage of the single-phase Boost power factor correction converter ΔU o and the output voltage given value U oref , determine the high and low inductance current coefficients K IH , K IL : ;

[0019] Step five: according to the peak inductance current reference i Lm and the high and low inductance current coefficients K IH , K IL , determine the discrete peak current i LmH , i LmL : ;

[0020] Step six: the average output voltage control loop principle is as follows: at the zero-crossing moment of the input voltage u in , compare the output voltage U o with the output voltage given value U oref , if the output voltage U o is less than the output voltage given value U oref , select i LmH as the inductance current peak given value i ref ; if the output voltage U o is greater than or equal to the output voltage given value U oref , select i LmL as the inductance current peak given value i ref : ;

[0021] Step seven: the inductance peak current control loop principle is as follows: compare the inductance current peak given value i ref with the inductance current i L , when the inductance current i L reaches the peak given value i ref , the comparator outputs a high level; the high level signal is connected to the reset end (R end) of the set-reset flip-flop (RS flip-flop), and the RS flip-flop outputs a low level signal v gs ;

[0022] Step eight: send the inductance current i L into the zero-crossing detector, when the inductance current i LWhen the zero-crossing detector output is zero, the high level signal is input into the set end (S end) of the set-reset flip-flop (RS flip-flop), and the RS flip-flop outputs a high level signal v gs ;

[0023] Step nine: using the signal v gs , the switch S of the single-phase Boost power factor correction converter is controlled to achieve the input power factor correction and fast transient response control target of the power factor correction converter working in the critical conduction mode of inductor current.

[0024] Preferably, the inductor current peak reference i Lm of the single-phase Boost power factor correction converter includes:

[0025] According to the input voltage amplitude U m , the output voltage given value U oref , the output current I o , the rectifier bridge output voltage u rec , the inductor current peak reference i Lm of the single-phase Boost power factor correction converter is determined: ;

[0026] Preferably, the high and low inductor current coefficients K IH , K IL of the fast transient high efficiency PFC discrete peak current control method include:

[0027] According to the output voltage average value ripple ΔU o and the output voltage given value U oref of the single-phase Boost power factor correction converter, the high and low inductor current coefficients K IH , K IL are determined: ;

[0028] Preferably, the discrete peak current i LmH , i LmL of the fast transient high efficiency PFC discrete peak current control method includes:

[0029] According to the inductor current peak reference i Lm and the high and low inductor current coefficients K IH , K IL of the single-phase Boost power factor correction converter, the discrete peak current i LmH , i LmL is determined: ;

[0030] Preferably, the inductor current peak setpoint i in the fast transient high-efficiency PFC discrete peak current control method is... ref ,include:

[0031] At input voltage u in At the zero crossing, the output voltage U o With the output voltage setpoint U oref Comparison, if the output voltage U o Less than the output voltage setpoint U oref Select i LmH As the peak value of the inductor current i ref If the output voltage U o Greater than or equal to the output voltage setpoint U oref Select i LmL As the given value of the peak inductor current i ref : ;

[0032] Preferably, the control signal v of the fast transient high-efficiency PFC discrete peak current control method gs ,include:

[0033] ① Set the peak inductor current value i ref With inductor current i L When comparing, when the inductor current i L Reaching the peak given value i ref When the comparator outputs a high level, this high-level signal is connected to the reset terminal (R terminal) of the set-reset flip-flop (RS flip-flop), and the RS flip-flop outputs a low-level signal v. gs ;

[0034] ② The inductor current i L The signal is fed into a zero-crossing detector when the inductor current i L When the zero-crossing signal drops to zero, the zero-crossing detector outputs a high level; this high-level signal is then connected to the set (S) terminal of the set-reset flip-flop (RS flip-flop), and the RS flip-flop outputs a high-level signal v. gs .

[0035] This invention proposes a fast transient high-efficiency PFC discrete peak current control method, as detailed below:

[0036] At the initial moment of the control cycle, the sampling circuit completes sampling of the input voltage, diode rectifier bridge output voltage, inductor current, output current, and output voltage. Based on the instantaneous value of the input voltage, its zero-crossing time and amplitude can be determined. Based on the input voltage amplitude, rectifier bridge output voltage, output current, and output voltage setpoint, the peak reference for the power factor correction converter inductor current can be determined. Based on the average output voltage ripple, the high and low inductor current coefficients can be determined. Based on the peak inductor current reference and the high and low inductor current coefficients, the discrete peak current can be determined. At the zero-crossing time of the input voltage, the average output voltage and the setpoint are compared to determine the discrete peak current. The method involves selecting and generating a peak inductor current setpoint. The peak inductor current setpoint is compared with the inductor current; when the inductor current reaches the peak setpoint, the comparator outputs a high level. This high-level signal is connected to the reset terminal of a set-reset trigger (RS trigger), causing the RS trigger to output a low-level signal. The inductor current is then fed to a zero-crossing detector; when the inductor current drops to zero, the zero-crossing detector outputs a high-level signal. This high-level signal is connected to the set terminal of the RS trigger, causing the RS trigger to output a high-level signal. The output signal of the RS trigger is used to control the switching transistor of the power factor correction converter, achieving discrete peak current control. This method achieves discrete peak current control of the output voltage and power factor correction on the input side of the converter without the need for an error amplifier. It eliminates the need for error amplifier parameters, simplifying implementation. Because the integrator stage in the error amplifier is eliminated, a fast transient response is achieved. Furthermore, since the single-phase BoostPFC converter operates in the inductor current critical conduction mode, the switching transistor can achieve zero-current soft-turn-on, reducing switching losses and improving efficiency. This invention has certain academic and engineering application value.

[0037] In addition, the present invention also provides a fast transient high-efficiency PFC discrete peak current control device, which has the same beneficial effects as the fast transient high-efficiency PFC discrete peak current control method described above. Attached Figure Description

[0038] Figure 1 This is a schematic diagram of the structure of a fast transient high-efficiency PFC discrete peak current control method and device provided in an embodiment of the present invention;

[0039] Figure 2 This is an embodiment of the present invention providing an operating condition (input voltage RMS value U). in =110V, input voltage frequency 50Hz, output voltage U o The instantaneous input voltage u of (400V, output load resistance R=1200Ω) in Time-domain simulation waveform;

[0040] Figure 3This is an embodiment of the present invention providing an operating condition (input voltage RMS value U). in =110V, input voltage frequency 50Hz, output voltage U o The instantaneous value of the input current i (V = 400V, output load resistance R = 1200Ω) in Time-domain simulation waveform;

[0041] Figure 4 This is an embodiment of the present invention providing an operating condition (input voltage RMS value U). in =110V, input voltage frequency 50Hz, output voltage U o The instantaneous output voltage u of the rectifier bridge (400V, output load resistance R=1200Ω) rec Time-domain simulation waveform;

[0042] Figure 5 This is an embodiment of the present invention providing an operating condition (input voltage RMS value U). in =110V, input voltage frequency 50Hz, output voltage U o The instantaneous value of the inductor current i (e.g., V = 400V, output load resistance R = 1200Ω) L Time-domain simulation waveform;

[0043] Figure 6 This is an embodiment of the present invention providing an operating condition (input voltage RMS value U). in =110V, input voltage frequency 50Hz, output voltage U o The instantaneous output voltage U of (400V, output load resistance R=1200Ω) o Time-domain simulation waveform;

[0044] Figure 7 This is an embodiment of the present invention providing an operating condition (input voltage RMS value U). in =110V, input voltage frequency 50Hz, output voltage U o =400V, output load resistance R=1200Ω) output voltage setpoint U oref and average value U oave Time-domain simulation waveform;

[0045] Figure 8 This is an embodiment of the present invention providing an operating condition (input voltage RMS value U). in =110V, input voltage frequency 50Hz, output voltage U o The discrete peak current i of (V=400V, output load resistance R=1200Ω) LmH i LmL Time-domain simulation waveform;

[0046] Figure 9 This is an embodiment of the present invention providing an operating condition (input voltage RMS value U). in =110V, input voltage frequency 50Hz, output voltage U o =400V, output load resistance R=1200Ω) The peak inductor current given by i ref Time-domain simulation waveform;

[0047] Figure 10 This is an embodiment of the present invention providing an operating condition (input voltage RMS value U). in =110V, input voltage frequency 50Hz, output voltage U o =400V, output load resistance R=1200Ω) The peak inductor current given by i ref Instantaneous value of inductor current i L and MOSFET control signal v gs Time-domain simulation waveform;

[0048] Figure 11 This is an embodiment of the present invention providing an operating condition (input voltage RMS value U). in =110V, input voltage frequency 50Hz, output voltage U o The MOSFET control signal v (400V, output load resistance R=1200Ω) is... gs and MOSFET current i d Time-domain simulation waveform;

[0049] Figure 12 This is an embodiment of the present invention providing a load switching condition (input voltage RMS value U) in =110V, input voltage frequency 50Hz, output voltage U o =400V, output resistance R jumps from 2400Ω to 1200Ω) instantaneous value of output current I o Time-domain simulation waveform;

[0050] Figure 13 This is an embodiment of the present invention providing a load switching condition (input voltage RMS value U) in =110V, input voltage frequency 50Hz, output voltage U o =400V, output resistance R jumps from 2400Ω to 1200Ω) average output voltage U oave and given value U oave Time-domain simulation waveform;

[0051] Figure 14 When using the traditional PI peak current control method, under a certain operating condition (the effective value of the input voltage U...), in=110V, input voltage frequency 50Hz, output voltage U o The instantaneous value of the input current i (V = 400V, output load resistance R = 1200Ω) in Time-domain simulation waveform;

[0052] Figure 15 When using the traditional PI peak current control method, under a load switching condition (input voltage RMS value U), in =110V, input voltage frequency 50Hz, output voltage U o =400V, output resistance R jumps from 2400Ω to 1200Ω) instantaneous value of output current I o Time-domain simulation waveform;

[0053] Figure 16 When using the traditional PI peak current control method, under a load switching condition (input voltage RMS value U), in =110V, input voltage frequency 50Hz, output voltage U o =400V, output resistance R jumps from 2400Ω to 1200Ω) average output voltage U oave and given value U oave The time-domain simulation waveform diagram. Detailed Implementation

[0054] The technical solutions of the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of the present invention, and not all embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those of ordinary skill in the art without creative effort are within the protection scope of the present invention.

[0055] The core of this invention is to provide a fast transient high-efficiency PFC discrete peak current control method: this method can achieve discrete peak current control of output voltage and power factor correction on the input side of the converter without the need for an error amplifier, and it is simple to implement without designing error amplifier parameters; since there is no need for the integration stage in the error amplifier, a fast transient response can be achieved; in addition, the converter switching transistor can achieve zero-current soft turn-on, with low switching losses and improved operating efficiency, which has certain academic and engineering application value.

[0056] The technical solutions of the embodiments of the present invention will be clearly and completely described below with reference to the accompanying drawings. Obviously, the described embodiments are only some embodiments of the present invention, and not all embodiments. Based on the embodiments of the present invention, all other embodiments obtained by those of ordinary skill in the art without creative effort are within the protection scope of the present invention.

[0057] To enable those skilled in the art to better understand the present invention, the present invention will be further described in detail below with reference to the accompanying drawings and specific embodiments.

[0058] When operating in the critical conduction mode of inductor current, single-phase Boost PFC converters typically employ one of two control methods: ① Peak current control method based on a PI controller: This method uses a voltage control loop as the outer loop, where the PI controller outputs the peak current command signal; the inner loop generates the control signal for the power switching devices based on the peak inductor current. ② Fixed on-time control method: This method eliminates the need for a current control loop, with the voltage control loop determining the on-time of the power switching devices. In both of these control methods, the integral element in the PI controller within the voltage control loop can slow down the converter's transient response.

[0059] To address the issue of slow transient response in single-phase Boost PFC converters operating in critical current conduction mode, the following method is employed:

[0060] ① Load current feedforward control: This method directly detects changes in the load current of the single-phase Boost PFC converter and feeds this signal forward to the controller. When the load changes abruptly, the controller can immediately adjust the conduction behavior of the switching transistors (e.g., temporarily increasing or decreasing the duty cycle) without waiting for the output voltage to change, thus injecting or reducing energy in advance. This method is one of the most direct and effective ways to improve transient response, significantly reducing output voltage overshoot and recovery time; however, the control delay caused by the integrator in the error amplifier is still unavoidable.

[0061] ② Variable Bandwidth Voltage Loop Control: Traditional CRM single-phase Boost PFC converters suffer from narrow voltage control loop bandwidth (typically 10~20Hz) to suppress double-frequency ripple in the output voltage. This method allows for dynamic variation of the voltage control loop bandwidth: in steady state, a low bandwidth is maintained to filter out double-frequency ripple; when a transient process is detected, the bandwidth can be temporarily increased to improve transient response speed. This method achieves a trade-off between steady-state accuracy and dynamic speed; however, it requires setting multiple bandwidths, increasing the difficulty of controller implementation; furthermore, oscillations are prone to occur when the controller switches between multiple bandwidths.

[0062] ③ Boundary Adaptive Control: During transients, the converter is temporarily deactivated from CRM mode and enters either Continuous Conduction Mode (CCM) or Discontinuous Conduction Mode (DCM). This allows the single-phase Boost PFC converter to overcome the upper limit of the switching frequency in CRM mode, thereby providing or absorbing more energy in a short time, enabling rapid recovery of the oscillating output voltage and improving transient response performance.

[0063] ④ Optimize the converter output capacitor: While meeting the steady-state ripple requirements of the output voltage, appropriately reduce the value of the converter output capacitor. A smaller capacitor will generate a larger voltage change rate under the same power disturbance. This change rate will provide a stronger feedback signal to the voltage control loop, triggering the control loop to correct faster. However, this method requires careful consideration of the output capacitor value to avoid excessive steady-state ripple.

[0064] ⑤ Add a parallel auxiliary circuit: Based on the single-phase boost PFC converter, add a parallel low-power converter (such as a Buck or Boost converter) triggered by a transient signal. During load changes, this parallel auxiliary circuit can quickly inject or absorb current from the PFC converter output, balancing energy and mitigating output voltage overshoot. This method can significantly improve the transient response performance of the single-phase boost PFC converter, but the auxiliary current increases system complexity and software / hardware costs.

[0065] Based on the above-mentioned research status, the embodiments of the present invention mainly provide a fast transient high-efficiency PFC discrete peak current control method and device.

[0066] Figure 1 This is a structural block diagram of a fast transient high-efficiency PFC discrete peak current control device provided in an embodiment of the present invention; Figure 1 Includes a single-phase Boost PFC converter and a discrete peak current control block diagram: The discrete peak current control block diagram consists of a voltage and current sampling module, an inductor current peak reference i Lm Calculation module, high and low inductor current coefficient K IH K IL Calculation module, discrete peak current i LmH i LmL The system consists of a calculation module, an average output voltage control loop, and an inductor peak current control loop; the average output voltage control loop comprises a comparator, a selector, and a discrete peak current i. LmH i LmL Composition: The inductor peak current control loop consists of a comparator, a zero-crossing detector, and an RS flip-flop.

[0067] Depend on Figure 1 The diagram shows a structural block diagram of a fast transient high-efficiency PFC discrete peak current control device. The steps of a fast transient high-efficiency PFC discrete peak current control method provided by this embodiment are as follows:

[0068] Step 1: At the beginning of each control cycle, sample the input voltage u of the single-phase Boost power factor correction converter. in The output voltage u of the diode rectifier bridge rec Inductor current i L Output current I o and output voltage U o ;

[0069] Step 2: Based on the input voltage u in Determine the zero-crossing time of the input voltage and the amplitude U of the input voltage. m ;

[0070] Step 3: Based on the input voltage amplitude U m Output voltage setpoint U oref Output current I o 1. Rectifier bridge output voltage u rec Determine the peak inductor current reference i for a single-phase Boost power factor correction converter. Lm :

[0071] ;

[0072] Step 4: Correct the average output voltage ripple ΔU of the single-phase Boost power factor correction converter. o and output voltage setpoint U oref Determine the high and low inductance current coefficients K. IH K IL :

[0073] ;

[0074] Step 5: Based on the peak inductor current reference i of the single-phase Boost power factor correction converter Lm and high and low inductance current coefficient K IH K IL Determine the discrete peak current i LmH i LmL :

[0075] ;

[0076] Step Six: The principle of the average output voltage control loop is as follows: When the input voltage u... in At the zero crossing, the output voltage U o With the output voltage setpoint Uoref Comparison, if the output voltage U o Less than the output voltage setpoint U oref Select i LmH As the given value of the peak inductor current i ref If the output voltage U o Greater than or equal to the output voltage setpoint U oref Select i LmL As the given value of the peak inductor current i ref :

[0077] ;

[0078] Step 7: The principle of the inductor peak current control loop is as follows: Set the inductor current peak value i... ref With inductor current i L When comparing, when the inductor current i L Reaching the peak given value i ref When the comparator outputs a high level, this high-level signal is connected to the reset terminal (R terminal) of the set-reset flip-flop (RS flip-flop), and the RS flip-flop outputs a low-level signal v. gs ;

[0079] Step 8: Convert the inductor current i L The signal is fed into a zero-crossing detector when the inductor current i L When the zero-crossing signal drops to zero, the zero-crossing detector outputs a high level; this high-level signal is then connected to the set (S) terminal of the set-reset flip-flop (RS flip-flop), and the RS flip-flop outputs a high-level signal v. gs ;

[0080] Step 9: Use the signal v output by the RS flip-flop gs By controlling the switching transistor S of a single-phase Boost power factor correction converter, the input power factor correction and fast transient response control objectives of the power factor correction converter operating in the critical conduction mode of inductor current are achieved.

[0081] The present invention also provides a fast transient high-efficiency PFC discrete peak current control device, which has the same beneficial effects as the fast transient high-efficiency PFC discrete peak current control method described above, and will not be repeated here.

[0082] Figures 2-11 This is a steady-state operating condition (input voltage RMS value U) provided in an embodiment of the present invention. in =110V, input voltage frequency 50Hz, output voltage U o Steady-state simulation results of a fast transient high-efficiency PFC discrete peak current control device (400V, output load resistance R=1200Ω).

[0083] Figures 2-11 The simulation results shown include the following circuit and controller parameters for the single-phase Boost PFC converter: the effective value of the input voltage U. in =110V, input voltage frequency 50Hz, output voltage U o =400V, output load resistance R=1200Ω, inductance L=0.6mH, output capacitance C=220uF, high and low inductor current coefficient K IH =1.01、K IL =0.99.

[0084] Figure 2 This is an embodiment of the present invention providing an operating condition (input voltage RMS value U). in =110V, input voltage frequency 50Hz, output voltage U o The instantaneous input voltage u of (400V, output load resistance R=1200Ω) in Time-domain simulation waveform; by Figure 2 It can be seen that the input voltage u in The amplitude is 155.54V and the frequency is 50Hz.

[0085] Figure 3 This is an embodiment of the present invention providing an operating condition (input voltage RMS value U). in =110V, input voltage frequency 50Hz, output voltage U o The instantaneous value of the input current i (V = 400V, output load resistance R = 1200Ω) in Time-domain simulation waveform; by Figure 3 It can be seen that the input current i in The amplitude is 3.5A and the frequency is 50Hz. in with u in They have the same phase and good sinusoidal properties.

[0086] Figure 4 This is an embodiment of the present invention providing an operating condition (input voltage RMS value U). in =110V, input voltage frequency 50Hz, output voltage U o The instantaneous output voltage u of the rectifier bridge (400V, output load resistance R=1200Ω) rec Time-domain simulation waveform; by Figure 4 It can be seen that u rec For u in The absolute value is 155.54V, the amplitude is 100Hz.

[0087] Figure 5 This is an embodiment of the present invention providing an operating condition (input voltage RMS value U). in=110V, input voltage frequency 50Hz, output voltage U o The instantaneous value of the inductor current i (e.g., V = 400V, output load resistance R = 1200Ω) L Time-domain simulation waveform; by Figure 5 It can be seen that the inductor current i L Peak discrete peak current i LmH i LmL The selection is made from the range specified in the original text.

[0088] Figure 6 This is an embodiment of the present invention providing an operating condition (input voltage RMS value U). in =110V, input voltage frequency 50Hz, output voltage U o The instantaneous output voltage U of (400V, output load resistance R=1200Ω) o Time-domain simulation waveform; by Figure 6 It can be seen that the instantaneous value of the output voltage U o It can follow its given value of 400V well, and there is a ripple of twice the power frequency at a frequency of 100Hz.

[0089] Figure 7 This is an embodiment of the present invention providing an operating condition (input voltage RMS value U). in =110V, input voltage frequency 50Hz, output voltage U o =400V, output load resistance R=1200Ω) output voltage setpoint U oref and average value U oave Time-domain simulation waveform; by Figure 7 It can be seen that the average output voltage U oave Output voltage setpoint U oref To achieve good follow-up.

[0090] Figure 8 This is an embodiment of the present invention providing an operating condition (input voltage RMS value U). in =110V, input voltage frequency 50Hz, output voltage U o The discrete peak current i of (V=400V, output load resistance R=1200Ω) LmH i LmL Time-domain simulation waveform; by Figure 8 It can be seen that there is a relationship i between discrete peak currents. LmH >i LmL .

[0091] Figure 9 This is an embodiment of the present invention providing an operating condition (input voltage RMS value U). in =110V, input voltage frequency 50Hz, output voltage U o=400V, output load resistance R=1200Ω) The peak inductor current given by i ref Time-domain simulation waveform; by Figure 9 It can be seen that the inductor current i L Peak discrete peak current i LmH i LmL The selection is made from the range specified in the original text.

[0092] Figure 10 This is an embodiment of the present invention providing an operating condition (input voltage RMS value U). in =110V, input voltage frequency 50Hz, output voltage U o =400V, output load resistance R=1200Ω) The peak inductor current given by i ref Instantaneous value of inductor current i L and MOSFET control signal v gs Time-domain simulation waveform; by Figure 10 It can be seen that when the instantaneous value of the inductor current i L Rise to peak inductor current given i ref At that time, the MOSFET control signal v gs When the inductor current becomes zero, the MOSFET is turned off; when the instantaneous value of the inductor current i... L When the voltage drops to zero, the MOSFET control signal v gs When the current changes to one, the MOSFET turns on, enabling control over the instantaneous value i of the inductor current. L Peak current control.

[0093] Figure 11 This is an embodiment of the present invention providing an operating condition (input voltage RMS value U). in =110V, input voltage frequency 50Hz, output voltage U o The MOSFET control signal v (400V, output load resistance R=1200Ω) is... gs and MOSFET current i d Time-domain simulation waveform; by Figure 11 It can be seen that the MOSFET control signal v gs When the MOSFET current i changes from zero to one, d A value equal to zero allows for zero-current turn-on of the MOSFET power switching device, reducing turn-on losses and improving operating efficiency.

[0094] Figures 12-13 This invention provides an embodiment of a load transient condition (input voltage RMS value U) in =110V, input voltage frequency 50Hz, output voltage U o=400V, output resistance R jumps from 2400Ω to 1200Ω), load transient simulation results of a fast transient high-efficiency PFC discrete peak current control device; where the high and low inductor current coefficients K IH =1.01、K IL =0.99.

[0095] Figure 14 When using the traditional PI peak current control method, under a certain operating condition (the effective value of the input voltage U...), in =110V, input voltage frequency 50Hz, output voltage U o The instantaneous value of the input current i (V = 400V, output load resistance R = 1200Ω) in Time-domain simulation waveform results.

[0096] Figures 15-16 When using the traditional PI peak current control method, under a certain load transient condition (input voltage RMS value U), in =110V, input voltage frequency 50Hz, output voltage U o The load transient simulation results are as follows: =400V, output resistance R jumps from 2400Ω to 1200Ω.

[0097] contrast Figure 13 and Figure 16 The simulation results show that when a load change occurs (the output resistance R changes from 2400Ω to 1200Ω), the average output voltage U, obtained by using the fast transient high-efficiency PFC discrete peak current control method provided in this embodiment of the invention, is significantly improved. oave It can quickly follow its given value U oref , and U oave The overshoot can be significantly reduced.

[0098] contrast Figure 12 and Figure 15 The simulation results show that when a load change occurs (output resistance R changes from 2400Ω to 1200Ω), the fast transient high-efficiency PFC discrete peak current control method provided in this embodiment of the invention can reduce the output current I. o The oscillation.

[0099] contrast Figure 3 and Figure 14 The simulation results show that the fast transient high-efficiency PFC discrete peak current control method provided in this embodiment of the invention, while ensuring the advantages of fast transient response, also achieves high efficiency in input current i. in Good sinusoidal properties can still be guaranteed, meaning steady-state performance can be ensured.

[0100] The above verification process demonstrates the effectiveness and feasibility of the fast transient high-efficiency PFC discrete peak current control method provided in this embodiment.

[0101] The foregoing has provided a detailed description of a fast transient high-efficiency PFC discrete peak current control method and apparatus provided by the present invention. The various embodiments in the specification are described in a progressive manner, with each embodiment focusing on its differences from other embodiments. Similar or identical parts between embodiments can be referred to interchangeably. For the apparatus disclosed in the embodiments, since it corresponds to the method disclosed in the embodiments, the description is relatively simple; relevant parts can be referred to in the method section. It should be noted that those skilled in the art can make several improvements and modifications to the present invention without departing from the principles of the invention, and these improvements and modifications also fall within the protection scope of the claims of the present invention.

Claims

1. A discrete peak current control method for a power factor correction converter, characterized by, The method comprises the following steps: S1, at the beginning of each control period, sample the input voltage u of the single-phase Boost power factor correction converter in , diode rectifier bridge output voltage u rec , inductor current i L , output current I o and output voltage U o ; S2, determining the input voltage zero-crossing time and input voltage amplitude U in based on the input voltage u m ; S3. determining a peak value reference i of the inductance current of the single-phase Boost power factor correction converter m , the output voltage given value U oref , the output current I o , the rectifier bridge output voltage u rec , the output voltage given value U Lm : ； S4, determining high, low inductance current coefficient K o and output voltage given value U oref , determining high, low inductance current coefficient K IH , K IL : ； S5. The inductor current peak reference i Lm and high, low inductor current coefficients K IH , K IL , determining a discrete peak current i LmH , i LmL : ； S6, setting average output voltage control loop: in input voltage u in Zero crossing moment, output voltage U o And output voltage given value U oref Comparison, if output voltage U o Less than output voltage given value U oref , select i LmH As inductance current peak value given value i ref ; If output voltage U o Greater than or equal to output voltage given value U oref , select i LmL As inductance current peak value given value i ref : ； S7, set inductance peak current control loop: the inductance current peak given value i ref is compared with the inductance current i L , when the inductance current i L reaches the peak given value i ref , the comparator outputs high level; the high level signal is connected to the reset end (R end) of the set-reset flip-flop (RS flip-flop), and the RS flip-flop outputs low level signal v gs ; S8, the inductor current i L is sent into a zero-crossing detector, when the inductor current i L falls to zero, the zero-crossing detector outputs a high level; the high level signal is sent into the set end of the RS flip-flop, the RS flip-flop outputs a high level signal v gs ; S9, the signal v output by the RS flip-flop gs The switch S of the single-phase Boost power factor correction converter is controlled to achieve the input power factor correction and fast transient response control target of the power factor correction converter operating in the critical inductor current conduction mode.

2. A discrete peak current control apparatus for a power factor correction converter, for use in the discrete peak current control method for a power factor correction converter as claimed in claim 1, characterized by, The method comprises a voltage and current sampling module, an input voltage zero-crossing time detection and amplitude calculation module, an inductance current peak reference calculation module, a high and low inductance current coefficient calculation module, a discrete peak current calculation module, an average output voltage control loop module, and an inductance peak current control loop module. The voltage and current sampling module is configured to: at the start of each control cycle, sample the input voltage u of the single-phase Boost power factor correction converter in , a diode rectifier bridge output voltage u rec , an inductor current i L , an output current I o , and an output voltage U o ; The input voltage zero-crossing time detection and amplitude calculation module is configured to determine the input voltage zero-crossing time and the input voltage amplitude U m according to the input voltage u in ​ The inductance current peak reference calculation module is configured to: determine the inductance current peak reference i of the single-phase Boost power factor correction converter according to input voltage amplitude U m , output voltage given value U oref , output current I o , rectifier bridge output voltage u rec . Lm The expression is . The high-low inductance current coefficient calculation module is configured to: determine high and low inductance current coefficients K o and K oref based on an output voltage average value ripple ΔU IH of a single-phase Boost power factor correction converter and an output voltage given value U IL , and the expression is: ； The discrete peak current calculation module is configured to: determine a discrete peak current i Lm and high and low inductance current coefficients K IH , K IL , determine a discrete peak current i LmH , i LmL , and an expression thereof is: ； The average output voltage control loop module includes a comparator and a selector, used for: controlling the input voltage u in At the zero crossing, the output voltage U o With the output voltage setpoint U oref Comparison, if the output voltage U o Less than the output voltage setpoint U oref Select i LmH As the given value of the peak inductor current i ref If the output voltage U o Greater than or equal to the output voltage setpoint U oref Select i LmL As the peak value of the inductor current i ref The expression is: ； The inductance peak current control loop module comprises a comparator, a zero-crossing detector and an RS flip-flop, and is used for: comparing an inductance current peak value i ref with a given value i L , and outputting a high level signal when the inductance current i L reaches the peak given value i ref ; inputting the high level signal into a reset end (R end) of a set-reset flip-flop (RS flip-flop), and outputting a low level signal v gs ; The inductance current i L is fed into a zero-crossing detector, which outputs a high level when the inductance current i L drops to zero; this high level signal is fed into the set terminal of an RS flip-flop, which outputs a high level signal v gs ; The RS flip-flop output signal v gs For: control the switch S of single-phase Boost power factor correction converter, realize the input power factor correction and fast transient response control target of power factor correction converter working in inductor current critical conduction mode, and realize zero current turn-on of converter power switch device, thereby reducing turn-on loss and improving converter working efficiency.

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