Thermal flow meter sensor drive circuit, thermal flow meter, and flow measurement method

Abstract

The present application relates to the technical field of fluid measurement, and discloses a thermal flowmeter sensor driving circuit, a thermal flowmeter and a flow measurement method, which comprise: a constant current source module connected to a series branch and used for providing a constant excitation current to the series branch; a virtual bridge arm module, two input ends of the virtual bridge arm module being connected to a first end and a second end respectively, and the virtual bridge arm module being used for generating and outputting a fixed reference voltage in response to a voltage across the series branch; and a signal control and collection module, which is used for enabling the voltage at a second input end of the signal control and collection module to be equal to the reference voltage in a steady state through internal negative feedback of the signal control and collection module, and outputting a voltage signal, wherein the magnitude of the voltage signal represents the imbalance degree of the resistance values of a first sensing resistor and a second sensing resistor. In the present application, all currents flow through the sensor, the signal strength is large, no additional amplification is needed, noise introduced by an amplification circuit is avoided, and the signal collected by an ADC is more stable.

Description

Technical Field

[0001] This invention relates to the field of fluid measurement technology, specifically to a thermal flow meter sensor drive circuit, a thermal flow meter, and a flow measurement method. Background Technology

[0002] Thermal flow meters measure fluid flow based on the principle of heat conduction. The capillary tube in the flow sensor has two sets of temperature-sensitive metal wires wound around it. By heating the two sets of metal wires to maintain a certain temperature, when fluid passes through, the thermal balance between the two sets of metal wires is broken. The heat of one set of metal wires is transferred to the other set of metal wires with the fluid, causing a change in the resistance difference between the two sets of metal wires. By measuring the change in resistance between the two sets of metal wires and making corresponding adjustments, the flow rate through the capillary tube can be obtained.

[0003] There are generally two methods for measuring the resistance change between two sets of metal wires: one is to collect the voltage at one end, that is, to directly collect the voltage value at the midpoint between the two sets of metal wires; the other is to use a Wheatstone bridge.

[0004] The first method is to directly acquire the voltage change between the two sets of metal wires. However, in order to ensure the accuracy of the flow meter and to make the fluid flowing into the flow meter pipe tend to be in a laminar flow state, most of the fluid entering the thermal flow meter will be diverted by the distributor in the pipe, and only a small portion will flow into the capillary tube in the sensor. At this time, the voltage change between the two sets of metal wires is relatively small when directly measured. For flow meters with larger flow rates, a high-resolution ADC acquisition chip is required. Subsequently, a signal amplification circuit needs to be added to amplify the voltage change between the two sets of metal wires, but this will introduce some additional noise, resulting in a relatively fluctuating voltage value acquired by the final control chip.

[0005] Another method is to use a Wheatstone bridge to measure the change in resistance between two sets of metal wires, such as... Figure 1As shown, the Wheatstone bridge consists of four resistors R1 to R4, with each pair forming an arm. For a thermal flow meter, the two sets of metal wires on its capillary tube form one arm (R1, R3), and two additional resistors (R2, R4) are needed for the other arm. The change in resistance between the metal wires R1 and R3 on the sensor can be measured by measuring the voltage Vout between points A and B. Similarly, when the resistance values ​​of the upper and lower arms of the R2 and R4 arm change, the voltage Vout between points A and B will also change. Since R2 and R4 are connected in parallel with R1 and R3, R2 and R4 will receive a portion of the current, resulting in some power consumption and heat. When the temperature changes, not only will the resistance values ​​of the metal wires R1 and R3 on the sensor change, but the resistance values ​​of the resistors R2 and R4 on the other arm will also change, leading to an increase in the temperature drift of the flow meter. At the same time, when the resistance values ​​of R2 and R4 change, the current flowing through the metal wires R1 and R3 on the sensor will also change. Currently, the solutions to this problem, both domestically and internationally, are to use a low-temperature drift resistor on the other bridge arm. This is relatively expensive. Furthermore, when using a low-temperature drift resistor, it is possible that one resistor on the other bridge arm will drift in the positive direction while the other will drift in the negative direction, which will lead to an increase in the temperature drift of the flow meter.

[0006] Therefore, there is an urgent need for a thermal flow meter sensor driving solution that can effectively suppress temperature drift and has a low cost. Summary of the Invention

[0007] The purpose of this invention is to overcome the shortcomings of the prior art and provide a thermal flow meter sensor driving circuit, a thermal flow meter, and a flow measurement method, thereby solving the problem of increased flow meter temperature drift caused by changes in resistance on the sensor driving bridge due to temperature.

[0008] This invention is achieved through the following technical solution:

[0009] A thermal flow meter sensor driving circuit is provided for driving and measuring a flow sensor comprising a first sensing resistor R6 and a second sensing resistor R12. The first sensing resistor R6 and the second sensing resistor R12 are connected in series to form a series branch. The end of the first sensing resistor R6 furthest from the second sensing resistor R12 is the first terminal A of the series branch, and the end of the second sensing resistor R12 furthest from the first sensing resistor R6 is the second terminal C of the series branch. The common connection terminal of the first sensing resistor R6 and the second sensing resistor R12 is an intermediate node B. The driving circuit includes:

[0010] A constant current source module is connected to the series branch and is used to provide a constant excitation current I1 to the series branch;

[0011] A virtual bridge arm module, wherein the two input terminals of the virtual bridge arm module are respectively connected to the first terminal A and the second terminal C, and are used to generate and output a fixed reference voltage Vd in response to the voltage across the series branch.

[0012] A signal control and acquisition module is provided, wherein the first input terminal of the signal control and acquisition module is connected to the output terminal of the virtual bridge arm module to obtain the reference voltage Vd, and the second input terminal of the signal control and acquisition module is connected to the intermediate node B; the signal control and acquisition module is used to make the voltage of the second input terminal of the signal control and acquisition module equal to the reference voltage Vd in steady state through internal negative feedback, and output a voltage signal Vadc, wherein the magnitude of the voltage signal Vadc represents the degree of imbalance between the resistance values ​​of the first sensing resistor R6 and the second sensing resistor R12.

[0013] As an optimization, the constant current source module includes a first operational amplifier U1.1, a first MOS transistor Q1, a first sampling resistor R1, a second sampling resistor R2, a first frequency compensation capacitor C1, and a second filter capacitor C2;

[0014] The drain of the first MOS transistor Q1 is connected to the power module through the first sampling resistor R1, and the source of the first MOS transistor Q1 is connected to the first terminal A to provide the excitation current I1 to the series branch.

[0015] The output terminal of the first operational amplifier U1.1 is connected to the gate of the first MOS transistor Q1. The non-inverting input terminal of the first operational amplifier U1.1 receives the reference voltage VREF. The inverting input terminal of the first operational amplifier U1.1 is grounded through a second resistor R2 in series.

[0016] The first frequency compensation capacitor C1 is connected between the output terminal and the inverting input terminal of the first operational amplifier U1.1, and the second filter capacitor C2 is connected between the non-inverting input terminal and ground of the first operational amplifier U1.1.

[0017] As an optimization, the virtual bridge arm module includes a summing and scaling circuit composed of operational amplifiers, which is used to sum and scale the voltages across the series branch to generate the reference voltage Vd.

[0018] As an optimization, the virtual bridge arm module includes a second operational amplifier U1.2, a third operational amplifier U1.3, and a fourth operational amplifier U1.4;

[0019] The second operational amplifier U1.2 and the third operational amplifier U1.3 are both configured as voltage followers. The non-inverting input of the second operational amplifier U1.2 is connected to the first terminal A, and the output of the second operational amplifier U1.2 is connected to the inverting input of the second operational amplifier U1.2. The non-inverting input of the third operational amplifier U1.3 is connected to the second terminal C, and the output of the third operational amplifier U1.3 is connected to the inverting input of the third operational amplifier U1.3.

[0020] The fourth operational amplifier U1.4 is configured as a non-inverting summing circuit. The non-inverting input terminal of the fourth operational amplifier U1.4 receives the output voltage of the second operational amplifier U1.2 through a fifth resistor R5 in series. The non-inverting input terminal of the fourth operational amplifier U1.4 receives the output voltage of the third operational amplifier U1.3 through an eleventh resistor R11 in series. At the same time, the inverting input terminal of the fourth operational amplifier U1.4 is connected to the output terminal of the fourth operational amplifier U1.4 through a fourth resistor R4, which serves as a feedback resistor. The output terminal of the fourth operational amplifier U1.4 outputs the reference voltage Vd as the output node D.

[0021] As an optimization, the fifth resistor R5 and the eleventh resistor R11 have the same resistance value; thus, the reference voltage Vd is half the sum of the voltage at the first terminal A and the voltage at the second terminal C.

[0022] As an optimization, the signal control and acquisition module includes a fourth operational amplifier U2.1; the fourth operational amplifier U2.1 is configured as a voltage follower, the non-inverting input of the fourth operational amplifier U2.1 is connected to the output D of the virtual bridge arm module, the output of the fourth operational amplifier U2.1 is used to output the buffered reference voltage Vd, and the inverting input of the fourth operational amplifier U2.1 is connected to the output of the fourth operational amplifier U2.1.

[0023] As an optimization, the signal control and acquisition module further includes a fifth operational amplifier U2.2 and a sixth operational amplifier U2.3;

[0024] The non-inverting input of the fifth operational amplifier U2.2 is connected to the output of the fourth operational amplifier U2.1 to receive the buffered reference voltage Vd; the inverting input of the fifth operational amplifier U2.2 is connected to the intermediate node B through the eighth resistor R8. At the same time, along the signal transmission direction, the two ends of the eighth resistor R8 are connected to the output of the fifth operational amplifier U2.2 through the third resistor R3 and the fourth capacitor in series, respectively.

[0025] The non-inverting input of the sixth operational amplifier U2.3 is connected to the output of the fourth operational amplifier U2.1 to receive the buffered reference voltage Vd; the inverting input of the sixth operational amplifier U2.3 is connected to the output of the fifth operational amplifier U2.2 through a series ninth resistor R9; at the same time, the inverting input of the sixth operational amplifier U2.3 is connected to the output of the sixth operational amplifier U2.3 through a parallel link consisting of a seventh resistor R7 and a third capacitor.

[0026] A ninth resistor R9 is connected between the output terminal of the fifth operational amplifier U2.2 and the inverting input terminal of the sixth operational amplifier U2.3; the output terminal of the sixth operational amplifier U2.3 outputs the voltage signal Vadc.

[0027] As an optimization, the signal control and acquisition module further includes a tenth resistor R10, a thirteenth resistor R13, a fifth capacitor C5, a sixth capacitor C6, and a seventh capacitor C7. The output of the sixth operational amplifier U2.3 is connected in series with the tenth resistor R10 to form a negative voltage signal output terminal ADC-. The output of the fourth operational amplifier U2.1 is connected in series with the thirteenth resistor R13 to form a positive voltage signal output terminal ADC+. The fifth capacitor C5 and the seventh capacitor C7 are connected in series between the positive output terminal ADC+ and the negative output terminal ADC-, and the series connection point of the fifth capacitor C5 and the seventh capacitor C7 is grounded. The two ends of the sixth capacitor C6 are respectively positioned between the positive output terminal ADC+ and the negative output terminal ADC-.

[0028] The present invention also discloses a thermal flow meter, including a flow sensor and a signal processing circuit. The flow sensor includes a first sensing resistor R6 and a second sensing resistor R12, and the signal processing circuit is the thermal flow meter sensor driving circuit as described above.

[0029] This invention also discloses a thermal flow meter flow measurement method based on the aforementioned thermal flow meter sensor drive circuit, comprising the following steps:

[0030] A constant current I1 is supplied to the first sensing resistor R6 and the second sensing resistor R12 of the sensor through a constant current source module;

[0031] A fixed reference voltage Vd is generated by the virtual bridge arm module. The reference voltage Vd is used to balance the system when the resistance values ​​of the first sensing resistor R6 and the second sensing resistor R12 are equal.

[0032] The signal control and acquisition module forces the voltage at the intermediate node B of the first sensing resistor R6 and the second sensing resistor R12 to follow the reference voltage Vd.

[0033] Acquire and measure the voltage signal Vadc output by the signal control and acquisition module to maintain the voltage following relationship;

[0034] The flow rate value flowing through the flow sensor is calculated based on the voltage signal Vadc.

[0035] Compared with the prior art, the present invention has the following advantages and beneficial effects:

[0036] This invention utilizes operational amplifiers to construct virtual bridge arms, completely replacing the temperature-sensitive physical reference resistors (R2, R4) in a traditional Wheatstone bridge, thus fundamentally eliminating the source of temperature drift in this part.

[0037] This invention features a full current flow through the sensor, resulting in a strong signal that requires no additional amplification. This avoids noise introduced by the amplification circuit and makes the signal acquired by the ADC more stable.

[0038] This invention eliminates the need for expensive, low-temperature drift precision resistors, thus reducing hardware costs.

[0039] This invention automatically maintains system balance through deep negative feedback of operational amplifiers, resulting in good dynamic response and a stable and reliable structure. Attached Figure Description

[0040] The accompanying drawings, which are included to provide a further understanding of embodiments of the invention and form part of this application, do not constitute a limitation thereof. In the drawings:

[0041] Figure 1 This is the circuit diagram of a traditional Wheatstone bridge;

[0042] Figure 2 This is a circuit diagram of a specific embodiment of the novel driving circuit for the thermal flow meter sensor of the present invention. Detailed Implementation

[0043] To make the objectives, technical solutions, and advantages of the present invention clearer, the present invention will be further described in detail below with reference to the embodiments and accompanying drawings. The illustrative embodiments and descriptions of the present invention are only used to explain the present invention and are not intended to limit the present invention.

[0044] The following combination Figure 2 The present invention will be described in detail below. Figure 2 The node identifiers A, B, C, D and Figure 1 The meanings of the identifiers in the background technology are different, and this is hereby explained.

[0045] This invention proposes a novel driving method for thermal flow meters to counteract the temperature drift caused by another arm of a Wheatstone bridge (i.e., R2 and R4 in a conventional bridge). The core of this method lies in using operational amplifier circuits to construct a virtual bridge arm to replace the physical resistive bridge arm.

[0046] Before proceeding with the explanation, it should be noted that the flow sensor of the Wheatstone bridge driven and measured using the driving circuit of the present invention includes a flow sensor with a first sensing resistor R6 and a second sensing resistor R12. The first sensing resistor R6 and the second sensing resistor R12 are connected in series to form a series branch. The end of the first sensing resistor R6 away from the second sensing resistor R12 is the first end A of the series branch, and the end of the second sensing resistor R12 away from the first sensing resistor R6 is the second end C of the series branch. The common connection end of the first sensing resistor R6 and the second sensing resistor R12 is the intermediate node B.

[0047] like Figure 2 As shown, the driving circuit of the present invention mainly includes three parts: a constant current source module, a virtual bridge arm module, and a signal control and acquisition module.

[0048] 1. Constant current source module

[0049] This module is used to provide a stable excitation current I1. A preferred embodiment includes a first operational amplifier U1.1, a first MOSFET Q1, a first sampling resistor R1, a second sampling resistor R2, a first frequency compensation capacitor C1, and a second filter capacitor C2.

[0050] Connections: The drain of the first MOSFET Q1 is connected to the power supply VCC through the first sampling resistor R1, and its source is connected to the first terminal A of the sensor branch, providing current I1. The output of the first operational amplifier U1.1 is connected to the gate of Q1. Its non-inverting input receives the reference voltage VREF, and its inverting input is grounded through the second resistor R2. The first frequency compensation capacitor C1 is connected between the output and the inverting input of U1.1 to stabilize the loop. The second filter capacitor C2 is connected between the non-inverting input of U1.1 and ground to filter out noise on VREF.

[0051] Working principle: Operational amplifier U1.1 controls the gate of Q1 to make the voltage across R2 equal to VREF, thereby setting a constant current I1=VREF / R2.

[0052] 2. Virtual bridge arm module

[0053] This module is used to generate a stable reference voltage Vd. A preferred embodiment includes a second operational amplifier U1.2, a third operational amplifier U1.3, and a fourth operational amplifier U1.4.

[0054] Connections: U1.2 and U1.3 are both configured as voltage followers. The non-inverting input of U1.2 is connected to point A, and the non-inverting input of U1.3 is connected to point C. U1.4 is configured as a non-inverting summing circuit, with its non-inverting input receiving the output voltage of U1.2 through resistor R5 and the output voltage of U1.3 through resistor R11. The inverting input of U1.4 is connected to its output D through feedback resistor R4.

[0055] Working principle: U1.2 and U1.3 buffer Va and Vc respectively. When the resistance values ​​of R5 and R11 are equal, a voltage of (Va + Vc) / 2 is obtained at the non-inverting input of U1.4. Since the inverting input of U1.4 is directly connected to the output through R4, it forms a voltage follower, therefore the output voltage Vd = (Va + Vc) / 2. This voltage is the virtual bridge arm midpoint voltage. Because it is generated by the operational amplifier circuit, it is not affected by resistor temperature drift.

[0056] 3. Signal Control and Acquisition Module

[0057] This module is used to process and output sensor signals. A preferred embodiment includes a fourth operational amplifier U2.1, a fifth operational amplifier U2.2, and a sixth operational amplifier U2.3.

[0058] U2.1 is configured as a voltage follower to buffer the Vd voltage output from the virtual bridge arm.

[0059] U2.2 and U2.3 constitute a complex control and filtering circuit. The non-inverting input of U2.2 is connected to the buffered Vd, and the inverting input is connected to the sensor's intermediate node B through R8. Its feedback path includes the parallel connection of R3 and C4, serving both amplification and filtering purposes. The non-inverting input of U2.3 is also connected to the buffered Vd, and the inverting input is connected to the output of U2.2 through R9. Its feedback path includes the parallel connection of R7 and C3.

[0060] Working principle: (Reference) Figure 2 The function of this module is implemented by operational amplifier U2 and its peripheral circuitry. U2.1 is configured as a voltage follower, which buffers the reference voltage Vd from the output terminal D of the virtual bridge arm and passes it to subsequent circuits.

[0061] Utilizing the "virtual short" principle of the op-amp, the voltage at the non-inverting input (pin 5) of U2.2 is equal to the voltage at the inverting input (pin 6) in steady state. Due to the buffering effect of U2.1, the voltage at the non-inverting input of U2.2 is Vd, thus forcing the voltage at its inverting input to also be Vd. The inverting input of U2.2 is connected to the intermediate node B of the sensor via resistor R8, forming a negative feedback loop. Its core function is to force the voltage Vb at point B to always be equal to the reference voltage Vd in steady state.

[0062] When the sensor is balanced, that is, when the resistance of R6 equals the resistance of R12, the bridge arm is naturally balanced, and Vb equals Vd. At this time, no feedback current is needed to maintain balance, and there is no voltage drop across the feedback resistor R3.

[0063] When fluid flows through and causes an imbalance in the resistance values ​​of R6 and R12, Vb will deviate from Vd. At this point, the negative feedback loop immediately activates: U2.2 and U2.3 work together, altering the current distribution through R6 and R12 by outputting or absorbing current, thereby pulling Vb back to a potential equal to Vd. The feedback current generated to maintain this balance will flow through resistor R3, producing a corresponding voltage drop across R3. .

[0064] After further processing and filtering by U2.3 and its peripheral circuits (such as R7, C3, R9, R10, etc.), the voltage signal ultimately forms a differential voltage signal Vadc between the output terminals ADC+ and ADC-. Therefore, the voltage Vadc finally acquired by the ADC is essentially the voltage drop across the feedback resistor R3. .

[0065] More specifically, since R9 equals R7, the output voltage of op-amp U2.3 at point H is: At the same time, by utilizing the virtual short principle of operational amplifiers, there is .

[0066] The voltage finally acquired by the ADC is:

[0067] .

[0068] , , , , They are respectively Figure 2 The voltage values ​​at points B, E, F, G, and H. , These represent the positive and negative voltages of the ADC chip. Same as Vb mentioned above.

[0069] That is, the voltage sampled by the ADC chip is the voltage drop across resistor R3. When the resistance of sensor wire R6 equals the resistance of sensor wire R12, the bridge arms remain balanced. At this time, the voltages at points D and B are equal to half the sum of the voltages at points A and C, and the voltage at point B is equal to the voltage at point F. The voltage acquired by the ADC is... =0; When there is flow, if the resistance value of sensor wire R6 is not equal to the resistance value of sensor wire R12, the bridge arm becomes unbalanced. At this time, the voltage at point D is still equal to half the sum of the voltages at points A and C, but the voltage at point B is not equal. Due to the virtual short of the op-amp, the op-amp will make the voltage at point B always keep the voltage at point D consistent. At this time, the op-amp will output a certain current or absorb part of the current to change the voltage drop across resistors R6 and R12, so that the voltage at point B and the voltage at point D are equal. Since the op-amp needs to output or absorb part of the current, a voltage drop will be generated across R3. By measuring the change in voltage drop across R3, the flow rate through the flow meter can be obtained through program conversion.

[0070] In other words, the voltage ultimately acquired by the ADC is Vadc. When the resistance value of sensor wire R6 equals the resistance value of R12, the bridge arms are balanced, and Vadc = 0. When flow occurs, the resistance values ​​of R6 and R12 are not equal, the bridge arms are unbalanced, and Vadc is not zero. By measuring the magnitude and polarity of Vadc, the flow rate through the flow meter can be calculated using a preset algorithm. The circuit of this invention uses an operational amplifier to construct another set of bridge arms of the Wheatstone bridge, effectively avoiding flow drift caused by resistor temperature drift when using resistors.

[0071] Traditional Wheatstone bridges rely on static comparison using resistive voltage division. When the sensor becomes unbalanced, the bridge outputs an imbalance voltage, passively reflecting the degree of imbalance; the system itself lacks the ability to actively restore balance. This invention generates a dynamic reference Vd through a virtual bridge arm and actively and forcibly pulls the potential of the sensor's midpoint B to be equal to Vd via a high-gain operational amplifier negative feedback loop (U2.2, U2.3, etc.). The current or voltage output to maintain this forced balance (ultimately manifested as Vadc) directly becomes the measurement signal. This achieves a paradigm shift from "passively detecting imbalance" to "actively maintaining balance and reading the maintaining force."

[0072] The reference voltage Vd output by the virtual bridge arm is Vd = f(Va, Vc) (e.g., (Va + Vc) / 2), and its value originates from the total voltage drop of the sensor's own branch. When changes in ambient temperature cause synchronous drift in the sensor resistors R6 and R12 (i.e., the resistance changes proportionally), Va and Vc will change proportionally, but Vd (as a function of them, such as the average value) will maintain its inherent proportional relationship with Va and Vc. Under the action of the servo loop, Vb is forced to be equal to Vd. Since Vb itself is the result of the voltage division of Va and Vc by R6 and R12, when Vd maintains its inherent relationship with Va and Vc, the system automatically and uniquely forces the voltage division ratio of R6 and R12 back to the equilibrium state (i.e., R6 = R12), regardless of their absolute resistance values. Therefore, as long as the effects of temperature changes on R6 and R12 are matched (which is the design basis of thermal flow meter sensors), under zero flow conditions, the system can automatically maintain the output Vadc at the theoretical zero point regardless of temperature changes, without any external temperature sensor or compensation algorithm.

[0073] The constant current source I1 flows entirely through the sensor branch without any shunt. The changes in Va and Vc caused by sensor imbalance are complete and fully captured by the virtual bridge arm. The output signal Vadc does not directly measure the offset of Vb (a small signal), but rather represents the corrective force applied by the servo loop to force Vb = Vd. In an ideal op-amp model, this corrective force (represented by the current flowing through feedback resistor R3) is linearly related to the imbalance of the sensor resistance (R6-R12). This linear relationship is determined by feedback resistor R3, etc., and is easy to control and maintain. Vadc directly represents the voltage of this linear corrective force, achieving a suitable level for ADC sampling without the need for additional differential amplification or gain stages, thus simplifying the signal chain.

[0074] The overall system accuracy depends primarily on: the stability of the constant current source I1; the accuracy of proportional calculations (such as averaging) in the virtual bridge arm, which is determined by the operational amplifier performance and the accuracy of the resistor ratios; and the accuracy of the resistors (such as R3) in the servo loop that determine the linear relationship between the correction force and the imbalance.

[0075] None of the aforementioned key accuracy factors require the use of resistors with high absolute accuracy and low temperature drift. The accuracy of the constant current source is determined by the reference voltage VREF and the sampling resistor R2; the accuracy of the virtual bridge arm and servo loop depends on the resistor matching and ratio stability, rather than absolute temperature drift. This allows for the use of ordinary, low-cost thin-film resistor pairs, achieving high-performance accuracy through a symmetrical PCB layout. The solution eliminates the need for expensive chips such as high-resolution DACs, digital potentiometers, or instrumentation amplifiers; the core components are all general-purpose operational amplifiers. Through a clever circuit architecture, the system accuracy dependency is shifted from expensive, low-temperature-drift components to easily implemented symmetrical design and ratio stability, breaking the convention that high precision inevitably leads to high cost.

[0076] In summary, this invention, by constructing an active servo balancing system based on a virtual bridge arm and using operational amplifier negative feedback as the actuator, produces a series of synergistic and beneficial effects:

[0077] It creates an inherent zero-point temperature drift self-stabilization mechanism, which can automatically maintain zero flow output zero point in the entire temperature range without the need for external temperature sensors and compensation algorithms.

[0078] It enables direct flow measurement with large signal and high linearity, transforms weak bridge imbalance signals into strong servo control signals that are linearly related to flow, and simplifies the signal conditioning circuit.

[0079] It achieves an excellent balance between system accuracy and manufacturing cost. Its accuracy relies on the easily guaranteed symmetry of the resistance ratio, rather than expensive absolute low-temperature drift components, providing a hardware foundation for the widespread use of high-performance thermal flow meters.

[0080] These effects stem from the unique circuit architecture of this invention, which are interconnected and together constitute a simple, stable, efficient, and low-cost thermal flow meter sensor drive solution.

[0081] It should be noted that, Figure 2 In the diagram, R represents a resistor, C represents a capacitor, U represents an operational amplifier, GND represents ground, and AIN+ and AIN- represent the positive output terminal of op-amp U2.1 and the negative input terminal of op-amp U2.2.

[0082] The specific embodiments described above further illustrate the purpose, technical solution, and beneficial effects of the present invention. It should be understood that the above description is only a specific embodiment of the present invention and is not intended to limit the scope of protection of the present invention. Any modifications, equivalent substitutions, improvements, etc., made within the spirit and principles of the present invention should be included within the scope of protection of the present invention.

Claims

1. A thermal flow meter sensor driving circuit for driving and measuring a flow sensor comprising a first sensing resistor R6 and a second sensing resistor R12, wherein, The first sensing resistor R6 and the second sensing resistor R12 are connected in series to form a series branch, and the end of the first sensing resistor R6 away from the second sensing resistor R12 is the first terminal A of the series branch, and the end of the second sensing resistor R12 away from the first sensing resistor R6 is the second terminal C of the series branch. The common connection terminal of the first sensing resistor R6 and the second sensing resistor R12 is the intermediate node B. The driving circuit includes: A constant current source module is connected to the series branch and is used to provide a constant excitation current I1 to the series branch; A virtual bridge arm module, wherein the two input terminals of the virtual bridge arm module are respectively connected to the first terminal A and the second terminal C, and are used to generate and output a fixed reference voltage Vd in response to the voltage across the series branch. A signal control and acquisition module is provided, wherein the first input terminal of the signal control and acquisition module is connected to the output terminal of the virtual bridge arm module to obtain the reference voltage Vd, and the second input terminal of the signal control and acquisition module is connected to the intermediate node B; the signal control and acquisition module is used to make the voltage at the second input terminal of the signal control and acquisition module equal to the reference voltage Vd in steady state through internal negative feedback, and output a voltage signal Vadc, wherein the magnitude of the voltage signal Vadc represents the degree of imbalance between the resistance values ​​of the first sensing resistor R6 and the second sensing resistor R12; The signal control and acquisition module includes a fourth operational amplifier U2.1; the fourth operational amplifier U2.1 is configured as a voltage follower, the non-inverting input of the fourth operational amplifier U2.1 is connected to the output D of the virtual bridge arm module, the output of the fourth operational amplifier U2.1 is used to output the buffered reference voltage Vd, and the inverting input of the fourth operational amplifier U2.1 is connected to the output of the fourth operational amplifier U2.

1. The signal control and acquisition module also includes a fifth operational amplifier U2.2 and a sixth operational amplifier U2.3; a ninth resistor R9 is connected between the output terminal of the fifth operational amplifier U2.2 and the inverting input terminal of the sixth operational amplifier U2.3; the output terminal of the sixth operational amplifier U2.3 outputs the voltage signal Vadc.

2. The thermal flow meter sensor driving circuit according to claim 1, characterized in that, The constant current source module includes a first operational amplifier U1.1, a first MOS transistor Q1, a first sampling resistor R1, a second sampling resistor R2, a first frequency compensation capacitor C1, and a second filter capacitor C2; The drain of the first MOS transistor Q1 is connected to the power module through the first sampling resistor R1, and the source of the first MOS transistor Q1 is connected to the first terminal A to provide the excitation current I1 to the series branch. The output terminal of the first operational amplifier U1.1 is connected to the gate of the first MOS transistor Q1. The non-inverting input terminal of the first operational amplifier U1.1 receives the reference voltage VREF. The inverting input terminal of the first operational amplifier U1.1 is grounded through a second resistor R2 in series. The first frequency compensation capacitor C1 is connected between the output terminal and the inverting input terminal of the first operational amplifier U1.1, and the second filter capacitor C2 is connected between the non-inverting input terminal and ground of the first operational amplifier U1.

1.

3. The thermal flow meter sensor driving circuit according to claim 1, characterized in that, The virtual bridge arm module includes a summing and scaling circuit composed of operational amplifiers, which is used to sum and scale the voltages across the series branch to generate the reference voltage Vd.

4. The thermal flow meter sensor driving circuit according to claim 3, characterized in that, The virtual bridge arm module includes a second operational amplifier U1.2, a third operational amplifier U1.3, and a fourth operational amplifier U1.4; The second operational amplifier U1.2 and the third operational amplifier U1.3 are both configured as voltage followers. The non-inverting input of the second operational amplifier U1.2 is connected to the first terminal A, and the output of the second operational amplifier U1.2 is connected to the inverting input of the second operational amplifier U1.

2. The non-inverting input of the third operational amplifier U1.3 is connected to the second terminal C, and the output of the third operational amplifier U1.3 is connected to the inverting input of the third operational amplifier U1.

3. The fourth operational amplifier U1.4 is configured as a non-inverting summing circuit. The non-inverting input terminal of the fourth operational amplifier U1.4 receives the output voltage of the second operational amplifier U1.2 through a fifth resistor R5 in series. The non-inverting input terminal of the fourth operational amplifier U1.4 receives the output voltage of the third operational amplifier U1.3 through an eleventh resistor R11 in series. At the same time, the inverting input terminal of the fourth operational amplifier U1.4 is connected to the output terminal of the fourth operational amplifier U1.4 through a fourth resistor R4, which serves as a feedback resistor. The output terminal of the fourth operational amplifier U1.4 outputs the reference voltage Vd as the output node D.

5. The thermal flow meter sensor driving circuit according to claim 4, characterized in that, The fifth resistor R5 and the eleventh resistor R11 have the same resistance value; thus, the reference voltage Vd is half the sum of the voltage at the first terminal A and the voltage at the second terminal C.

6. The thermal flow meter sensor driving circuit according to claim 1, characterized in that, The non-inverting input of the fifth operational amplifier U2.2 is connected to the output of the fourth operational amplifier U2.1 to receive the buffered reference voltage Vd; the inverting input of the fifth operational amplifier U2.2 is connected to the intermediate node B through the eighth resistor R8. At the same time, along the signal transmission direction, the two ends of the eighth resistor R8 are connected to the output of the fifth operational amplifier U2.2 through the third resistor R3 and the fourth capacitor in series, respectively. The non-inverting input of the sixth operational amplifier U2.3 is connected to the output of the fourth operational amplifier U2.1 to receive the buffered reference voltage Vd; the inverting input of the sixth operational amplifier U2.3 is connected to the output of the fifth operational amplifier U2.2 through a series ninth resistor R9; at the same time, the inverting input of the sixth operational amplifier U2.3 is connected to the output of the sixth operational amplifier U2.3 through a parallel link consisting of a seventh resistor R7 and a third capacitor.

7. The thermal flow meter sensor driving circuit according to claim 6, characterized in that, The signal control and acquisition module further includes a tenth resistor R10, a thirteenth resistor R13, a fifth capacitor C5, a sixth capacitor C6, and a seventh capacitor C7. The output of the sixth operational amplifier U2.3 is connected in series with the tenth resistor R10 to form a negative voltage signal output terminal ADC-. The output of the fourth operational amplifier U2.1 is connected in series with the thirteenth resistor R13 to form a positive voltage signal output terminal ADC+. The fifth capacitor C5 and the seventh capacitor C7 are connected in series between the positive output terminal ADC+ and the negative output terminal ADC-, and the series connection point of the fifth capacitor C5 and the seventh capacitor C7 is grounded. The two ends of the sixth capacitor C6 are respectively located between the positive output terminal ADC+ and the negative output terminal ADC-.

8. A thermal flow meter, comprising a flow sensor and a signal processing circuit, wherein the flow sensor includes a first sensing resistor R6 and a second sensing resistor R12, characterized in that, The signal processing circuit is the thermal flow meter sensor driving circuit as described in any one of claims 1 to 7.

9. A method for measuring flow rate of a thermal flow meter based on the thermal flow meter sensor driving circuit according to any one of claims 1 to 7, characterized in that, Includes the following steps: A constant current I1 is supplied to the first sensing resistor R6 and the second sensing resistor R12 of the sensor through a constant current source module; A fixed reference voltage Vd is generated by the virtual bridge arm module. The reference voltage Vd is used to balance the system when the resistance values ​​of the first sensing resistor R6 and the second sensing resistor R12 are equal. The signal control and acquisition module forces the voltage at the intermediate node B of the first sensing resistor R6 and the second sensing resistor R12 to follow the reference voltage Vd. Acquire and measure the voltage signal Vadc output by the signal control and acquisition module to maintain the voltage following relationship; The flow rate value flowing through the flow sensor is calculated based on the voltage signal Vadc.

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