Switching amplifier

Abstract

This switching amplifier is provided with a push-pull structure of a narrow-band filter on the output side, and is provided with: a partial resonance circuit that suppresses a first surge voltage due to a square wave; and an LCR series resonator that suppresses a second surge voltage caused by a harmonic frequency component, the waveform shape of the drain-source voltage Vds of the switching element being set to a trapezoidal waveform. As a result, surge voltages of a first surge voltage due to a square wave and a second surge voltage due to a harmonic frequency component are suppressed, the frequency characteristic of the output of the switching amplifier is broadened, and the output frequency is varied.

Description

Technical Field

[0001] This invention relates to switching amplifiers with switching models such as D-stage, E-stage, F-stage, reverse F-stage, and EF-stage, and particularly to switching amplifiers suitable for variable frequency harmonic power supplies with wide-bandgap output frequency adjustment. Background Technology

[0002] A switching amplifier generates a square wave through switching action. By modulating the pulse width of the generated square wave according to the input signal level, the output is amplified. As a circuit configuration for a switching amplifier, it increases the output current by inverting the operation of two switching elements, alternately performing current outflow and current inflow actions. Therefore, a large-capacity output can be obtained by applying a push-pull circuit.

[0003] In the switching amplifier of the D-class amplifier involved in push-pull, the drain-source voltage waveform generated by the switching action is square wave. The drain-source voltage waveform is distorted (surge) in the square wave waveform due to leakage inductance. In order to make the drain-source voltage waveform square wave, it is known to increase the coupling of the output transformer to reduce leakage inductance (Non-Patent Document 1).

[0004] Existing technical documents

[0005] Non-patent literature

[0006] Non-Patent Literature 1: Microsemi PPG Application Note1812 / 13.56 MHz, Class DPush-Pull, 2KW RF Generator with Microsemi DRF1300 Power MOSFET Hybrid / By GuiChoi 2011 September Summary of the Invention

[0007] The problem that the invention aims to solve

[0008] In a switching amplifier, during the switching action that generates a square wave, a surge voltage (hereinafter referred to as the first surge voltage) can be generated due to the leakage inductance of the output transformer, etc.

[0009] Furthermore, the output waveform of the output transformer in a switching amplifier contains harmonic frequency components in addition to the fundamental frequency components. As a harmonic countermeasure to remove these harmonic frequency components from the output, a narrowband filter such as an LPF can be installed on the secondary side of the output transformer. The narrowband filter allows the fundamental frequency components to pass through while blocking the harmonic frequency components. These harmonic frequency components are reflected back to the switching element side by the narrowband filter. The returned harmonic frequency components act as a surge voltage (hereinafter referred to as the second surge voltage) on the switching element. In VHF band switching amplifiers using switching elements with bandwidths up to several hundred MHz, such as LDMOS, the problem of the second surge voltage becomes more pronounced due to the fact that the frequency of the harmonic frequency components reflected back by the narrowband filter is also within the switching element's frequency band.

[0010] Figure 14 is a diagram illustrating the second surge voltage involved in harmonic frequency components. In the circuit example of a switching amplifier in a push-pull D-class amplifier, Figure 14A This represents a circuit example that does not have a narrowband filter. Figure 14B This represents a circuit example equipped with a narrowband filter.

[0011] Figure 14A The switching amplifier 101a shown includes: a switching section 111 with two switching elements 111A and 111B; an output transformer 112 connected between the two drains of the switching elements 111A and 111B; and a DC power supply 113 connected to the center tap of the primary winding 112a of the output transformer 112. The switching amplifier 101a uses the inverted gate signals Pg from the signal sources 110A and 110B to cause the switching elements 111A and 111B to switch alternately, thereby forming a push-pull circuit 114. Current is transmitted through the push-pull circuit 114 to the secondary winding 112b of the output transformer 112. The transmitted current is supplied to the load 117 connected to the output terminal 116.

[0012] exist Figure 14A In the switching amplifier 101a, the drain-source voltages VdsA and VdsB of the switching elements 111A and 111B are switched, and the current Inf0, which is a square wave and a harmonic frequency component, is transferred to the load 117 and consumed. Because the output waveform is square, this configuration is unsuitable for use as a harmonic power supply in semiconductor manufacturing devices that require a sinusoidal output waveform.

[0013] on the other hand, Figure 14BThe switching amplifier 101b sets the output waveform to a sine wave by providing a narrowband filter 115 on the secondary side of the output transformer 112. Although the structure with this narrowband filter 115 can obtain a sinusoidal output voltage, it cannot maintain the drain-source voltages VdsA and VdsB in a square wave waveform. The inventors of this application discovered that the main reason for the inability to maintain a square wave waveform is that the current Inf0 containing the harmonic frequency components of the drain-source voltages VdsA and VdsB is reflected back to the switching elements 111A and 111B by the narrowband filter 115.

[0014] In VHF band amplifiers, wide-bandwidth switching elements, such as LDMOS, are used to achieve wide frequency response. Therefore, in wideband switching amplifiers with broad output frequency characteristics, it's impossible to actively incorporate narrowband filters on the secondary side of the output transformer to prevent harmonic frequency components from reflecting back. Consequently, there's a problem of not being able to simultaneously achieve both wide bandwidth and sinusoidal output waveform.

[0015] Therefore, in a VHF band amplifier, during switching operation, there are two surge voltages: a first surge voltage caused by the leakage inductance of the output transformer, etc., and a second surge voltage caused by the harmonic frequency components reflected by the narrowband filter. These surge voltages become the main reason for the broadbanding of the output frequency of the harmonic power supply using the switching amplifier.

[0016] To broadband the frequency response of the switching amplifier's output and obtain a stable output even when the output frequency is set to be variable, the drain-source voltage Vds of the switching element must be a trapezoidal waveform that suppresses the first surge voltage caused by the square wave and the second surge voltage caused by the harmonic frequency components.

[0017] The present invention addresses the aforementioned problems and aims to suppress surge voltages caused by a first surge voltage due to a square wave and a second surge voltage due to harmonic frequency components, thereby widening the frequency characteristics of the output of the switching amplifier and making the output frequency variable.

[0018] Methods for solving problems

[0019] The switching amplifier involved in this invention is a push-pull switching amplifier with a narrowband filter on the output side. The switching amplifier includes: a partial resonant circuit that suppresses a first surge voltage caused by a square wave; and an LCR series resonator that suppresses a second surge voltage caused by harmonic frequency components. The waveform shape of the drain-source voltage Vds of the switching element is set to a trapezoidal waveform, thereby widening the frequency response of the output of the switching amplifier and making the output frequency variable.

[0020] (Drain-source voltage Vds with a trapezoidal waveform)

[0021] The trapezoidal waveform has a sloping shape during voltage rise and fall, within the interval requiring a specified time width, and a waveform with a fixed peak value during the interval from voltage rise to fall.

[0022] The tilted waveform suppresses surge voltage or harmonic frequency components of the drain-source voltage Vds of the switching elements generated during voltage rise and fall operations. The waveform with a fixed peak value maintains a constant drain-source voltage Vds even when scanning in a wide bandwidth with a variable output frequency, thus keeping the output gain constant.

[0023] (Structure of a switching amplifier)

[0024] The switching amplifier involved in this invention has the following structural elements:

[0025] (a) A transformer-coupled push-pull circuit, which consists of a pair of switching elements that alternately switch on or off by input signals that are opposite to each other, a primary winding that receives the output signals of the switching elements at both ends, a secondary winding that extracts the output signals, and a transformer that has an external power supply connected to the center tap of the primary winding.

[0026] (b) A narrowband filter connected to the output of the secondary winding of the transformer;

[0027] (c) A pair of partially resonant circuits, which consist of the inductive component on the drain side of each switching element and the capacitive component between the drain and source of each switching element.

[0028] (d) LCR series resonator, which consists of a series circuit structure of inductance, capacitance and resistance connected between the drains of each switching element.

[0029] The switching element of the switching amplifier according to the present invention can be adapted to a MOSFET, and in the case of a power MOSFET, it can be adapted to an LDMOS or VDMOS with a DMOS structure.

[0030] (Push-pull circuit)

[0031] A push-pull circuit alternately turns a pair of switching elements on and off using input signals of opposite phase. Through this alternating switching action, each switching element is alternately connected to an external power supply via the center tap of the primary winding of the transformer, and inductive currents flow alternately in opposite directions into the primary winding. This inductive current is coupled through the transformer to carry the load current in the secondary winding. The inductive current flowing through the primary winding alternately flows in opposite directions. Since the inductive current on the primary winding side is equivalent to alternately carrying out and receiving current on the secondary winding side, the received load current flows through the secondary winding.

[0032] The load current induced by the secondary winding contains fundamental and harmonic frequency components corresponding to the switching frequency of the switching element. Due to the characteristics of the push-pull circuit, the even-order harmonic frequency components are canceled out by each other, thus resulting in only odd-order harmonic frequency components.

[0033] (Narrowband filter)

[0034] A narrowband filter allows the fundamental frequency component of the load current in the secondary winding to pass through while blocking harmonic frequency components. These harmonic frequency components are reflected by the narrowband filter and returned to the switching element side. The harmonic frequency components returned to the switching element become a surge voltage (second surge voltage) relative to the switching element.

[0035] (LCR series resonator)

[0036] An LCR series resonator consists of a series circuit of an inductor, a capacitor, and a resistor connected between the drains of a pair of switching elements. The resonant frequency of the series circuit is set in conjunction with the frequency of the harmonic components of the load current.

[0037] (Parallel connection of LCR series resonators)

[0038] By setting the LCR series resonators to be connected in parallel, the resonant frequency can be varied by multiple series circuits, thus broadening the bandwidth of the harmonic frequency components of the load current.

[0039] The LCR series resonator, which is formed by connecting two series circuits in parallel, has a first series circuit and a second series circuit. The first series circuit has a resonant frequency at a low frequency relative to the fundamental frequency f0 of the high-frequency output from the secondary side of the transformer, and the second series circuit has a resonant frequency at a high frequency relative to the harmonic frequency n×f0 of the high-frequency output.

[0040] When an LCR series resonator is composed of multiple series circuits connected in parallel, the number of series circuits connected in parallel is not limited to two and can be any number.

[0041] (Suppression of the second surge voltage involved in LCR series resonators)

[0042] The harmonic frequency components reflected by the narrowband filter flow to ground via the LCR series resonator and the switching element in the ON state. At this time, the harmonic frequency components are consumed and attenuated by the resistance of the series circuit. This attenuation of the harmonic frequency components suppresses the second surge voltage caused by the harmonic frequency components applied to the switching element. With the second surge voltage suppressed, the peak value variation of the drain-source voltage Vds of the switching element is suppressed and becomes approximately constant.

[0043] (Partial resonant circuit)

[0044] A partial resonant circuit is constructed using the inductive component on the drain side of the switching element and the capacitive component between the drain and source of the switching element. A partial resonant circuit is configured as a pair of separate switching elements.

[0045] The inductive component (Ls) of each resonant circuit is any one or any combination of the series inductance (Ls1) connected in series between the switching element and one end of the transformer, the leakage inductance (Lleak) of the output transformer, and the wiring inductance (Lline) between the switching element and one end of the transformer.

[0046] The capacitance component (Ctotal) of each part of the resonant circuit is either the output parasitic capacitance (Coss) of the switching element or the combined capacitance of the output parasitic capacitance (Coss) of the switching element and the additional capacitance (Cp) connected in parallel to the switching element.

[0047] When the switching element changes from the ON state to the OFF state, the charge of the capacitive component (Ctotal) that was discharged in the ON state is charged through the first resonance of the partial resonance based on the primary current of the transformer. During this charging, the drain-source voltage Vds of the switching element increases through the time constant τr determined by the inductive component (Ls) and the capacitive component (Ctotal).

[0048] On the other hand, when the switching element changes from the off state to the on state, the charge of the capacitive component (Ctotal) that was charged in the off state is discharged through a second resonance based on the partial resonance of the primary current of the transformer. During this discharge, the drain-source voltage Vds of the switching element is reduced by the time constant τf determined by the inductive component (Ls) and the capacitive component (Ctotal).

[0049] During the dead time, the capacitive component (Ctotal) of the first resonance is connected in series with the capacitive components (Ctotal) of two switching elements. On the other hand, since the capacitive component (Ctotal) of the second resonance is only the capacitive component (Ctotal) of one switching element, the time constant τr of the first resonance is smaller than the time constant τf of the second resonance, resulting in a faster voltage rise. Through the first and second resonances, the drain-source voltage Vds becomes a trapezoidal waveform.

[0050] (Suppression of the first surge voltage involved in partial resonant circuits and LCR series resonators)

[0051] By partially resonating the primary current of the transformer, the drain-source voltage Vds of the switching element increases or decreases during the dead time through the time constants τr and τf, thus suppressing the surge voltage that would be generated in the drain-source voltage Vds.

[0052] Furthermore, excluding the dead time, the charging current of the first resonance and the discharging current of the second resonance, based on partial resonance, flow between the two drains via the LCR series resonator. The charging and discharging current flowing through the LCR series resonator is consumed and attenuated by the resistance Rr of the LCR series resonator. During the period excluding the dead time, since the charging and discharging current based on partial resonance increases and decreases through the time constants τr and τf, and is consumed and attenuated by the LCR series resonator, the generation of the first surge voltage in the drain-source voltage Vds is suppressed.

[0053] Therefore, since the first surge voltage caused by the square wave is suppressed by the partial resonant circuit and the LCR series resonator, the waveform of the drain-source voltage Vds becomes a trapezoidal wave.

[0054] (Resonant frequency and bandwidth of a partial resonant circuit)

[0055] The resonant frequency of a pair of partial resonant circuits is in the range from the fundamental frequency f0 of the high-frequency output from the secondary side of the transformer to the third harmonic frequency 3×f0. Through the partial resonant circuit, the impedance of the frequency range where the impedance increases decreases, and at the same time, a specified time constant is set for the rise and fall of the drain-source voltage Vds of the LDMOS.

[0056] (Resonant frequency and bandwidth of LCR series resonator)

[0057] When the fundamental frequency of the high-frequency output from the secondary side of the transformer is set to f0, the resonant frequency of the LCR series resonator is the harmonic frequency n×f0 (n is an integer). The bandwidth of the resonant frequency is set to the frequency range of (n×f0-min~n×f0-max) according to the variable frequency range (f0-min~f0-max) of the fundamental frequency f0.

[0058] (The function of the switching amplifier in this invention)

[0059] The switching amplifier involved in this invention achieves the following secondary effects by suppressing surge voltage, widening the frequency characteristics of the switching amplifier output, and setting the output frequency to be variable.

[0060] • Leakage countermeasures without output transformer

[0061] In existing technical literature, methods have been disclosed for increasing the coupling of the output transformer to reduce leakage inductance in order to obtain a square-wave drain-source voltage waveform. To reduce leakage inductance, a magnetic core is prepared as the material for the output transformer, and a twisted wire winding is wound around this core. This is a general method for increasing coupling to reduce leakage inductance. Furthermore, there are methods for selecting a material with a high relative permeability μ for the core. However, since the core material loss increases proportionally to the relative permeability μ, it is difficult to design it to withstand continuously high output voltages.

[0062] According to the present invention, an amplifier capable of continuously outputting high power can be realized without the need for a core material and twisted wire structure with relatively high permeability as a leakage countermeasure for the output transformer.

[0063] Generally speaking, even when using a coreless air-core output transformer with high leakage inductance and a planar transformer with poor coupling in the winding structure, a D-class push-pull amplifier with fewer output harmonics can be achieved by inserting an LPF (low-pass filter) on the secondary side of the transformer.

[0064] • Since the peak value of the drain-source voltage Vds can be made approximately the same across the entire broadband range, the power gain can be fixed.

[0065] When using this invention, since the waveform of the drain-source voltage Vds always has the same peak value, output fluctuations can be suppressed, and power gain fluctuations caused by significant frequency variations in AB or C-class amplifiers can be suppressed, thereby achieving a flat output characteristic.

[0066] • Even with increased leakage inductance, a stable trapezoidal Vds waveform can be obtained, thus enabling broadband operation.

[0067] • By inserting an LPF (low-pass filter) on the secondary side of the output transformer, output harmonics can be reduced.

[0068] • Suppressing surge voltage caused by partial resonance can be achieved by ensuring that the drain-source voltage Vds of the switch does not exceed the withstand voltage of the switching element.

[0069] Invention Effects

[0070] As described above, according to the present invention, surge voltages caused by a first surge voltage due to a square wave and a second surge voltage due to harmonic frequency components can be suppressed, and the frequency characteristics of the output of the switching amplifier can be widened so that the output frequency becomes variable. Attached Figure Description

[0071] Figure 1 This is a diagram illustrating the first structure of the switching amplifier of the present invention.

[0072] Figure 2 This is a diagram illustrating the second structure of the switching amplifier of the present invention.

[0073] Figure 3A This is a graph used to compare the transmission characteristics of LCR series resonators and LC series resonators.

[0074] Figure 3B This is a diagram of the equivalent circuits used to compare LCR series resonators and LC series resonators.

[0075] Figure 4 This is a diagram illustrating the signals of various parts of the switching amplifier of the present invention.

[0076] Figure 5A This diagram illustrates an example of operation in each section of the switching amplifier of the present invention.

[0077] Figure 5B This diagram illustrates an example of operation in each section of the switching amplifier of the present invention.

[0078] Figure 6A This diagram illustrates an example of operation in each section of the switching amplifier of the present invention.

[0079] Figure 6B This diagram illustrates an example of operation in each section of the switching amplifier of the present invention.

[0080] Figure 7A This diagram illustrates an example of operation in each section of the switching amplifier of the present invention.

[0081] Figure 7B This diagram illustrates an example of operation in each section of the switching amplifier of the present invention.

[0082] Figure 8A This diagram illustrates an example of operation in each section of the switching amplifier of the present invention.

[0083] Figure 8B This diagram illustrates an example of operation in each section of the switching amplifier of the present invention.

[0084] Figure 9 This is a diagram illustrating the trapezoidal waveform of the drain-source voltage of a switching element.

[0085] Figure 10 The diagram shows an example waveform when the fundamental frequency f0 is 45MHz.

[0086] Figure 11 The diagram shows an example waveform when the fundamental frequency f0 is 35MHz.

[0087] Figure 12A This is a diagram used to illustrate the broadbanding of the output involved in an LCR series resonator.

[0088] Figure 12B This is a diagram used to illustrate the broadbanding of the output involved in an LCR series resonator.

[0089] Figure 13A This is a Smith chart used to illustrate the relationship between the frequency band of partial resonance and the frequency band of high impedance.

[0090] Figure 13B This is a frequency response diagram used to illustrate the relationship between the frequency band of partial resonance and the frequency band of high impedance.

[0091] Figure 14A This is a diagram used to illustrate the second surge voltage involved in harmonic frequency components.

[0092] Figure 14B This is a diagram used to illustrate the second surge voltage involved in harmonic frequency components. Detailed Implementation

[0093] (1) Simplified structure of a switching amplifier

[0094] Hereinafter, a schematic structure of the switching amplifier of the present invention will be described using... Figure 1 , Figure 2 Let me explain.

[0095] Figure 1 This represents a first structural example with an LCR series resonator. Figure 2 This represents a second structural example with two LCR series resonators connected in parallel.

[0096] (1-1: First structural example)

[0097] In addition to the structure consisting of an external power supply composed of a switch section 11, an output transformer 12, a DC power supply 13, and a switch amplifier with a narrowband filter 15, the first example of the switching amplifier also includes an LCR series resonator 20 and partial resonant circuits 21 (21A, 21B).

[0098] The switching unit 11 has two switching elements. A MOSFET is used as the switching element, and a power MOSFET is used for high current applications. A DMOS such as an LDMOS or VDMOS is used as the power MOSFET. Here, an LDOMOS is used as an example of the switching element, and LDMOS (11A) and LDMOS (11B) are used to represent the two switching elements of the switching unit 11.

[0099] The gate of LDMOS (11A) receives a gate signal Pg from signal source 10A, switching the on / off state between the source terminal (drain-grounded) and the gate terminal. Similarly, the gate of LDMOS (11B) receives a gate signal Pg from signal source 10B, switching the on / off state between the source terminal (drain-grounded) and the gate terminal. Because the gate signals Pg from signal sources 10A and 10B are inverted, the outputs of LDMOS (11A) and LDMOS (11B) are out of phase.

[0100] An output transformer 12 is connected between the drain terminals of LDMOS (11A) and LDMOS (11B). A DC power supply 13 is connected to the center tap of the output transformer 12. By setting the outputs of LDMOS (11A) and LDMOS (11B) to be inverted, a push-pull circuit 14 is formed. Through the push-pull circuit 14, the secondary winding 12b of the output transformer 12 alternately performs the operation of current outflow and current inflow, transmitting the load current as the output current.

[0101] A narrowband filter 15, such as an LPF, is installed between the secondary winding 12b of the output transformer 12 and the output terminal 16. The narrowband filter 15 blocks the harmonic frequency n×f0 contained in the load current of the output current, allowing the fundamental frequency f0 to pass through the output terminal 16.

[0102] In the push-pull circuit 14, since the even-order harmonic frequency components cancel each other out, the harmonic frequency n×f0 flowing to the secondary winding 12b becomes the odd-order harmonic frequency. Moreover, the higher the order, the smaller the peak value of the harmonic frequency. Therefore, in the design of the narrowband filter 15, since it has the frequency characteristic of blocking the third harmonic frequency 3×f0, which is the harmonic frequency n×f0, the harmonic frequency components can be effectively blocked.

[0103] Regarding the switching amplifier involved in this invention, as a structure for suppressing the second surge voltage caused by the harmonic frequency components contained in the load current of the output current, it is characterized by having an LCR series resonator 20. As a structure for suppressing the first surge voltage caused by the high frequency components contained in the output voltage of the LDMOS, i.e., the drain-source voltage Vds, which is a switching element, it is characterized by having an LCR series resonator 20 and partial resonant circuits 21 (21A, 21B).

[0104] (LCR series resonator)

[0105] The LCR series resonator 20 is configured as a series circuit of inductor Lr, capacitor Cr and resistor Rr, and is connected between the drain terminals of LDMOS (11A) and LDMOS (11B), and is also connected in parallel to the primary winding 12a of the output transformer 12.

[0106] The resonant frequency of the series circuit of the LCR series resonator 20 is set in accordance with the frequency of the harmonic frequency components of the load current.

[0107] In a push-pull circuit, the harmonic frequency components of the load current are odd-order harmonics. Since the peak values ​​of harmonics above the fifth order are smaller than those of the third harmonic, the resonant frequency of the series circuit is approximately three times the fundamental frequency f0 and its bandwidth [f0-min, f0-max]. Furthermore, considering the deviation of the resonant constant, it is set to a frequency approximately two to four times the fundamental frequency f0. Additionally, f0-min is the lower limit frequency of the load current, and f0-max is the upper limit frequency of the load current.

[0108] The resistor Rr, which is part of the series circuit, achieves sufficient attenuation at the center frequency of the third harmonic. To achieve a wide attenuation effect across the entire bandwidth, it is set to, for example, 0.5 Ω [ohm] to 15 Ω [ohm]. Furthermore, the resistor Rr is set such that its resistance value viewed from the primary side of the output transformer is similar to the load impedance of the fundamental frequency.

[0109] When the turns ratio of the output transformer is set to m:n, the primary side resistance is set to R1, and the secondary side resistance is set to R2, then R1 = (m / n). 2 The relationship is ×R2. In this relationship, for example, when m=1, n=2, and R2=50 [ohm], since the primary side resistance R1 becomes 12.5 [ohm], 15 [ohm] is set as the upper limit. Furthermore, as the fundamental frequency increases, the load impedance suitable for LDMOS characteristics decreases, so 0.5 [ohm] is set as the lower limit.

[0110] The LCR series resonator 20 attenuates the harmonic frequency components reflected back to the LDMOS by the narrowband filter. Therefore, since the peak value of the LDMOS drain-source voltage is set to a fixed value, the output gain characteristics can be kept constant even when the frequency of the load current fluctuates in the broadband, resulting in a stable output in the broadband.

[0111] (Partial resonant circuit)

[0112] The partial resonant circuit 21 has a partial resonant circuit 21A on the LDMOS (11A) side and a partial resonant circuit 21B on the LDMOS (11B) side.

[0113] The partial resonant circuit 21A is formed by connecting the inductive component Ls-A on the drain side of the LDMOS (11A) and the capacitive component CA of the LDMOS (11A) in series.

[0114] The inductance component Ls-A is any one or any combination of the following: the series inductance Ls1-A connected in series between the LDMOS (11A) and one end of the output transformer 12; the leakage inductance Lleak of the output transformer 12; and the wiring inductance Lline-A between the LDMOS (11A) and one end of the output transformer 12. The series inductance Ls1-A is connected as needed, corresponding to the resonant frequency of the partial resonant circuit 21A.

[0115] The capacitance component CA of the LDMOS(11A) uses the parasitic capacitance coss-A of the LDMOS(11A). In addition, the total capacitance Ctotal-A can be formed by the additional capacitance Cp-A connected in parallel between the drain and source terminals of the LDMOS(11A) and the parasitic capacitance Coss-A.

[0116] Similar to the partial resonant circuit 21A, the partial resonant circuit 21B is composed of a series circuit of the series inductance component Ls-B on the drain side of the LDMOS (11B) and the capacitance component CB of the LDMOS (11B).

[0117] The inductance component Ls-B is any one or any combination of the following: the series inductance Ls1-B connected in series between the LDMOS (11B) and the other end of the output transformer 12; the leakage inductance Lleak of the output transformer 12; and the wiring inductance Lline-B between the LDMOS (11B) and the other end of the output transformer 12. The series inductance Ls1-B is connected as needed, corresponding to the resonant frequency of the partial resonant circuit 21B.

[0118] The capacitance component CB of the LDMOS(11B) uses the parasitic capacitance Coss-B of the LDMOS(11B). Alternatively, it can use the combined capacitance Ctotal-B, formed by an additional capacitance Cp-B connected in parallel between the drain and source terminals of the LDMOS(11B) and the parasitic capacitance Coss-B. The additional capacitances Cp-A and Cp-B can also be either lumped constants or distributed constants.

[0119] The resonant frequencies of the partial resonant circuits 21A and 21B are in the range from the fundamental frequency f0 of the high-frequency output from the secondary side of the output transformer 12 to the third harmonic frequency 3×f0. Through the partial resonant circuits, the impedance of the frequency range where the impedance is high is reduced, and at the same time, a specific time constant is set for the rise and fall of the drain-source voltage Vds of the LDMOS.

[0120] The partial resonant circuit is tilted in such a way that the rise and fall of the drain-source voltage Vds form a trapezoidal waveform with a specified time width, suppressing high-frequency components generated during the rise and fall. At this time, the partial resonant circuits 21A and 21B partially resonate near the frequency region where the impedance between the drains is high, thereby reducing the impedance.

[0121] (Series resonator and partial resonant circuit)

[0122] The resonant frequency of a series resonator and a partial resonant circuit differs from that of a resonant suppression circuit. The resonant frequency of a series resonator is used to suppress surge voltage by reducing the harmonic frequency components contained in the high-frequency output. On the other hand, the resonant frequency of a partial resonant circuit is determined by setting the drain-source voltage Vds to have a trapezoidal waveform with sloping rise and fall characteristics, thereby reducing impedance and suppressing surge voltage at frequencies that are considered high impedance.

[0123] When the series resonator resonates, the impedance becomes only the resistance Rr. This resistance Rr, along with the inductive component Ls and the capacitive component C of part of the resonant circuit, constitute a parallel damping circuit. The attenuation rate ζ of this parallel damping circuit is ζ = (1 / (2×Rr))×(Ls / C). The smaller the resistance Rr, the larger the attenuation rate ζ, thereby increasing the damping effect.

[0124] The resonant frequency of the series resonator is set as the bandwidth of the harmonic frequency [n×f0min~n×f0max] in conjunction with the variable frequency range [f0min~f0max] that allows the fundamental frequency f0 to be varied. Therefore, when the frequency of the high-frequency output is changed, multiple resonant frequencies will be set according to the changed frequency.

[0125] On the other hand, the resonant frequency of a partial resonant circuit is set such that the drain-source voltage Vds forms a trapezoidal waveform with a sloping portion, within a frequency range near the frequency where high impedance occurs. The inductive component Ls and the capacitive component C are set according to the set resonant frequency. Even if the high-frequency output frequency changes, the frequency range for setting the resonant frequency will not change significantly. Therefore, even if the high-frequency output frequency changes, it is not necessary to change the resonant frequency according to the changed frequency; setting only a single resonant frequency is sufficient.

[0126] (1-2: Second structural example)

[0127] The second structural example of the switching amplifier 1 is the same as the first structural example, and includes a switching section 11, an output transformer 12, a DC power supply 13, a narrowband filter 15, an LCR series resonator 20, and partial resonant circuits 21 (21A, 21B).

[0128] Regarding the switching amplifier 2 of the second structural example, two LCR series resonators 20A and 20B are connected in parallel between the drain terminals of LDMOS (11A) and LDMOS (11B) and the primary winding 12a of the output transformer 12.

[0129] The first LCR series resonator 20A and the second LCR series resonator 20B have the same series circuit as the LCR series resonator 20 in the first configuration example. The first LCR series resonator 20A is composed of a series circuit of inductor Lr1, capacitor Cr1 and resistor Rr1, and the second LCR series resonator 20B is composed of a series circuit of inductor Lr2, capacitor Cr2 and resistor Rr2.

[0130] The first LCR series resonator 20A and the second LCR series resonator 20B are set by making the resonant frequencies of the series circuit different, thereby widening the frequency band that attenuates the harmonic frequency components of the load current.

[0131] The first LCR series resonator 20A has a resonant frequency at a lower frequency than the harmonic frequency n×f0 of the high-frequency output, which is the fundamental frequency f0 of the high-frequency output from the secondary side of the transformer. The second LCR series resonator 20B has a resonant frequency at a higher frequency than the harmonic frequency n×f0 of the high-frequency output. By shifting the resonant frequencies of the parallel-connected LCR series resonators, the bandwidth of the attenuated harmonic frequency is broadened.

[0132] Here, although a structure with two LCR series resonators connected in parallel is shown, the number of LCR series resonators connected in parallel is not limited to two and can be any number. The number of LCR series resonators connected in parallel and the frequency values ​​of each resonant frequency are set according to the bandwidth of the output harmonic frequencies. The suppression of harmonic frequencies related to the LCR series resonator 20 will be described later.

[0133] (1-3: Comparison of LCR series resonator and LC series resonator)

[0134] Regarding the surge voltage suppression effect of LCR series resonators, a comparison is made with that of LC series resonators. Figure 3A This indicates the attenuation characteristics of a structure with two LC series resonators connected in parallel. Figure 3A The attenuation characteristics are represented by Figure 3B The equivalent circuit shown represents the characteristics of a structure with two LCR series resonators connected in parallel.

[0135] Figure 3A The solid lines in the diagram represent the transmission characteristics of an LCR series resonator, while the dashed lines represent the transmission characteristics of an LC series resonator; both are set to resonate at the same frequency. Figure 3A The diagram shows the characteristics represented by the attenuation rate with respect to frequency; the further down the vertical axis, the greater the attenuation rate.

[0136] Because the drain-source voltage Vds contains odd-order harmonics, the resonant frequency of the LCR series resonant wave or LCR series resonator is designed to resonate at a frequency that is approximately 2 to 4 times the frequency of the fundamental frequency f0 and its bandwidth (f0-min to f0-max), which is about 3 times the fundamental frequency f0. Figure 3A In this context, the attenuation frequency range is set such that the impedance of the series resonator is considered to be below 50Ω [ohm], and the attenuation rate is below -6dB when the amplitude becomes 1 / 2.

[0137] By using two series resonators with different frequency ranges, the frequency range of the resonant frequency can be widened. For widening this frequency range, an LC series resonator is unsuitable; an LCR series resonator is preferred.

[0138] according to Figure 3A The dashed line illustrates the conduction characteristics of an LR series resonator without resistance Rr, exhibiting no attenuation effect in frequencies containing the third harmonic ((2~4)×f0). In such frequency bands where attenuation is insufficient, frequency components of the band appear in the drain-source voltage Vds, becoming a major cause of surge voltage. When the surge voltage exceeds the withstand voltage of components such as LDMOS, it also becomes a primary reason for component failure.

[0139] On the other hand, according to Figure 3A The solid line shows the transmission characteristics of an LCR series resonator with resistance Rr. It has a sufficient attenuation rate even in frequencies containing the third harmonic ((2~4)×f0), and can have an attenuation effect over a wide frequency range.

[0140] (2) Example of the operation of a switching amplifier

[0141] Regarding the operating example of the switching amplifier of the present invention, using Figure 4 Figure 8 will be used for illustration. Figure 4 The signals represent the various parts of the switching amplifier. Figures 5 to 8 show examples of operation in each section of the switching amplifier.

[0142] Figure 4 A represents the gate-source voltage VgsA of the LDMOS (11A). Figure 4 B represents the gate-source voltage VgsB of the LDMOS (11B). Figure 4 C represents the drain-source voltage VdsA of the LDMOS (11A). Figure 4 D represents the drain-source voltage VdsB of the LDMOS (11B). Figure 4 E represents the current IdsA flowing from the LDMOS (11A) into the primary winding 12a of the output transformer 12, and the current IdsB flowing from the LDMOS (11B) into the primary winding 12a of the output transformer 12. Furthermore, current IdsA is represented by a solid line, and current IdsB by a dashed line.

[0143] Regarding LDMOS (11A) and LDMOS (11B), the gate-source voltages VgsA and VgsB are switched on and off with a critical voltage boundary. This is achieved by switching the gate-source voltages VgsA and VgsB in opposite phases to each other, thus performing a push-pull operation. To avoid the period where both LDMOS (11A) and (11B) constituting the push-pull circuit are simultaneously on, a dead time interval is provided where the gate-source voltages of both LDMOS (11A) and (11B) become 0V and both LDMOS (11A) and (11B) become off during the switching of their respective gate-source voltages.

[0144] The following explanation will divide the operation examples of LDMOS (11A) and LDMOS (11B) into 8 intervals, T1 to T8.

[0145] Intervals T1 and T5 are the dead time intervals when both LDMOS (11A) and (11B) are in the off state. Intervals T2, T3, and T4 are the intervals when LDMOS (11A) is in the off state and LDMOS (11B) is in the on state. Intervals T6, T7, and T8 are the intervals when LDMOS (11A) is in the off state and LDMOS (11B) is in the on state.

[0146] Intervals T1 and T2 are migration intervals where the drain-source voltage VdsA of LDMOS(11A) rises when LDMOS(11A) is turned off. Intervals T5 and T6 are migration intervals where the drain-source voltage VdsB of LDMOS(11B) rises when LDMOS(11B) is turned off.

[0147] Intervals T4 and T5 are migration intervals in which the drain-source voltage VdsA of LDMOS (11A) decreases due to the resonance of LDMOS (11A) and the turn-off of LDMOS (11B). Intervals T8 and T1 are migration intervals in which the drain-source voltage VdsB of LDMOS (11B) decreases due to the resonance of LDMOS (11B) and the turn-off of LDMOS (11A).

[0148] (Interval T1)

[0149] Interval T1 is the dead time interval during which both LDMOS (11A) and LDMOS (11B) are in the off state. Additionally, interval T1 is the time interval during which LDMOS (11A) changes from the on / off state to the off state.

[0150] use Figure 5A Explain the operating state of interval T1. When the LDMOS (11A) is turned off, in the partial resonant circuit 21A on the LDMOS (11A) side, the current IdsA flows towards the LDMOS (11A) side due to the energy of the primary winding 12a. The capacitive component CA of the partial resonant circuit 21A is charged by the current IdsA. As the capacitive component CA is charged, the drain-source voltage VdsA rises by the time constant τr determined by the LC constant of the capacitive component CA and the inductive component Ls-A.

[0151] By increasing the drain-source voltage VdsA over a time constant τr, the rise time of VdsA can be delayed. This delay reduces harmonic frequency components caused by the rapid rise of VdsA, thus suppressing surge voltage at the rise time.

[0152] At this point, the capacitive component CA of partial resonant circuit 21A and the capacitive component CB of partial resonant circuit 21B are connected in series. Through this series connection, the LC constant becomes smaller, and the rise time tr of the drain-source voltage VdsA is slightly shorter than the fall time tf.

[0153] On the other hand, in the partial resonant circuit 21B on the LDMOS (11B) side, the current IdsA of the partial resonant circuit 21A is attracted, and the current IdsB flows out from the LDMOS (11B) side toward the DC power supply 13 side. Through the outflow of this current IdsB, the charge stored in the capacitance component CB of the partial resonant circuit 21B is discharged.

[0154] As the capacitor component CB is discharged, the drain-source voltage VdsB decreases toward 0V through the time constant τf determined by the LC constant of the capacitor component CA and the series inductance component Ls1-B.

[0155] (Interval T2)

[0156] Interval T2 is the interval in which LDMOS(11A) is in the off state and LDMOS(11B) is in the on state.

[0157] use Figure 5B Explain the operating state of interval T2. The LDMOS (11A) is in the off state. Through the resonance of the partial resonant circuit 21A on the LDMOS (11A) side, the charging voltage of the drain-source voltage VdsA increases through the time constant τr, while the current IdsA decreases.

[0158] Similar to interval T1, even in interval T2, the drain-source voltage VdsA rises through the time constant τr, thereby delaying the rise of VdsA. This reduces the high-frequency components generated by the rapid rise of VdsA, suppressing the surge voltage at the rise moment. Through intervals T1 and T2, the rising waveform of the drain-source voltage VdsA becomes a trapezoidal waveform.

[0159] Because the LDMOS (11B) is turned on, the drain-source voltage VdsB is maintained at 0V. The current IdsB of the partial resonant circuit 21B switches from the current direction of interval T1 and flows from the DC power supply 13 toward the ground side.

[0160] (Interval T3)

[0161] Interval T3 is the interval in which LDMOS(11A) is in the off state and LDMOS(11B) is in the on state.

[0162] use Figure 6AExplain the operating state of interval T3. Since LDMOS (11B) is in the ON state, current IdsB begins to flow towards LDMOS (11B). However, since LDMOS (11A) is in the OFF state, the charge of the capacitance component CA of LDMOS (11A) is maintained in the charging state, and the peak value of the drain-source voltage VdsA is maintained at approximately a fixed value.

[0163] Subsequently, on the LDMOS (11A) side, current IdsA is attracted by current IdsB and flows from LDMOS (11A) towards DC power supply 13. The drain-source voltage VdsA, together with the resonance based on the partial resonant circuit 21A, oscillates through the resonance of the inductive component of the primary winding 12a of the output transformer 12. During the middle of the resonant oscillation according to interval T3, the peak value of the drain-source voltage VdsA is approximately fixed, and the surge voltage generated by the rapid change in load current is suppressed by the partial resonance.

[0164] In the later part of interval T3, the charge of the capacitive component CA of the LDMOS (11A) becomes discharged, and the peak value of the drain-source voltage VdsA tends to decrease.

[0165] In the secondary winding 12b of the output transformer 12, current IdsB flows to the primary winding 12a, thereby flowing through the load current Iload accompanied by harmonic frequency components. The load current Iload flows from the output terminal 16 to the load 17 through the narrowband filter 15. On the other hand, the harmonic frequency components are reflected by the narrowband filter 15, flow through the output transformer 12 to the LCR series resonator 20, and then to the LDMOS (11B) in the ON state. At this time, the resistance Rr of the LCR series resonator 20 is reduced due to the harmonic frequency components, and the surge voltage generated by the harmonic frequency components is suppressed.

[0166] (Interval T4)

[0167] Similar to interval T3, interval T4 is the interval where LDMOS (11A) is in the off state and LDMOS (11B) is in the on state.

[0168] use Figure 6B The operating state of interval T4 is explained. In interval T4, the drain-source voltage VdsA, following the later part of interval T3, shows a decreasing tendency due to the resonance of the inductive component of the primary winding 12a of the output transformer 12, and the surge voltage is suppressed by the resonance of the partial resonant circuit 21A.

[0169] (Interval T5)

[0170] Interval T5 is the dead time interval during which both LDMOS (11A) and LDMOS (11B) are in the off state, and LDMOS (11B) changes from the on-off state to the off state.

[0171] use Figure 7A Explain the operating state of interval T5. When the LDMOS (11B) is turned off, in the partial resonant circuit 21B on the LDMOS (11B) side, the current IdsB flows towards the LDMOS (11B) side through the energy of the primary winding 12a.

[0172] In the partial resonant circuit 21A on the LDMOS (11A) side, the current IdsB from the partial resonant circuit 21B is attracted, and the current IdsA flows out from the LDMOS (11A) side toward the DC power supply 13 side. Through the outflow of this current IdsA, the charge accumulated in the capacitance component CA of the partial resonant circuit 21A is discharged. As the capacitance component CA is discharged, the drain-source voltage VdsA decreases toward 0V over time constant τf.

[0173] On the other hand, the capacitive component CB of the partial resonant circuit 21B is charged by the current IdsB. As the capacitive component CB is charged, the drain-source voltage VdsB rises by the time constant τr determined by the LC constant of the capacitive component CB and the inductive component Ls-B.

[0174] By increasing the drain-source voltage VdsB over a time constant τr, the rise time of VdsB can be delayed. This delay reduces the high-frequency components caused by the rapid rise of VdsB, thus suppressing surge voltage at the rise time.

[0175] At this point, the capacitive component CB of the partial resonant circuit 21B is connected in series with the capacitive component CA of the partial resonant circuit 21A. Through this series connection, the LC constant becomes smaller, and the rise time tr of the drain-source voltage VdsB is slightly shorter than the fall time tf.

[0176] (Interval T6)

[0177] Interval T6 is the interval in which LDMOS(11A) is in the ON state and LDMOS(11B) is in the OFF state.

[0178] use Figure 7B Explain the operating state of interval T6. Since the LDMOS (11A) is turned on, the drain-source voltage VdsA is maintained at 0V. The current IdsA of the partially resonant circuit 21A switches from the current direction of interval T5 and flows from the DC power supply 13 toward the ground side.

[0179] When the LDMOS (11B) is in the off state, the charging voltage of the drain-source voltage VdsB increases through the resonance of the partial resonant circuit 21B on the LDMOS (11B) side, and the current IdsB decreases.

[0180] Similar to interval T5, even in interval T6, the drain-source voltage VdsB rises through the time constant τr, thereby delaying the rise of VdsB. This reduces the high-frequency components generated by the rapid rise of VdsB, suppressing the surge voltage at the rise moment. Through intervals T5 and T6, the rising waveform of the drain-source voltage VdsB becomes a trapezoidal waveform.

[0181] (Interval T7)

[0182] Interval T7 is the interval in which LDMOS(11B) is turned on when LDMOS(11A) is turned on.

[0183] use Figure 8A Explain the operating state of interval T7. Since LDMOS (11A) is in the ON state, current IdsA begins to flow towards LDMOS (11A). However, since LDMOS (11B) is in the OFF state, the charge of the capacitance component CB of LDMOS (11B) is maintained in the charging state, and the peak value of the drain-source voltage VdsB is maintained at approximately a fixed value.

[0184] Subsequently, on the LDMOS (11B) side, current IdsB is attracted by current IdsA and flows from the LDMOS (11AB) towards the DC power supply 13 side. The drain-source voltage VdsB resonates with the partial resonant circuit 21B, and simultaneously oscillates through the inductive component of the primary winding 12a of the output transformer 12. During the middle of the resonant oscillation accompanying interval T7, the peak value of the drain-source voltage VdsB is approximately fixed, and the surge voltage generated by the rapid change in load current is suppressed by the partial resonance.

[0185] In the later part of interval T7, the charge of the capacitive component CB of the LDMOS (11B) begins to discharge, and the peak value of the drain-source voltage VdsB tends to decrease. In the secondary winding 12b of the output transformer 12, the current IdsA flows to the primary winding 12a, thereby flowing through the load current Iload accompanied by harmonic frequency components. The load current Iload flows from the output terminal 16 to the load 17 through the narrowband filter 15. On the other hand, the harmonic frequency components are reflected by the narrowband filter 15, flow through the output transformer 12 to the LCR series resonator 20, and then to the LDMOS (11A) in the on state. At this time, the resistance Rr of the LCR series resonator 20 is reduced due to the harmonic frequency components, and the surge voltage generated by the harmonic frequency components is suppressed.

[0186] (Interval T8)

[0187] Similar to interval T7, interval T8 is the interval where LDMOS (11A) is in the off state and LDMOS (11B) is in the on state.

[0188] use Figure 8B Explain the operating state of interval T8. In interval T8, the drain-source voltage VdsB, following the later part of interval T7, shows a decreasing tendency through the resonance of the inductive component of the primary winding 12a of the output transformer 12, while the surge voltage is suppressed through the resonance of the partial resonant circuit 21B.

[0189] (3) Trapezoidal waveform

[0190] This invention describes the trapezoidal waveform of the drain-source voltage of the switching element involved in the present invention. Figure 9 The trapezoidal waveform shown is the drain-source voltage VdsA of the LDMOS (11A) between interval T1 and interval T5, representing the two resonances involved in the partial resonant circuit 21 and the LCR series resonator 20, as well as the surge voltage suppression involved in the two resonances.

[0191] (Suppression of the first surge voltage)

[0192] The partial resonant circuit 21 reduces the high-frequency components in the load current in the intervals T1 to T5 and T6 to T8, thereby suppressing the first surge voltage generated by the high-frequency components.

[0193] The load current flows as current IdsA in the partial resonant circuit 21A and as current IdsB in the partial resonant circuit 21B.

[0194] In intervals T1 and T2, current IdsA charges the capacitive component CA of part of resonant circuit 21A in both intervals T1 and T2, while current IdsB discharges the capacitive component CB of part of resonant circuit 21B in interval T1.

[0195] In interval T3, the current IdsA will change the capacitance component CA of part of the resonant circuit 21A from the charging state to the discharging state.

[0196] In intervals T4 and T5, current IdsA discharges the capacitive component CA of part of resonant circuit 21A in intervals T1 and T2, and current IdsB charges the capacitive component CB of part of resonant circuit 21B in interval T5.

[0197] In intervals T1, T2, T4, and T5, the partial resonant circuit 21, through charging and discharging, sets the rise and fall of the drain-source voltage Vds into a sloping trapezoidal shape with rise time tr and fall time tf, reducing the high-frequency components contained in the current Ids and suppressing the first surge voltage generated by the high-frequency components. In interval T3, the partial resonant circuit 21, accompanied by resonance, reduces the high-frequency components contained in the current Ids through the resistive portion of the load impedance and suppresses the first surge voltage generated by the high-frequency components.

[0198] (Suppression of the second surge voltage)

[0199] In interval T3, the partial resonant circuit 21 reduces the harmonic frequency components contained in the load current and suppresses the second surge voltage generated by the harmonic frequency components.

[0200] In interval T3, the load current flows to the second side of the output transformer 12. The load current contains a frequency component of the fundamental frequency f0 and a frequency component of the harmonic frequency n×f0. The LCR series resonator 20 is set to a resonant frequency corresponding to the harmonic frequency n×f0, thereby extracting the frequency component of the harmonic frequency n×f0 reflected by the narrowband filter, which is consumed and attenuated by the resistor Rr. Thus, the LCR series resonator 20 suppresses the second surge voltage generated by the harmonic frequency component.

[0201] (4) Broadband

[0202] The broadbanding of the output of the switching amplifier according to the present invention will be described below. Broadbanding in partial resonance and broadbanding in an LCR series resonator will be shown below.

[0203] (Broadband of partial resonance)

[0204] For broadbanding of the output based on partial resonance, using Figure 10 and Figure 11 Let me explain. Figure 10 This shows a waveform example when the fundamental frequency f0 is 45MHz. Figure 11 This is an example of a waveform when the fundamental frequency f0 is 35MHz.

[0205] in addition, Figure 10 A and Figure 11 A represents the currents IdsA and IdsB. Figure 10 B and Figure 11 B represents the drain-source voltages VdsA and VdsB. Figure 10 C and Figure 11 C represents the gate-source voltages VgsA and VgsB. Figure 10 D and Figure 11 D represents the voltages VR1 and VR2 across the resistance Rr of the LCR DC resonator.

[0206] The resonant frequency of the partial resonant circuit is set to be in the range of 1 to 3 times the fundamental frequency f0, preferably about 1.5 to 2.5 times. It is desirable that when the resonant frequency of the partial resonant circuit is set within the above frequency range, even when the fundamental frequency f0 is variable, the combined losses, including switching losses and the increased chance of losses due to higher frequencies, are still suitable for the output of the switching amplifier.

[0207] Figure 10 Waveform examples and Figure 11 The waveform example shows an example where the adjustment frequency is variable between 35MHz and 45MHz when the fundamental frequency f0 is set to 40MHz.

[0208] At a frequency of 35MHz, the waveform of the residual drain-source voltage Vds before the LDMOS switches from the off state to the on state through partial resonance has a slight voltage rise.

[0209] However, although the switching losses per cycle increase due to this voltage rise, the impact on the overall loss is small because of the low frequency.

[0210] On the other hand, at a frequency of 45MHz, the drain-source voltage Vds is supplemented by the gate-source voltage Vgs of the LDMOS near its lowest point, thus enabling switching via ZVS and reducing switching losses per cycle. Although the frequency increases, the impact on the overall loss is small due to the reduction in switching losses. When the partial resonant frequency exceeds three times the fundamental frequency f0, the residual drain-source voltage Vds increases further, leading to a decrease in overall efficiency.

[0211] (Broadband of LCR series resonator)

[0212] The broadbanding of the output based on the LCR series resonator is illustrated using Figure 12. Figure 12 shows the resonance curves expressed as admittance or current.

[0213] Figure 12A The resonance curve represents the case where an LCR series resonator is constructed from a series circuit. Figure 12B This represents the resonance curve when two series circuits connected in parallel form an LCR series resonator.

[0214] Figure 12A The curve on the left represents the bandwidth of the variable frequency band, passing through frequencies near the maximum value of the resonance curve, and setting the lower limit frequency f0-min and upper limit frequency f0-max relative to the fundamental frequency f0.

[0215] Compared to the bandwidth of a variable frequency band, Figure 12A The curve on the right is the resonance curve of the series circuit in the harmonic band when a series circuit is used to form an LCR series resonator. The resonance curve of the series circuit in this harmonic band is set with a lower limit frequency n×f0-min and an upper limit frequency n×f0-max relative to the harmonic frequency n×f0 (n is an integer) in the frequency range where the series resonator of the series circuit in the harmonic bandwidth is below 50Ω [ohm] (set as reference 1) and the current flowing to the series resonator is more than 1.

[0216] The lower limit frequency n×f0-min and the upper limit frequency n×f0-max define the frequency range of the harmonic band. This frequency range is, for example, set to be ±1.5% or more relative to the harmonic frequency n×f0, totaling 3% or more. By setting this harmonic band, even when the frequency varies within the variable band, the admittance or current relative to the 50Ω [ohm] reference becomes more than 1 times, and high loss (attenuation) broadband can be expected.

[0217] Figure 12B The curve on the left and Figure 12A Similarly, the curve on the left represents the bandwidth of the variable frequency band, passing through the frequency near the maximum value of the resonance curve, and setting the lower limit frequency f0-min and upper limit frequency f0-max relative to the fundamental frequency f0.

[0218] Compared to the resonant curve of the variable frequency band, Figure 12B The two curves on the right are the resonance curves of the series circuit in the harmonic frequency band when two series circuits constitute an LCR series resonator. The resonance curves of this series circuit in the harmonic frequency band use two... Figure 12A The resonance curve of the series circuit in the harmonic frequency band to the right is configured to cause the center frequencies f1 and f2 to deviate.

[0219] The resonance curve of a series circuit in one harmonic band is set to a center frequency f1 that is shifted towards the lower frequency side, and the resonance curve of a series circuit in another harmonic band is set to a center frequency f2 that is shifted towards the higher frequency side. Using the two resonance curves, the lower limit frequency f0-min and the upper limit frequency f0-max relative to the harmonic frequency n×f0 (n is an integer) are set by making the current flowing through the series resonator of the combined harmonic band more than 1 times the frequency.

[0220] The lower limit frequency n×f0-min and upper limit frequency n×f0-max, set as the frequency range of the harmonic band, are for example set to be ±3% or more relative to the harmonic frequency n×f0, totaling 6% or more. By using two series circuits, the frequency range of the harmonic band can be widened. With this harmonic band setting, even when the frequency varies within the variable band, and the admittance or current becomes more than double that of the 50Ω [ohm] reference, high loss (attenuation) can still be expected, thus achieving broadband performance.

[0221] (Bandwidth of partial resonance and bandwidth of high impedance)

[0222] Figure 13 illustrates the relationship between the bandwidth of partial resonance and the bandwidth of high impedance. Figure 13A This represents the Smith chart. Figure 13B Frequency response diagram representing the attenuation rate.

[0223] Figure 13A The Smith chart presents the standing wave ratio (SWR) circle S, representing the reflection state between the load and the impedance. As the frequency is scanned, the impedance is represented by a trajectory on the SWR circle S, turning right from the low-frequency side towards the high-frequency side. Figure 13A In the standing wave ratio (SWR) circle S shown, in the frequency range from the fundamental frequency f0 to the harmonic frequency n×f0, there is a high impedance bandwidth fb between the fundamental frequency f0 and the harmonic frequency n×f0, where the impedance increases.

[0224] Therefore, the frequency range from the fundamental frequency f0 to the harmonic frequency n×f0 is used as the output frequency range. The switching amplifier, with its high impedance bandwidth fb and harmonic frequency n×f0, is used as the location where the impedance increases. High impedance becomes the main cause of surge voltage.

[0225] To suppress surge voltage in the high-impedance bandwidth fb, the impedance in the frequency range near fb is reduced. This invention uses a partially resonant circuit to set a partial resonant bandwidth FB2, thereby reducing the impedance and suppressing surge voltage in the high-impedance bandwidth fb.

[0226] To suppress surge voltages within a frequency range centered on the harmonic frequency n×f0, and to reduce the impedance in this frequency range, this invention uses an LCR series resonator to set the bandwidth FB3 of the LCR series resonator, thereby reducing the impedance and suppressing surge voltages within the frequency range centered on the harmonic frequency n×f0.

[0227] Figure 13B Expressed in terms of frequency characteristics of attenuation rate Figure 13A The Smith chart shows that the partial resonant bandwidth FB2 corresponds to the high impedance bandwidth fb, and the LCR series resonator bandwidth FB3 corresponds to the frequency range centered on the harmonic frequency n×f0. Surge voltage is suppressed by setting high attenuation rates in both the partial resonant bandwidth FB2 and the LCR series resonator bandwidth FB3.

[0228] Industrial applicability

[0229] The switching amplifier involved in this invention can be applied to harmonic power supplies (RF generators) used in semiconductor manufacturing equipment and liquid crystal panel manufacturing equipment.

[0230] Explanation of reference numerals in the attached figures

[0231] 1, 2 Switching Amplifiers

[0232] 10A signal source

[0233] 10B signal source

[0234] 11 Switching Section

[0235] 12 Output Transformer

[0236] 12a primary winding

[0237] 12b secondary winding

[0238] 13 DC power supply

[0239] 14 Push-Pull Circuit

[0240] 15 Narrowband Filters

[0241] 16 Output terminals

[0242] 17 Load

[0243] 20, 20a, 20b LCR series resonators

[0244] Resonant circuits 21, 21A, and 21B

[0245] 101a, 101b Switching Amplifiers

[0246] 110A and 110B signal sources

[0247] 111 Switch Section

[0248] 111A and 111B switching elements

[0249] 112 Output Transformer

[0250] 112a Primary winding

[0251] 112b secondary winding

[0252] 113 DC power supply

[0253] 114 Push-Pull Circuit

[0254] 115 Narrowband Filter

[0255] 116 Output Terminal

[0256] 117 Load

[0257] C, CA, CB capacitance components

[0258] Coss-A and Coss-B parasitic capacitances

[0259] Additional capacitors for Cp-A and Cp-B

[0260] Cr, Cr1, Cr2 capacitors

[0261] Combined capacitors of Ctotal-A and Ctotal-B

[0262] FB2 partial resonant bandwidth

[0263] FB3 LCR series resonator bandwidth

[0264] Ids, IdsA, IdsB Current

[0265] Iload load current

[0266] Lline-A and Lline-B wiring inductors

[0267] Lr, Lr1, Lr2 inductors

[0268] Lleak leakage inductance

[0269] Ls, Ls-A, Ls-B Inductance Components

[0270] Ls1-A and Ls1-B series inductors

[0271] Pg gate signal

[0272] R1 primary side resistor section

[0273] Resistors Rr, Rr1, and Rr2

[0274] S-shaped standing wave ratio (SWR) circle

[0275] Intervals T1, T2, T3, T4, T5, T6, T7, T8

[0276] Vds, VdsA, VdsB Drain-Source Voltages

[0277] Vgs, VgsA, VgsB Gate-Source Voltages

[0278] f0 fundamental frequency

[0279] f0-max upper limit frequency

[0280] f0-min lower limit frequency

[0281] center frequencies of f1 and f2

[0282] fb high impedance bandwidth

[0283] τf and τr are time constants.

Claims

1. A switching amplifier, characterized in that, have: A transformer-coupled push-pull circuit consists of a pair of switching elements that alternately switch on / off via input signals that are in opposite phases, a primary winding that receives the output signals of the switching elements at both ends, a secondary winding that extracts the output signals that are in opposite phases, and an output transformer connected to an external power supply at the center tap of the primary winding. A narrowband filter is connected to the output terminal of the secondary winding of the output transformer; A pair of partially resonant circuits, which consist of the inductive component on the drain side of each switching element and the capacitive component between the drain and source of each switching element; The LCR series resonator is composed of a series circuit of inductance, capacitance and resistance connected between the drain terminals of each of the switching elements.

2. The switching amplifier according to claim 1, characterized in that, The inductive component of each resonant circuit is any one or any combination of the following: the series inductance connected in series between each switching element and one end of the output transformer, the leakage inductance of the output transformer, and the wiring inductance between each switching element and one end of the output transformer. The capacitance component of each resonant circuit is either the output parasitic capacitance of the switching element or the combined capacitance of the output parasitic capacitance of the switching element and the additional capacitance connected in parallel to the switching element.

3. The switching amplifier according to claim 1, characterized in that, The resonant frequency of each resonant circuit is set to the range from the fundamental frequency f0 of the high-frequency output from the secondary side of the output transformer to the third harmonic frequency 3×f0. The impedance of the resonant circuit is set to decrease within the frequency range that makes the impedance increase through the partial resonant circuit, and the time constants of the rise and fall of the drain-source voltage of the switching element are set to the set values.

4. The switching amplifier according to claim 1, characterized in that, Relative to the fundamental frequency f0 of the high-frequency output from the secondary side of the output transformer, the resonant frequency of the LCR series resonator is the harmonic frequency n×f0, where n is an integer. The bandwidth of the resonant frequency is set to a frequency range of n×f0-min to n×f0-max based on the variable frequency range of the fundamental frequency f0 from f0-min to f0-max.

5. The switching amplifier according to claim 1, characterized in that, The switching amplifier has multiple LCR series resonators with different resonant frequencies connected in parallel.

6. The switching amplifier according to claim 5, characterized in that, The switching amplifier has a first LCR series resonator and a second LCR series resonator. The first LCR series resonator has a resonant frequency at a low frequency relative to the fundamental frequency f0 of the high-frequency output from the secondary side of the output transformer, and the second LCR series resonator has a resonant frequency at a high frequency relative to the harmonic frequency n×f0 of the high-frequency output.

7. The switching amplifier according to any one of claims 1 to 6, characterized in that, The switching element is a DMOS.

Images