Sensorless control method for permanent magnet synchronous motor based on high frequency injection

Abstract

This invention discloses a sensorless control method for permanent magnet synchronous motors based on high-frequency injection, belonging to the field of sensorless control technology for permanent magnet synchronous motors. It constructs an input combination of a high-frequency square wave injection signal and a PWM triangular carrier signal; converts the high-frequency square wave injection signal in the input combination with the rotor position estimate into an α-β stationary coordinate system voltage, and then generates an inverter drive signal through an SVPWM module driven by the PWM triangular carrier signal; it acquires the stator three-phase current response signal, processes it to obtain a high-frequency current component, obtains the d-q rotating coordinate system fundamental frequency current, and inputs it to the current loop; based on the high-frequency current component, it obtains a high-frequency current envelope signal, inputs it to a PI-type phase-locked loop, calculates the rotor position estimate and speed estimate, and feeds them back to the speed loop and current loop, dispersing fixed-frequency harmonic concentrated energy and reducing audible noise and electromagnetic interference during zero-low speed operation of the motor.

Description

Technical Field

[0001] This invention relates to the field of sensorless control technology for permanent magnet synchronous motors, specifically a sensorless control method for permanent magnet synchronous motors based on high-frequency injection. Background Technology

[0002] In new energy vehicles and other applications, there is a strong demand for stable operation and precise start-up of motors at zero and low speeds. However, traditional control relies on mechanical position sensors, which are prone to damage and failure under complex operating conditions. This can directly lead to loss of control at zero and low speeds, causing safety accidents and increasing costs and electromagnetic interference risks. Therefore, sensorless control technology in the zero and low speed range has become a core research direction in the industry. In the zero and low speed range of permanent magnet synchronous motors, the fundamental signal is weak, rendering conventional observation methods ineffective. High-frequency signal injection is the mainstream practical solution in this range. However, existing conventional high-frequency square wave injection techniques mostly use injection signals with fixed frequency and amplitude, coupled with PWM carrier waves with fixed parameters. During operation, this generates concentrated high-frequency harmonics, causing electromagnetic interference and high-frequency audible noise. Furthermore, the fixed parameters cannot adapt to sudden load changes, directly resulting in a sharp drop in rotor position estimation accuracy and poor system robustness in the zero and low speed range. It is difficult to simultaneously meet the dual requirements of low noise, high precision, and adaptive control under zero and low speed operation. Summary of the Invention

[0003] To address the shortcomings of existing technologies, this invention provides a sensorless control method for permanent magnet synchronous motors based on high-frequency injection, aiming to improve the noise and accuracy problems in sensorless control of permanent magnet synchronous motors based on traditional fixed high-frequency signal injection.

[0004] To achieve the above objectives, the present invention provides the following technical solution: a sensorless control method for a permanent magnet synchronous motor based on high-frequency injection, comprising the following steps: Construct an input combination of a high-frequency square wave injection signal and a PWM triangular carrier signal; In the low-speed operating range of the permanent magnet synchronous motor, the high-frequency square wave injection signal in the input combination is combined with the rotor position estimate to convert it into an α-β stationary coordinate system voltage, and then the inverter drive signal is generated by the SVPWM module driven by the PWM triangular carrier signal. The inverter drive signal is input to the permanent magnet synchronous motor, the stator three-phase current response signal is collected, the stator three-phase current response signal is processed to obtain the high-frequency current component, the fundamental frequency current of the dq rotating coordinate system is obtained based on the high-frequency current component, and input to the current loop. The high-frequency current envelope signal is obtained based on the high-frequency current component. The high-frequency current envelope signal is input into the PI-type phase-locked loop. The rotor position estimate and speed estimate are calculated through coordinate transformation and fed back to the speed loop and current loop.

[0005] The present invention has the following beneficial effects: 1. This invention employs a strategy of periodically varying high-frequency injected square wave signals combined with randomly varying PWM triangular carrier signals to disperse fixed-frequency harmonics and concentrate energy, thereby reducing audible noise and electromagnetic interference during low-speed motor operation and optimizing the operating environment.

[0006] 2. This invention addresses the problem of weak fundamental wave signals at zero and low speeds by dynamically modulating the injected signal, combined with filter demodulation and phase-locked loop observer calculation, to accurately calculate the rotor position information of the permanent magnet synchronous motor, thereby improving the rotor position estimation accuracy in the zero and low speed range.

[0007] 3. This invention enhances the robustness and adaptability of the system under load change conditions by dynamically adjusting the injection frequency, adapting to load fluctuations and sudden changes in operating conditions, avoiding high-frequency response distortion, and ensuring system response speed and closed-loop stability.

[0008] Of course, any product implementing this invention does not necessarily need to achieve all of the advantages described above at the same time. Attached Figure Description

[0009] Figure 1 This is a flowchart of the sensorless control method for a low-noise permanent magnet synchronous motor based on adaptive high-frequency injection, as described in this invention.

[0010] Figure 2 This is a structural block diagram of the overall control system of the present invention.

[0011] Figure 3 This is a spectrum diagram of a fixed-frequency current.

[0012] Figure 4 It is a current spectrum diagram under the random carrier frequency vector control and periodic high-frequency injection strategy. Detailed Implementation

[0013] Please see Figure 1 and Figure 2 This invention provides a technical solution: a sensorless control method for a permanent magnet synchronous motor based on high-frequency injection, comprising the following steps: S1. Construct an input combination of a high-frequency square wave injection signal and a PWM triangular carrier signal.

[0014] The high-frequency square wave injection signal includes a periodically changing high-frequency square wave injection signal and a randomly changing high-frequency square wave injection signal; the PWM triangular carrier signal includes a periodically changing PWM triangular carrier signal and a randomly changing PWM triangular carrier signal.

[0015] Among them, the frequency of the periodically changing high-frequency square wave injection signal The calculation formula is: ; In the formula, The center frequency of the periodically varying high-frequency injected square wave signal is... The periodic perturbation frequency of a periodically varying high-frequency injected square wave signal. ω is the periodic angular frequency of the high-frequency injected square wave signal, and t is the motor running time.

[0016] The frequency of the randomly varying high-frequency square wave injection signal The calculation formula is: ; In the formula, The center frequency of the injected signal is a randomly varying high-frequency square wave. The random perturbation frequency of the signal injected into the randomly varying high-frequency square wave. Output a random integer from -1, 0, or 1.

[0017] The frequency of the periodically changing PWM triangular carrier signal The calculation formula is: ; In the formula, The center frequency of the periodically changing PWM triangular carrier signal. The periodic perturbation frequency of the periodically changing PWM triangular carrier signal. The angular frequency of the periodic variation of the PWM triangular carrier signal.

[0018] The frequency of the randomly varying PWM triangular carrier signal The calculation formula is: ; In the formula, The center frequency of the randomly varying PWM triangular carrier signal, The random perturbation frequency of the randomly varying PWM triangular carrier signal.

[0019] The input combinations include: ① Input combination of periodically varying high-frequency square wave injection signal and randomly varying PWM triangular carrier signal; ② Input combination of periodically varying high-frequency square wave injection signal and periodically varying PWM triangular carrier signal; ③ Input combination of randomly varying high-frequency square wave injection signal and periodically varying PWM triangular carrier signal; ④ Input combination of randomly varying high-frequency square wave injection signal and randomly varying PWM triangular carrier signal.

[0020] This embodiment defines multiple types of adaptive input combinations of high-frequency square wave injection signals and PWM triangular carrier signals, and further clarifies the frequency calculation methods for various periodically and randomly varying high-frequency square wave injection signals and PWM triangular carrier signals. The combination of the two can achieve multi-mode adaptive matching of the injection signal and the carrier signal, effectively dispersing the concentrated high-frequency harmonic energy brought about by fixed frequency injection and fixed carrier parameters, reducing audible noise and electromagnetic interference during the zero-low speed operation of the motor. At the same time, the dynamic modulation of the injection signal and carrier signal frequencies can adapt to the load change conditions, avoid high-frequency response distortion, improve the rotor position estimation accuracy and control system robustness of the permanent magnet synchronous motor in the zero-low speed range, and solve the technical problem that the existing fixed parameter high-frequency injection scheme is difficult to balance low noise, high precision and adaptive control under zero-low speed operation.

[0021] This invention uses the input combination of a periodically varying high-frequency square wave injection signal and a randomly varying PWM triangular carrier signal as an example for illustration. The implementation process is the same for other input combinations, and will not be repeated here.

[0022] S2. In the low-speed operating range of the permanent magnet synchronous motor, the high-frequency square wave injection signal in the input combination is combined with the rotor position estimate and converted into an α-β stationary coordinate system voltage, and then the inverter drive signal is generated by the SVPWM module driven by the PWM triangular carrier signal.

[0023] S201, Obtain the fundamental d-axis voltage and the high-frequency square wave voltage corresponding to the high-frequency square wave injection signal. Adding them together, we obtain the total voltage along the d-axis in the dq rotating coordinate system. Simultaneously obtain the total q-axis voltage in the dq rotating coordinate system. .

[0024] S202, with the estimated rotor position To change the angle, the total voltage along the d-axis is... q-axis total voltage Transforming from the dq rotating coordinate system to the α-β stationary coordinate system yields the corresponding α-axis voltage in the α-β stationary coordinate system. β-axis voltage .

[0025] The specific calculation formula is as follows: .

[0026] S203, Obtain DC voltage The DC voltage The α-axis voltage in the α-β stationary coordinate system β-axis voltage Input the SVPWM module; the SVPWM module then sequentially executes the first step using the coordinate transformation formula. , , Will In a stationary coordinate system, the two-phase voltage is converted into a three-phase voltage (a, b, c). Then, based on the sign of the three-phase voltage, the flags a, b, and c are obtained, and the formula is used to... Determine the sector (1-6) where the voltage vector is located, and then combine the sector, DC voltage, and switching cycle. ,pass , Calculate and determine the duration of action of the two effective voltage vectors by sector. , ,like Then, the modulation is proportionally over-limited, and then... The zero-vector action time was calculated, and a seven-segment symmetrical PWM strategy was finally adopted. Allocate vector time and assign values ​​to the three-phase duty cycles Ta, Tb, and Tc according to sector rules to control the inverter switches.

[0027] S204. Taking the three-phase duty cycles Ta, Tb, and Tc, and the PWM triangular carrier as inputs, each phase duty cycle is compared point-by-point with the triangular carrier: when the duty cycle is greater than the carrier, a high level is output; when it is less than the carrier, a low level is output. Simultaneously, complementary pulses for the lower bridge arm are generated through logical inversion, ultimately outputting the three-phase original PWM pulses to drive the three-phase inverter bridge arms, achieving the inverter output of the target voltage vector.

[0028] S205. Perform complementary logic processing and dead-time processing on the three-phase original PWM pulses in sequence, and output the inverter drive signal; output the inverter drive signal to the three-phase inverter to drive the switching devices of the three-phase inverter to operate.

[0029] In this embodiment, a high-frequency square wave injection signal is superimposed with the fundamental d-axis voltage to construct the total d-axis voltage in the low-speed operating range of the permanent magnet synchronous motor. Combined with the real-time rotor position estimation value, a synchronous rotation transformation is completed to obtain the α-β two-phase stationary coordinate system voltage. Then, the SVPWM module is driven by a PWM triangular carrier signal matched with the injection signal to complete pulse width modulation, pulse generation, and inverter drive signal output. At the same time as the motor outputs the fundamental torque, a high-frequency excitation signal adapted to the carrier signal is accurately injected into the motor stator winding, providing a stable and effective high-frequency current response basis for subsequent rotor position and speed calculation, and ensuring the accuracy of rotor position estimation in the zero-low speed range of the motor without sensor control.

[0030] The process of sequentially performing complementary logic processing and dead-time processing on the three-phase original PWM pulses is as follows: The logic is inverted sequentially on the upper PWM pulses of the three-phase original PWM transistors to generate complementary pulses for the corresponding lower PWM transistors to ensure that the initial states of the bridge arms are reversed. Then, a fixed dead time is inserted into the complementary pulses of the upper and lower PWM transistors of each phase. At the moment of switching, the upper and lower PWM transistors are turned off simultaneously, reserving the switching buffer time of the devices to completely avoid bridge arm shoot-through. Finally, a three-phase six-channel safe PWM drive pulse with dead time is output.

[0031] S3. Input the inverter drive signal into the permanent magnet synchronous motor, collect the stator three-phase current response signal, process the stator three-phase current response signal to obtain the high-frequency current component, obtain the dq rotating coordinate system fundamental frequency current based on the high-frequency current component, and input it into the current loop.

[0032] S301, Response signal to the stator three-phase current , , Perform the Clark transformation to obtain the α-axis current in the α-β stationary coordinate system. β-axis current ; Specifically, the calculation formula is as follows: .

[0033] S302, the α-axis current β-axis current An input high-pass filter is used, the cutoff frequency of which is set between the fundamental frequency and the high-frequency injection frequency. The high-frequency current component in the α-β stationary coordinate system is extracted after high-pass filtering. .

[0034] S303, Regarding the high-frequency current component Perform the Park transformation to obtain the d-axis high-frequency current in the dq rotating coordinate system. Shaft high frequency current ; Specifically, the calculation formula is as follows: .

[0035] S304. The d-axis high-frequency current at the current time and the previous time. q-axis high-frequency current The average values ​​were taken separately, and the fundamental frequency current of the d-axis in the dq rotating coordinate system was extracted by low-pass filtering. q-axis fundamental frequency current .

[0036] Specifically, the calculation formula is as follows: ; In the formula, This represents the high-frequency current along the d-axis at the current moment. This represents the high-frequency d-axis current at the previous moment. This represents the q-axis high-frequency current at the current moment. This represents the q-axis high-frequency current at the previous moment.

[0037] d-axis fundamental frequency current q-axis fundamental frequency current Input to the current loop.

[0038] This embodiment sequentially performs Clark transformation, high-pass filtering with a cutoff frequency set between the fundamental frequency and the high-frequency injection frequency, Park transformation, and low-pass filtering on the stator three-phase current response signal of the permanent magnet synchronous motor driven by the inverter drive signal. On the one hand, the coordinate system transformation of the current signal can be completed through Clark transformation and Park transformation, which is adapted to the coordinate system system of the permanent magnet synchronous motor vector control, greatly simplifying the calculation of subsequent signal processing and improving the real-time performance of the control system. On the other hand, the high-frequency current component containing rotor position information and the fundamental frequency torque current component can be accurately separated through parameter-matched high-pass filter, effectively filtering out the interference of low-frequency fundamental component on the high-frequency position observation signal, providing a clean high-frequency current input for subsequent rotor position and speed calculation, and ensuring the accuracy of rotor position estimation in the zero-low speed range of the motor.

[0039] Meanwhile, by using a low-pass filtering method that averages the current at adjacent sampling times, high-frequency interference is efficiently filtered out with a minimal algorithm structure, and a pure fundamental frequency current is extracted and input into the current loop. This reduces the computational delay of the filtering stage and ensures the stability and response speed of the current closed-loop control. It achieves effective decoupling between high-frequency position observation and fundamental torque closed-loop control, and solves the technical problems of incomplete separation of fundamental and high-frequency current components and difficulty in balancing position estimation accuracy and current closed-loop control performance in existing high-frequency injection sensorless control schemes.

[0040] S4. Obtain the high-frequency current envelope signal based on the high-frequency current component, input the high-frequency current envelope signal into the PI-type phase-locked loop, calculate the rotor position estimate and speed estimate through coordinate transformation, and feed them back to the speed loop and current loop.

[0041] S401. The high-frequency square wave injection signal delayed by one sampling period is used as the demodulation reference signal; S402, the current high-frequency current component , The difference between the value from the previous sampling period and the value from the previous sampling period is calculated to obtain the high-frequency current difference along the α axis. β-axis high-frequency current difference ; S403, using rotor position estimation value Generate quadrature demodulation reference signal , Substitute both into the formula to calculate the rotor position error signal. Ultimately, a high-frequency current envelope signal containing only the rotor position error is obtained.

[0042] The high-frequency current envelope signal is output as the modulation signal for rotor position observation.

[0043] S404. Using the high-frequency current envelope signal as the position error feedback signal, the PI-type phase-locked loop observer performs closed-loop regulation through its built-in PI regulator: proportional term Fast response position deviation, integral term To eliminate steady-state error, the total PI output is processed by a discrete integrator. The rotor electrical angle estimate is obtained by calculation. Simultaneously, the electric angular velocity is extracted, and the estimated rotational speed is obtained after low-pass filtering and rotational speed conversion. ; S405, The rotor position estimate is... These parameters are fed back to the voltage transformation stage of the α-β stationary coordinate system as angle parameters for the corresponding coordinate transformation. S406, The estimated rotational speed value Feedback is sent to the speed loop, relative to the given speed. The difference is used to obtain the speed deviation signal. The input variable parameter anti-integral saturation PI regulator passes through the proportional term. Integral terms Then, through the discrete integral stage The q-axis current setpoint was obtained by calculation. The input current loop performs regulation.

[0044] This embodiment uses a high-frequency square wave injection signal delayed by one sampling period as the demodulation reference signal. The high-frequency current component is calculated by multiplying the difference between adjacent sampling periods with the demodulation reference signal to accurately extract the high-frequency current envelope signal containing rotor position error information. Then, a PI-type phase-locked loop observer is used to calculate the rotor position and speed. The calculated rotor position and speed estimates are fed back to the coordinate transformation stage, speed loop, and current loop, respectively. On the one hand, there is no need to construct an additional complex demodulation carrier; synchronous demodulation is completed with a reference signal completely synchronized with the injection signal, significantly simplifying the demodulation algorithm complexity. Simultaneously, the difference calculation between adjacent sampling periods effectively eliminates static bias and unrelated harmonic interference, significantly improving the signal-to-noise ratio of the high-frequency current envelope signal. This provides an accurate error feedback source for rotor position observation, ensuring the accuracy of rotor position and speed calculations in the zero-low speed range of the motor. On the other hand, the closed-loop adjustment of the PI-type phase-locked loop observer achieves zero-steady-state-error rapid tracking of the rotor position deviation, ensuring the dynamic response performance and steady-state accuracy of position and speed observations.

[0045] Simultaneously, by providing real-time feedback of rotor position estimates to the coordinate transformation loop and closed-loop feedback of speed estimates to the speed and current loops, the entire link of synchronous updates for injection signal orientation, current transformation, and speed and current closed-loop control is realized. This constructs a complete sensorless vector adaptive closed-loop control system for permanent magnet synchronous motors, solving the technical problems of complex demodulation algorithms, low position observation accuracy, and poor robustness of existing high-frequency injection sensorless control schemes. It is compatible with the preceding adaptive high-frequency injection and carrier modulation loops, ultimately achieving stable operation of the motor in the zero-low speed range with low noise, high precision, and high robustness.

[0046] By comparing the current spectrum at a fixed frequency ( Figure 3 ) and the current spectrum diagram under the random carrier frequency vector control and periodic high-frequency injection strategy ( Figure 4 As can be seen, the sensorless control strategy for low-noise permanent magnet synchronous motors based on adaptive high-frequency injection of the present invention effectively suppresses the audible noise caused by fixed high-frequency injection, while improving the estimation accuracy of rotor position under low-speed conditions, providing a more robust and lower-noise adaptive control scheme for sensorless control of permanent magnet synchronous motors.

[0047] An electronic device includes: a processor; and a memory storing computer program instructions, which, when executed by the processor, cause the processor to perform the sensorless control method for a low-noise permanent magnet synchronous motor based on adaptive high-frequency injection as described above.

[0048] A computer-readable storage medium for storing a program that, when executed by a processor, implements the sensorless control method for a low-noise permanent magnet synchronous motor based on adaptive high-frequency injection as described above.

[0049] Those skilled in the art will understand that embodiments of the present invention can be provided as methods, systems, or computer program products. Therefore, the present invention can take the form of a completely hardware embodiment, a completely software embodiment, or an embodiment combining software and hardware aspects. Furthermore, the present invention can take the form of a computer program product embodied on one or more computer-usable storage media (including, but not limited to, disk storage, CD-ROM, optical storage, etc.) containing computer-usable program code.

[0050] This invention is described with reference to flowchart illustrations and / or block diagrams of systems, apparatus (systems), and computer program products according to embodiments of the invention. It will be understood that each block of the flowchart illustrations and / or block diagrams, and combinations of blocks in the flowchart illustrations and / or block diagrams, can be implemented by computer program instructions. These computer program instructions can be provided to a processor of a general-purpose computer, special-purpose computer, embedded processor, or other programmable data processing apparatus to produce a machine, such that the instructions, which execute via the processor of the computer or other programmable data processing apparatus, generate instructions for implementing the flowchart illustrations and / or block diagrams. Figure 1 One or more processes and / or boxes Figure 1 A device that provides the functions specified in one or more boxes.

[0051] These computer program instructions may also be stored in a computer-readable storage medium that can direct a computer or other programmable data processing device to function in a particular manner, such that the instructions stored in the computer-readable storage medium produce an article of manufacture including instruction means, which are implemented in a process Figure 1 One or more processes and / or boxes Figure 1 The function specified in one or more boxes.

[0052] These computer program instructions may also be loaded onto a computer or other programmable data processing equipment to cause a series of operational steps to be performed on the computer or other programmable equipment to produce a computer-implemented process, thereby providing instructions that execute on the computer or other programmable equipment for implementing the process. Figure 1 One or more processes and / or boxes Figure 1 The steps of the function specified in one or more boxes.

[0053] Although preferred embodiments of the invention have been described, those skilled in the art, upon learning the basic inventive concept, can make other changes and modifications to these embodiments. Therefore, the appended claims are intended to be interpreted as including both the preferred embodiments and all changes and modifications falling within the scope of the invention.

[0054] Obviously, those skilled in the art can make various modifications and variations to this invention without departing from its spirit and scope. Therefore, if these modifications and variations fall within the scope of the claims of this invention and their equivalents, this invention also intends to include these modifications and variations.

Claims

1. A sensorless control method for permanent magnet synchronous motors based on high-frequency injection, characterized in that, Includes the following steps: Construct an input combination of a high-frequency square wave injection signal and a PWM triangular carrier signal; In the low-speed operating range of the permanent magnet synchronous motor, the high-frequency square wave injection signal in the input combination is combined with the rotor position estimate to convert it into an α-β stationary coordinate system voltage, and then the inverter drive signal is generated by the SVPWM module driven by the PWM triangular carrier signal. The inverter drive signal is input to the permanent magnet synchronous motor, the stator three-phase current response signal is collected, the stator three-phase current response signal is processed to obtain the high-frequency current component, the fundamental frequency current of the dq rotating coordinate system is obtained based on the high-frequency current component, and input to the current loop. The high-frequency current envelope signal is obtained based on the high-frequency current component. The high-frequency current envelope signal is input into the PI-type phase-locked loop. The rotor position estimate and speed estimate are calculated through coordinate transformation and fed back to the speed loop and current loop.

2. The sensorless control method for a permanent magnet synchronous motor based on high-frequency injection according to claim 1, characterized in that, The high-frequency square wave injection signal includes periodically changing high-frequency square wave injection signals and randomly changing high-frequency square wave injection signals. The PWM triangular carrier signal includes a periodically changing PWM triangular carrier signal and a randomly changing PWM triangular carrier signal. The input combinations include: The input combination of a periodically varying high-frequency square wave injection signal and a randomly varying PWM triangular carrier signal; The input combination of a periodically varying high-frequency square wave injection signal and a periodically varying PWM triangular carrier signal; The input combination of a randomly varying high-frequency square wave injection signal and a periodically varying PWM triangular carrier signal; The input combination of a randomly varying high-frequency square wave injection signal and a randomly varying PWM triangular carrier signal.

3. The sensorless control method for a permanent magnet synchronous motor based on high-frequency injection according to claim 2, characterized in that, The frequency of the periodically varying high-frequency square wave injection signal The calculation formula is: ； In the formula, The center frequency of the periodically varying high-frequency injected square wave signal is... The periodic perturbation frequency of a periodically varying high-frequency injected square wave signal. The periodic angular frequency of the high-frequency injected square wave signal is t, where t is the motor running time. The frequency of the randomly varying high-frequency square wave injection signal The calculation formula is: ； In the formula, The center frequency of the injected signal is a randomly varying high-frequency square wave. The random perturbation frequency of the signal injected into the randomly varying high-frequency square wave. Output a random integer from -1, 0, or 1; The frequency of the periodically varying PWM triangular carrier signal The calculation formula is: ； wherein is the center frequency of the periodically varying PWM triangular carrier signal, is the periodic perturbation frequency of the periodically varying PWM triangular carrier signal, is the periodic variation angular frequency of the PWM triangular carrier signal; The frequency of the randomly varying PWM triangular carrier signal The calculation formula is: ； In the formula, The center frequency of the randomly varying PWM triangular carrier signal, The random perturbation frequency of the randomly varying PWM triangular carrier signal.

4. The sensorless control method for a permanent magnet synchronous motor based on high-frequency injection according to claim 1, characterized in that, The process of converting the high-frequency square wave injection signal combined with the rotor position estimate into an α-β stationary coordinate system voltage is as follows: The fundamental d-axis voltage is obtained and added to the high-frequency square wave voltage corresponding to the high-frequency square wave injection signal to obtain the total d-axis voltage in the dq rotating coordinate system; at the same time, the total q-axis voltage in the dq rotating coordinate system is obtained. Using the estimated rotor position as the transformation angle, the total voltage of the d-axis and the total voltage of the q-axis are transformed from the dq rotating coordinate system to the α-β stationary coordinate system, thereby obtaining the α-axis voltage and β-axis voltage in the α-β stationary coordinate system.

5. The sensorless control method for a permanent magnet synchronous motor based on high-frequency injection according to claim 1, characterized in that, The process of generating six inverter drive signals using the SVPWM module driven by the PWM triangular carrier signal is as follows: The DC voltage is acquired, and the DC voltage, the α-axis voltage and the β-axis voltage in the α-β stationary coordinate system are input into the SVPWM module. The SVPWM module sequentially performs three-phase voltage calculation, sector determination, vector action time solution and symmetrical PWM calculation, and outputs the three-phase duty cycle. The three-phase duty cycle is compared point by point with the PWM triangular carrier signal to obtain the original three-phase PWM pulse; The three-phase original PWM pulses are sequentially processed with complementary logic and dead-time processing to output the inverter drive signal.

6. The sensorless control method of a high-frequency injection-based permanent magnet synchronous motor according to claim 5, characterized in that, The process of sequentially performing complementary logic processing and dead-time processing on the three-phase raw PWM pulses is as follows: The logic is sequentially inverted on the upper PWM pulses of the three-phase original transistors to generate complementary pulses for the corresponding lower transistors so that the initial state of the bridge arms is reversed. Then, a fixed dead time is inserted into the complementary pulses of the upper and lower transistors of each phase. At the moment of switching, the upper and lower transistors are turned off simultaneously, reserving the switching buffer time of the devices, and outputting the three-phase six-channel safety inverter drive signal with dead time.

7. The sensorless control method for a permanent magnet synchronous motor based on high-frequency injection according to claim 1, characterized in that, The process of processing the stator three-phase current response signal to obtain the high-frequency current component is as follows: Perform Clark transformation on the stator three-phase current response signal to obtain the α-axis current and β-axis current in the α-β stationary coordinate system; The α-axis current and β-axis current are input into a high-pass filter. The cutoff frequency of the high-pass filter is set between the fundamental frequency and the high-frequency injection frequency. The high-frequency current component in the α-β stationary coordinate system is extracted by the high-pass filter.

8. The sensorless control method for a permanent magnet synchronous motor based on high-frequency injection according to claim 1, characterized in that, The process of obtaining the fundamental frequency current of the dq rotating coordinate system based on high-frequency current components is as follows: Perform a Park transform on the high-frequency current components to obtain the d-axis high-frequency current and q-axis high-frequency current in the dq rotating coordinate system; The average values ​​of the d-axis high-frequency current and q-axis high-frequency current at the current time and the previous time are respectively taken, and the d-axis fundamental frequency current and q-axis fundamental frequency current in the dq rotating coordinate system are extracted by low-pass filtering.

9. The sensorless control method for a permanent magnet synchronous motor based on high-frequency injection according to claim 1, characterized in that, The process of obtaining the high-frequency current envelope signal based on the high-frequency current component is as follows: The high-frequency square wave injection signal delayed by one sampling period is used as the demodulation reference signal; The difference between the current high-frequency current component and the value of the previous sampling period is calculated to obtain the high-frequency current difference along the α-axis and the high-frequency current difference along the β-axis. The high-frequency current difference along the α-axis and the high-frequency current difference along the β-axis are multiplied by the demodulated reference signal to obtain a high-frequency current envelope signal containing rotor position error information. The high-frequency current envelope signal is output as the modulation signal for rotor position observation.

10. The sensorless control method for a permanent magnet synchronous motor based on high-frequency injection according to claim 1, characterized in that, The process of inputting the high-frequency current envelope signal into a PI-type phase-locked loop and calculating the rotor position estimate and speed estimate through coordinate transformation is as follows: Using the high-frequency current envelope signal as the position error feedback signal, the PI regulator built into the PI-type phase-locked loop observer performs closed-loop regulation to track the rotor position deviation. The rotor position estimate is obtained by solving the integral element, and the speed estimate is output through the PI-type phase-locked loop observer. The rotor position estimates are fed back to the α-β stationary coordinate system voltage transformation stage as angle parameters for the corresponding coordinate transformation. The estimated speed is fed back to the speed loop, and the difference between the estimated speed and the given speed of the speed loop is compared. The speed deviation signal obtained from the comparison is input to the current loop to perform adjustment.

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