Composite bidirectional WPT system with constant current and constant voltage output characteristics

Abstract

This invention relates to the field of power transmission technology, specifically disclosing a composite bidirectional WPT system with constant current and constant voltage output characteristics. The system includes an E# class single-transistor circuit, a coupled resonant network, and a Class E class single-transistor circuit. Both the E# and Class E class single-transistor circuits use NMOS transistors, and their sources are connected in series. During forward transmission, the E# class circuit inverts and the Class E class circuit rectifies, outputting a constant voltage. During reverse transmission, the Class E class circuit inverts and the E# circuit rectifies, outputting a constant current. The system achieves synchronous rectification by acquiring coil voltage or current and detecting zero-crossing using a comparator. It determines hard switching or under-resonance based on the zero-crossing state of the drain-source voltage of the switching transistors before the rising edge of the drive signal, adjusting the phase shift angle accordingly to achieve zero-voltage conduction. Simultaneously, it performs dual closed-loop cascade control of power and efficiency. The outer power loop adjusts the phase shift angle to meet power requirements, while the inner efficiency loop searches for the optimal efficiency point. This invention achieves bidirectional constant current and constant voltage output without adding an additional converter.

Description

Technical Field

[0001] This invention relates to the field of wireless transmission technology, and in particular to a composite bidirectional WPT system with constant current and constant voltage output characteristics. Background Technology

[0002] Wireless Power Transfer (WPT) technology is a technology that enables energy transfer without physical contact, offering advantages such as convenience, flexibility, and high security. Currently, experts and scholars both domestically and internationally have conducted related research in application areas such as electric vehicle charging stations, wireless charging for drones, underwater robots, and medical electronic devices, achieving numerous theoretical and applied results in magnetic field coupling, electric field coupling, and microwave wireless power transfer.

[0003] Magnetic field coupled WPT (Wave Transmission Power Transfer) systems primarily employ three inverter topologies: full-bridge, half-bridge, and single-transistor. In high-power power transfer applications, S-type or P-type inverter topologies based on half-bridge and full-bridge are currently widely used. However, these circuits suffer from high cost, complex control strategies, and cumbersome peripheral circuit design. Full-bridge topologies experience lower current and voltage stress on the switching transistors, making them suitable for high-power applications, but the large number of switching transistors leads to complex drive circuits and bridge arm shoot-through issues. Traditional bidirectional single-transistor WPT systems require only one MOSFET, offering advantages such as low loss, high reliability, convenient driving, low cost, and easy zero-voltage turn-on, making them widely applicable in scenarios requiring constant voltage output. Research on Class E is relatively comprehensive. Class E single-transistor wireless power transfer systems offer advantages such as good soft-switching characteristics and high power output, and can achieve constant current output characteristics in a simpler way, exhibiting stable output characteristics within a certain range, thus improving the system's power transfer capability to some extent. Furthermore, the E# class single-transistor topology has a high sinusoidal voltage and current, and a low total harmonic distortion (THD).

[0004] Scholars have conducted research on related issues: Some literature studies reducing voltage stress on switching transistors in isolated UHF DC-DC converters based on Class E resonant topologies, providing in-depth analysis of their current and impedance characteristics and offering corresponding stress suppression strategies. Other literature proposes a wireless power transfer system based on a novel single-transistor E# class inverter, consisting of a power transfer channel and a signal transmission channel based on high-frequency carrier injection, achieving synchronous energy and signal transmission. Still other literature proposes an improved E# single-transistor topology based on a WPT system, which adjusts the phase of the switching transistor's drive signal to release the energy stored inside the resonant element earlier or later, thus correcting the transistor back to a soft-switching state. This phase-shifting control strategy ensures the circuit is always in optimal operating condition. Finally, another literature proposes a control strategy based on a Class E WPT system at UHF frequencies, ensuring high transmission efficiency even when the coupling distance changes within a certain range.

[0005] Existing research has limitations regarding the output characteristics of WPT systems. The systems proposed in the aforementioned literature can only provide a single constant voltage or constant current output characteristic, and cannot flexibly switch the output mode according to the energy transfer direction. Meanwhile, due to the inherent characteristics of the diode in the switching transistor, traditional single-transistor circuits exhibit reverse conduction during the turn-off period, forming a discharge loop. This limits the implementation of phase-shift control strategies. Summary of the Invention

[0006] This invention provides a composite bidirectional WPT system with constant current and constant voltage output characteristics. The technical problem it solves is that traditional single E or E# type bidirectional topologies can only achieve a single output characteristic of constant voltage (CV) or constant current (CC), which is not easy to meet the requirements of topology constant current and constant voltage (CC-CV) output characteristics under changing scenarios.

[0007] To address the above technical problems, this invention provides a composite bidirectional WPT system with constant current and constant voltage output characteristics, comprising an E# class single-transistor circuit, a coupled resonant network, and a Class E class single-transistor circuit; the E# class single-transistor circuit includes a first DC power supply. First resonant inductor First coupling capacitor First resonant capacitor First switching transistor Second switching transistor The coupled resonant network includes a first coil. Second coil ;

[0008] The Class E single-transistor circuit includes a second DC power supply. Second resonant inductor Second resonant capacitor Second coupling capacitor Third switching transistor and the fourth switching transistor The first to fourth switching transistors are all NMOS transistors.

[0009] First resonant inductor First coupling capacitor First resonant capacitor Sequentially connected in series with the first DC power supply Between the positive and negative terminals; the first coil Parallel to the first coupling capacitor The two ends; the first switching transistor and the second switching transistor After being connected in series, they are all connected in parallel to the first resonant capacitor. The two ends; the first switching transistor The source is connected to the second switch. The source of the first switch transistor The drain is connected to the first coupling capacitor. With the first resonant capacitor The common terminal of the second switch transistor The drain is connected to the first DC power supply. The negative electrode;

[0010] Second resonant inductor Second resonant capacitor Sequentially connected in series with the second DC power supply Between the positive and negative terminals; the second coil and the second coupling capacitor After being connected in series, they are all connected in parallel to the second resonant capacitor. The two ends; the third switch transistor and the fourth switching transistor After being connected in series, they are all connected in parallel to the second resonant capacitor. The two ends; the third switch transistor The source is connected to the fourth switch. The source of the third switch. The drain of the second resonant inductor is connected. With the second resonant capacitor The common terminal of the fourth switch transistor The drain is connected to the second DC power supply. The negative electrode.

[0011] Preferably, the parameter settings of the E# class single-transistor circuit satisfy the following relationship:

[0012] ,

[0013] in, The operating angular frequency of the system. Is with The corresponding frequency, The AC load resistor at the system receiver. For quality factor, To compensate for inductance Compensation capacitor The input resonant frequency between It is a multiple of the frequency.

[0014] Preferably, the first coil satisfy:

[0015] .

[0016] Preferably, the parameter settings of the Class E single-transistor circuit satisfy the following relationship:

[0017] .

[0018] Preferably, the second coil satisfy:

[0019] .

[0020] Preferably, during forward transmission of the system, the first switch transistor... The second switch is kept normally open. For frequency Duty cycle The PWM wave drive signal; the second coil is acquired. The voltage across the terminals is compared at zero-crossing points by an operational amplifier to obtain a square wave signal, which drives the third switch. and the fourth switching transistor Synchronous rectification.

[0021] Preferably, during system reverse transmission, the third switch transistor The fourth switch is kept normally open. For frequency Duty cycle The PWM wave drive signal; the first coil is acquired. The current is compared with zero to obtain a square wave signal, which drives the first switch transistor. and the second switching transistor Synchronous rectification.

[0022] Preferably, during forward transmission of the system, the first switch transistor is sampled. and the second switching transistor Overall drain-source voltage and with the second switching transistor The drive signals are compared to determine the second switching transistor. If zero-voltage turn-on is achieved within the preset zero-voltage turn-on margin, then the second switching transistor is fixed. The driving signal is either delayed or advanced; otherwise, it is adjusted either by lag or lead.

[0023] Preferably, during reverse transmission, the third switch transistor is sampled. and the fourth switching transistor Overall drain-source voltage and with the fourth switching transistor The drive signals are compared to determine the fourth switch. If zero-voltage conduction is achieved within the preset zero-voltage conduction margin, then the fourth switch is fixed. The driving signal is either delayed or advanced; otherwise, it is adjusted either by lag or lead.

[0024] Preferably, the system performs power-efficiency dual closed-loop cascade control:

[0025] The power outer loop determines the current output power; if it is insufficient, the phase shift angle is adjusted.

[0026] The efficiency inner loop compares the transmission efficiency under different phase shift angles and determines the search direction and step size based on the comparison results; the PI controller adjusts the phase shift angle; the near-zero state of the drain-source voltage of the switching transistor is detected synchronously; when the output power meets the requirements, the transmission efficiency is stable, and the zero-voltage conduction condition is met, the current phase shift angle is maintained; if the load, coil spacing, or mutual inductance changes, the search is restarted.

[0027] The composite bidirectional WPT system with constant current and constant voltage output characteristics provided by this invention consists of an E# class single-transistor circuit, a coupled resonant network, and a Class E class single-transistor circuit. Both switching transistors are NMOS transistors with source-to-source connections to eliminate the body diode discharge loop and provide a basis for phase-shift control. During forward transmission, the E# class circuit inverts and the Class E class circuit rectifies, resulting in a constant voltage output. During reverse transmission, the roles are reversed, resulting in a constant current output. Both bidirectional operations achieve synchronous rectification by acquiring coil voltage or current signals for zero-crossing comparison. Simultaneously, the drain-source voltage of the switching transistors and the timing of the drive signals are detected in real time. Based on the zero-voltage conduction margin, the system determines whether the switching is hard-switching or under-resonance and adjusts the phase shift angle to ensure zero-voltage conduction. On the basis of ZVS, a power-efficiency dual closed-loop cascade control is further implemented. The power outer loop adjusts the phase shift angle to meet power requirements, while the efficiency inner loop searches for the optimal efficiency point and restarts optimization when the load and mutual inductance change. Thus, without adding an additional converter, bidirectional constant current and constant voltage output and efficiency improvement are adaptively achieved. Attached Figure Description

[0028] Figure 1 This is a circuit topology diagram of a composite bidirectional WPT system with constant current and constant voltage output characteristics provided in an embodiment of the present invention.

[0029] Figure 2 This is an equivalent circuit model diagram of the system provided in this embodiment of the invention during forward and reverse transmission; wherein, Figure 2 (a) corresponds to the forward transmission equivalent model, and (b) corresponds to the reverse transmission equivalent model.

[0030] Figure 3 This is a block diagram of the overall system control strategy provided in the embodiments of the present invention;

[0031] Figure 4 This is a schematic diagram of the adaptive load phase-shifting control strategy provided in an embodiment of the present invention when the load is greater than the optimal load;

[0032] Figure 5 This is a schematic diagram of the adaptive load phase-shifting control strategy provided in an embodiment of the present invention when the load is less than the optimal load;

[0033] Figure 6 This is a schematic diagram of the adaptive mutual inductance phase-shifting control strategy provided in an embodiment of the present invention when the mutual inductance is greater than the optimal value;

[0034] Figure 7 This is a schematic diagram of the adaptive mutual inductance phase-shifting control strategy provided in an embodiment of the present invention when the mutual inductance is less than the optimal value;

[0035] Figure 8 This is a graph showing the changes in system output power and transmission efficiency with load during forward transmission, as provided in an embodiment of the present invention.

[0036] Figure 9 This is a graph showing the changes in system output power and transmission efficiency with load during reverse transmission, as provided in an embodiment of the present invention.

[0037] Figure 10 This is a test diagram of the constant voltage characteristic of the system output voltage during forward transmission as the load changes, provided in an embodiment of the present invention.

[0038] Figure 11 This is a test diagram of the constant current characteristic of the system output current changing with the load during reverse transmission, provided in an embodiment of the present invention.

[0039] Figure 12 This is a waveform diagram of the drain-source voltage of the switching transistor when the forward transmission is less than the optimal load (hard switching state) provided by an embodiment of the present invention.

[0040] Figure 13This is a waveform diagram of the drain-source voltage of the switching transistor after phase shift control (soft switching state) when the forward transmission is less than the optimal load, provided by an embodiment of the present invention.

[0041] Figure 14 This is a waveform diagram of the drain-source voltage of the switching transistor (under-resonance state) when the forward transmission is provided and the load is greater than the optimal load, according to an embodiment of the present invention.

[0042] Figure 15 This is a waveform diagram of the drain-source voltage of the switching transistor after phase shift control (soft switching state) when the forward transmission is provided and the load is greater than the optimal load, according to an embodiment of the present invention.

[0043] Figure 16 This is the drain-source voltage waveform of the switching transistor (hard switching state) when forward transmission is provided by an embodiment of the present invention and the mutual inductance is less than the optimal value.

[0044] Figure 17 This is a waveform diagram of the drain-source voltage of the switching transistor after phase shift control (soft switching state) when the forward transmission is provided and the mutual inductance is less than the optimal value, according to an embodiment of the present invention.

[0045] Figure 18 This is the drain-source voltage waveform of the switching transistor (under-resonance state) when forward transmission and mutual inductance are greater than the optimal value, as provided in the embodiment of the present invention.

[0046] Figure 19 This is a waveform diagram of the drain-source voltage of the switching transistor after phase shift control (soft switching state) when the forward transmission and mutual inductance are greater than the optimal value, provided by an embodiment of the present invention. Detailed Implementation

[0047] The embodiments of the present invention are described in detail below with reference to the accompanying drawings. The embodiments are given for illustrative purposes only and should not be construed as limiting the present invention. The accompanying drawings are for reference and illustration only and do not constitute a limitation on the scope of patent protection of the present invention, because many changes can be made to the present invention without departing from the spirit and scope of the present invention.

[0048] This invention provides a composite bidirectional WPT system with constant current and constant voltage output characteristics, the circuit topology of which is shown in the figure below. Figure 1 As shown, it includes Class E single-transistor circuits, coupled resonant networks, and Class E single-transistor circuits.

[0049] E# class single-transistor circuit includes a first DC power supply. First resonant inductor First coupling capacitor First resonant capacitor First switching transistor Second switching transistor The coupled resonant network includes a first coil. Second coil Class E single-transistor circuits include a second DC power supply. Second resonant inductor Second resonant capacitor Second coupling capacitor Third switching transistor and the fourth switching transistor The first through fourth switching transistors are all NMOS transistors.

[0050] The specific connection relationships of each component are as follows: First resonant inductor First coupling capacitor First resonant capacitor Sequentially connected in series with the first DC power supply Between the positive and negative terminals. First coil. Parallel connection to the first coupling capacitor The two ends. The first switching transistor. Second switching transistor After being connected in series, they are all connected in parallel to the first resonant capacitor. The two ends. Among them, the first switching transistor The source is connected to the second switching transistor. The source, the first switch transistor The drain is connected to the first coupling capacitor. With the first resonant capacitor The common terminal, the second switching transistor The drain is connected to the first DC power supply. The negative terminal. The second resonant inductor. Second resonant capacitor Sequentially connected in series with the second DC power supply Between the positive and negative terminals. Second coil. Second coupling capacitor After being connected in series, they are all connected in parallel to the second resonant capacitor. The two ends. The third switching transistor. and the fourth switching transistor After being connected in series, they are all connected in parallel to the second resonant capacitor. The two ends. Among them, the third switch transistor. The source is connected to the fourth switch. The source, the third switch The drain is connected to the second resonant inductor. With the second resonant capacitor The common terminal, the fourth switch The drain is connected to the second DC power supply. The negative electrode.

[0051] Forward transmission is defined as the energy supply from a Class E single-transistor circuit to a Class E single-transistor circuit via a coupled resonant network, abbreviated as: E# → Class E, energy flows from... Figure 1 Transmission from the left to the right, first DC power supply Input, second DC power supply Output. At this time, the first DC power supply... The output DC voltage passes through the first switching transistor Second switching transistor Inverter and first resonant inductor First coupling capacitor First resonant capacitor Resonance compensation is achieved through the first coil. To the second coil (At this time, the first coil) Second coil The coupling and mutual inductance between them are expressed as ) Transmits high-frequency alternating current; second coil After receiving, it passes through the second resonant inductor Second resonant capacitor Second coupling capacitor Resonance compensation and third switch Fourth switching transistor The rectifier generates an output DC power supply which is then directed to the second DC power supply. powered by.

[0052] The reverse transmission direction is defined as the power supply from a Class E single-transistor circuit to an E# single-transistor circuit via a coupled resonant network, abbreviated as Class E→E#, where energy flows from... Figure 1 Transmission from the right side to the left side, second DC power supply Input, first DC power supply Output. At this time, the second DC power supply... The output DC voltage passes through the third switching transistor Fourth switching transistor Inverter and second resonant inductor Second resonant capacitor Second coupling capacitor Resonance compensation is achieved through the second coil. To the first coil (At this time, the second coil) and the first coil The coupling and mutual inductance between them are expressed as ) Transmits high-frequency alternating current; First coil After receiving, through the first resonant inductor First coupling capacitor First resonant capacitor Resonance compensation and the first switching transistor Second switching transistor The rectification produces an output DC power supply to the first DC power source. powered by.

[0053] When the system is operating in the forward direction, the first switching transistor Keep it constantly on, second switch transistor For frequency kHz, duty cycle The PWM wave drive signal has an operating angular frequency denoted as . Second switching transistor During the turn-off period, the input current is equal to that of the first resonant capacitor. Charging, second switching transistor Drain-source voltage Rising from zero; in the second switching transistor Before conduction, the first resonant capacitor Discharge to zero, drain-source voltage The voltage is reduced to zero, thus achieving zero-voltage conduction. The second coil is then sampled. The voltage across the terminals is compared at zero-crossing points by an operational amplifier to obtain a square wave signal, which drives the third switching transistor. Fourth switching transistor Synchronous rectification: In this case, the Class E circuit acts as a rectifier, and the system enters constant voltage output mode.

[0054] When operating in reverse, the third switching transistor In this embodiment, the fourth switch is kept normally open. For the same frequency kHz, duty cycle The PWM wave drive signal has an operating angular frequency denoted as . (and (Same numerical value). Fourth switching transistor. Second resonant capacitor during turn-off Charging and discharging, the voltage is reduced to zero before conduction to achieve zero-voltage conduction. Data is collected from the first coil. The current, after being compared at zero, drives the first switching transistor. Second switching transistor Synchronous rectification: When the E# circuit acts as a rectifier, the system enters constant current output mode.

[0055] Compared to traditional single-transistor topologies, this circuit employs a composite MOSFET source-to-source connection. This connection method ensures that the drive signal uses the common ground of the two switching transistor sources, avoiding the body diode and parallel capacitor forming a discharge circuit. This design also creates the prerequisites for implementing the phase-shift control strategy, ensuring the parallel capacitor of the switching transistors is connected. and Phase shift control ensures that the energy stored in the parallel resonant capacitor of the switching transistor is released before the drive signal becomes high, thus ensuring that the system is always in a soft-switching state.

[0056] Figure 2 This is the equivalent circuit model for the system during forward and reverse transmission. Figure 2 (a) corresponds to forward transmission. Figure 2 (b) corresponds to the reverse transmission, where For forward transmission from the first DC power supply Looking at the equivalent load resistance inside, For reverse transmission from the second DC power supply Looking at the equivalent load resistance inside, , These represent the AC input voltage and current of a Class E single-transistor circuit during forward transmission, respectively. , These represent the equivalent AC load and current of a Class E single-transistor circuit during forward transmission. , This refers to the AC input voltage and current of a Class E single-transistor circuit during reverse transmission. , These represent the equivalent AC load and current of a single-transistor E# class circuit during reverse transmission. For the first coil The parallel equivalent internal resistance, For the second coil The series equivalent internal resistance.

[0057] During forward transmission, based on circuit equivalence and resonance conditions, the E# type inverter (second switching transistor) The duty cycle is The operating angular frequency is To achieve zero-voltage conduction, the resonance condition must be met:

[0058] (1)

[0059] in, for amplitude, This is a positive phase shift angle. This condition ensures that the second switching transistor... During the turn-off period, the first resonant capacitor connected in parallel The stored energy is released just before the circuit is turned on.

[0060] During reverse transmission, the Class E inverter (fourth switch) The duty cycle is The operating angular frequency is To achieve zero-voltage conduction, the resonance condition must be met:

[0061] (2)

[0062] Analyzing the system's forward operating mode, the active power reaches its maximum when the system is at ideal resonance. The equivalent input power at the transmitter is then calculated at this resonance point. for:

[0063] (3)

[0064] Receiver equivalent output power for:

[0065] (4)

[0066] in, and These represent the equivalent impedances of the transmitter and receiver during forward transmission:

[0067] (5)

[0068] Finally, the forward transmission efficiency of the system is obtained. The expression is:

[0069] (6)

[0070] Introducing coefficients The amplitude of the transmitting coil voltage is obtained. With input voltage Relationship:

[0071] (7)

[0072] When duty cycle When the coefficient is 0.5 The general empirical value is 3.16.

[0073] The derivation process for the reverse transmission of the system is similar to that for the forward transmission. Analyzing the system's reverse operating mode, when the system is at ideal resonance and the active power reaches its maximum, the equivalent input power at the transmitter is calculated at this resonance point. for:

[0074] (8)

[0075] Receiver equivalent output power for:

[0076] (9)

[0077] in, , Let the equivalent impedances of the transmitter and receiver be represented respectively during reverse transmission, and their expressions are:

[0078] (10)

[0079] Finally, the reverse transmission efficiency of the system is obtained. The expression is:

[0080] (11)

[0081] It can be seen that the transmission efficiency of the system in both directions is closely related to the load and mutual inductance. When the load deviates from the optimal matching value, the efficiency decreases accordingly; when the mutual inductance deviates from the design value, the coil coupling changes, which also causes efficiency degradation. Therefore, in practical applications, a phase-shifting control strategy is needed to dynamically adjust the system operating point to maintain high transmission efficiency over a wide range of loads and mutual inductances.

[0082] To verify the constant voltage output characteristic of the system during forward operation, a formula for the output voltage needs to be established. If this formula does not include the output resistance... ( The equivalent resistance indicates that the system operates independently of the load during forward operation and exhibits constant voltage output characteristics. Through derivation, the system output voltage during forward transmission is... The expression is:

[0083] (12)

[0084] When the system is ideal and at resonance, the internal resistance of the coil is negligible, then: .

[0085] To obtain the equivalent load With load The analytical relation is used, and ideal assumptions are made for the model:

[0086] The resonant current at the receiving end is approximately sinusoidal; the rectifier switch switches synchronously according to the current zero-crossing point; neglecting the losses of the rectifier components, the following derivation is obtained: .

[0087] Finally, the output voltage is obtained. The expression is:

[0088] (13)

[0089] Since the right side of the equation in formula (13) consists of constants, this part can be considered a constant value. Therefore, the derived output voltage... The expression does not include output load. This item indicates and This is irrelevant, thus verifying that the system has constant voltage source output characteristics during forward transmission.

[0090] In practical circuits, the equivalent resistance of a coil cannot be ignored. When the load... When the resistance is relatively small, the equivalent impedance at the receiving end is... Output load cannot be ignored. Changes in resistance will significantly affect the output current. Therefore, to enable the topology to operate in constant voltage output mode, the load should be set within a larger resistance range.

[0091] When the system is in reverse operation, the output current is... ( The equivalent resistance The current (i.e., the system output circuit) has only one loop, therefore Expanding on this, we get:

[0092] (14)

[0093] Input voltage and duty cycle When fixed, voltage It does not change with the output load. When the system is at ideal resonance, we have:

[0094] (15)

[0095] When the model approaches ideality, there is It is also easy to know the equivalent output load. With output load They are positively correlated, therefore the output current is obtained by formula (14). With output load It is irrelevant; it is a constant current source.

[0096] Overall control block diagram as follows Figure 3 As shown. During forward transmission, an open-loop reference drive is first established. The first switching transistor on the primary side... Keeping it constantly on, the microcontroller sends a signal to the second switch. Provide frequency kHz (corresponding angular frequency) Positive duty cycle The fixed square wave signal, the second switching transistor High-frequency chopping is performed. The secondary side then acquires data via a second coil. The voltage across the terminals is zero-crossed by the TLV3501 high-speed comparator, generating a square wave signal synchronized with the zero-crossing point, which is then used as the third switching transistor. and the fourth switching transistor The reference drive enables synchronous rectification. The reason for using a zero-crossing voltage signal is that, in the forward direction, the Class E circuit is equivalent to a constant voltage source, and the switching of the rectifier diode at the zero-crossing voltage point can minimize losses.

[0097] Zero-voltage conduction margin detection is based on the first and second switching transistors. , Overall drain-source voltage (Switching transistor) When the circuit is in operation for an extended period, the on-state voltage drop is very small; this voltage actually reflects... The drain-source voltage (DLV) is divided and fed into a comparator. The comparator output is connected to the input capture port of the microcontroller. The microcontroller determines the drain-source voltage. Whether the drain-source voltage has dropped to near zero before the rising edge of the drive signal determines whether the system is currently in a hard-switching or underresonant state. The circuit is not at zero and is in a hard-switching state, requiring control of the second switching transistor. phase shift angle Make the second switching transistor The drive signal is delayed to extend the discharge time; conversely, it is in an under-resonant state and needs to be turned on ahead of time to shorten the charging time and reduce voltage stress.

[0098] Due to electromagnetic interference and comparator jitter in actual designs, if the goal is solely to "detect zero voltage at the rising edge of the drive," any minute disturbance could cause the actual turn-on time to fall before the drain-source voltage has completely dropped to zero, resulting in hard switching. Therefore, this design introduces a soft-switching margin. The following section explains the phase shift judgment conditions of the system:

[0099] 1) Soft switching state

[0100] Before the rising edge of the drive signal arrives, the drain-source voltage of the switching transistor has already dropped to near zero, and a preset soft-switching margin is maintained between the near-zero state and the rising edge of the drive signal; simultaneously, the peak drain-source voltage during turn-off is normal, without obvious voltage spikes or bounces. At this point, the drain-source voltage has returned to zero and the soft-switching margin is within a reasonable range, indicating that:

[0101] The energy in the parallel resonant capacitor of the switching transistor has been fully released;

[0102] The switching transistor can achieve zero-voltage conduction;

[0103] The reserved soft-switching margin is quite appropriate.

[0104] Therefore, the current phase shift angle is maintained, and no leading or lagging adjustments are made.

[0105] 2) Lag adjustment

[0106] When the rising edge of the drive signal arrives, it is detected that the drain-source voltage of the switching transistor has not yet dropped to near zero; or although the drain-source voltage is close to zero, its zeroing time is too close to the rising edge of the drive signal, and the preset soft-switching margin has not been reached.

[0107] This indicates that the energy in the parallel resonant capacitor of the switching transistor has not been fully released, and the switching transistor still bears a certain drain-source voltage when it is turned on. The system is in a hard-switching state or close to a hard-switching state.

[0108] At this point, the drive signal should be delayed to delay the switching transistor's conduction, extend the discharge time of the resonant capacitor, ensure that the drain-source voltage drops to near zero before conduction, and form sufficient soft-switching margin.

[0109] 3) Proactive adjustment

[0110] The drain-source voltage of the switching transistor has dropped to near zero prematurely before the drive signal arrives, and the duration of the near-zero state significantly exceeds the preset soft-switching margin; or the peak value of the drain-source voltage is detected to be too high during the turn-off period of the switching transistor, with obvious spikes or voltage rebound.

[0111] This indicates that the current drive timing is too lagging, or the system is in an underresonant state. Although the drain-source voltage may already be zero when the drive rises, the peak voltage across the switch is high, resulting in significant voltage stress on the device, and therefore it is not the optimal soft-switching state.

[0112] At this point, the drive signal should be appropriately advanced to shorten the turn-off charging and discharging process of the switching transistor, reduce the peak drain-source voltage and voltage stress, and restore the soft-switching margin to a reasonable range.

[0113] Figure 4 As shown An adaptive load phase-shift control strategy based on optimal load resistance is employed. When the drive signal goes high, if the energy stored in the transmitter's drive switch has not yet been fully released (the corresponding voltage value is not zero), the system will enter a hard-switching state. To ensure the switch has sufficient energy release margin, the drive signal needs to be delayed by a corresponding conduction angle (if it's forward transmission, then delayed by...). If it is reverse transmission, then there will be a lag. This extends the discharge time of the parallel resonant capacitor of the switching transistor, ensuring that the voltage across the switching transistor drops to zero before conduction, thereby effectively improving the soft-switching range of the load.

[0114] Figure 5 As shown The adaptive load phase-shift control strategy is employed. At this point, the peak voltage stored in the parallel resonant capacitor of the switching transistor is too high (i.e., the detected voltage value is greater than the preset value), and the switching transistor is in an under-resonant state. To ensure sufficient energy release time for the switching transistor, the drive signal needs to be controlled to conduct with a leading conduction angle (if it is forward transmission, then leading). If it is a reverse transmission, then it is ahead. This reduces the peak voltage when the switching transistor is turned off, effectively reducing the voltage stress on the device.

[0115] Figure 6 As shown An adaptive mutual inductance phase-shifting control strategy (optimal coupling mutual inductance). When the coupling distance decreases, the mutual inductance... As the resistance increases, the equivalent resistance increases, and the detected voltage changes accordingly. When the signal is consistent, the system enters a hard-switching state. To ensure the switching transistor has sufficient energy release margin, the drive signal needs to be controlled to lag the conduction angle (if it is forward transmission, then lag). If it is reverse transmission, then there will be a lag. This extends the discharge time of the parallel resonant capacitor, ensuring that the voltage across the switching transistor drops to zero before it is turned on, thereby effectively improving the soft-switching range of the load.

[0116] Figure 7 The figure shows M < M opt An adaptive mutual inductance phase-shifting control strategy. As the coupling distance increases, the mutual inductance... As the resistance decreases, the equivalent resistance decreases, and the detected voltage at this point is similar to that of R. o <R opt Consistent. At this point, the system enters an underresonant state. The drive signal needs to be controlled to lead the conduction angle (if it's forward transmission, then lead). If it is a reverse transmission, then it is ahead. This provides sufficient energy release margin for the switching transistor, reduces the peak voltage when the switching transistor is turned off, and effectively reduces the voltage stress on the device.

[0117] In the proposed control strategy, the design of the soft-switching margin is particularly critical: if the soft-switching margin is too small, the switching transistor will not be able to fully release its energy before the drive signal arrives, and the system will enter a hard-switching state, significantly increasing switching losses; if the soft-switching margin is too large, the drain-source voltage amplitude of the switching transistor will passively increase, leading to increased device voltage stress and affecting system reliability. Therefore, the soft-switching margin is set to 0.08.

[0118] Under the premise of ensuring zero-voltage conduction, the system further implements power-efficiency dual closed-loop cascade control. The outer power loop first determines whether the current output power meets the set requirements. If the output power is insufficient, the controller adjusts the phase shift angle to gradually bring the system into the operating range that meets the power requirements.

[0119] Within the feasible power range, the efficiency inner loop further compares the transmission efficiency at different phase shift angles. When the adjusted efficiency improves, the search continues in the current direction; when the efficiency decreases, the search direction is changed and the adjustment step size is reduced. The PI controller is used to smooth the phase shift angle adjustment process, reducing repeated adjustments caused by efficiency fluctuations, and gradually stabilizing the system near a higher efficiency operating point.

[0120] During the search for the optimal transmission point, the system synchronously detects the voltage. The near-zero state. After voltage division, the signal is connected to a high-speed comparator, and the comparator output signal is sent to the microcontroller's input capture port. The program determines whether the switching transistor meets the zero-voltage turn-on condition based on the feedback edge.

[0121] When the output power meets the requirements, the transmission efficiency stabilizes, and the ZVS condition is met, the system maintains the current phase shift angle. If the load, coil spacing, or mutual inductance changes, the controller restarts the search process, allowing the system to re-enter a working state that meets the power constraints and has higher efficiency under the new operating conditions.

[0122] The control logic for reverse transmission is symmetrical to that for forward transmission, but the key sampling objects are different. In open-loop reference drive, the third switch on the primary side... Normally on, fourth switching transistor With frequency kHz (corresponding angular frequency) ) and reverse duty cycle Chopper; the secondary side collects data from the first coil. The current is detected at zero-crossing point by a comparator to obtain a square wave signal synchronized with the current's zero-crossing point, which serves as the first switching transistor. Second switching transistor The reference drive. This is because the E# circuit is equivalent to a constant current source when reversed, and the rectifier diode needs to switch at the current zero crossing to achieve natural commutation.

[0123] Zero-voltage conduction margin detection is based on the third and fourth switching transistors. , Overall drain-source voltage (Switching transistor) When the circuit is in operation for an extended period, the on-state voltage drop is very small; this voltage actually reflects... The drain-source voltage (DLV) is divided and fed into a comparator. The comparator output is connected to the input capture port of the microcontroller. The microcontroller determines the drain-source voltage. Whether the drain-source voltage has dropped to near zero before the rising edge of the drive signal determines whether the system is currently in a hard-switching or underresonant state. The circuit is not at zero and is in a hard-switching state, requiring control of the fourth switch. phase shift angle Make the fourth switch transistor The drive signal is delayed to extend the discharge time; conversely, it is in an underresonant state and needs to be turned on ahead of time to shorten the charging time and reduce voltage stress.

[0124] Under the premise of ensuring zero voltage conduction, the system further implements power-efficiency dual closed-loop cascade control, which is the same as the forward process and will not be described here.

[0125] Based on existing literature on E# circuits (research on bidirectional E# type WPT systems with wide load range capabilities), the design of component parameters needs to satisfy the aforementioned E# circuit resonance conditions. This embodiment sets... Then there is Compensating inductor L t1 Compensation capacitor C t1 The input resonant frequency between It is usually taken as the power frequency of the circuit. of Times:

[0126] (16)

[0127] Compensation capacitor The expression is:

[0128] (17)

[0129] Compensating inductor The expression is:

[0130] (18)

[0131] Resonant capacitor The expression is:

[0132] (19)

[0133] Where Q is the quality factor, and Q=5 is set.

[0134] Resonant inductor The expression is:

[0135] (20)

[0136] Based on existing literature on Class E circuit derivations (Parameter Analysis and Design of Magnetic Resonance Wireless Transmission System for ZVS Class E Inverters), for the parameter design of Class E circuits, when the system achieves ZVS, the duty cycle of the switching transistor is set to 0.5, resulting in the following system parameters:

[0137] Compensating inductor The range of values ​​for is:

[0138] ,(twenty one)

[0139] Compensation capacitor Represented as:

[0140] .(twenty two)

[0141] Resonant capacitor Represented as:

[0142] .(twenty three)

[0143] Resonant inductor The range of values ​​for is:

[0144] .(twenty four)

[0145] To verify the output characteristics of the proposed system and the rationality of the parameter design method, the operating frequency was set to 200kHz and the duty cycle to 0.5. The component parameters were designed according to the above formula, and the main component parameters of the system are shown in Table 1. An experimental platform was then built.

[0146] Table 1 Parameters of the main components of the system

[0147]

[0148] Figure 8 To illustrate the relationship between system power, transmission efficiency, and load during forward transmission, the forward input voltage is 30V, the optimal coupling coefficient k=0.4, and the load varies from 5-30Ω. Figure 8 It can be seen that the system output efficiency first increases and then decreases with increasing load, and the optimal efficiency point is reached at the lowest load. The maximum power point load was reached at 85.32%, corresponding to an input power of 57.27W. The efficiency is 80.8%, corresponding to an efficiency of 98.34W.

[0149] Figure 9 This describes the relationship between system power, transmission efficiency, and load during reverse transmission, with a reverse input voltage of 30V, an optimal coupling coefficient k=0.3, and a load varying from 1-10Ω. Figure 9 It can be seen that the system output efficiency first increases and then decreases with increasing load, and at the optimal output load... The maximum power point load reached 84.6%, corresponding to an input power of 53.61W; The time efficiency is 81.63%, corresponding to an efficiency of 69.8W.

[0150] To verify the constant voltage characteristic of the circuit during forward operation, the relationship between the output voltage and the load was tested with an input voltage of 10V. The results are as follows: Figure 10 As shown. From Figure 10 It can be seen that when the load is in a low resistance range ( When the load increases, the output voltage increases almost linearly; when the load is in a high resistance range ( When the voltage rises, the upward trend gradually slows down and stabilizes at around 7V. This verifies that the proposed circuit achieves a relatively stable constant voltage output within a certain resistance range.

[0151] To verify the constant current characteristic of the circuit when it operates in reverse, the relationship between the output current and the load was tested with an input voltage of 10V. The results are as follows: Figure 11 As shown. From Figure 11 It can be seen that when the load is in a low resistance range ( When the load increases, the output current fluctuates relatively little, decreasing slightly around 2A. According to the power calculation formula, the output current stabilizes within a certain range as the load increases. Only when the output power reaches its peak does the output current become unstable and gradually decrease. This verifies that the proposed circuit achieves relatively stable constant current output within a certain resistance range.

[0152] Figure 12 for The drain-source voltage waveform of the switching transistor at (optimal load resistance) is shown below. The voltage across the primary-side switching transistor. This is the drive voltage for the primary-side switching transistor. From... Figure 12 It can be seen that the circuit is in a hard-switching state at this time. The instant the switching transistor changes from off to on, the voltage stored in the parallel resonant capacitor C of the switching transistor... f1 The energy in the circuit is not fully released, which increases switching losses and reduces system efficiency. The proposed control strategy achieves... Figure 13 The voltage waveform across the primary-side switch after phase-shift control is shown. Figure 13 It can be seen that the drive signal is delayed in conduction, which causes the drain-source voltage of the switching transistor to be higher before it is turned on. The value has been reduced to 0. After correction, the circuit operates in soft-switching mode, achieving ZVS.

[0153] Figure 14 for The waveform of the drain-source voltage of the switching transistor is shown. At this time, the circuit is in an underresonant state, the peak voltage of the switching transistor increases, and the voltage stress on the device increases. Through the proposed control strategy, the following is obtained: Figure 15 The voltage waveform across the primary-side switch is shown after phase-shift control. The drive signal leads the switching, providing sufficient energy release margin for the switch. The corrected circuit operates in soft-switching mode, achieving ZVS.

[0154] Figure 16 for The drain-source voltage waveform of the switching transistor. Due to the reduced coupling distance, the mutual inductance... As the resistance increases, the equivalent resistance also increases. At this point, the circuit enters a hard-switching state. To ensure the switching transistor has sufficient energy release margin, the drive signal needs to be delayed during conduction. The proposed control strategy achieves this. Figure 17 The voltage waveform across the primary-side switch after phase-shift control is shown. The drive signal is delayed, resulting in a lower drain-source voltage across the switch before it is turned on. The value has been reduced to 0. After correction, the circuit operates in soft-switching mode, achieving ZVS.

[0155] Figure 18 for The drain-source voltage waveform of the switching transistor. Due to the increased coupling distance, the mutual inductance... As the voltage decreases, the equivalent resistance decreases. At this point, the circuit enters an underresonant state. The peak voltage of the switching transistor increases, increasing the voltage stress on the device. Through the proposed control strategy, the following is obtained: Figure 19 The voltage waveform across the primary-side switch is shown after phase-shift control. The drive signal leads the switching, providing sufficient energy release margin for the switch. The corrected circuit operates in soft-switching mode, achieving ZVS.

[0156] In summary, this invention addresses the inherent limitations of traditional single-transistor bidirectional WPT systems, which suffer from limited output characteristics and difficulty in simultaneously meeting constant voltage and constant current requirements. It proposes a composite E#-Class E bidirectional wireless power transfer system. Through the coordinated configuration of E# and Class E circuits and a source-to-source switching transistor structure, the system simultaneously provides bidirectional power supply capabilities with both forward constant voltage and reverse constant current outputs without adding an additional converter. Furthermore, this paper proposes an adaptive phase-shift control strategy, which adjusts the forward phase shift angle... and reverse phase shift angle Real-time adjustment enables the system to maintain zero-voltage conduction of the switching transistor within a wide load range (5~30 Ω) and a wide coupling coefficient range (0.1~0.6), effectively suppressing hard switching losses and voltage stress. Simultaneously, based on ZVS (Zero Voltage Switching), power-efficiency dual closed-loop cascade control is implemented, achieving optimal efficiency tracking while ensuring output power. Test results on a 100 W experimental platform show that the highest efficiency reaches 85.3% in the forward transmission and 84.6% in the reverse transmission, with clearly defined constant voltage and constant current output ranges, and ZVS is achieved under varying load and mutual inductance. This invention provides a feasible technical solution for low-to-medium power bidirectional wireless charging scenarios (such as electric vehicle auxiliary batteries, drones, and medical devices) that is structurally simple, flexible in control, and offers switchable output characteristics.

[0157] The above embodiments are preferred embodiments of the present invention, but the embodiments of the present invention are not limited to the above embodiments. Any changes, modifications, substitutions, combinations, or simplifications made without departing from the spirit and principle of the present invention shall be considered equivalent substitutions and shall be included within the protection scope of the present invention.

Claims

1. A composite bidirectional WPT system with constant current and constant voltage output characteristics, characterized in that, This includes Class E single-transistor circuits, coupled resonant networks, and Class E single-transistor circuits; the Class E single-transistor circuit includes a first DC power supply. First resonant inductor First coupling capacitor First resonant capacitor First switching transistor Second switching transistor ; The coupled resonant network includes a first coil. Second coil The Class E single-transistor circuit includes a second DC power supply. Second resonant inductor Second resonant capacitor Second coupling capacitor Third switching transistor and the fourth switching transistor The first to fourth switching transistors are all NMOS transistors. First resonant inductor First coupling capacitor First resonant capacitor Sequentially connected in series with the first DC power supply Between the positive and negative terminals; the first coil Parallel to the first coupling capacitor The two ends; the first switching transistor and the second switching transistor After being connected in series, they are all connected in parallel to the first resonant capacitor. The two ends; the first switching transistor The source is connected to the second switch. The source of the first switch transistor The drain is connected to the first coupling capacitor. With the first resonant capacitor The common terminal of the second switch transistor The drain is connected to the first DC power supply. The negative electrode; Second resonant inductor Second resonant capacitor Sequentially connected in series with the second DC power supply Between the positive and negative terminals; the second coil and the second coupling capacitor After being connected in series, they are all connected in parallel to the second resonant capacitor. The two ends; the third switch transistor and the fourth switching transistor After being connected in series, they are all connected in parallel to the second resonant capacitor. The two ends; the third switch transistor The source is connected to the fourth switch. The source of the third switch. The drain of the second resonant inductor is connected. With the second resonant capacitor The common terminal of the fourth switch transistor The drain is connected to the second DC power supply. The negative electrode.

2. The composite bidirectional WPT system with constant current and constant voltage output characteristics according to claim 1, characterized in that, The parameter settings of the E# class single-transistor circuit satisfy the following relationship: ， in, The operating angular frequency of the system. Is with The corresponding frequency, The AC load resistor at the system receiver. For quality factor, To compensate for inductance Compensation capacitor The input resonant frequency between It is a multiple of the frequency.

3. The composite bidirectional WPT system with constant current and constant voltage output characteristics according to claim 2, characterized in that, First coil satisfy: 。 4. The composite bidirectional WPT system with constant current and constant voltage output characteristics according to claim 2, characterized in that, The parameter settings of the Class E single-transistor circuit satisfy the following relationship: 。 5. The composite bidirectional WPT system with constant current and constant voltage output characteristics according to claim 4, characterized in that, The second coil satisfy: 。 6. The composite bidirectional WPT system with constant current and constant voltage output characteristics according to any one of claims 1 to 5, characterized in that: During forward transmission of the system, the first switch transistor The second switch is kept normally open. For frequency Duty cycle The PWM wave drive signal; the second coil is acquired. The voltage across the terminals is compared at zero-crossing points by an operational amplifier to obtain a square wave signal, which drives the third switch. and the fourth switching transistor Synchronous rectification.

7. The composite bidirectional WPT system with constant current and constant voltage output characteristics according to claim 6, characterized in that: During system reverse transmission, the third switch transistor The fourth switch is kept normally open. For frequency Duty cycle The PWM wave drive signal; the first coil is acquired. The current is compared with zero to obtain a square wave signal, which drives the first switch transistor. and the second switching transistor Synchronous rectification.

8. The composite bidirectional WPT system with constant current and constant voltage output characteristics according to claim 6, characterized in that: During forward transmission, the first switch transistor is sampled. and the second switching transistor Overall drain-source voltage and with the second switching transistor The drive signals are compared to determine the second switching transistor. If zero-voltage turn-on is achieved within the preset zero-voltage turn-on margin, then the second switching transistor is fixed. The driving signal is either delayed or advanced; otherwise, it is adjusted either by lag or lead.

9. The composite bidirectional WPT system with constant current and constant voltage output characteristics according to claim 6, characterized in that: During reverse transmission, the third switch transistor is sampled. and the fourth switching transistor Overall drain-source voltage and with the fourth switching transistor The drive signals are compared to determine the fourth switch. If zero-voltage conduction is achieved within the preset zero-voltage conduction margin, then the fourth switch is fixed. The driving signal is either delayed or advanced; otherwise, it is adjusted either by lag or lead.

10. The composite bidirectional WPT system with constant current and constant voltage output characteristics according to claim 1, characterized in that, The system performs power-efficiency dual closed-loop cascade control: The power outer loop determines the current output power; if it is insufficient, the phase shift angle is adjusted. The efficiency inner loop compares the transmission efficiency under different phase shift angles and determines the search direction and step size based on the comparison results; the PI controller adjusts the phase shift angle; the near-zero state of the drain-source voltage of the switching transistor is detected synchronously; when the output power meets the requirements, the transmission efficiency is stable, and the zero-voltage conduction condition is met, the current phase shift angle is maintained; if the load, coil spacing, or mutual inductance changes, the search is restarted.

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