Inductive proximity switch with reduced temperature dependence

Abstract

Inductive proximity switch (1) with a transmitting coil (2), two anti-serially connected receiving coils (3) for generating a receiving signal and a calibration coil (4) which is coupled to one of the receiving coils (3) by means of a transformer, wherein the calibration coil (4) is connected to a controllable resistor (9) which is part of a control loop which regulates the receiving signal to zero, characterized in that the circuit of the calibration coil (4) has a current measuring unit (10).

Description

The invention relates to an inductive proximity switch according to the features of the preamble of claim 1 and of the method according to claim 2. inductive proximity switches are used as contactless electronic switching devices, especially in automation technology.They contain a transmitting coil which generates an electromagnetic magnetic field that can be influenced by a metallic trigger. The influence of the magnetic field by the metallic trigger is evaluated and an electronic switching stage is actuated when a threshold value is exceeded.Switching devices of this type are manufactured and sold in a wide variety of designs, including those by the applicant.In this case, both the activation of the transmitting coil and the evaluation of the influence of the metallic trigger can take place in different ways.In many cases, the transmitting coil is a component of an oscillator which is influenced by the metallic trigger and whose amplitude and / or frequency change is evaluated.In addition to the widespread sinusoidal control of the transmitting coil and the evaluation of frequency and amplitude changes, control with a short rectangular pulse is known. In this case, the transmitting coil is not part of an oscillator, but rather is supplied with strong voltage or current pulses. The echo triggered by the eddy currents induced in the metallic trigger is evaluated. This evaluation can be carried out directly at the transmitting coil as well as at a magnetically coupled receiving coil.Since this is a closed resonant circuit or a closed coil circuit, only little energy is radiated. The interaction with the metallic trigger (target) is limited to the near field. It decreases approximately 3.5 times the power of the switching distance.In order to still be able to detect slight interactions with the metallic trigger, it is advantageous to compensate the signal in the unaffected state and to evaluate only the changes caused by the trigger.For this purpose, preferably two receiving coils are operated in differential circuit. The design is chosen such that one of the two coils is more strongly influenced by the target than the other.By zeroing in the unaffected state, an extremely sensitive arrangement is obtained, which is also referred to as a differential transformer (LVDT=linear variable differential transformer).The differential transformer is balanced in such a way that the signals of the two receiving coils cancel one another out in the unaffected state.The better this adjustment is achieved, the higher the sensor signal can be amplified without any over-control occurring. Since the magnetic field decreases very rapidly with increasing distance, one soon comes into regions where the temperature response, in particular of the copper windings, but also of the other materials and components involved, produce effects of the order of magnitude of the sensor signal to be expected. Therefore, higher switching intervals can only be achieved if the temperature dependence of the arrangement over the operating temperature range can be compensated.Since this equilibrium can be disturbed during production, but also by the installation situation, subsequent balancing of the differential transformer is desirable both in the factory and in later operation.In DE 10 2007 014 343 A1, it is proposed to connect a receiving coil to trimmable resistors, with the aid of which the switching distance is adjusted. These are controlled by an evaluation unit (microcontroller), which is also connected to a temperature sensor, so that temperature compensation can be carried out.It is considered disadvantageous here that the temperature compensation consists of a relatively long process from the measurement by the temperature sensor, the calculation of the correction values in the microcontroller, the subsequent digital output of the correction values and the digital-to-analog conversion until the trimmable resistors are actuated. Linearization of, for example, -25° C. to +70° C. appears problematic in view of this long chain.WO 2007 / 012502 A1 proposes a control loop with a controlled system comprising the sensor. Controlled variable is, inter alia, the amplitude of the sensor circuit.The regulation takes place at the transmitter end, i.e. the amplitude is kept constant against the attenuation disturbance variable.The energy required for adjusting the amplitude is supplied via an adjustable resistor. The control variable for this resistance is decoupled as a variable for the instantaneous attenuation and thus as a measured variable.A disadvantage here is that the adjustable resistor, preferably a transistor, has both a temperature characteristic and a non-linear characteristic curve.EP 0 058 076 A2 discloses a metal detector device using an oscillator coil and detector coils with compensation signals and phase sensitive detectors to correct drifts and maintain a balanced state. Such an approach, while effective for metal detection, may be complex, which may result in slower response times and a higher number of components. Therefore, it is desirable to provide a simpler and more efficient method that reduces complexity and improves responsiveness.EP 0 304 272 A2 discloses an inductive proximity switch in which a measuring coil and a reference coil are connected in series. However, such designs may be sensitive to imbalance between the coils, which may reduce sensitivity and accuracy. Therefore, it is advantageous to provide an inductive proximity sensor that minimizes the effects of coil imbalance.DE 10 2010 002 201 A1 discloses an inductive proximity switch in which the control circuit and the receiving circuit are decoupled in order to reduce undesired interactions. However, since the control current of the control transistor flows through the receiving coil, temperature dependencies may occur. Therefore, it is advantageous to provide an inductive proximity switch that reduces temperature dependence.The object of the invention is to further improve this method. The characteristic curve is to be linearized and the temperature dependence is to be reduced.This object is achieved according to the features of claims 1 and 2.The essential idea of the invention is to measure the manipulated variable directly instead of its temperature-related control variable. Thus, the resistance of the actuator or its load current and not its control voltage is measured.The essential advantage is that the temperature response of the controlled system can be permitted, because the control variable is measured directly on the control element in the form of the damping resistance.Since the actuator is generally a transistor, neither its control voltage nor its control current, but rather the path resistance or the load current should be measured and output.The invention is explained in more detail below with reference to the drawings.The following are shown: FIG. 1 is a schematic diagram of an inductive proximity switch according to the invention, FIG. 2 shows an embodiment with a field effect transistor as an actuator.FIG. 1 shows an inductive proximity switch 1 with a transmitting coil 2, two receiving coils 3, a balancing coil 4 and a further winding 5 symbolizing the switching flag or the target.The transmitting coil 2 is supplied by a high-frequency generator 6. The two antiseries-connected receiving coils 3 are connected to the synchronous rectifier 7. The rectified signal passes through the low-pass filter 8 and controls the variable resistor 9, which extracts energy from the system via the balancing coil 4 until the signal at the synchronous rectifier 7 becomes zero, i.e. the balance between the target 5 and the balancing coil 4 is established.The control current flowing through the controllable resistor 9 (control element) located in the circuit of the adjusting coil 4 is a measure of the damping. It is measured by the current measuring unit 10 and can be output as an analog signal.Finally, it should be noted that a structure according to the invention can also be realized with two transmitting coils and one receiving coil.FIG. 2 shows a detailed exemplary embodiment. The sinusoidal transmit signal is generated by the 74HC4053 dual channel analog multiplexer MUX1. For the frequency f, the relationship: f=1 / (2π×√LS×(C 1×C 2) / (C 1+C 2)) holds.The transmitting coil LS is connected to the two receiving coils L 1 and L 2 via the transformer coupling factors M 1 and M 2. These are adjusted so that the voltages generated in the receiving windings L1 and L2 are equal in the unloaded state.The center tap of the two receiving coils is at half the operating voltage. The other ends of the two antiseries-connected receiving coils are connected to the inputs of the operational amplifier OV1. Because of the lack of negative feedback, this operates as a comparator and a pulse shaper.As the balance is disturbed by reducing the resistance Rx connected to the target winding 5 to be symbolically understood, the output of OV1 supplies a square wave signal phase-synchronous with the oscillator. Comparator OV2 also provides a square wave signal phase synchronous with the oscillator. The multiplexer MUX2 operates as a phase sensitive rectifier. The operational amplifier OV3 integrates the signal supplied by the MUX2 and thus controls the transistor T1, a junction FET. Transistor T1 corresponds to the controllable resistor 9 in FIG. 1 and draws so much energy from the differential transformer until equilibrium is restored. The measured variable is the drain-source resistance R DS of T1 or the drain current I DS.The measurement is carried out using the operational amplifier OPV 4, which embodies the current measurement unit 10 mentioned in FIG. 1.For DC voltage, its amplification is Vu=-RA / Ra*, where Ra*=Ra+R Cu+ R DS.(RCu: copper resistance of the coil, RDS: drain-source resistance RDS of T1)Ri and Ci determine the integration time. For DC voltage they are of no concern. The direct voltage present across the discharge capacitor C HF appears around the above-mentioned value. This is amplified by a factor at the output of OPV4 as the useful signal. Neglecting the copper resistance of the coil, one obtains: R DS( T1)=Uref×RA / UA-Ra.The following applies to the output voltage UA: UA=Uref×RA / (R DS+ Ra).For the current, one obtains: I DS( T1)=(UA-Uref) / RA.List of reference characters1 Inductive proximity switch 2 transmitting coil 3 receiving coils 4 balancing coil, balancing coil 5 target winding, symbolizing the target 6 high-frequency generator 7 synchronous rectifier 8 low-pass filter 9 controllable resistor 10 current measuring unit

Claims

Inductive proximity switch (1) having a transmitting coil (2), two antiseries-connected receiving coils (3) for generating a received signal, and a balancing coil (4) which is transformer-coupled to one of the receiving coils (3), wherein the balancing coil (4) is connected to a controllable resistor (9) which is part of a control loop which regulates the received signal to zero, characterized in that the circuit of the balancing coil (4) has a current measurement unit (10).Method for operating an inductive proximity switch according to Claim 1, characterized in that the current flowing in the balancing coil (4) is measured and evaluated.

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