Amplifier circuit and communication device for transmitting an RF output signal

Abstract

Amplifier circuit (300; 400) having the following features: a power amplifier (110) configured to amplify an RF input signal (105) based on a supply voltage to obtain an RF output signal (115); and an operating point controller (320; 420) configured to control an operating point of the power amplifier (110), wherein the operating point control (320; 420) has the following features: an impedance determination device (322; 422) configured to determine a measure of a load impedance of a load coupled to an output of the power amplifier (110); and a DC-DC converter (326; 430) configured to adjust the supply voltage of the power amplifier (110) characterized by the fact that the operating point controller (320; 420) further comprises a mapping unit (324) configured to map the measure of the load impedance to a mapped voltage such that the mapped voltage is based on a phase of a reflection factor determined by the load impedance; wherein the DC-DC converter (326; 430) is configured to adjust the supply voltage of the power amplifier (110) based on the mapped voltage; and wherein the mapping unit (324) is configured to provide the mapped voltage such that an adjacent channel leakage power ratio value of the RF output signal (115) is within a predetermined range for a plurality of load impedances resulting in reflection factors of equal magnitude.

Description

Embodiments of the invention relate to an amplifier circuit and a mobile communication device for adjusting an operating point of a power amplifier. Further embodiments of the invention relate to power amplifier operating point optimization based on a measured antenna impedance.Conventional methods for adjusting an operating point of a power amplifier include, for example, methods for reducing an operating point current of the power amplifier based on a detected output power level. Such a reduction in operating point current is performed by using a conventional operating point control approach. Operating point control comprises, for example, a control of a driver for changing the supply voltage of the power amplifier on the basis of specific input parameters. Conventional mobile communication devices including such a work point control are based on wideband code division multiple access (WCDMA), for example. WCDMA describes a multiple access method, while Universal Mobile Telecommunications System (UMTS) is the standard based thereon.The generic US 2007 / 0 026 838 A1 describes an RF power amplifier circuit which measures an impedance deviation at the output of the circuit and correspondingly adjusts the operating parameters of the power amplifier in order to suppress distortions in the output signal.US 2004 / 0 075 494 A1 describes a power amplifier which adjusts the bias voltage of the power amplifier in accordance with a respective one of different output power stages.US 5 216 379 A describes a power amplifier. The impedance of a load may be either determined by the power amplifier or may be predetermined by the user. The bias voltage of the power amplifier is adjusted according to the measured or input impedance value.US 6 018 650 A relates to a communication device for a cellular communication network including a gain control circuit which samples the output power level and adjusts the gain of the RF transmitter adjusting based on the digitized sampled output power level so that the output power level of the RF transmitter approaches a predetermined output power level. The gain may also be adjusted in response to the received RSSI to conserve the battery.JP 2006-319 508 A discloses a compact RF power amplifier with excellent linearity, which is independent of the impedance of the load. The power amplifier includes a plurality of serially coupled amplifier stages coupled to a matching network that detects a voltage swing in the output signal of the amplifier stages and correspondingly adjusts the gain of the input stage of the power amplifier.The object of the present invention is to provide an amplifier circuit and communication apparatus for transmitting an RF output signal.The object is achieved by the features of the independent claims. Further developments are found in the dependent claims.Embodiments of the invention provide an amplifier circuit, the amplifier circuit comprising a power amplifier for amplifying an RF input signal to obtain an RF output signal, and a bias control for controlling a bias of the power amplifier. The operating point controller is configured to determine a measure of a load impedance of a load coupled to an output of the power amplifier and provide an operating point control signal for adjusting the operating point of the power amplifier based on the determination of the measure of the load impedance.Embodiments of the invention provide an amplifier circuit, wherein the amplifier circuit comprises a power amplifier for amplifying an RF input signal based on a supply voltage to obtain an RF output signal, and a bias controller for controlling a bias of the power amplifier. The operating point controller comprises an impedance determination device for determining a measure of a load impedance of a load coupled to an output of the power amplifier, and a mapping unit for mapping the measure of the load impedance to a mapped voltage n, such that the mapped voltage is based on a phase of a reflection factor determined by the load impedance. The operating point controller further comprises a DC-DC converter configured to adjust the supply voltage of the power amplifier on the basis of the depicted voltage. The mapping unit is configured to provide the mapped voltage such that a adjacent channel leakage power ratio (ACLR) value of the RF output signal is within a predetermined range for a plurality of load impedances resulting in reflection factors of equal magnitude.Embodiments of the invention provide an amplifier circuit, the amplifier circuit comprising means for amplifying an RF input signal to obtain an RF output signal and means for controlling an operating point of the power amplifier. The means for controlling is configured to determine a measure of a load impedance of a load coupled to an output of the means for amplifying and to provide a working point control signal for adjusting the working point of the means for amplifying based on the determination of the measure of the load impedance.Embodiments of the invention provide a mobile communication device for transmitting an RF output signal. The mobile communication device includes a baseband generator for generating a baseband signal, and an RF signal generator for generating an RF signal based on the baseband signal. The apparatus further comprises a power amplifier for receiving the RF signal from the RF signal generator and amplifying the received RF signal to obtain an RF output signal, and a bias controller for controlling a bias of the power amplifier. The operating point controller is configured to determine a measure of a load impedance of a load coupled to an output of the power amplifier and provide an operating point control signal to adjust the operating point of the power amplifier based on the determination of the measure of the load impedance. The communication device further comprises an antenna for transmitting the RF output signal.Preferred exemplary embodiments of the present invention are explained in more detail below with reference to the attached drawings. The following are shown: FIG. 1 is a block diagram of an embodiment of an amplifier circuit including a bias control; FIG. 2 is a block diagram of another embodiment of an amplifier circuit that includes operating point control with various look-up tables; FIG. 3 shows a block diagram of a further exemplary embodiment of an amplifier circuit which comprises a working point controller having an impedance determination device, a mapping unit and a DC-DC converter; FIG. 4 is a block diagram of another embodiment of an amplifier circuit that includes a bias control with an impedance information look-up table; FIG. 5 shows a graph of an exemplary dependence of a adjacent channel leakage power ratio (ACLR) value on a phase of a reflection factor; FIG. 6 shows a graph of an exemplary dependence of a supply voltage on a phase of a reflection factor; FIG. 7 shows a graph of an exemplary dependence of a battery current on a phase of a reflection factor; FIG. 8 is a block diagram of another embodiment of an amplifier circuit including a bias controller for receiving frequency information; and FIG. 9 is a block diagram of an embodiment of a mobile communication device including the embodiment of the amplifier circuit of FIG. 1.Hereinafter, operating conditions and requirements of some mobile communication devices to which the present invention can be applied will be described. Some embodiments according to the invention provide good performance under the conditions discussed below.Mobile terminals often have to cope with changing environmental conditions. The operating temperature range is usually between -10°C and 55°C (according to 3GPP), while the supply voltage ranges are usually between 3.0V and 4.3V. The latter is determined by the battery discharge characteristic and the voltage drop during a transmission operation. In addition, the radiated power of a mobile terminal greatly depends on the antenna conditions, e.g., free space, speech position (antenna covered by hand or close to the head). The different antenna conditions result in different load impedances acting at a power amplifier output. In some cases, the power amplifier may need to cope with a wide range of load impedances. Antenna mismatch causes, for example, problems of spectral growth (spectral growth) in transmission schemes with a non-constant envelope such as UMTS and LTE (long term evolution). Both standards have severe limits for leakage power on adjacent channels, so-called ACLR requirements (ACER=adjacent channel leakage power ratio).Therefore, there is a need for an approach to adjusting a power amplifier operating point that allows a balanced trade-off between improved ACLR performance, low power consumption, and computational complexity.Embodiments of the invention achieve the compromise just mentioned by measuring a load impedance of a load coupled to an output of the power amplifier and by adjusting a working point control signal of the power amplifier based on the determination of the load impedance measurement. In this way, it is possible to reduce ACLR degradation present in a mismatch, so that ACLR performance can be maintained at comparatively low cost and / or comparatively low power consumption.FIG. 1 shows a block diagram of an amplifier circuit 100 comprising a bias controller 120. As shown in FIG. 1, the amplifier circuit 100 includes a power amplifier 110 and a bias controller 120. Here, the power amplifier 110 is configured to amplify an RF input signal 105 to obtain an RF output signal 115. In addition, the operating point controller 120 is configured to control an operating point of the power amplifier 110. The RF input signal 105 may have a plurality of frequencies in specific frequency bands, such as defined by the UMTS standard (or may be switchable between a plurality of frequencies). Referring to the embodiment of FIG. 1, the operating point controller 120 is configured to measure a measure of a load impedance of a load coupled to an output of the power amplifier 110 and provide an operating point control signal 125 for adjusting the operating point of the power amplifier 110 based on the determination of the measure of the load impedance. Such a measure of the load impedance is, for example, a quantity that depends on the load impedance. The RF output signal 115 obtained at the output of the amplifier circuit 100 represents an amplified version of the RF input signal 105.FIG. 2 shows a block diagram of an amplifier circuit 200 that includes a bias controller 220 with various look-up tables (224-1, 224-2...). Here, the amplifier circuit 200 of FIG. 2 has substantially the same blocks as the amplifier circuit 100 of FIG. 1. Moreover, the operating point controller 220 of the amplifier circuit 200 shown in FIG. 2 may correspond to the operating point controller 120 of the amplifier circuit 100 shown in FIG. 1. Referring to the embodiment of FIG. 2, the operating point controller 220 may include a look-up table 222 configured to store a plurality of operating point voltage values associated with respective values of a reflection factor for a plurality of load impedances, such that the operating point voltage values are dependent on a phase and amplitude (an amount) of respective reflection factors for reflection factors of a same amount (e.g., for reflection factors leading to the same SWV). The operating point controller 220 is configured to retrieve a single operating point voltage value from the look-up table 222, wherein the single operating point voltage value may correspond to a value of a reflection factor determined by the load impedance. Here, the operating point controller 220 is configured to provide the operating point control signal 125 to adjust the operating point of the power amplifier 110 based on the single operating point voltage extracted from the look-up table 222.It is further shown in FIG. 2 that the operating point controller 220 may include different look-up tables 224- 1, 224- 2,... for a plurality of frequencies of the RF input signal 105. Here, the various look-up tables (LUTI) 224- 1, 224- 2,... are with 'LUTI: Freq. 1', 'LUT2: Freq. 2'.., indicating that these different frequencies (Freq. 1, Freq. 2... ) of the RF input signal 105. The look-up tables 224-1, 224-2 may be considered as part tables of the look-up table 222, as shown in Figure 2. However, the lookup tables 224-1, 224-2 may alternatively be substituted for the lookup table 222. Referring to the embodiment of FIG. 2, each of the look-up tables (or sub-tables) 224- 1, 224- 2,... may be configured to store a plurality of operating point voltage values associated with corresponding values of a reflection factor for a plurality of load impedances such that the operating point voltage values are dependent on a phase of corresponding reflection factors for reflection factors of an equal amount.Additionally, the operating point controller 220 may be further configured to retrieve a single operating point voltage value from a selected look-up table associated with a current frequency of the RF input signal 105. In an embodiment, the single operating point voltage value corresponds to a phase of a reflection factor determined by the load impedance for the current frequency of the RF input signal 105. It should be noted that the current frequency of the RF input signal 105 is indicated in one embodiment by the frequency information 205 received from the operating point controller 220 having the various look-up tables or lookup sub-tables (224-1, 224-2...). In the embodiment of FIG. 2, the operating point controller 220 is configured to provide the operating point control signal 125 to adjust the operating point of the power amplifier 110 based on the single operating point voltage value extracted from the selected look-up table.FIG. 3 shows a block diagram of an embodiment of an amplifier circuit 300 comprising a working point controller 320 having an impedance determination device 322, a mapping unit 324 and a DC-DC converter 326. Here, the amplifier circuit 300 of FIG. 3 has substantially the same blocks as the amplifier circuit 100 of FIG. 1. Therefore, identical blocks having similar implementations and / or functions are denoted by the same reference numerals. Moreover, the operating point controller 320 and a power amplifier supply voltage 325, Vcc of the amplifier circuit 300 shown in FIG. 3 correspond to the operating point controller 120 and the operating point control signal 125 of the amplifier circuit 100 shown in FIG. 1. Referring to the embodiment of FIG. 3, the amplifier circuit 300 comprises a power amplifier 110 for amplifying an RF input signal 105 based on the power amplifier supply voltage 325, Vcc, to obtain an RF output signal 115. It can also be seen in FIG. 3 that the operating point controller 320 of the amplifier circuit 300 has an impedance determination device 322, a mapping unit 324 and a DC-DC converter 326. Here, the impedance determination device 322 is configured to determine a measure 321, r L, of a load impedance of a load coupled to an output of the power amplifier 110, for example in the form of a reflection factor FL. The mapping unit 324 is configured to map the measure 321 of the load impedance to a mapped voltage 323, Vramp, such that the mapped voltage Vramp depends on a phase and amplitude of the reflection factor F L determined by the load impedance. Here, r L is only a different representation of the load impedance than an S parameter that takes into account a predetermined reference impedance. DC-DC converter 326 is configured to adjust supply voltage 325 of power amplifier 110 based on depicted voltage 323. In the embodiment of FIG. 3, the mapping unit 324 is configured to provide the mapped voltage 323 such that a adjacent channel leakage power ratio (ACLR) value of the RF output signal 115 is within a predetermined range for a plurality of load impedances resulting in reflection factors of equal magnitude and different phase.FIG. 4 shows a block diagram of another embodiment of an amplifier circuit 400 that includes a bias controller 420 with an impedance information look-up table 424 (LUT B). As shown in FIG. 4, the amplifier circuit 400 includes a power amplifier 410, a bias controller 420, and a directional coupler 406. Here, the power amplifier 410 and the operating point controller 420 of the amplifier circuit 400 shown in FIG. 4 may correspond to the power amplifier 110 and the operating point controller 120 of the amplifier circuit 100 shown in FIG. 1. In the embodiment of FIG. 4, a baseband generator 402 (baseband generator 'BB') and an RF signal generator 404 ('RF signal generation') are also shown. In particular, the baseband generator 402 is configured to generate a baseband signal 403, s(t), while the RF signal generator 404 is configured to generate an RF signal 405 on the basis of the baseband signal 403, s(t). It can further be seen in FIG. 4 that the power amplifier 410 is configured to receive the RF signal 405 from the RF signal generator 404 in order to obtain an RF output signal 415. Here, the RF signal 405 received by the power amplifier 410 and the RF output signal 415 output by the power amplifier 410 as shown in the embodiment of FIG. 4 correspond to the RF input signal 105 received by the power amplifier 110 and the RF output signal 115 output by the power amplifier 110 as shown in the embodiment of FIG. 1. Moreover, the power amplifier 410 of FIG. 4 comprises a power amplifier unit 412 connected to an RF output circuit.The directional coupler 406 of the amplifier circuit 400 is coupled to the output of the power amplifier unit 412 or to the output of the RF output circuit such that it can be used to perform a reflection factor measurement depending on a load impedance.Referring to the embodiment of FIG. 4, the operating point controller 420 of the amplifier circuit 400 includes an impedance determiner 422, an impedance information lookup table 424 (LUT B), a lookup table 426 (LUT A), a first digital / analog converter (DAC) 428- 1, a second digital / analog converter (DAC) 428- 2, and a DC-DC converter 430. Here, the impedance determiner 422 is denoted by 'antenna impedance Ft determine l while the impedance information look-up table 424 and the look-up table 426 are denoted by 'LUT B: r L- store data l and 'LUT A: Vcq=f(TL); Vcc=f(TL)', respectively. In addition, the lookup table 426 of FIG. 4 may correspond to the lookup table 222 of FIG. 2, while the impedance determiner 422 and DC-DC converter 430 of FIG. 4 may correspond to the impedance determiner 322 and DC-DC converter 326 of FIG. 3.According to the exemplary embodiment of FIG. 4, the directional coupler 406 is configured to provide a measurement signal 407, which represents a complex load impedance, for example, and to forward the provided measurement signal 407 to the impedance determination device 422 of the operating point controller 420. The impedance determination device 422 can in turn be configured to determine a measure 421, r L, of the (complex-valued) load impedance, for example a complex-valued reflection factor FL. Here, the measure 421, FL of the load impedance obtained by the impedance determination means 422 as shown in FIG. 4 may correspond to the measure 321, r L, of the load impedance obtained by the impedance determination means 322 as shown in FIG. 3.In the embodiment of FIG. 4, the impedance information lookup table 424 of the operating point controller 420 may be configured to store a plurality of measures of load impedance for corresponding frequencies of the RF input signal 405 based on the measured impedance information 421, FL. In addition, the operating point controller 420 is configured to retrieve a single measure 425 of load impedance from the impedance information look-up table 424 (LUT B). Here, the single measure 425 of the load impedance may correspond to a frequency of the RF input signal 405, such as in a frequency hopping mode. In addition, the operating point controller 420 is configured to provide the operating point control signal to adjust the operating point of the power amplifier 410 based on the single measure 425 of the load impedance taken from the impedance information look-up table 424. Accordingly, a currently measured impedance value 421 may be used as the impedance measure 425 if the lookup table 424 does not have a stored previously measured impedance value, and otherwise a stored previously measured impedance value from the lookup table 424 may otherwise be used as the impedance measure 425.As shown in the embodiment of FIG. 4, the operating point controller 420 is configured to retrieve first and second digital operating point voltage values 427- 1, 427- 2 from the look-up table 426 based on the single measure 425 of the load impedance retrieved from the impedance information look-up table 424 or based on measured impedance information 421. In an embodiment, the first and second digital operating point voltage values 427- 1, 427- 2 represent voltages Vcc or Vcq derived from a functional dependence on a measure of the load impedance and the reflection factor, respectively.The first and second digital operating point voltage values 427- 1, 427- 2 taken from the look-up table 426 are converted into a first analog operating point voltage value 429- 1, Vramp and a second analog operating point voltage value 429- 2, Vcq by the first and second digital / analog converters 428- 1 and 428- 2, respectively. The DC-DC converter 430 of the operating point controller 420 is configured to adjust the supply voltage 435, Vcc, of the power amplifier 412 based on the first analog operating point voltage value 429- 1, Vramp. Here, the first analog operating point voltage value 429- 1 and the supply voltage 435 in the exemplary embodiment of FIG. 4 substantially correspond to the depicted voltage 323 and the supply voltage 325 in the exemplary embodiment of FIG. 3, respectively.In other words, the DC-DC converter 430 is configured to adjust a supply voltage 435 of the power amplifier 410 based on a mapped voltage 429- 1 determined by a lookup table 426 (LUT A) entry.Thus, as shown in FIG. 4, in one embodiment, the operating point controller 420 is configured to provide a first operating point control signal to adjust a supply voltage 435, Vcc, of the power amplifier 410. In addition, the operating point controller 420 is configured to provide a second operating point control signal to adjust an input side operating point voltage 429- 2, Vcqto adjust a quiescent current of the power amplifier 410.In particular, in the embodiment of FIG. 4, the power amplifier 412 is configured to amplify the RF input signal 405 based on the supply voltage 435 and a quiescent current set by an input-side operating point voltage 429- 2 (wherein this input-side operating point voltage may set, for example, a grid bias or base bias of an amplifier transistor).FIG. 5 shows a graph 500 of an example dependence of a adjacent channel leakage power ratio (ACLR) value 505 on a phase 501 of a reflection factor. In particular, in FIG. 5, the ACLR overload phase for Vcc= constant=3.7 V and 'Vcc servo controlled 4 is shown. Here, the reflection factor may correspond to the reflection factor 321, 421 obtained from the impedance determination device 322 or 422 in the exemplary embodiments of FIGS. 3 and 4, respectively. Since the reflection factor is substantially determined by the load impedance of a load coupled to an output of the power amplifier, the phase 501 of the reflection factor may also be referred to as a "load phase.". The adjacent channel leakage power ratio (ACLR) value 505 (in dC) is typically defined as the ratio of the transmitted power, for example, the power in the UMTS transmission channel selected for communication, to the power in the adjacent channel (or the inverse thereof). Thus, the ACLR value 505 represents a measure of power leakage flowing into the adjacent channel, with a smaller ACLR value (in dC) substantially corresponding to better ACLR performance.According to embodiments of the invention, the power amplifier may be configured, for example, as a single ended power amplifier, such that, for a constant supply voltage 502 (e.g. Vcc=3.7 V) of the power amplifier, a dependence of the ACLR value 505 of the RF output signal on the phase 501 (in degrees) of a reflection factor for a plurality of load impedances leading to reflection factors of the same magnitude (i.e. for a given SWV), as is illustrated, for example, in the graph 500 of FIG. 5 by a first curve 507- 1, has, for example, a single peak 509 in an entire phase angle range 511 between -180 and 180°. Note here that in the graph 500 of FIG. 5, the battery voltage 506, Vbatt, used to operate the amplifier circuit has an example value of 3.7 V.It is further noted here that in the graphs 500, 600, and 700 of FIGS. 5, 6, and 7, the dependencies (i.e., ACLR value, supply voltage, and battery current) are specifically shown for an example standing wave ratio (SWV) value of 3:1. The standing wave ratio is usually defined as the ratio of a maximum voltage (Vmax) and a minimum voltage (Vmin) of a standing wave, and it depends on the magnitude of the reflection factor (FL) determined by the load impedance, as is well known to those skilled in the art. Here, the maximum voltage Vmax substantially corresponds to the sum of the forward propagating wave voltage V F at the output of the power amplifier and the reflected wave voltage V R reflected by the load impedance of a load coupled to the output of the power amplifier (i.e., Vmax=V F+ VR), while the minimum voltage V m j n substantially corresponds to the difference between the forward propagating wave voltage V F and the reflected wave voltage V R (i.e., Vmin=V F- V R).The fact that the SWV value is constant in the graphs of FIGS. 5-7 means that the corresponding dependencies are associated with a plurality of load impedances which result in reflection factors of the same amount.In embodiments related to FIG. 5, the operational point controller of the amplifier circuit is configured to provide a variable supply voltage 435, 504, denoted 'Vcc servo controlled 4 to the power amplifier such that the ACLR value 505, which is based on the phase 501 of corresponding reflection factors for reflection factors of the same magnitude, is within a predefined range within the entire phase angle range 511. Such an ACLR value 505, which is based on the phase 501, is represented in the graph 500 of FIG. 5 by a second curve 507- 2, which shows a nearly constant behavior over the entire phase angle range. In particular, the second curve 507- 2 of the graph 500 of FIG. 5 may have ACLR values lying, for example, in the predefined range between -37 and -35 dB, while the variable supply voltage 435, 504 may preferably be adjusted to obtain constant ACLR values of approximately -36 d. Accordingly, the entries of the look-up table 426 may be chosen to set the voltage 435 to be approximately constant (e.g., varied by less than ±2 dB or even by less than ±1 dB) over the entire phase range from -180° to +180° based on the phase of the reflectance factor FL (represented by information 425).FIG. 6 shows a graph 600 of an exemplary dependence of a supply voltage 435, 605 on a phase 501 of a reflection factor (for example the reflection factor r L). More specifically, in FIG. 6, Vcc is shown over load phase if Vcc is servo controlled to obtain a constant ACLR value. In the graph 600 of FIG. 6, a curve 607 is shown, which represents the supply voltage 435, 605, Vcc (in V) as a function of the phase 501 of corresponding reflection factors (for example the reflection factor FL) for reflection factors of the same amount (SVW=3:1) over the entire phase angle range 511. Here, the supply voltage 605 shown in the graph 600 of FIG. 6 may correspond to the supply voltage 435, 504 (Vcc servo controlled) provided to the power amplifier as described with reference to FIG. 5. The supply voltage Vcc as shown in FIG. 6 may be provided, for example, by the cooperation of the look-up table 426, the DAC 428- 1, and the DC-DC converter 430.In the graph 600 of FIG. 6, it can be seen that the curve 607 showing the supply voltage 435, 605 as a function of the phase 501 of a reflection factor for a plurality of load impedances which lead to reflection factors of the same magnitude, for example by a peak 609 in a first phase angle region 610 forA phase angle between -70° and -50° and characterized by a valley 611 in a second phase angle region 620 for phase angles between 60° and 90°.In embodiments related to FIG. 6, a variation ratio of the supply voltage 605 over the entire phase angle range 511 is, for example, less than 1.5:1, wherein the variation ratio can be defined as the ratio of the maximum value 613 and the minimum value 615 of the curve 607.FIG. 7 shows a graph 700 of an exemplary dependence of a battery current 705 on a phase 501 of a reflection factor. In particular, in FIG. 7, the battery current over load phase is shown for Vcc= constant=3.7 V and 'Vcc servo controlled 4. The graph 700 of FIG. 7 has substantially the same components as the graph 500 of FIG. 5 ; therefore, identical components are denoted by the same reference numerals. In the graph 700 of FIG. 7, a first and a second curve 707- 1, 707- 2 representing the current 705 (in A) as a function of the phase 501 (load phase) are shown in the entire phase angle range 511. Here, phase 501 of graph 700 shown in FIG. 7 corresponds to reflection factors for reflection factors of the same magnitude as for graphs 500 and 600 of FIGS. 5 and 6 (SWV=3:1). In addition, the first curve 707-1 corresponds to a constant supply voltage 502 of 3.7 V, while the second curve 707-2 corresponds to the variable supply voltage 504 (Vcc servo controlled). The battery voltage 506 for the amplifier circuit is set to an exemplary value of 3.7V. It can clearly be seen in FIG. 7 that in the case of using the variable or servo controlled supply voltage 504, the battery current 705 consumed by the power amplifier can be reduced considerably at least over a significant phase range compared to the case of using the constant supply voltage 502. Accordingly, in the graph 700 of FIG. 7, the second curve 707- 2 is at least partially below the first curve 707- 1. This applies in particular to phase angles of more than about -80°. Thus, the power consumption of the power amplifier can be reduced by adjusting the supply voltage of the power amplifier based on the load phase to obtain an ACLR value of the RF output signal that is within a predetermined range.To recapitulate the embodiment described with reference to FIG. 4, the load impedance of a load, for example an antenna (antenna 408 of FIG. 4 ) coupled to the output of the power amplifier, or the antenna impedance may be measured using a directional coupler. However, for embodiments of the invention, any method that provides phase and magnitude of antenna impedance may be applicable. The use of a directional coupler is one implementation, but there are other approaches that can also provide the load impedance. A DC-DC converter may be used for setting the supply voltage Vcc of the power amplifier. According to further exemplary embodiments of the present invention, the DC-DC converter can also be realized as a step-down-only actuator, a step-up-only actuator or as a step-down step-up actuator. The supply voltage Vcc applied to the power amplifier determines the linear output performance of the power amplifier. The higher the supply voltage, the higher the maximum linear output power of the power amplifier. However, a high supply voltage also means a high battery current due to a lower conversion ratio of the DC-DC converter. Therefore, the supply voltage is preferably set to a value that should be as low as possible to save battery power, but high enough to ensure good ACLR performance under all conditions. The look-up table (LUT A) may be used to set the operating point of the power amplifier depending on the load phase. The impedance information look-up table (LUT B) may be used to store the measured load impedances, e.g., in dependence on transmission channel or frequency information (e.g., to be able to quickly adjust the power amplifier operating point in the event of a frequency change).The basic concept of the invention is summarized below. FIG. 5 shows ACLR performance over load phase in a SWV 3:1 load mismatch in two different cases. In the first case, the power amplifier supply voltage Vcc is constant at 3.7 V, and in the second case, the supply voltage Vcc is optimized to maintain an exemplary ACLR target of -36 dC over load phase.If the supply voltage Vcc is constantly at 3.7 V or any other value within the battery voltage range, the ACLR performance depends significantly on the load phase angle. This is typical of all linear power amplifiers. Only the characteristics differ depending on the power amplifier architecture.Unbalanced power amplifiers have a maximum over phase, as shown in FIG. 5. Here, the maximum occurs at -120 degrees, whereas balanced amplifiers have two maximum values due to the 90 degree hybrid. In Figure 5, it can be seen that the ACLR value for some phase angles is much better than needed if Vcc is held constant. This typical behavior can be used to decrease the supply voltage Vcc depending on the phase, which means that the Vcc voltage can be decreased at the expense of an ACLR reserve. This is also shown in FIG. 5 as the case "Vcc servo controlled". Here, "Vcc servo-controlled" means that the power amplifier supply voltage Vcc is set at each phase angle so that the ACLR value is constant (in this example, - 36 dC). This adjustment of the supply voltage Vcc can be achieved, for example, by selecting the lookup table (LUT A) entries depending on the load phase such that the ACLR is approximately constant (e.g., varied by less than ±2 dB or even by less than ±1 dB) over the entire phase range from -180° to +180°.FIG. 6 shows the corresponding Vcc relationship (e.g., the Vcc behavior created by the cooperation of the look-up table 426, the DAC 428-1, and the DC-DC converter 430 in the embodiment of FIG. 4) over a phase angle. The Vcc variation is more than 1 V over a phase angle. phase angles corresponding to the so-called high impedance range of the power amplifier (-60 degrees in this example) may, for example, require a high supply voltage that prevents voltage saturation of the power amplifier; phase angles corresponding to the low impedance phase range of the power amplifier (60... 90 degrees) allow for a lower supply voltage.FIG. 7 shows the effect on battery current. If Vcc is servo controlled for a constant ACLR according to an embodiment of the invention, the battery current can be greatly reduced, which is an advantage of the present invention.The proper Vcc setting may require knowledge of the load impedance. If the load impedance is known, the Vcc voltage may be set to the lowest possible value high enough to maintain a particular target ACLR performance as shown in FIG. 6. This method also substantially reduces the 50 ohm current that is important for an evaluation method. In the case of Vcc = constant, the battery current of 50 ohms for this type of power amplifier is approximately 370 mA. In the case of Vcc being "servo controlled" to an ACLR target of -36 dC, the battery current at Vbatt=3.7 V is 297 mA. The required Vcc to achieve an ACLR value of -36 dB at 50 ohms is, for example, 2.8 V. If the battery voltage is 3.7 V, the conversion ratio is 3.7 V / 2.8 V = 1.32. For example, by eliminating the power margin not needed at 50 ohms, the battery current can be decreased by more than 20%, which would result in best category power consumption performance.FIG. 8 shows a block diagram of another embodiment of an amplifier circuit 800 comprising a bias controller 820 for receiving frequency information 805. The amplifier circuit 800 of FIG. 8 has substantially the same blocks as the amplifier circuit 100 of FIG. 1. therefore, identical blocks having similar implementations and / or functions are denoted by the same reference numerals. However, in the embodiment of FIG. 8, the operating point controller 820 of the amplifier circuit 800 is configured to perform the following steps. First, a maximum operating point control signal is provided to set the operating point of the power amplifier 110 to a maximum level before an initial time slot of a sequence of time slots. Subsequently, the measure F L of the load impedance is determined. Finally, a working point control signal 825 different from the maximum working point control signal is provided to adjust the working point of the power amplifier 110 based on the determination of the measure FL of the load impedance for a subsequent time slot of the sequence of time slots.For example, the operating point controller 820 is configured to perform the steps described with reference to the embodiment of FIG. 8 for each frequency of a hopping sequence. Here, each of the frequencies of the hopping sequence may be indicated by frequency information 805 received from the operating point controller 820, such as in a frequency hopping mode. The frequency information 805 received by the operating point controller 820 of FIG. 8 can correspond to the frequency information 205 received by the operating point controller 220 of FIG. 2. Thus, in the embodiments of FIGS. 2 and 8, the operating point controller 220 or 820 is configured to provide the operating point control signal 125, 825 to the power amplifier 110 based on the frequency information 205 and 805, respectively.According to the embodiment of FIG. 8, the operating point controller 820 of the amplifier circuit 800 is configured to store measures of the load impedance for a plurality of frequencies and reuse the stored measures of the load impedance upon a return to a frequency previously used in the hopping sequence.According to further exemplary embodiments, the procedure described above with reference to FIG. 8 can have the following steps. First, if the antenna impedance is unknown (e.g., prior to a first slot or frequency change), the operating point of the power amplifier is set to a high level that ensures adequate ACLR performance independent of the antenna impedance. During a next transmission slot, the antenna impedance is then measured. The operating point voltage for the next slot is then established as a function of the measured impedance. In embodiments, the operating point control can change the supply voltage and / or the quiescent current of the power amplifier. This is also referred to as dual operating point control. In the embodiment described above with reference to FIG. 4, only (or preferably) the Vcc level is changed. It is noted here that the dual operating point control further reduces power consumption, but increases complexity.If the mobile device also operates in a frequency hopping mode, the steps of the just mentioned procedure can be applied to each hopping frequency independently of one another. This can be achieved from a dedicated table (e.g., LUT B in the embodiment of Fig. 4) storing the impedance data for each channel.Referring again to the embodiment of Figure 4, the relationship between the load impedance and the operating point condition is contained in the look-up table LUT A. This look-up table may in some cases be needed for each band supported by the mobile device and may be determined by laboratory measurements. For each band, there should be several frequency points in some cases to compensate for the frequency response of the power amplifier. This is particularly the case for bands comprising a duplexer. It is not necessary to have a table for each output power level. A table normalized with respect to the operating point of 50 Ohm is sufficient.In another embodiment, the same procedure described above may be applied to 2G mobile devices to reduce maximum current in the event of a mismatch. The reduction of the maximum current is a serious problem due to enormous heat dissipation, especially for smartphones. If the antenna impedance is in the low impedance region of the power amplifier (phase region where the power amplifier draws most current), the output power may be reduced to keep dissipated heat below a certain value, If the 2G power amplifier were powered by a DC-DC converter, the supply voltage may be reduced to reduce battery current. By evaluating the load impedance, the heat dissipation can be easily limited, which is important for the phone suppliers.FIG. 9 shows a block diagram of an embodiment of a mobile communication device 900 including the embodiment of the amplifier circuit 100 of FIG. 1. As shown in FIG. 9, the mobile communication device 900 for transmitting an RF output signal may include a baseband generator 910, an RF signal generator 920, a power amplifier 110, a bias controller 120, and an antenna 930. Here, the power amplifier 110 and the operating point controller 120 may constitute the amplifier circuit 100 as shown in the embodiment of FIG. 1. Therefore, similar blocks in FIG. 9 having similar implementations and / or functions to those in FIG. 1 are denoted by the same reference numerals. Moreover, the baseband generator 910 and the RF signal generator 920 of the mobile communication device 900 shown in FIG. 9 may correspond to the baseband generator 402 and the RF signal generator 404 shown in the embodiment of FIG. 4, while the baseband signal 915, s(t) and the RF signal 925 obtained from the baseband generator 910 and the RF signal generator 920 correspond to the baseband signal 403, s(t) and the RF signal 405 obtained from the baseband generator 402 and the RF signal generator 404. In the embodiment of FIG. 9, the baseband generator 910 is configured to generate a baseband signal 915. The RF signal generator 920 may be configured to generate an RF signal 925 based on the baseband signal 915. The power amplifier 110 is configured to receive the RF signal 925 from the RF signal generator 920 and amplify the received RF signal to obtain an RF output signal 115. The operating point controller 120 is configured to control an operating point of the power amplifier 110. Here, the operating point controller 120 is configured to determine the measure r L( for example a reflection factor) of a load impedance of a load coupled to an output of the power amplifier 110 and provide an operating point control signal 125 to adjust the operating point of the power amplifier 110 based on the determination of the measure FL of the load impedance. The antenna 930 is configured to transmit the RF output signal 115 as a transmitted RF output signal 935.Some embodiments according to the invention provide better performance than conventional 3G mobile devices using an isolator to maintain good ACLR performance in the case of antenna mismatch. The isolator solved the problem of linearity degradation in the case of antenna mismatch, but acted considerably on size and cost. The increasing number of bands increased the drawback of the isolator approach in terms of cost and size. Thus, the isolator has been removed from most designs and replaced with other approaches that are also intended to provide load insensitive behavior. Today, balanced power amplifiers are the most important category of load insensitive power amplifier solutions. Depending on the power amplifier supplier, there are some design variants, but all implementations rely on a 90 degree hybrid as a core element to reduce load sensitivity. A great disadvantage of each balanced power amplifier is that the load insensitivity is gained at the cost of lower power amplifier efficiency due to additional losses caused by the hybrid network. The peak efficiency of a balanced amplifier is usually in the range 35-37%, whereas a unbalanced power amplifier achieves more than 40%.Some embodiments according to the invention provide a better compromise between efficiency and complexity than amplifiers having more reserve. It has been found that it is often a less effective approach to using a single ended amplifier with a larger reserve of linear output power. Due to the additional linear performance, ACLR degradation is reduced in the event of a mismatch. The advantages compared to a balanced power amplifier are less complex hardware, enabling more cost effective and smaller size solutions. However, the effect on efficiency is even more severe than in the case of a balanced power amplifier if the same ALCR performance is assumed for both architectures in the case of mismatch.Some embodiments according to the invention provide a better compromise between efficiency and complexity than conventional software-based solutions. Apart from the hardware-based (HW-based) solutions described above, there are some software (SW) solutions that also aim to improve the ACLR in the event of a mismatch and that have less impact on the power amplifier's efficiency, cost, or complexity. Good examples are predistortion and adaptive operating point setting. The disadvantage here is a considerably higher software complexity, which hitherto hinders use in mobile devices.Although some aspects have been described in connection with a device, it is understood that these aspects also represent a description of the corresponding method, wherein a block or a device corresponds to a method step or a feature of a method step. Analogously, aspects described in connection with a method step likewise represent a description of a corresponding block or item or feature of a corresponding device. Some or all of the method steps may be performed by (or by using) a hardware device, for example a microprocessor, a programmable computer or an electronic circuit. In some exemplary embodiments, some or more of the most important method steps can be carried out by such a device.Depending on certain implementation requirements, embodiments of the invention may be implemented in hardware or in software. The implementation can be carried out using a digital storage medium, for example a floppy disk, a DVD, a Blu-ray, a CD, a ROM, a PROM, an EPROM, an EEPROM or a FLASH memory, on which electronically readable control signals are stored, which cooperate (or are capable of) cooperating with a programmable computer system such that the respective method is carried out. Therefore, the digital storage medium may be computer readable.Some embodiments according to the invention comprise a data carrier having electronically readable control signals capable of interacting with a programmable computer system such that one of the methods described herein is carried out.In general, embodiments of the present invention can be implemented as a computer program product having a program code, wherein the program code is operative to perform one of the methods when the computer program product runs on a computer. The program code can be stored, for example, on a machine-readable carrier.Other embodiments include the computer program for performing any of the methods described herein, wherein the computer program is stored on a machine readable carrier.In other words, an exemplary embodiment of the method according to the invention is thus a computer program which has a program code for carrying out one of the methods described herein when the computer program runs on a computer.A further embodiment of the methods according to the invention is thus a data carrier (or a digital storage medium or a computer-readable medium) on which the computer program for carrying out one of the methods described herein is recorded.The data carrier, digital storage medium, or recorded medium is typically tangible and / or non-transitory.A further exemplary embodiment of the method according to the invention is thus a data stream or a sequence of signals which represents or represent the computer program for carrying out one of the methods described herein. The data stream or sequence of signals may be configured, for example, to be transferred over a data communication link, for example, over the Internet.A further embodiment comprises a processing device, for example a computer or a programmable logic device, configured or adapted to perform one of the methods described herein.A further embodiment comprises a computer on which the computer program for carrying out one of the methods described herein is installed.A further embodiment according to the invention comprises an apparatus or a system which is designed to transmit a computer program for carrying out at least one of the methods described herein to a receiver (for example electronically or optically). The receiver may be, for example, a computer, a mobile device, a storage device, or similar device. The apparatus or system may, for example, comprise a file server for transmitting the computer program to the recipient.In some embodiments, a programmable logic device (e.g., a field programmable gate array, an FPGA) may be used to perform some or all of the functionalities of the methods described herein. In some embodiments, a field programmable gate array may cooperate with a microprocessor to perform any of the methods described herein. Generally, the methods are preferably performed by any hardware device.The above-described embodiments are merely illustrative of the principles of the present invention. It is to be understood that modifications and variations of the arrangements and details described herein will be apparent to others skilled in the art. Therefore, it is intended that the invention be limited only by the scope of the following claims and not by the specific details presented herein by way of description and explanation of the embodiments.In summary, embodiments of the invention provide an attractive solution to a problem occurring in all mobile terminals based on, for example, 8PSK7GMSK (8-phase shift keying / Gaussian minimum shift keying), WCDMA or LTE. Embodiments of the present invention are mostly related to WCDMA, but could also be applied to solve similar problems in other fields.Embodiments of the invention provide a methodology that combines low software complexity with minimal (or at least quite low) impact on hardware architecture, enabling excellent power consumption and ACLR performance in the event of a mismatch.It should be noted that knowing the antenna impedance or measuring it in a mobile device is important and a prerequisite for embodiments of this invention. In addition, it is expected that many mobile devices will in the future aid in measuring antenna impedance for other reasons. Such reasons may include, for example, setting up an antenna tuner or performing a self-test, for example during production, to check whether the antenna is properly assembled, etc.Compared to the balanced power amplifier approach, embodiments of the invention have the advantages that a less complex power amplifier can be realized, thereby allowing for a smaller size and lower cost thereof, and that a higher peak efficiency at 50 ohms can be achieved.In addition, compared to pre-distortion, embodiments of the invention have the advantages of providing less complex software algorithms, short settling time, and a forward transmission implementation. Here it is pointed out that in case of a short settling time only one measurement point is needed, while the forward transmission implementation can be made robust, thus having no stability problems.In essence, embodiments of the invention provide a concept for setting the operating point of a power amplifier as a function of the measured load impedance. In some embodiments, a directional coupler may be used and the ACLR may be kept relatively constant across a phase. In some other embodiments, other approaches such as pre-distortion could also be used, which would require a plurality of slots to settle the ACLR value.

Claims

An amplifier circuit (300; 400) comprising: a power amplifier (110) configured to amplify an RF input signal (105) based on a supply voltage to obtain an RF output signal (115); and a bias controller (320; 420) configured to control a bias of the power amplifier (110), the bias controller (320; 420) comprising: impedance determiner (322; 422) configured to determine a measure of a load impedance of a load coupled to an output of the power amplifier (110); and a DC-DC converter (326; 430) configured to adjust the supply voltage of the power amplifier (110), characterized in that the bias controller (320; 422); 420 ) further comprising an imaging unit (324) configured to map the measure of the load impedance to a mapped voltage such that the mapped voltage is based on a phase of a reflection factor determined by the load impedance; wherein the DC-DC converter (326; 430) is configured to adjust the supply voltage of the power amplifier (110) based on the mapped voltage; and wherein the imaging unit (324) is configured to provide the mapped voltage such that a adjacent channel leakage power ratio value of the RF output signal (115) is within a predetermined range for a plurality of load impedances resulting in reflection factors of equal magnitude.The amplifier circuit (300; 400) of claim 1, wherein the operating point controller (320; 420) is configured to provide a first operating point control signal (125) as the operating point control signal to adjust a supply voltage of the power amplifier (110).The amplifier circuit (300; 400) of claim 2, wherein the operating point controller (320; 420) is configured to further provide a second operating point control signal as the operating point control signal (125) to adjust an input side operating point voltage to adjust a quiescent current of the power amplifier (110).The amplifier circuit (300; 400) of claim 2 or 3, wherein the power amplifier (110) is configured to amplify the RF input signal (105) based on the supply voltage and a quiescent current.The amplifier circuit (300; 400) of any of claims 1 to 4, wherein the operating point controller (320; 420) comprises a look-up table configured to store a plurality of operating point voltage values associated with respective values of a reflection factor for a plurality of load impedances such that the operating point voltage values are dependent on a phase and amplitude of respective reflection factors for reflection factors of equal magnitude.Amplifier circuit (300; 400) according to claim 5, wherein the operating point controller (320; 420) is configured to extract a single operating point voltage value from the look-up table, wherein the single operating point voltage value corresponds to a value of a reflection factor determined by the determined load impedance, and wherein the operating point controller is configured to provide the operating point control signal (125) to adjust the operating point of the power amplifier (110) based on the single operating point voltage value extracted from the look-up table.The amplifier circuit (300; 400) of any of claims 1 to 6, wherein the operating point controller (320; 420) each comprises different look-up tables for a plurality of different frequencies of the RF input signal (105), each of the look-up tables being configured to store a plurality of operating point voltage values associated with corresponding values of a reflection factor for a plurality of load impedances such that the operating point voltage values are dependent on a phase of corresponding reflection factors for reflection factors of equal magnitude.Amplifier circuit according to claim 7, wherein the operating point controller is configured to extract a single operating point voltage value from a selected look-up table associated with a current frequency of the RF input signal (105), the single operating point voltage value corresponding to a phase of a reflection factor determined by the determined load impedance for the current frequency of the RF input signal (105), and wherein the operating point controller (320, 420) is configured to provide the operating point control signal (125) to adjust the operating point of the power amplifier (110) based on the single operating point voltage value extracted from the selected look-up table.The amplifier circuit of any of claims 1 to 8, wherein the operating point controller (320; 420) further comprises a DC-DC converter (326; 430) configured to adjust a supply voltage of the power amplifier (110) based on a mapped voltage determined by a lookup table entry.The amplifier circuit (300; 400) of any of claims 1 to 9, wherein the operating point controller (320; 420) comprises an impedance information look-up table configured to store a plurality of measures of load impedance for corresponding frequencies of the RF input signal (105).The amplifier circuit (300; 400) of claim 10, wherein the operating point controller (320; 420) is configured to extract a single measure of a load impedance from the impedance information look-up table, wherein the single measure of the load impedance corresponds to a frequency of the RF input signal (105) in a frequency hopping mode, and wherein the operating point controller is configured to provide the operating point control signal (125) to adjust the operating point of the power amplifier (110) based on the single measure of the load impedance extracted from the impedance information look-up table.Amplifier circuit (300; 400) according to any of claims 1 to 11, wherein the power amplifier (110) is configured such that for a constant supply voltage of the power amplifier (110), a dependence of a adjacent channel leakage power ratio value of the RF output signal (115) on a phase of a reflection factor for a plurality of load impedances leading to reflection factors of equal magnitude has a single peak in an entire phase angle range.The amplifier circuit (300; 400) of any of claims 1 to 12, further comprising a directional coupler (406) configured to perform a reflection factor measurement based on the load impedance.The amplifier circuit (300; 400) of any of claims 1 to 13, wherein the operating point controller (320; 420) is configured to provide a maximum operating point control signal to set the operating point of the power amplifier (110) to a maximum level prior to an initial time slot of a sequence of time slots, and wherein the operating point controller 320; 420 is further configured to subsequently determine the measure of the load impedance and provide an operating point control signal (125) different from the maximum operating point control signal to adjust the operating point of the power amplifier (110) based on the determination of the measure of the load impedance for a subsequent time slot of the sequence of time slots.The amplifier circuit (300; 400) of claim 14, wherein the operating point controller (320; 420) is configured to perform the step of providing a maximum operating point control signal to set the operating point of the power amplifier (110) to a maximum level before an initial time slot of the sequence of time slots, the step of subsequently determining the measure of the load impedance, and the step of providing an operating point control signal (125) different from the maximum operating point control signal to set the operating point of the power amplifier (110) based on the determination of the measure of the load impedance for the subsequent time slot of the sequence of time slots for each frequency of a hopping sequence.The amplifier circuit (300; 400) of claim 15, wherein the operating point controller (320; 420) is configured to store measures of the load impedance for a plurality of frequencies and reuse the stored measures of the load impedance upon a return to a frequency previously used in the hopping sequence.A communication device (900) for transmitting an RF output signal (115), the communication device (900) comprising: a baseband generator (402, 910) configured to generate a baseband signal (403, 915); an RF signal generator (404, 920) configured to generate an RF signal (105, 925) based on the baseband signal (403, 915); and an amplifier circuit (300, 400) according to any one of claims 1 to 16.

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