Inductive proximity switch and method for operating an inductive proximity switch

Abstract

Inductive proximity switch (100) with a resonant circuit (10) with a resonant circuit coil (12), with an oscillator amplifier (20) and with an output stage (30) that interacts with the resonant circuit (10) and / or the oscillator amplifier (20) to provide an output signal (32) depending on an influence, in particular a damping, of the resonant circuit (10) by a target (T) to be detected, characterized by that a measuring device (50) is available for measuring an inductance (L) of the resonant circuit coil (12) and that a compensation device (60) is provided which interacts with the measuring device (50) and is designed to adjust the resonant circuit (10), the oscillator amplifier (20) and / or the output stage (30) as a function of a measured inductance (L) of the resonant circuit coil (12) in order to compensate for, in particular temperature-related, changes in the resonant circuit (10) which are due to, in particular temperature-dependent, changes in the inductance (L).

Description

[0001] The present invention relates, in a first aspect, to an inductive proximity switch according to the preamble of claim 1. In a second aspect, the invention relates to a method for operating an inductive proximity switch according to the preamble of claim 24.

[0002] An inductive proximity switch of this type is described, for example, in DE 10 2013 202 573 B3 and comprises the following components: a resonant circuit with a resonant circuit coil, an oscillator amplifier and an output stage that interacts with the resonant circuit and / or the oscillator amplifier to provide an output signal depending on an influence, in particular a damping, of the resonant circuit by a target to be detected.

[0003] A generic method for operating an inductive proximity switch of the type described above is also described in DE 10 2013 202 573 B3. In this method, an output signal is generated and provided by an output stage due to an influence, in particular a damping, of the resonant circuit by a target to be detected.

[0004] The measuring principle of an inductive proximity switch generally consists of a coil driven by an oscillator radiating an alternating magnetic field into a monitoring area, and the interaction between this alternating magnetic field and a target object being detected being measured. One effect that occurs and can be evaluated for measurement is that the alternating magnetic field induces eddy currents in a metallic target, resulting in losses that draw energy from the oscillator.

[0005] In this most common evaluation method, which can be used particularly with the proximity switch according to the invention, the damping of the oscillator by the approaching target is evaluated. The proximity switch activates at a specific distance, the so-called switching distance, when the damping caused by the approaching target becomes so great that the oscillation amplitude falls below a threshold value or ceases. Such an evaluation can be implemented with relatively simple electronic means.

[0006] A general goal for inductive proximity switches is to ensure that the switching distance is as large and well-defined as possible.

[0007] The measured variable in proximity switches is generally the change in the resonant impedance of the resonant circuit as a monitored target approaches the switch. The resonant impedance Rp is approximately given by Rp = L / (C*Rs), where L is the inductance of a resonant circuit coil, C is the capacitance of the resonant circuit, and Rs is the resistance of the resonant circuit coil at the resonant frequency. For the achievable accuracy of the switching distance, it must be considered that the resistance Rs, in particular, but also the inductance of the resonant circuit coil, exhibit a significant temperature dependence. Furthermore, changes in inductance, in particular, are sometimes neither reproducible nor reversible and can depend, among other things, on the potting materials used and their history. This means that such effects cannot be corrected by factory calibration of the proximity switch.The magnitude of the observed change in resonant impedance limits the maximum achievable switching distance because the temperature dependencies of the relevant physical properties of the components used lead to significant, non-negligible changes in the resonant impedance. For this reason, only relatively small switching distances are currently achievable, and these distances are fundamentally dependent on the device design.

[0008] Component tolerances, particularly absolute electrical and mechanical tolerances as well as manufacturing tolerances (i.e., relative tolerances), play a role here. Deviations also arise from the dependence of electrical parameters on manufacturing tolerances, for example, positioning tolerances of the winding relative to the core.

[0009] Furthermore, the quality of the components used must be considered. For example, the aging of components and materials can lead to drifts in electrical parameters. Finally, aging processes can also alter and degrade existing connections, such as joints, which in turn leads to changes or drifts in the relevant electrical parameters.

[0010] For example, in the "M 18, shielded" design, the resonant impedance changes by only about 1% when the target is approached from a great distance and the switching distance is three times the standard switching distance sn. In contrast, with this design, the change when approaching the target up to the standard switching distance is approximately 20% of the resonant impedance.

[0011] This implies that the resistance and inductance must be measured with a relative accuracy of better than one per mille if the change in the standard switching distance with temperature is to be kept below 10 percent.

[0012] The proximity switch described in DE 10 2013 202 573 B3 uses a lock-in amplifier to measure the copper resistance of the coil system of an oscillator. The measured coil resistance is used to influence a threshold value of a downstream comparator via the temperature, thereby achieving temperature compensation.

[0013] An inductive proximity switch in which temperature compensation is carried out using a thermally coupled temperature sensor is described in DE 39 31 892 A1.

[0014] A so-called temperature self-compensation with a bifilar coil system is described in EP 70 796 A1 and EP 319 470 A1.

[0015] EP 813 306 B1 discloses a system in which a voltage proportional to the copper resistance of the coil system is generated and this voltage is used to keep the break-up point of the oscillator oscillation essentially independent of temperature by adjusting the loop gain of an inductive oscillator.

[0016] German patent DE 41 42 680 A1 describes an arrangement in which a synchronous rectifier is used to measure either the real or the imaginary part of a coil impedance. By selecting the correct phase angle, either the real or the imaginary part can be measured. Temperature compensation is intended to be possible by selecting a suitable phase angle, i.e., a suitable mixture of the real and imaginary parts.

[0017] In GB 2 040 054 A an inductive proximity switch is described in which the inductance L serves as a measure of the approach of a target and is measured using a clever method.

[0018] DE 195 38 575 C2 relates to an inductive proximity sensor in which the change in the impedance of a measuring coil is used as a measure of the distance to a target. To increase the switching speed, it is proposed to measure the phase shift between the applied or induced alternating voltage and the flowing alternating current at a sensor coil directly during each half-cycle or multiple of each half-cycle of current and voltage using electronic means.

[0019] One object of the invention is to provide an inductive proximity switch and a method for operating an inductive proximity switch in which improved compensation of temperature effects is possible. This object is achieved by the inductive proximity switch with the features of claim 1 and by the method with the features of claim 24.

[0020] Advantageous embodiments of the proximity switch according to the invention and preferred variants of the method according to the invention are described below, in particular in connection with the dependent claims and the figures.

[0021] The inductive proximity switch of the type described above is further developed according to the invention in that a measuring device is provided for measuring an inductance of the resonant circuit coil and that a compensation device is provided which cooperates with the measuring device and is configured to adjust the resonant circuit, the oscillator amplifier and / or the output stage as a function of a measured inductance of the resonant circuit coil in order to compensate for changes in the resonant circuit that are due to changes in the inductance.

[0022] The method for operating an inductive proximity switch of the type described above is further developed according to the invention in that the inductance of the resonant circuit coil is measured with the measuring device and that the resonant circuit, the oscillator amplifier and / or the output stage is adjusted as a function of a value of the measured inductance of the resonant circuit coil in order to compensate for, in particular temperature-related, changes in the resonant circuit which are due to, in particular temperature-dependent, changes in the inductance.

[0023] Changes in the resonant circuit can be temperature-related, meaning they can be caused by a change in temperature. Changes in inductance can also be temperature-dependent or temperature-induced, meaning they can be caused by a change in temperature.

[0024] An inductive proximity switch, as described here, is a sensor in which a coil driven by an oscillator, in particular a resonant circuit coil, emits an alternating magnetic field into a monitoring area. The interaction between this alternating magnetic field and an object to be detected in the monitoring area, also referred to as the target, serves as the measured variable. This interaction can, in particular, involve the damping of the resonant circuit by the target.

[0025] In this description, a resonant circuit is understood to be an electromagnetically oscillating system that includes at least one resonant circuit coil. Typically, a resonant circuit comprises the resonant circuit coil and a capacitor.

[0026] The term "resonant circuit coil" refers in particular to a single coil, but also generally to a coil system consisting of a plurality of single coils.

[0027] The oscillator amplifier, which can basically consist of known electronic components, serves to maintain a vibration or oscillation of the resonant circuit.

[0028] The output stage is the electronic device that ultimately provides an analog or binary output signal. This output signal can be derived from the oscillator amplifier or the resonant circuit itself. The output stage is electrically or electronically connected to the resonant circuit and / or the oscillator amplifier in some way and interacts with them accordingly.

[0029] The output signal is a signal that is representative of the result of a measurement of the inductive proximity switch according to the invention. It is provided at least at some point within the inductive proximity switch. It does not necessarily have to be output externally. It is also possible that it is further processed internally and communicated externally in processed form.

[0030] The influence of a target on the resonant circuit is understood to mean any form of influence by a target, especially a metallic target, on the properties of the resonant circuit. This can include, among other things, a shift in the frequency of the resonant circuit, a damping of the resonant circuit, and / or inductive coupling based on the transformer principle.

[0031] Damping of the resonant circuit refers to the extraction of energy from the resonant circuit by a target, particularly a metallic one, which is to be detected. Such damping can occur, for example, due to eddy currents induced in the metallic target by the alternating magnetic field of the inductive proximity switch, causing ohmic losses in the target.

[0032] For the purposes of this description, the term "compensation device" refers to a device, in particular a digital control device, which receives measurement data from a measuring device, for example, a lock-in amplifier device, to which it is functionally connected, in particular electrically or electronically, and processes this data in a suitable and defined manner. A digital compensation device and a digital lock-in amplifier device can also form a single physical unit, for example, within a microcontroller or a comparable logical and, in particular, programmable component.

[0033] According to the invention, the compensation device is designed to adjust the resonant circuit, the oscillator amplifier and / or the output stage as a function of a measured inductance of the resonant circuit coil in order to compensate for, in particular temperature-related, changes in the resonant circuit which are due to, in particular temperature-dependent, changes in the inductance.

[0034] The resonant circuit can be adjusted, for example, by manipulating the quality factor of the resonant circuit using adjustable resistors.

[0035] Compensation or balancing means, in particular, changing the oscillator amplifier and / or the output stage depending on the measured data so that the switching distance remains as constant as possible.

[0036] A key aspect of the present invention can be considered to be the determination or measurement of the inductance of the resonant circuit coil and the consideration of the measured inductance during compensation, which may in particular be temperature compensation.

[0037] The invention recognizes that the inductance L of the resonant circuit coil exhibits significant temperature dependencies, meaning it can drift considerably with temperature. One reason for this can be, for example, the influence of thermal expansion of a casting resin on a ferrite core. Consequently, the amplitude of an LC oscillator, which is typically evaluated, is also subject to temperature fluctuations in the inductance. A particular advantage of the proximity switch according to the invention is that temperature compensation can be significantly improved because any drift of the inductance L with temperature is also taken into account during compensation.

[0038] In comparison to DE 39 31 892 A1, a thermally coupled temperature sensor is not necessary, so that the shortcomings of the coupling and temperature gradients cannot have a detrimental effect in the present invention.

[0039] The manufacture of a bifilar coil system, as used in the solutions described in EP 70 796 A1 and EP 319 470 A1, is complex. The inductive proximity switch of the present invention does not require such a bifilar coil system and can instead be constructed with a conventional coil system with two terminals.

[0040] The primary purpose of the present invention is to compensate for thermal changes in inductance. In prior art compensation methods, such as those using temperature sensors, manufacturing tolerances of the coil inductance typically affect temperature compensation. In particular, non-reproducible changes in coil inductance cannot be compensated. However, this is not a problem with the compensation method proposed by the present invention, as such influences on the inductance can also be taken into account.

[0041] In principle, the present invention can also compensate for incipient core saturation in low-frequency magnetic fields.

[0042] Theoretically, in the compensation method of the proximity switch and the method according to the invention, a ferrite near the sensor would not cause any damping of the resonant circuit because the inductance L increases, since this effect occurs at, for example, 2 kHz just as it does at 200 kHz and is therefore compensated.

[0043] In a particularly preferred embodiment of the inductive proximity switch according to the invention, a current source is provided for supplying the resonant circuit with a time-varying test current, and the measuring device is set up to measure the inductance of the resonant circuit coil based on a voltage drop caused by the test current.

[0044] Accordingly, variants of the inventive method are advantageous in which the resonant circuit is supplied with a time-varying test current and the inductance of the resonant circuit coil is measured with the measuring device on the basis of a voltage drop caused by the test current.

[0045] Using a test current allows for a simple yet accurate determination of the inductance.

[0046] In principle, any known devices can be used as a power source to apply a time-varying test current to the resonant circuit.

[0047] In principle, the invention is already realized if only the inductance of the resonant circuit is determined and compensation is performed based on the measurement result. Particularly advantageous variants of the inductive proximity switch according to the invention are characterized, however, by the fact that the measuring device is also configured to measure the resistance of the resonant circuit coil based on the voltage drop caused by the test current, and that the compensation device is further configured to adjust the resonant circuit, the oscillator amplifier, and / or the output stage as a function of a measured resistance of the resonant circuit coil in order to compensate for changes in the resonant circuit that result from changes in the resistance of the resonant circuit coil, particularly at the resonant circuit frequency.

[0048] These preferred embodiments correspond to variants of the method according to the invention, which are characterized in that the resistance of the resonant circuit coil is measured with the measuring device on the basis of a voltage drop caused by the test current and that the resonant circuit, the oscillator amplifier and / or the output stage are adjusted as a function of a value of the measured resistance of the resonant circuit coil in order to compensate for, in particular temperature-related, changes in the resonant circuit which are due to, in particular temperature-dependent, changes in the resistance of the resonant circuit coil, especially at the resonant circuit frequency.

[0049] These designs essentially aim to compensate for temperature-related changes in the resonant circuit that result from temperature-related changes in the resistance. However, changes in the resonant circuit that are not temperature-related and / or result from non-temperature-related changes in the resistance can also be compensated or balanced.

[0050] When the present application refers to measuring the resistance and / or inductance of the resonant circuit coil based on a voltage drop caused by the test current, this means that measured quantities are determined which are in a clear functional relationship with the resistance or inductance. Measuring in this sense does not necessarily mean that the resistance and / or inductance are actually determined and, as such, are specifically held at a particular location within the inductive proximity switch in any process step.

[0051] Significant advantages can be achieved by determining both the real and imaginary parts of a resonant circuit impedance measured via a voltage drop. The resistance of the resonant circuit coil can then be derived from the real part of the impedance across which a voltage drop is measured, and the inductance of the resonant circuit coil can be obtained from the imaginary part of the impedance across which a voltage drop is measured.

[0052] In particularly preferred embodiments of the inductive proximity switch and the method according to the invention, the measuring device is implemented by a lock-in amplifier. For the purposes of this description, a lock-in amplifier is understood to be a device that implements the functionality of at least one lock-in amplifier. A lock-in amplifier is a measuring device with which a noisy signal can be rectified in a frequency-selective and phase-selective manner. A lock-in amplifier is therefore particularly well suited for measuring signals with a known frequency spectrum, especially periodic signals that are highly noisy, and can thus be used particularly advantageously to monitor the inductance of the resonant circuit during ongoing oscillator operation.

[0053] This variant of the inductive proximity switch according to the invention is also particularly resistant to interference with higher harmonics of the measurement frequency. Compared to simple multiplication by a square wave voltage, as described, for example, in DE 10 2013 202 573 B3, these variants of the proximity switch and the method according to the invention combine the advantages of a true lock-in measurement, in which multiplication is performed with both a sine and a cosine function, followed by averaging. This means, for example, that a resistance measurement independent of the imaginary part of the resonant circuit impedance can be performed, with the advantages of the typically high signal-to-noise ratio of a lock-in measurement.

[0054] A further particular advantage of this variant of the present invention is that, in the event that the test current contains higher Fourier components, the signal components caused by these higher Fourier components can also be measured with the lock-in amplifier device by appropriately selecting the correlation frequency. In principle, different Fourier components or Fourier coefficients can be measured simultaneously. This feature of the present invention can be used, for example, to perform plausibility checks or to increase interference immunity.

[0055] In contrast to DE 41 42 680 A1, where the measurement for temperature compensation is performed at the same measurement frequency at which the target is detected, the proximity switch and method according to the invention allow for a separate measurement of the resistance and inductance of the resonant circuit coil, independent of the target itself. This is because the test current used for temperature compensation in the present invention is selected to be so low-frequency that the measurement of the target itself is only negligibly affected. More precisely, the test current induces only negligible eddy currents in the target. Therefore, the compensation of temperature effects is both simpler and more accurate in the present invention.

[0056] In principle, other measuring devices are also possible. For example, the relationship can be used to... ν=1 / (2πLC) Information about the inductance can also be obtained by monitoring or measuring the resonant circulating frequency v.

[0057] Furthermore, to determine the inductance, a test current can be applied to the coil of the resonant circuit, and the time response of the self-induced voltage can be measured after switching the test current off or on. The information about the inductance is then obtained via the time constant τ = L / R of an RL circuit.

[0058] Finally, information about the inductance can also be obtained by adjusting an AC bridge, in particular a Maxwell-Wien bridge.

[0059] In principle, for the realization of the invention, it is sufficient if both the resistance and the inductance of the resonant circuit coil are determined using the measuring device, in particular the lock-in amplifier. However, because the measurement of a voltage drop caused by a specific test current is highly dependent on the magnitude of the test current itself, the accuracy of the temperature compensation can be further increased in embodiments of the proximity switch according to the invention in which the measuring device, in particular the lock-in amplifier, is also configured to determine the test current applied to the resonant circuit.

[0060] The test current can be determined or measured using generally known methods. In a relatively simple embodiment of the proximity switch according to the invention, a measuring resistor is provided for measuring the test current, and the measuring device, in particular the lock-in amplifier device, is configured to measure the test current based on a voltage drop across the measuring resistor.

[0061] At large switching intervals, the resonant impedance of the coil system typically changes by only a few percent or even just a few parts per thousand. For sufficiently accurate temperature compensation, which requires a measurement accuracy of less than 0.1% for the resistance and inductance of the resonant circuit coil, the amplitude and phase of the test current must therefore be known with sufficient precision. When measuring the test current with a lock-in amplifier, the amplitude and phase of the test current can advantageously be measured in parallel with the voltage drop across the resonant circuit impedance. This significantly reduces the requirements for the stability and reproducibility of the test current. The test current through the resonant circuit can be measured, in particular, as the voltage drop across a measuring resistor Ri.

[0062] For a lock-in amplifier device, it is essential that the desired functionality of at least one lock-in amplifier is provided. For example, the lock-in amplifier device can have a first lock-in amplifier used to measure the resistance and inductance of the resonant circuit coil, and a second lock-in amplifier used to measure the test current through the resonant circuit.

[0063] In a particularly preferred embodiment of the proximity switch according to the invention, which is characterized by a reduced number of components, the lock-in amplifier device has exactly one lock-in amplifier for alternately measuring the resistance and / or inductance of the resonant circuit coil on the one hand and the test current on the other.

[0064] This embodiment corresponds to a variant of the inventive method in which the lock-in amplifier device is operated in multiplex mode, in which the resistance and / or inductance on the one hand and the test current on the other hand are determined alternately.

[0065] In principle, the present invention can be implemented using analog lock-in amplifiers. However, the lock-in amplifier device of the inductive proximity switch according to the invention preferably comprises at least one digital lock-in amplifier.

[0066] In principle, the present invention can be implemented with separately constructed measuring devices, in particular separate lock-in amplifier devices, compensation devices, and / or output stages. A particularly compact design can be achieved in embodiments where the measuring devices, in particular the lock-in amplifier devices, the compensation devices, and / or the output stage, are implemented in a single microcontroller.

[0067] In principle, any known components can be used as a power source to provide the test current. A controllable power source is particularly preferred, and a function generator can be used to control this power source.

[0068] In a particularly preferred embodiment of the proximity switch according to the invention, the test current is periodic. However, the test current does not necessarily have to be periodic, i.e., it does not necessarily have to have a fixed fundamental frequency and discrete frequency components.

[0069] In principle, the applied test current can also contain non-harmonic frequency components. This is then referred to as dual / multi-frequency excitation. A determined variation of the measurement frequency, i.e., the frequency of the test current, or possibly a statistical distribution (a key term here is "spread spectrum") of the test current frequencies is also possible. The advantages of these variants with a variable test current frequency are that the susceptibility of the measurement to interference coupled into the measurement frequency itself is reduced, and the plausibility of the measurement can be verified.

[0070] Test measurements have shown that the lower the frequency of the test current, the less dependent the measurement of the ohmic resistance of the resonant circuit coil is on the distance of a target from the proximity switch. Particularly advantageous embodiments of the proximity switch according to the invention are characterized in that the fundamental frequency of the test current is 100 Hz to 5 kHz, preferably 1 kHz to 3 kHz, and most preferably 1.5 kHz to 2.5 kHz. At such low frequencies, AC losses play a subordinate role, so that the determined resistance essentially corresponds to the ohmic resistance of the coil.

[0071] The resonant frequency of the resonant circuit can be, for example, on the order of values ​​above 100 kHz. It is particularly preferred that the resonant frequency of the resonant circuit is at least ten times higher than the fundamental frequency of the periodic test current.

[0072] Since the fundamental frequency of the test current can be significantly lower than the resonant frequency, the measured resistance may differ from the effective resistance at the resonant frequency by a small AC loss component. This difference may need to be taken into account for optimal temperature compensation.

[0073] The test current causes a voltage drop across an impedance Z of the resonant circuit, also known as the resonant circuit impedance Z. After amplification and, if necessary, signal matching, this voltage drop can be measured as a voltage U1(ω) using a complex transfer function G(ω) that depends on the angular frequency ω of the test current. The impedance Z(ω) of the resonant circuit depends on the angular frequency ω of the test current and is given by Ohm's law as follows: Z(ω)=U1(ω)G1(ω)I(ω)≈Rs(ω)+jωL≈RCu+jωL

[0074] It was assumed that the measurement frequency ω is sufficiently far from the resonant frequency of the resonant circuit and that the copper losses, i.e. the losses caused by the ohmic resistance of the resonant circuit coil, dominate the losses of the coil system.

[0075] U1(t) is a periodic function and can therefore be expanded into a series: U1(t)=a1.02+∑k=1∞((a1,k⋅cos(kω0t))+(b1,k⋅sin(kω0t))).

[0076] The real and imaginary parts of U1(t) at frequencies kω0 (k>1 and integer) are given by the even and odd Fourier coefficients a1k and b1k, respectively: a1k=2T∫0T(U1(t)⋅cos(kω0t))dt b1k=2T∫0T(U1(t)⋅sin(kω0t))dt, with T = 2π / ω0.

[0077] These Fourier coefficients can be measured or calculated by the lock-in amplifier, particularly a digital one. For coefficient a1k, this is done by multiplying by a cosine function, and for coefficient b1k, by multiplying by a sine function, followed by averaging. Essentially, the voltage drop is measured, and the Fourier coefficients are calculated from its time course. In this sense, it is a measurement of the respective Fourier coefficients.

[0078] In particularly preferred variants of the inductive proximity switch according to the invention, the lock-in amplifier device is configured to determine, for a fundamental frequency of the periodic test current, at least one, in particular the first, even Fourier coefficient and at least one, in particular the first, odd Fourier coefficient of the voltage drop caused by the periodic test current across the impedance of the resonant circuit.

[0079] Procedurally, variants are advantageous in which the lock-in amplifier device calculates at least one, in particular the first, even Fourier coefficient and at least one, in particular the first, odd Fourier coefficient of the voltage drop caused by the test current across the impedance of the resonant circuit for a fundamental frequency of a periodic test current.

[0080] With regard to determining the test current, method variants are advantageous in which, for determining the test current, a voltage drop caused by the test current across the measuring resistor is supplied to the lock-in amplifier device, in which the lock-in amplifier device calculates at least one, in particular the first, even Fourier coefficient and at least one, in particular the first, odd Fourier coefficient of the voltage drop caused by the test current across the measuring resistor at a fundamental frequency of a periodic test current, and in which, in particular, the test current is determined from the calculated Fourier coefficients.

[0081] Similarly, in further preferred embodiments of the inductive proximity switch according to the invention, the lock-in amplifier device is configured to determine, at a fundamental frequency of the periodic test current, at least one, in particular the first, even Fourier coefficient and at least one, in particular the first, odd Fourier coefficient of the voltage drop caused by the periodic test current across the measuring resistor. Here, too, determining the Fourier coefficients means that they are calculated based on a measured voltage drop, more precisely, based on the time course of a measured voltage.

[0082] In principle, it may be sufficient to evaluate (i.e., calculate) only the first even and the first odd Fourier coefficient. However, a significant advantage of using a lock-in amplifier is that, in principle, Fourier coefficients at higher harmonics can also be evaluated, either alternatively or additionally. This improves noise immunity and allows for plausibility checks.

[0083] From Fourier coefficients measured in the above-explained understanding of the term "measurement", the inductance L and, if applicable, the resistance Rs of the resonant circuit coil at a measurement frequency kω0 can then be calculated as follows, where ω0 is a fundamental frequency of the periodic test current and k is an integer. L=a1k2+b1k2G1.0(kω0)I0(kω0)sin[atan(a1k / b1k)−φI(kω0)−φG1(kω0)] Rcu=a1k2+b1k2G1.0(kω0)I0(kω0)cos[atan(a1k / b1k)−φI(kω0)−φG1(kω0)]

[0084] where I0(kω0) and φ I (kω0) the necessarily known amplitude and phase of the measuring current used and G 1,0 (kω0) and φ G1 (kω0) are the amplitude response and the phase shift of the signal matching.

[0085] Variants of the method according to the invention, in contrast to known solutions, allow for separate measurement of the resistance on the one hand and the inductance of the coil system on the other. This means that both drifts in resistance, i.e., the copper conductivity, and drifts in inductance with temperature can be specifically compensated for.

[0086] In particularly preferred embodiments of the inductive proximity switch according to the invention, the lock-in amplifier device is accordingly configured to determine the resistance and inductance of the resonant circuit coil using the Fourier coefficients which were determined based on the measured voltage drop across the resonant circuit impedance and the measuring resistance.

[0087] The compensation of drift effects, in particular temperature effects, can be further improved in embodiments of the inductive proximity switch and the method according to the invention, in which the frequency of the resonant circuit is also measured in parallel with the measurement of the inductance of the resonant circuit.

[0088] It is therefore particularly preferred if the inductive proximity switch according to the invention has a further measuring device for measuring a frequency of the resonant circuit.

[0089] In this context, a method according to the invention is particularly advantageous in that a frequency of the resonant circuit is also measured, that the measured frequency of the resonant circuit is evaluated with regard to a drift of a capacitance of the resonant circuit, and that the resonant circuit, the oscillator amplifier and / or the output stage are adjusted as a function of a value of the capacitance of the resonant circuit in order to compensate for, in particular temperature-related, changes in the resonant circuit which are due to, in particular temperature-dependent, changes in the capacitance of the resonant circuit.

[0090] The additional information about the frequency of the resonant circuit makes it possible, at least when no target is present, to make statements about existing changes or drifts in the resonant circuit capacitance.

[0091] Accordingly, preferred method variants are characterized in that the resistance and inductance of the resonant circuit coil are determined from at least one, in particular the first, even Fourier coefficient and at least one, in particular the first, odd Fourier coefficient of the voltage drop across the impedance of the resonant circuit using the measured test current.

[0092] Here too, determining essentially means calculating, especially according to the formulas given above.

[0093] The output stage can be of a generally known nature. For example, the output stage can comprise a transistor stage, in particular a push-pull stage. In particularly preferred embodiments, the output stage comprises a rectifier, in particular an amplifying rectifier, and a comparator. The comparator can, in particular, have an adjustable switching threshold.

[0094] In principle, it is possible for the output signal to be a signal that varies continuously with distance. However, variants of the inductive proximity switch according to the invention in which the output signal is a binary signal are particularly preferred.

[0095] Compensation itself can be achieved in various ways. For example, the loop gain of the oscillator amplifier can be adjusted. Another compensation option for oscillators where the amplitude doesn't drop abruptly but continuously when damped is a temperature-dependent adjustment of the amplitude measured via a rectifier, particularly one that amplifies the signal. Finally, the switching threshold of a comparator can also be changed to compensate for or offset temperature effects.

[0096] Therefore, variants of the method according to the invention are particularly preferred in which, when adjusting the oscillator amplifier and / or the output stage, a loop gain of the oscillator amplifier, a gain of a rectifier and / or a switching threshold of a comparator are set as a function of a measured resistance and / or a measured inductance.

[0097] This method can be expediently implemented with an embodiment of the proximity switch according to the invention, characterized in that the compensation device has a memory for storing values ​​for the resistance and / or inductance of the resonant circuit coil as well as at least values ​​for a loop gain of the oscillator amplifier, for a gain of a rectifier and / or for a switching threshold of a comparator, which are each to be set as a function of a measured resistance and / or a measured inductance, and that the compensation device is configured to set the stored values ​​for the gain of the oscillator amplifier, for the gain of the rectifier and / or for the switching threshold of the comparator according to the assignments stored in the memory.

[0098] In principle, it is also possible that, given a specific measured value of inductance or a specific combination of measured values ​​of inductance and resistance of the resonant circuit coil and / or the test current, adjustment values ​​can be calculated, on the basis of which the resonant circuit, the oscillator amplifier and / or the output stage are then adjusted.

[0099] Further advantages and features of the present invention are described with reference to the accompanying figures. These show: Fig. 1: a schematic representation of an inductive proximity switch according to the invention and Fig. 2: an embodiment of a proximity switch according to the invention.

[0100] Equivalent and equivalent components are usually provided with the same reference symbols in the figures.

[0101] A proximity switch 100 according to the invention is shown schematically in Fig. Figure 1 shows this proximity switch 100. It serves to detect a target T, in particular a metallic one, which, as schematically indicated, is located at a distance d from the proximity switch 100.

[0102] The proximity switch 100 comprises as essential components a resonant circuit 10 with a resonant circuit coil 12, an oscillator amplifier 20, a measuring device 50, which in the example shown is a lock-in amplifier device, and a compensation device 60.

[0103] The resonant circuit coil 12 can in principle also be a coil system consisting of several individual coils.

[0104] Furthermore, an output stage 30 is provided, which in the example shown interacts with the oscillator amplifier 20 and serves to provide an output signal 32 depending on an influence, in particular a damping, of the resonant circuit 10 by the target T to be detected.

[0105] It is also possible, in principle, for the output stage 30 to be additionally or alternatively connected to the resonant circuit 10, and for the output signal 32 to be derived from a physical state of the resonant circuit, for example, an oscillation amplitude. In the example shown, the output signal 32 is provided at a physical output of the output stage 30. However, this is not necessary for the realization of the invention. It is sufficient if the output signal 32 is provided somewhere in the proximity switch 100. Furthermore, in the illustrated variant, a current source 70 is provided for applying a time-varying test current 72 to the resonant circuit 10. In the illustrated embodiment, this current source 70 is implemented by an adjustable current source 51 and a function generator 53. The function generator 53 supplies a synchronization signal to the lock-in amplifier device via a line 57.

[0106] The lock-in amplifier device is operatively connected to the resonant circuit 10, which in Fig. Figure 1 schematically illustrates a connection between the resonant circuit 10 and the lock-in amplifier device. The lock-in amplifier device can also be configured to measure the resistance of the resonant circuit coil 12 based on a voltage drop across an impedance Z of the resonant circuit 10 caused by the test current 72.

[0107] The compensation device 60 works together with the lock-in amplifier device and is designed according to the invention to variably adjust either the oscillator amplifier 20 or the output stage 30 or both depending on a measured resistance of the resonant circuit coil 12 in order to compensate for changes in the resonant circuit 10 with temperature, which are due to temperature-dependent changes in the resistance.

[0108] In the example shown, the lock-in amplifier device is implemented by a digital lock-in amplifier 56, which can be implemented in a unit with the compensation device 60 and / or the output stage 30, for example in a microcontroller.

[0109] The lock-in amplifier device is configured according to the invention to measure the inductance L of the resonant circuit coil 12. In the illustrated embodiment, this is done on the basis of the voltage drop across a measuring resistor Ri caused by the test current 72. Finally, the compensation device 60 is additionally configured to adjust either the oscillator amplifier 20 or the output stage 30, or both, in order to compensate for, in particular, temperature-dependent changes in the resonant circuit 10, which are due to, in particular, temperature-dependent changes in the inductance L.

[0110] This compensation, depending on the measured values ​​for the resistance and inductance L of the resonant circuit coil 12, is particularly preferably carried out such that a switching distance, i.e., the distance at which the inductive proximity switch 100 switches from a first value to a second value when the target T approaches, remains as constant as possible within a specified temperature range. For example, it may be required that the switching distance does not drift by more than 10% in a temperature range of -40 °C to +85 °C or -25 °C to +75 °C.

[0111] This setting can be achieved with respect to the oscillator amplifier 20 via line 67, for example, by changing the loop gain V of the oscillator amplifier 20 depending on the measured resistance and / or inductance L of the resonant circuit coil 12. Alternatively or additionally, the compensation device 60 can also selectively change parameters of the output stage 30, such as gains and / or threshold values, via line 65, depending on a measured resistance and / or inductance L of the resonant circuit coil 12.

[0112] For this purpose, the compensation device 60, for example in a microcontroller or a comparable logic device, can have a memory 62 for storing values ​​for the resistance and / or the inductance L of the resonant circuit coil 12. Corresponding parameters can be stored in the memory 62 for these measured resistance and inductance L values. These parameters are to be set at the oscillator amplifier 20 and / or the output stage 30 when the corresponding resistance and inductance L values ​​are measured, so that the desired temperature compensation is ultimately achieved. These parameters can include values ​​for the loop gain of the oscillator amplifier 20, values ​​for the gain Vg of a rectifier 36 (see [reference to relevant section]). Fig. 2) and / or values ​​for a switching threshold S of a comparator 34 (see below) Fig. 2) act, which are each to be set depending on a measured resistance and a measured inductance L. The compensation device 60 can now be configured to set the stored values ​​for the gain V of the oscillator amplifier 20, for the gain Vg of a rectifier 36 and / or for the switching threshold of the comparator 34 according to the assignments stored in the memory 62.

[0113] In principle, it is also possible that, given a specific measured value of inductance L or a specific combination of measured values ​​of inductance and resistance of the resonant circuit coil and / or the test current, adjustment values ​​can be calculated, on the basis of which the resonant circuit, the oscillator amplifier and / or the output stage are then adjusted.

[0114] For example, a new calibration value for a currently existing combination of L and Rs can be calculated based on the values ​​of Rs and L stored during calibration at room temperature.

[0115] A more specific embodiment of an inductive proximity switch 100 according to the invention is described below with reference to Fig. 2 explained. The essential components here are again a resonant circuit 10, an oscillator amplifier 20, a lock-in amplifier device as a measuring device 50, a current source 70 for providing a time-varying test current 72, a compensation device 60 and an output stage 30.

[0116] The resonant circuit 10 consists of the one in Fig. The embodiment shown in Figure 2 consists of a resonant circuit coil 12 with an inductance L, a capacitor 14 with a capacitance C, and a resistive component 16 with a resistance. The representation of the resonant circuit 10 in Figure 2 is shown in Figure 2. Fig. Figure 2 is to be understood schematically in the sense of an equivalent circuit diagram. As a good approximation, one can generally assume that the resistance is essentially given by the copper resistance Rcu of the resonant circuit coil 12.

[0117] The oscillator amplifier 20 is connected to the resonant circuit 10 via a node between the capacitor 14 and the resonant circuit coil 12 and has an amplifier 21 and a resistor 22 in a feedback circuit.

[0118] In the illustrated embodiment, the output stage 30 comprises an amplifying rectifier 36 and a comparator 34. One input of the amplifying rectifier 36 is connected to an output of the amplifier 21. One output of the amplifying rectifier 36 is connected to an input of the comparator 34. Finally, a transistor stage is located at one output of the comparator 34, via which the output signal 32 is provided.

[0119] The lock-in amplifier device features in the Fig. In the example shown, two separate lock-in amplifiers are used: a first lock-in amplifier 52, which is used to measure the resistance and inductance L of the resonant circuit coil 12, and a second lock-in amplifier 54 for measuring the test current 72.

[0120] In the illustrated embodiment, the current source 70 for providing the time-varying test current 72 comprises a controllable current source 51, which is driven by a function generator 53. The function generator 53 can, for example, supply a periodic signal with a low fundamental frequency. In this context, low frequency means that the test current impressed into the resonant circuit 10 does not significantly excite the resonance of the resonant circuit 10, whose resonant frequency is typically above 100 kHz. The fundamental frequency of the test current is typically on the order of about 100 Hz to a few kHz. However, the periodic signal need not necessarily be a sinusoidal signal; that is, the time-varying test current 72 can contain significant components of higher harmonics.For example, the stream can have a rectangular shape, a triangular shape, or even a shape described as a rounded rectangular shape. The latter refers to a rectangular shape in which the corners of the rectangles are flattened or rounded.

[0121] The time-varying test current 72 is supplied to the resonant circuit 10 via the node between the resonant circuit coil 12 and the capacitor 14, to which the oscillator amplifier 20 is also connected. The function generator 53 also supplies a synchronization signal to the first lock-in amplifier 52 and the second lock-in amplifier 54 via a line 57.

[0122] A potential applied at the circuit node between the resonant circuit coil 12 and the capacitor 14 is fed to an input of the first lock-in amplifier 52 via a first signal conditioning device 92, which comprises a bandpass filter 93 and an amplifier 94 as its essential components. The bandpass filter 93 serves to block the actual oscillator oscillation, the frequency of which can be on the order of 100 kHz to 200 kHz. For sensors with a metallic end face, for example, an oscillator frequency of 10 kHz would also be possible. In principle, the oscillator frequency can also be higher than 200 kHz. The blocking does not need to be arbitrarily effective, but only sufficient to ensure that the amplitude of the signal supplied to the first lock-in amplifier 52 does not overload an analog-to-digital converter located there. Only a component varying with a frequency or the frequencies of the time-varying test current 72 should be allowed to pass through.This frequency, or these frequencies, can be in the range of a few kHz.

[0123] The circuit node between the resistive component 16 and the capacitor 14 is raised to a defined potential by means of an operational amplifier 74. The value of this potential is determined by a voltage divider formed by resistors 75 and 76. For example, this defined potential can be 600 mV. This causes the voltage at the output of the operational amplifier 74 to oscillate between approximately 500 mV and 600 mV. This ensures that the measurement range of the analog-to-digital converter in the second lock-in amplifier 54 is not exceeded. A measuring resistor Ri is located between the circuit node between the resistive component 16 and the capacitor 14 and an output of the operational amplifier 74. This resistor is used to measure the time-varying test current 72.A potential applied to the output of the operational amplifier 74 is fed via a second signal conditioning device 95, which, like the first signal conditioning device 92, comprises a bandpass filter 96 and an amplifier 95 as essential components, to an input of the second lock-in amplifier 54. The bandpass filter again serves to block the actual oscillator oscillation and to allow the frequency or frequencies of the time-varying test current 72 to pass through.

[0124] In a preferred embodiment of the invention, the first lock-in amplifier 52 determines, as explained above, based on the voltage supplied to its input via the first signal conditioning device 92, at least one even Fourier coefficient, in particular the first even Fourier coefficient a1k, and at least one odd Fourier coefficient, in particular the first odd Fourier coefficient b1k, of the voltage drop caused by the periodic test current 72 across the resonant circuit impedance Z. The first lock-in amplifier 52 can be a digital lock-in amplifier, so that when the term "determine" is used here, it means that the first lock-in amplifier 52 calculates the corresponding Fourier coefficients based on the voltage supplied to its input.These Fourier coefficients, in particular the Fourier coefficients a1k and b1k, are then transmitted via connecting lines located in . Fig. The values ​​shown schematically in Figure 2 are fed to the compensation device 60. In the compensation device 60, the resistance and inductance L of the resonant circuit coil 12 can, in principle, be calculated from the Fourier coefficients.

[0125] Particularly preferably, the second lock-in amplifier 54 further determines, on the basis of the voltage supplied to its input via the second signal conditioning device 95, at least one even Fourier coefficient, in particular the first even Fourier coefficient ak2 and at least one odd Fourier coefficient, in particular the first odd Fourier coefficient bk2 of the voltage drop caused by the periodic test current 72 across the measuring resistor Ri, at a fundamental frequency of the periodic test current 72.

[0126] The second lock-in amplifier 54 can also be a digital lock-in amplifier, so that when we speak of determining, we mean that the second lock-in amplifier 54 calculates the corresponding Fourier coefficients based on the voltage applied to its input. These Fourier coefficients, in particular the Fourier coefficients a2k and b2k, are also transmitted via connecting lines that are in Fig.The currents shown schematically in Figure 2 are fed to the compensation device 60. In the compensation device 60, the time-varying test current 72 can, in principle, be calculated from the Fourier coefficients. The test current 72 measured in this way is particularly preferably taken into account when calculating the resistance and inductance L of the resonant circuit coil 12. The accuracy of the measurement of the inductance L and the resistance of the resonant circuit coil 12, and thus the overall temperature compensation, can be significantly improved by measuring the test current 72.

[0127] In principle, it is also possible that the first lock-in amplifier 52 and the second lock-in amplifier 54 are implemented by one and the same digital lock-in amplifier, which are operated in multiplex mode, i.e., alternating in rapid succession.

[0128] The compensation device 60 includes a memory 62 in which values ​​for the resistance and inductance L of the resonant circuit coil 12 are stored. Corresponding to these values ​​or pairs of values, parameter settings can be stored in memory 62. These settings can be adjusted for specific pairs of values ​​for the resistance and inductance L of the resonant circuit coil 12 at the oscillator amplifier, the amplifying rectifier 36, and the comparator 34. Specifically, for example, a loop gain V of the oscillator amplifier 20 can be set via a line 65, a gain Vg of the amplifying rectifier 36 can be set via a line 64, and finally, a threshold of the comparator 34 can be set via a line 63.

[0129] As described above, the values ​​to be set can also be calculated based on measured values ​​of the inductance L, the resistance and / or the test current.

[0130] In principle, the logic or target setting can be chosen arbitrarily. A particularly common and useful target setting is that the switching interval, i.e., the interval at which the proximity switch 100 switches as a day T approaches, changes as little as possible with the temperature.

[0131] The present invention provides a novel inductive proximity switch and a novel method for operating inductive proximity switches. Because, according to the invention, the inductance of a resonant circuit coil, in particular in addition to its resistance, is measured using a measuring device, especially a lock-in amplifier, and taken into account when compensating for temperature effects, significant improvements in the compensation of temperature variations can be achieved. The accuracy of the evaluation can be further increased considerably if a time-varying test current, which is used to measure the resistance and inductance of the resonant circuit coil, is itself also measured, particularly using a lock-in amplifier. Reference symbol list 10 Oscillating circuit 12 Resonant circuit coil 14 Capacitor of the resonant circuit 10 16 Resistance of the resonant circuit 10 20 oscillator amplifiers 21 amplifiers 22 Resistance in the feedback circuit of the amplifier 21 30 Output stage 32 Output signal 34 Comparator 36 amplifying rectifier 50 measuring device 51 controllable power source 52 first lock-in amplifier 53 Function Generator 54 second lock-in amplifier 56 digital lock-in amplifiers 57 Synchronization line 60 Compensation device 62 storage locations 63 Line from the compensation device 60 to the rectifier 36 64 Line from the compensation device 60 to the comparator 34 65 Line from the compensation device 60 to the oscillator amplifier 20 67 Line from the compensation device 60 to the output stage 30 70 Power source 72 Test current 74 operational amplifiers 75 resistor for voltage divider 76 Resistor for voltage divider 80 microcontrollers 92 first signal conditioning device 93 bandpass filters 94 amplifiers 95 second signal conditioning unit 96 bandpass filters 97 amplifiers 100 Inductive proximity switches d Distance of the target T from the inductive proximity switch 100 G1(ω) Transfer function of the voltage drop across the resonant circuit impedance Z G2(ω) Transfer function of the voltage drop across measuring resistor Ri L Inductance of the resonant circuit coil Ri measuring resistor Rcu ohmic resistance of the resonant circuit coil 12 S Switching threshold of the comparator 34 T target to be detected V Loop gain of the oscillator amplifier 20 Vg Rectifier gain 36 Z resonant circuit impedance

Claims

[1] Inductive proximity switch (100) with a resonant circuit (10) with a resonant circuit coil (12), with an oscillator amplifier (20) and with an output stage (30) that interacts with the resonant circuit (10) and / or the oscillator amplifier (20) to provide an output signal (32) depending on an influence, in particular a damping, of the resonant circuit (10) by a target (T) to be detected, characterized by , that a measuring device (50) is available for measuring an inductance (L) of the resonant circuit coil (12) and that a compensation device (60) is provided which interacts with the measuring device (50) and is designed to adjust the resonant circuit (10), the oscillator amplifier (20) and / or the output stage (30) as a function of a measured inductance (L) of the resonant circuit coil (12) in order to compensate for, in particular temperature-related, changes in the resonant circuit (10) which are due to, in particular temperature-dependent, changes in the inductance (L). [2] Inductive proximity switch (100) according to claim 1, characterized by , that a current source (70) is available for supplying the resonant circuit (10) with a time-varying test current (72) and that the measuring device (50) is set up to measure the inductance (L) of the resonant circuit coil (12) on the basis of a voltage drop caused by the test current (72). [3] Inductive proximity switch (100) according to claim 2, characterized by, that the inductance (L) of the resonant circuit coil (12) is obtained from the imaginary part of a resonant circuit impedance (Z), across which a voltage drop is measured. [4] Inductive proximity switch (100) according to claim 2 or 3, characterized by , that the measuring device (50) is also configured to measure an effective resistance of the resonant circuit coil (12) based on the voltage drop caused by the test current (72) and that the compensation device (60) is also designed to adjust the resonant circuit (10), the oscillator amplifier (20) and / or the output stage (30) as a function of a measured resistance of the resonant circuit coil (12) in order to compensate for, in particular temperature-related, changes in the resonant circuit (10) which are due to, in particular temperature-dependent, changes in the resistance. [5] Inductive proximity switch (100) according to one of claims 1 to 4, characterized by , that the measuring device (50) is a lock-in amplifier device. [6] Inductive proximity switch (100) according to claim 4 or 5, characterized by , that the resistance of the resonant circuit coil (12) is obtained from the real part of a resonant circuit impedance (Z), across which a voltage drop is measured. [7] Inductive proximity switch (100) according to one of claims 2 to 6, characterized by , that the measuring device (50) is also designed to determine the test current (72) with which the resonant circuit (10) is supplied. [8] Inductive proximity switch (100) according to one of claims 2 to 7, characterized by , that a measuring resistor (Ri) is available for measuring the test current (72) and that the measuring device (50) is set up to measure the test current (72) based on a voltage drop across the measuring resistor (Ri). [9] Inductive proximity switch (100) according to one of claims 5 to 8, characterized by , that the lock-in amplifier device comprises a first lock-in amplifier (52) which serves to measure the resistance and inductance (L) of the resonant circuit coil (12), and that the lock-in amplifier device has a second lock-in amplifier (54) which is used to measure the test current (72) through the resonant circuit (10). [10] Inductive proximity switch (100) according to one of claims 5 to 8, characterized by , that the lock-in amplifier device has exactly one lock-in amplifier (56) for alternately measuring the resistance and inductance (L) of the resonant circuit coil (12) on the one hand and the test current (72) on the other. [11] Inductive proximity switch (100) according to one of claims 5 to 10, characterized by that the lock-in amplifier device includes at least one digital lock-in amplifier. [12] Inductive proximity switch (100) according to any one of claims 1 to 11, characterized by , that the measuring device (50), the compensation device (60) and / or the output stage (30) are implemented in a microcontroller (80). [13] Inductive proximity switch (100) according to one of claims 2 to 12, characterized by , that the power source (70) for providing the test current (72) is a controllable power source (51) and that a function generator (53) is available for controlling the controllable power source (51). [14] Inductive proximity switch (100) according to one of claims 2 to 13, characterized by , that the test current (72) is periodic. [15] Inductive proximity switch (100) according to one of claims 2 to 14, characterized by , that the fundamental frequency of the test current (72) is 100 Hz to 5 kHz, preferably 1 kHz to 3 kHz and particularly preferably 1.5 kHz to 2.5 kHz. [16] Inductive proximity switch (100) according to claim 14 or 15, characterized by , that the resonant frequency of the resonant circuit (10) is at least ten times greater than the fundamental frequency of the periodic test current (72). [17] Inductive proximity switch (100) according to any one of claims 2 to 16, characterized by , that the lock-in amplifier device is configured to determine, at one or the fundamental frequency of the periodic test current (72), at least one, in particular the first, even Fourier coefficient and at least one, in particular the first, odd Fourier coefficient of the voltage drop caused by the periodic test current (72) across the impedance (Z) of the resonant circuit (12). [18] Inductive proximity switch (100) according to any one of claims 2 to 17, characterized by, that the lock-in amplifier device is configured to determine, at one or the fundamental frequency of the periodic test current (72), at least one, in particular the first, even Fourier coefficient and at least one, in particular the first, odd Fourier coefficient of the voltage drop caused by the periodic test current (72) across the measuring resistor (Ri). [19] Inductive proximity switch (100) according to one of claims 17 or 18, characterized by , that the lock-in amplifier device is set up to determine the resistance and inductance (L) of the resonant circuit coil (12) using the Fourier coefficients determined from the measured voltage drop across the resonant circuit impedance (Z) and the measuring resistance. [20] Inductive proximity switch (100) according to any one of claims 1 to 19, characterized by, that a further measuring device for measuring a frequency of the resonant circuit (10) is available. [21] Inductive proximity switch (100) according to any one of claims 1 to 20, characterized by , that the output stage (30) has a rectifier (36), in particular an amplifying rectifier, and / or a comparator (34). [22] Inductive proximity switch (100) according to one of claims 1 to 21, characterized by , that the output signal (32) is a binary signal. [23] Inductive proximity switch (100) according to one of claims 1 to 22, characterized by , that the compensation device (60) has a memory (62) for storing values ​​for the resistance and / or inductance (L) of the resonant circuit coil (12) as well as at least values ​​for a loop gain (V) of the oscillator amplifier (20), for a gain (Vg) of a rectifier (36) and / or for a switching threshold (S) of one or the comparator (34), which are each to be set depending on one or the measured resistance and / or the measured inductance (L), and that the compensation device (60) is configured to adjust the stored values ​​for the gain (V) of the oscillator amplifier (20), for the gain (Vg) of the rectifier (36) and / or for the switching threshold (S) of the comparator (34) according to the assignments stored in the memory (62). [24] Method for operating an inductive proximity switch (100) according to any one of claims 1 to 23, wherein, due to an influence, in particular a damping, of the resonant circuit (10) by a target (T) to be detected, an output signal (32) is generated and provided by an output stage (30), characterized by , that the inductance (L) of the resonant circuit coil (12) is measured with the measuring device (50) and that the resonant circuit (10), the oscillator amplifier (20) and / or the output stage (30) is adjusted as a function of a value of the measured inductance (L) of the resonant circuit coil (12) in order to compensate for, in particular temperature-related, changes in the resonant circuit (10) which are due to, in particular temperature-dependent, changes in the inductance (L). [25] Method according to claim 24, characterized by , that the resonant circuit (10) is subjected to a time-varying test current (72) and that the inductance (L) of the resonant circuit coil (12) is measured with the measuring device (50) on the basis of a voltage drop caused by the test current (72). [26] Method according to claim 25, characterized by , that the resistance of the resonant circuit coil (12) is measured with the measuring device (50) on the basis of a voltage drop caused by the test current (72) and that the resonant circuit (10), the oscillator amplifier (20) and / or the output stage (30) are adjusted as a function of a value of the measured resistance of the resonant circuit coil (12) in order to compensate for, in particular temperature-related, changes in the resonant circuit (10) which are due to, in particular temperature-dependent, changes in the resistance, [27] Method according to any one of claims 24 to 26, characterized by , that the measuring device (50) is a lock-in amplifier device. [28] Method according to claim 27, characterized by , that the lock-in amplifier device calculates at least one, in particular the first, even Fourier coefficient and at least one, in particular the first, odd Fourier coefficient (b1k) of the voltage drop (U(t)) caused by the test current (72) across the impedance (Z) of the resonant circuit (10) at a fundamental frequency of a periodic test current (72). [29] Method according to claim 27 or 28, characterized by , that, in order to determine the test current (72), a voltage drop caused by the test current (72) across the measuring resistor (Ri) is supplied to the lock-in amplifier device, that the lock-in amplifier device calculates at least one, in particular the first, even Fourier coefficient (a1k) and at least one, in particular the first, odd Fourier coefficient (b1k) of the voltage drop caused by the test current (72) across the measuring resistor (Ri) at or the fundamental frequency of the periodic test current (72), and that in particular the test current (72) is determined from the calculated Fourier coefficients. [30] Method according to any one of claims 27 to 29, characterized by , that the resistance and inductance (L) of the resonant circuit coil (12) are determined from at least one, in particular the first, even Fourier coefficient (a1k) and at least one, in particular the first, odd Fourier coefficient (b1k) of the voltage drop across the impedance of the resonant circuit using the measured test current (72). [31] Method according to any one of claims 24 to 30, characterized by , that, in addition, a frequency of the resonant circuit (10) is measured, that the measured frequency of the resonant circuit (10) is evaluated with regard to a drift of a capacitance (C) of the resonant circuit (10) and that the resonant circuit (10), the oscillator amplifier (20) and / or the output stage (30) are adjusted as a function of a value of the capacitance (C) of the resonant circuit (10) in order to compensate for, in particular temperature-related, changes in the resonant circuit (10) which are due to, in particular temperature-dependent, changes in the capacitance (C) of the resonant circuit (10). [32] Method according to any one of claims 24 to 31, characterized by , that the measuring device (50), in particular the lock-in amplifier device, is operated in multiplex mode, in which the resistance and / or inductance (L) on the one hand and the test current (72) on the other hand are determined alternately. [33] Method according to any one of claims 24 to 32, characterized by , that when adjusting the oscillator amplifier (20) and / or the output stage (30), a loop gain (V) of the oscillator amplifier (20), a gain (Vg) of a rectifier (36) and / or a switching threshold of a comparator (34) are set as a function of a measured resistance and / or a measured inductance (L).

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