Method for determining the phase currents in a multi-phase H-bridge for controlling a PMSM using a single shunt resistor

Abstract

Method for measuring the phase currents of a motor with n motor phases (U, V, W), where n is a positive integer greater than or equal to 3, comprising the steps - Generating a PWM signal for each of the n motor phases (U, V, W), • where the PWM signals have a common PWM period (T P ) and • where each of the PWM signals is generated at a duty cycle different from 0% and 100% within a common PWM period (T P ) has a rising edge and a falling edge and • where the phase voltages for the corresponding n motor phases (U, V, W) are generated using the PWM signals and • where each of these PWM signals has a duty cycle and • where each of these PWM signals is a left-aligned PWM signal with a left temporal alignment side or a right-aligned PWM signal with a right temporal alignment side and • where all PWM signals are aligned in the same way and • where in the case of left-aligned PWM signals, the PWM signals have a minimum time offset (ΔT min ) of the rising edge signals within the PWM period (T P ) to each other and in the case of right-aligned PWM signals, the PWM signals have a minimum temporal offset (ΔT min ) of the falling edge signals within the PWM period (T P ) to each other; - specifying the measuring windows (1, 2, 3, 4) and specifying the associated measuring times (SP1, SP2) depending on the specified or permitted duty cycles; - Specifying or allowing a duty cycle for each motor phase of the n motor phases (U, V, W) by specifying or allowing a duty cycle for the respective associated PWM signal of this respective motor phase, • whereby the duty cycle is limited so that the measurement is possible at the specified measuring times; - Modulation of the PWM signals with the respective specified or permitted duty cycle of the respective PWM signals; - Measurement of n-1 motor phase currents of the n motor phase currents (I U , I V , I W ) by measuring the shunt current (I S ) by a common shunt resistor (R SS ) or a quantity derived from it (V Shunt ) in these valid measurement windows.

Description

generic term

[0001] The invention relates to a method for determining the phase currents in a multi-phase H-bridge for controlling a PMSM. Introduction and state of the art

[0002] The control of permanent magnet synchronous machines (PMSMs) is now almost exclusively carried out using three-phase H-bridges. For this purpose, we refer first to Fig. 1. Each of the three motor phases (U, V, W) shown here as an example can be controlled either via a high-side switch (HS U , HS V , HS W ) with the supply voltage (V BAT ) or via a low-side switch (LS U , LS V , LS W ) can be connected to the ground potential (GND) as a reference potential. In addition, the Fig. 1 can be switched to high impedance. By controlling it with pulse-width modulated signals (PWM), any voltage between the reference potential (GND) and the value of the supply voltage (V BAT ) at each of the motor phases (U, V, W) (see Fig. 1).

[0003] Field-oriented control (FOC) is a method for commutation of permanent magnet synchronous motors (PMSMs) in which the periodically changing input and output variables are converted to DC voltage and DC current variables. In particular, it offers very high dynamics and high electrical efficiency by recalculating the output vector for each individual period of the PWM signals (PWM period T P ) as well as the possibility of field weakening operation.

[0004] The basic prerequisite for the FOC algorithm, which is generally known from the state of the art, is knowledge of the three motor phase currents (I U , I V , I W ) at any time. For this reason, two shunt resistors are currently used for the measurement (see Fig. 2). In the example of Fig. 2, a first amplifier (V U ) the first motor phase current (I U ) by means of a first shunt resistor (R SU ) in the current path of the low-side switch (LS U ) of the half bridge (HS U , LS U ) for the first motor phase (U) and a second amplifier (V V ) the second motor phase current (I V ) by means of a second shunt resistor (R SV ) in the current path of the low-side switch (LS V ) of the half bridge (HS V , LS V ) for the second motor phase (V). Whenever in the example the Fig. 2 one of the motor phases (U, V) through one of these two low-side switches (LS U , LS V ) is connected to the reference potential (GND), the motor phase current (I U , I V ) of this motor phase (U, V) through the respective shunt resistor (R SU , R SV ) of the respective motor phase (U, V) and generates a respective voltage drop there, which can be measured. Since the motor (as a sufficient approximation) represents a node in the Kirchhoff sense, the third motor phase current (I W ) can be easily calculated from the two current values ​​of the two measured motor phase currents (I U , I V ) can be calculated.

[0005] This device of Fig. 2 from the prior art has the disadvantage that two shunt resistors (R SU , R SV ) and thus also two operational amplifiers (V U , V V) are required to implement this measuring method. Since considerable motor phase currents (I U , I V ) via the respective shunt resistors (R U , R V ) must be dissipated against the reference potential (GND), the shunt resistors (R SU , R SV ) are either provided discretely externally within an integrated circuit or are implemented within such an integrated circuit with a very large chip area requirement. The amplifiers (V U , V V ) are relatively complex components. This is particularly problematic in the automotive sector.

[0006] In principle, it is possible to use a single shunt (R SS ) all three motor phase currents (I U , I V , I W ) to reconstruct (see Fig. 3). Here, knowledge of the switching states of the low-side switches (LS U , LS V , LS W) and the high-side switch (HS U , HS V , HS w ) is exploited. Of the four possible switching states of each half-bridge, only three states are permitted, since the high-side switch and the low-side switch of a half-bridge may never be active at the same time.

[0007] In such a configuration, the motor phase currents (I U , I V , I W ) directly to the common shunt resistor (R SS ) can be measured (see Fig. 4). In the Fig. An example switching state is shown in Figure 4. The high-side switch (HS U ) of the first motor phase (U) is closed in this exemplary switching state, whereby the first half-bridge (HS U , LS U ) the motor phase current (I U ) of the first motor phase (U) into the motor (not shown). The low-side switch (LS V) of the second motor phase (V) is closed, whereby the second half-bridge (HS V , LS V ) the second motor phase current (I V ) of the second motor phase (V) from the motor (not shown). The low-side switch (LS W ) of the third motor phase (W) is closed, whereby the third half-bridge (HS W , LS W ) the third motor phase current (I W ) of the third motor phase (W) is taken from the motor (not shown). Due to Kirchhoff's node rule, the following applies: I U +I V +I W =0A. Therefore, across the common shunt resistor (R SS ) due to the shunt current flowing through it (I S ), which in this switching state example corresponds to the first motor phase current (I U ), a shunt voltage (V Shunt ) corresponding to the first motor phase current (I U) of the first motor phase (U). Similar drawings are possible for the other current supply cases. As can easily be seen, the shunt current (I S ) by the common shunt resistance (R SS ) in the example of Fig. 4 the first motor phase current (I U ). It is a general convention to define motor phase currents as positive when they flow into the motor. Inverting the switch positions of the example (LS U closed, HS V closed, HS W closed, all other switches HS U , LS V , LS W opened), would -I U by the common shunt resistance (R SS ) flow.

[0008] The main problem with this method is that such a switching state vector (hereinafter also referred to simply as vector) cannot be applied at every possible duty cycle to every PWM period long enough to reduce the shunt voltage across the common shunt resistor (R SS ) to be able to measure. Limiting factors here include the inductance of the common shunt resistor (R SS ), the settling time of the common amplifier (V S ), the time that the common amplifier (V S ) a subsequent analog-to-digital converter (ADC) is required for the measured value acquisition (sample time). These critical conditions regarding measurability exist: a. Generally below a minimum modulation amplitude, as well as b. whenever the applied voltages to two of the three motor phases (U, V, W) are approximately equal.

[0009] Microchip's prior art Application Note 1299 provides a detailed explanation of this topic. As mentioned above, knowledge of the motor phase currents (I U , I V , I W ) but a condition for the applicability of the said FOC algorithm.

[0010] The current state of the art is to control the motor using a frequency related to the PWM period (T P ) for each motor phase (U, V, W) center-aligned PWM modulation, in which the PWM blocks are generated from the temporal center of the respective PWM period (T P ) can be shifted appropriately. This influences the time length of the output vectors so that the measurement of the shunt voltage V Shunt at the common shunt resistance (R SS) without changing the effective duty cycle of the respective half-bridge. Please also refer to Microchip Application Note AN1299.

[0011] This procedure is in Fig. 5 shown. Fig. 5a shows an example of the switching states of the low-side switches (LS U , LS V , LS W ) with block commutation of the three motor phases (U, V, W) based on the control voltages at the control terminals of these low-side switches (LS U , LS V , LS W ) before shifting the time blocks within the PWM period (T P ) as a function of time t over the course of a PWM period (T P ). A higher level symbolizes a closed low-side switch and a lower level symbolizes an open low-side switch. Fig. 5b shows an example of the switching states of the low-side switches (LS U , LS V , LSW ) during block commutation of the three motor phases (U, V, W) after exemplary shifting of the center position of the previously centered block commutation of the control signal of the second low-side switch (LS V ) of the middle motor phase (V) within the PWM period (T P ).

[0012] Before the shift, the measurement window (ΔT SDT ) accordingly Fig. 5a is very short. Here, it is assumed that the measurement technology used is not capable of measuring the temperature within this short time window (ΔT SDT ) from the state of the art.

[0013] After the postponement according to Fig. 5b is the measurement window (ΔT SDTimp ) by the hatched area compared to the original measurement window (ΔT SDT) is extended in time. Here, it is assumed, for example, that if the hatched area is sufficiently large, the measuring technology used is now able to measure in this extended time window (ΔT SDTimp ) from the state of the art.

[0014] In this process of Fig. 5, however, depending on the rotor angle, two different motor phase currents are present at the common shunt resistor (R SS ) measurable. This leads to a more or less significant jump in the FOC control input variables the moment two phase currents other than those in the previous PWM period become "visible" (due to current ripple, entrained / corrected measurement errors, etc.). These errors, or their correction, cause audible noise in some motors and also reduce the system's efficiency.

[0015] The need to switch currents cannot be avoided at large amplitudes. The temporal shift of PWM blocks within the PWM period (T P ) has another audible effect: At the moment of shifting, the average output voltage is determined by the shift of the PWM blocks within the PWM period (T P ) itself distorts the signal. This injects corresponding error currents into the motor, which are audible. Therefore, it is important to avoid moving the PWM blocks as much as possible while still having sufficient measurement windows available.

[0016] This means that information about at least two of the three motor phase currents (I U , I V , I W ) is required to control the motor correctly and to extract the motor state correctly.

[0017] When measuring current with the common shunt resistor (R SS) depends on the shunt current (I S ) by this common shunt resistance (R SS ) thus from the switch position of the switches (HS U , HS V , HS w , LS U , LS V , LS W ) and a minimum time is required for this current measurement, since the motor phase currents (I U , I V , I W ) must settle before the measurement. Thus, in principle, one can only determine a motor phase current of the motor phase currents (I U , I V , I W ) meaningfully if at least the state of one half-bridge is different from all the others. In this model, there are only two states per half-bridge. (Output of the half-bridge with the positive supply voltage (V BAT ) connected = ON, output of the half bridge connected to the reference potential (GND) = OFF).

[0018] This means that (see Fig. 6) only two times two states (T 11 , T 21 , T22 , T 12 ) in which the motor phase currents (I U , I V , I W ) at the common shunt resistor (R SS ) can be measured. In certain states, individual intervals (T 11 , T 21 , T 22 , T 12 ) no longer have a temporal extension, because e.g. T 02 is larger. However, it should always be possible to determine the motor phase currents (I U , I V , I W ) in order to operate the FOC control continuously. Fig. The signal curve shown in Figure 6 has the PWM period T P and can be divided into different time periods (T 01 , T 11 , T 21 , T 02 , T 22 , T 12 , T 03 ) in which certain switches are closed and other switches are open.

[0019] The Fig. 7 to 10 show the different switch positions of the switches (HS U , HS V , HS W , LS U , LS V , LS W ) in the individual intervals Fig. 6. Fig. 7 shows the switching states in the intervals T 01 and T 03 . Fig. 8 Shows the switching states in the interval T 02 . Fig. 9 shows the switching states in the intervals T 11 and T 12 . Fig. 10 shows the switching states in the intervals T 21 and T 22 .

[0020] It should be noted that when transitioning from one switch position to another, it is important to prevent both switches of a half-bridge, i.e. the low-side switch and the high-side switch, from being closed at the same time. This would lead to very high cross-currents and even the destruction of the power section. To prevent this, a so-called dead time is typically inserted when transitioning from one switch position to another, during which both switches are open. However, taking the dead time into account in all figures and the description would make it very confusing. Therefore, the dead time is only mentioned here, and it is always assumed in this disclosure that such a dead time is also still present during control.

[0021] Since a vector representation is often used in conjunction with space vector modulation, this should also be briefly mentioned here. A total of six different switch positions exist in which a voltage other than zero is applied to the motor. These switch position combinations are shown in Fig. 11 as vectors. Additionally, there are two zero vectors in which a voltage of zero is applied to the motor. The corresponding switch positions are shown in Fig. 7 and Fig. 8 shown. Fig. 9 and Fig. 10 each correspond to a vector in Fig. 11. In space vector modulation, a central, rotating voltage vector is applied to the motor by switching back and forth between a neighboring pair of vectors and at least one zero vector. This allows any central voltage vector to be generated that lies within the area spanned by the two neighboring vectors.

[0022] A widely used method for measuring motor phase current is the aforementioned sum current measurement in the ground path. With this measurement, two motor phase currents (I U , I V , I W ). In the case shown above ( Fig. 6) is measured with the sum current measurement of the shunt current (I S ) in the interval T 11 and T 12 the third motor phase current (I W ) of the third motor phase (W). In the interval T 21 and T 22the first motor phase current (I U ) with a negative sign -I U measured. The third motor phase current (I V ) is then determined by the node set I U +I W +I U =0A. The motor is therefore considered as a node. The measured values ​​of the motor phase currents (I U , I V , I W ) are required to determine the current amplitude and the phase position of the resulting current vector for the FOC control.

[0023] The previously mentioned possibility (Microchip AN1299) to shift the blocks so that during a PWM period (T P) to provide at least one time range in which a motor phase current flows and can be measured requires a dynamic shift. This shift distorts the average voltage at the motor terminals. During operation, it may be necessary to make a sudden, large shift, which is typically audible.

[0024] Fig. 12 shows a scheme for shifting the PWM blocks according to Microchip AN1299 with a first temporal measuring point SP1 and a second temporal measuring point SP2.

[0025] The problem with this method based on the Microchip AN1299 is that the time shift must be canceled at a certain point in time. This creates a voltage error for the duration of a PWM cycle. Fig. Figure 13 shows a schematic example of the occurrence of a temporary voltage error that occurs when the PWM block offset for a motor phase (U, V, W) is canceled at a given time. This voltage error causes, among other things, a brief error in the motor current, which is therefore audible and perceived as disturbing.

[0026] From DE 10 2010 001 181 A1 (claim 1) a method is known in which three successive PWM intervals are generated shifted relative to one another in such a way that measurement windows of sufficient temporal size are obtained, characterized in that the clock patterns are shifted in ascending order of duty cycle, the switching element with the shortest duty cycle being switched on first.

[0027] This sorting results in the following: every time the order of the PWM blocks needs to be changed due to the sorting described in DE 10 2010 001 181 A1, an incorrect average voltage is generated at the swapped PWM blocks for exactly one clock cycle. This results in audible noise.

[0028] DE 10 2015 219 200 A1 presents a method for measuring the phase currents of a multi-phase motor with a shunt resistor, in which the necessary measurement windows are created by inserting a corresponding pulse pattern. However, it should be noted that the insertion itself causes an error in the average output voltage. The motor current, in turn, is largely generated by integration over this error. This also means that if the pulse pattern itself is very short, it leads to a longer-lasting error in the motor current. This error, in turn, is audible in the form of noise, even if the current error is in the per-mille range. A complete compensation of the error, e.g., by an attached complementary pulse sequence, is also not possible, since, for example,The voltage drop across the resistive elements in the arrangement, even when the rotor is stationary, leads to different current increases in the measurement interval and the compensation interval, so that even with a compensation pulse sequence, a residual error in the current remains, usually audible. Thus, the method according to DE 10 2015 219 200 A1 also leads to audible noise in its own way. The aim of this application is to avoid this noise.

[0029] DE 10 2005 062 759 A1 also dynamically modifies the PWM control patterns. Section

[0009] of DE 10 2005 062 759 A1 discloses "[...] that the clock patterns are shifted in time or changed when a measurement window falls below a predetermined size, such that a measurement window of sufficient size is formed, essentially no noise perceptible to humans is generated, and simultaneously, using a current measuring resistor, all phase currents can be measured. [...]" As already explained above, the dynamic shift claimed in the technical teaching of DE 10 2005 062 759 A1 inevitably leads to audible noise. Once such a shift is applied, the only option is to reduce the noise generated by the shift, either through a compensating pulse sequence or through the dynamics of the shift itself. This is also indicated by the formulation "[...]] essentially no noise perceptible by humans [...]" in the text of DE 10 2005 062 759 A1 in its section

[0009] . However, in the technical teaching presented here, such noises generated by the PWM generation are completely avoided. Task

[0030] The invention is therefore based on the object of creating a solution which does not have the above disadvantages of the prior art and has further advantages.

[0031] This object is achieved by a method according to claim 1. Solution to the task

[0032] Part of the proposal is now not to use center-aligned PWM, but rather a method similar to Microchip's application note AN1299. First procedure

[0033] A method is developed to determine the magnitude of the phase currents (I U , I V , I W) in a multi-phase, in particular three-phase H-bridge, for controlling a BLDC motor and in particular a permanent magnet synchronous machine (PMSM) by field-oriented control (FOC), wherein PWM signals are generated for the H-bridge, which allow the determination of all, in particular, three motor phase currents using only a single common shunt resistor (R SS ) is possible.

[0034] In order to meet the requirements for current measurement in that it is measured at each PWM period (T P) is completely possible and does not produce any measurement-related jumps over time, the invention does not use the concept of center-aligned PWM for modulating the control of the motor phases (U, V, W). Instead, the temporal measurement windows for the current measurement are used as the starting point for generating the PWM signals. The exemplary case in which all three phases (LS U , LS V , LS W ) a duty cycle of 50% is output, is in Fig. 14. The falling edges (falling edge of the half-bridge, ie rising edge of the control signal of the associated low-side switch) have a fixed time interval (ΔT min ) to each other to measure the motor phase currents (I U , I V , I W ) (see areas 1 and 2). The duty cycle of the individual half-bridges is controlled exclusively by the temporal position of the falling edges (see Fig. 14) relative to the temporal position of the rising edges within the PWM period (T P ). Since the order of the rising edges always remains the same, the same two motor phase currents of the motor phase currents (I U , I V , I W ) so that no measurement-related jumps occur. Of course, the concept could also be inverted and the measurement windows could be generated between the rising edges.

[0035] The edges of the PWM signals for controlling the switches are generated in a PWM controller. The PWM controller typically includes a PWM counter with a PWM counter value. The PWM counter counts, for example, a PWM base clock pulse, typically starting with a PWM counter value of, for example, 0 at the beginning of the PWM period (T P ) at a time arbitrarily t=0s up to a PWM counter end value at the PWM period end of the PWM period (TP ) at time t=T P from the beginning of the period at t=0s. The PWM controller then generates the PWM signals to control the switches within a PWM period depending on the current PWM counter reading. At certain predefined counter readings, the timer compare values, the state of one or more half-bridges is switched. The timer compare values ​​thus specify the positions of the edges and thus the duty cycle and the temporal position of the blocks within a PWM period. Other implementations of the PWM controller are expressly conceivable.

[0036] Since the current flow of the motor phase currents (I U , I V , I W ) into the motor only the differences of the average phase voltages (V U , U V , U W) are relevant, the rising edges for each individual PWM cycle can be shifted as required, as long as the time intervals between them are not changed. This corresponds to a common offset of the Timer Compare values ​​of the rising edges. The maximum symmetrical control is thus effectively even greater than before. In practice, by shifting the star point potential, the applied voltages are optimally distributed between the reference potential (GND) and the operating voltage (V BAT ) is "adjusted." In the extreme case of maximum modulation, this results in a modulation with increased controllability, in which the starpoint voltage changes over time. The controllability corresponds to that of a so-called flat-bottom modulation.

[0037] Furthermore, it should be noted that the proposed measurement method introduces a systematic error due to current ripple. Depending on the time of the rising edge of a motor phase whose current is being measured, the measurement takes place at different points along the current waveform within a PWM period.

[0038] However, this error creates a) no sudden changes and behaves b) absolute to the amplitude of the motor phase current.

[0039] The latter means that in the typically interesting case of a high load and thus a large current amplitude, the relative error is very small. Furthermore, with the appropriate performance of a microcontroller that executes the FOC control, for example, this error can be completely calculated and thus compensated, since the winding resistance and inductance of a motor phase must be known anyway to calculate a sensorless FOC, and these are easy to measure.

[0040] The subject of the invention is therefore a method for PWM generation for a three-phase half-bridge, which makes it possible to measure all three motor phase currents (I U , I V , I W ) using a single shunt measurement on a common shunt resistor (R SS ) in each PWM period (T P). The essence of the invention is to generate, in a first embodiment, a type of left-aligned PWM, in which the edges are not exactly left-aligned, but are offset by a temporal offset of the windows by at least the minimum measurement window size (ΔT min ) the measuring windows for the motor phase current measurement are reliably generated in such a way that a measurement of the motor phase currents (I U , I V , I W ) is always possible in these minimum measurement windows.

[0041] The essence of the invention is also to generate, in a second embodiment, a type of right-sided PWM (English: right-aligned PWM), in which the edges are not arranged exactly right-aligned, but by a temporal offset of the windows at least by the temporal minimum measurement window size (ΔT min) the measuring windows for the motor phase current measurement are reliably generated in such a way that a measurement of the motor phase currents (I U , I V , I W ) is always possible in these minimum measurement windows.

[0042] This first proposal thus provides an implementation of a single-shunt FOC control (FOC control with a single common shunt resistor (R SS ), Fig. 3) which, in terms of noise, efficiency and stability, exceeds the performance of a dual-shunt FOC control (FOC control with a first shunt resistor (e.g. R SU ) and with a second shunt resistor (e.g. R SV ), Fig. 2). One aspect of the invention is the concept for PWM generation, which allows the reconstruction of all three motor phase currents (I U , I V , I W ) at any time with only a single common shunt resistor (R SS ) is possible.

[0043] Advantages of this first proposal are: a) Silent operation of PMSM with single-shunt FOC becomes possible. b) The lower circuit complexity results in a cost reduction compared to dual-shunt FOC solutions ( Fig. 2). c) The proposed solution is particularly efficient. It allows for complete implementation on a 16-bit CPU with a 48 MHz clock speed, a program size of approximately 9 kbytes, and a CPU load of approximately 60%.

[0044] This first proposal thus improves the usual methods for PWM generation of a single-shunt FOC control in that all three motor phase currents (I U , I V , I W ) of a PMSM (but also of a BLDC motor) in each PWM period (T P) can be measured without generating measurement-related jumps. Thus, in terms of efficiency, dynamics, and motor noise, this proposed implementation achieves the performance of a dual-shunt FOC control, which requires one more shunt resistor and one more amplifier than the invention.

[0045] The method described here first uses a different approach than the method described in Microchip’s Application Note 1299. It is used here in the example of Fig. 14 uses a left-aligned PWM whose PWM blocks for the individual motor phases (U, V, W) are time-aligned against each other by a constant minimum measurement window size (ΔT min ), within which the switching sequence shown is always fulfilled. As already mentioned, this achieves two significant advantages over the rest of the state of the art: • No dynamic change of the displacement of the PWM blocks is required. This eliminates the described noise and • the measurement of the motor phase currents using the analog-to-digital converter (ADC) and the shunt resistor (R SS ) and the amplifier (V S ) always occurs at the same temporal positions in the PWM pattern within a PWM period (T P ) and always the same two phase currents of the three phase currents (I U , I V , I W ) measured.

[0046] This simplifies the evaluation and can also lead to advantages in the behavior of the observer. The disadvantage is that in the example, the first motor phase (U) is switched on for at least the minimum switch-on time interval (ΔT Umin ) (see Fig. 14) must remain switched on. In the example of the Fig. 14 would be the duration of this minimum switch-on time interval (ΔT Umin ) of the first motor phase (U) is twice the length of the minimum time (ΔT min ). This results in a minimum duty cycle, which cannot be undercut due to the process. Limiting the duty cycle also automatically limits the voltage, which is used by the process as the phase voltage (V U , V V , V W ) can be output at each motor phase (U, V, W). This is dominated by the transient response of the current measurement and the measurement speed of the analog-to-digital converter (ADC), which is not shown. This duty cycle limitation can easily lead to a 10% voltage drop compared to the other methods, which, however, have the disadvantage of increased noise. Refined second method

[0047] In the first method described above, the blocks are switched unilaterally at the respective beginning of the respective PWM period (see Fig. 11) (left-aligned PWM) or alternatively at the respective end of the respective PWM period (T P ) (right aligned PWM) and then offset from each other by a constant, preferably equal value, preferably at least by the time length of the minimum measurement window size (ΔT min ), shifted against each other. This creates two measurement windows, the length of which is crucial for stabilizing the values ​​and successfully measuring the two motor phase currents (I U , I V , I W ) is sufficient. This situation is for the case of positioning at the beginning of the PWM period (T P ) in Fig. 14 shown as an example.

[0048] The main disadvantage of this initially described method was that a duty cycle of 100% could not be achieved. A motor phase voltage could therefore neither be fully switched on nor fully switched off. As a result, the full motor phase voltage equal to the value of the positive supply voltage (V BAT ) is applied to the motor. This means that the full motor power cannot be utilized and the efficiency is reduced. One could now try to solve the problem by adjusting the temporal positioning of the blocks within the PWM period (T P ) is switched appropriately. Such switching requires shifting the blocks against each other again during operation at a very large duty cycle close to 100%. However, this generates noise, which should generally be avoided.

[0049] The idea of ​​this refined second method is the left- or right-sided temporal orientation of the blocks within the PWM period (T P ) and the minimum temporal shift by a constant, preferably equal value, preferably with the temporal length (ΔT min ) the minimum measurement window size, from the first described procedure and instead of the previously rejected option due to noise, now changing the measurement times for carrying out the measurements.

[0050] The refined second method described here dispenses with the second advantage of the first method. The measurement of the two motor phase currents (I U , I V , I W ) by means of the analog-to-digital converter (ADC), which is not shown in the figures, and the common shunt resistor (R SS ) and the amplifier (V S) no longer always occurs at the same time position in the PWM pattern of the PWM period (T P ). By doing so, this second, modified method is then able to set the maximum output voltage of the half-bridges to the value of the positive supply voltage (V BAT ) and the measured motor phase currents (I U , I V , I W ) without generating noise due to the described temporary voltage errors resulting from temporal block shifts during operation.

[0051] If the waveform of one of the motor phase voltages (U, V, W) in the first method reaches a 100% duty cycle or a 0% duty cycle, a measurement interval is lost. The inventive finding of the refined second method is that while one measurement window is lost, another measurement window opens up that can be used instead. A) The idea is that in one PWM period (T P ) several potential time windows for the measurement are specified and, depending on the specified duty cycle, suitable measurement windows are selected from this plurality of measurement windows, preferably by the PWM controller which controls the PWM signal generation. B) A suitable measurement window exists when a. Firstly, at least a minimum measurement window size (ΔT min) which is sufficient to ensure temporal stability of the measurement result of the motor phase current to be measured (I U , I V , I W ) from the beginning of the measurement window to the measurement time within the measurement window by allowing the measuring device to oscillate and b. Secondly, for the duration of the measurement of the motor phase current to be measured (I U , I V , I W ) the state of all three half-bridges is not the same and does not change. C) There are always two measurement windows per PWM period (T P ) was selected. D) The number of potential measurement windows provided may be greater than or equal to four.

[0052] It should be noted that a motor typically has n phases with n ∈ N (N={1, 2, 3... to infinity}). Such a motor then has n-1 currents to be measured. Therefore, n-1 temporal measurement windows are required.

[0053] In the following, the PWM control and the selection of the sampling times for the motor phase current measurement for the normal case as well as the resulting special cases for an exemplary three-phase motor are described as examples.

[0054] Fig. 15a shows the normal case of positioning the measurement times within a PWM period (T P ). The two measurement points (SP1, SP2) are indicated by the two black arrows.

[0055] Fig. 15b shows the positioning of the measurement times within a PWM period (T P ) at the lower control limit for the first motor phase (U). The two measurement points (SP1, SP2) are indicated by the two arrows.

[0056] Fig. 15c shows the positioning of the measurement times within a PWM period (T P) at the upper control limit for the first motor phase (U). The two measurement points (SP1, SP2) are indicated by the two arrows.

[0057] Fig. 15d shows the positioning of the measurement times within a PWM period (T P ) at the lower control limit for the second motor phase (V). The two measurement points (SP1, SP2) are indicated by the two arrows.

[0058] Fig. 16a shows the positioning of the measurement times within a PWM period (T P ) at the upper control limit for the second motor phase (V). The two measurement points (SP1, SP2) are indicated by the two arrows.

[0059] Fig. 16b shows the positioning of the measurement times within a PWM period (T P ) at the lower control limit for the third motor phase (W). The two measurement points (SP1, SP2) are indicated by the two arrows.

[0060] Fig. 16c shows the positioning of the measurement times within a PWM period (T P ) at the upper control limit for the third motor phase (W). The two measurement points (SP1, SP2) are indicated by the two arrows.

[0061] The exemplary rising edges of the individual PWM channels (motor phases U, V, W) are also preferably delayed by a constant time (ΔT min ) shifted against each other, as in Fig. This figure also shows the normal case, which prevails at sufficiently low voltage amplitudes. This case corresponds exactly to the situation of the Fig. 14, which is present in the first method. Therefore, at sufficiently low voltage amplitudes (typically <90% of the maximum amplitude), no difference is visible from the first-described method. This difference only becomes apparent when the first-described method reaches the voltage limit for a motor phase. This is evident in the other illustrated Fig. 15b to 15d and 16a to 16c. As already mentioned, the solution pursued here does not use any limitation of the duty cycle or any dynamic temporal shifting of the PWM blocks during operation. Instead, the PWM blocks are shifted at a constant time by at least the minimum measurement window size (ΔT min) to each other and either a left-aligned or right-aligned PWM is used. The positive properties of the solution pursued here are exclusively due to the constant shift of the PWM blocks by a constant period of time, the minimum measurement window size (ΔT min ), in combination with a suitable choice of measurement times (SP1, SP2 in the Fig. 15 and Fig. 16) for the total current at the common shunt resistor (R SS ) was concluded. In Fig. the PWM signal of the first motor phase (U) reaches its lower control limit, so that the measuring intervals after Fig. no longer exist. The current through the common shunt resistor (R SS ) will then be added to the Fig. 15b with black arrows, the measured values ​​are measured at the measuring times (SP1, SP2) shown in Figure 15b. The result is two different measured values ​​for two different motor phase currents than in the normal case, from which the third motor phase current can always be calculated. Further special cases at maximum control are shown in the Fig. 15c to 15d and 16a to 16c. Further special cases can be constructed, but the same applies: Due to the constant temporal shift of the modulation of the half-bridge PWM signals relative to each other, two suitable measurement intervals can always be found for two different motor phase currents. The essence of the implementation thus lies in the "intelligence" of the case-specific selection of the measurement times depending on the duty cycles of the PWM signals of the three motor phases (U, V, W).

[0062] The Fig. Figure 17 shows four exemplary measurement windows for a three-phase motor, numbered 1 to 4. The following table can be used to indicate in which two of these four potential measurement windows measurements are taken and under which conditions. TV1 TV2 TV3 Messung 1 Messung 2 Offsetmessung 0<TV1<1 0<TV2<1 0<TV3<1 2 3 1 / 4 0 0<TV2<1 0<TV3<1 3 4 1 / 2 0 1 0<TV3<1 3 4 - 0 0<TV2<1 1 2 3 - 1 0<TV2<1 0<TV3<1 2 3 4 1 0 0<TV3<1 3 4 - 1 0<TV2<1 0 2 3 - 0<TV1<1 0 0<TV3<1 3 4 1 0<TV1<1 0 1 1 2 - 0<TV1<1 1 0<TV3<1 1 2 4 0<TV1<1 1 0 1 2 -

[0063] A 0 in the TV1 column means that the first duty cycle (TV1) of the PWM signal of the first motor phase (U) is below a first lower threshold value (SW uU ) lies.

[0064] A 1 in the TV1 column means that the first duty cycle (TV1) of the PWM signal of the first motor phase (U) is above a first upper threshold value (SW oU ) lies.

[0065] A 0 in the TV2 column means that the second duty cycle (TV2) of the PWM signal of the second motor phase (V) is below a second lower threshold (SW uV ) lies.

[0066] A 1 in the TV2 column means that the second duty cycle (TV2) of the PWM signal of the second motor phase (V) is above a second upper threshold (SW oV ) lies.

[0067] A 0 in the TV3 column means that the third duty cycle (TV3) of the PWM signal of the third motor phase (W) is below a third lower threshold (SW uW ) lies.

[0068] A 1 in the TV3 column means that the third duty cycle (TV3) of the PWM signal of the third motor phase (W) is above a third upper threshold value (SW oW ) lies.

[0069] The number in the Measurement 1 column indicates in which of the four measurement windows the Fig. 17 prefers the first phase current measurement of the total current through the common shunt resistor (R SS ) of the two measurements to be performed within the PWM period (T P ) takes place.

[0070] The number in the Measurement 2 column indicates in which of the four measurement windows the Fig. 17 prefers the second phase current measurement of the total current through the common shunt resistor (R SS ) of the two measurements to be performed within the PWM period (T P ) takes place.

[0071] In the table, an additional column for the offset measurements has been added as a third measurement. In the intervals specified there, the value of the common shunt resistance (R SS ) measured shunt current (I S ) Zero amperes. In this currentless state, it is then possible to evaluate the measuring system itself. This third measurement is therefore not used to measure the motor phase currents (I U , I V , I W ), but for measuring error detection and correction. The measurement error determined in this way can then be used, for example, to correct the offset of the amplifier (V S). "1 / 4" means that this additional, optional third offset measurement can be performed in measurement window 1 and / or measurement window 4. If nothing is noted for this additional third offset measurement, then there is currently no time interval suitable for offset measurement for the respective combination. The offset measurement can still be performed this way, since the offset is a very slowly changing quantity, and the special combinations in which no offset measurement is possible typically only prevail for a short time.

[0072] Fig. 18 shows 5 measuring windows for an example 4-phase motor. Characterization of the invention

[0073] The invention thus relates to a method for measuring the motor phase currents of a motor with n motor phases (U, V, W), where n is a positive integer greater than or equal to 3.

[0074] The method comprises generating a PWM signal for each n motor phases (U, V, W), wherein the PWM signals have a common PWM period (T P ). Each of the PWM signals has a duty cycle different from 0% and 100% within a common PWM period (T P ) a rising edge and a falling edge. Using these PWM signals, the phase voltages for the corresponding n motor phases (U, V, W) are determined by means of the half bridges (HS U , LS U ), (HS V , LS V ), (HS W , LS W )) are generated, which are controlled by these PWM signals. Each of these PWM signals has a duty cycle (TW1, TW2, TW3). Thus, the output signals of the half-bridges ((HS U , LS U ), (HS V , LS V ), (HS W , LS W)) have the same duty cycles (TW1, TW2, TW3). Each of these PWM signals is a left-aligned PWM signal with a left temporal alignment side or a right-aligned PWM signal with a right temporal alignment side, whereby all PWM signals are aligned in the same way, for example, either all left-aligned or all right-aligned. "Equally aligned" here means that the edges on the alignment side are offset by the temporal length of the minimum measurement window size (ΔT min ) are typically fixedly shifted relative to each other. PWM blocks that are shifted so constantly relative to each other are still considered to be unidirectional. The PWM blocks of the Fig. 14 are therefore still considered left-aligned for the purposes of this disclosure. In the case of left-aligned PWM signals, the PWM signals therefore fall below a minimum temporal offset (ΔT min ) of the rising edge signals within the PWM period (T P ) to each other. In the case of right-aligned PWM signals (see e.g. Fig. 19) the PWM signals fall below a minimum time offset (ΔT min ) of the falling edge signals within the PWM period (T P) to each other. In the method, the potential measurement windows (1, 2, 3, 4) are determined or already specified by design. This can also be done by programming or setting parameters of the PWM control. Typically, potential measurement times within these potential measurement windows are also determined or specified with the measurement windows. As a rule, the measurement corresponds to a gate pulse during which the current values ​​are determined. This is considerably smaller than the measurement window itself. If only these gate pulses are specified for the measurement, the measurement windows that result from the structure of the PWM signals and their arrangement among each other are naturally selected. The step of determining the measurement windows can be limited to specifying or determining the measurement times without departing from the scope of this disclosure.For the operation of the motor, duty cycles are specified, determined, or permitted for each motor phase of the n motor phases (U, V, W) during this operation. This is done by specifying or permitting such a duty cycle (TW1, TW2, TW3) for the respective associated PWM signal of this respective motor phase. The PWM signals for controlling the half-bridges and thus the motor phase voltages are modulated by the half-bridges with the respective specified or permitted duty cycle (TW1, TW2, TW3) of the respective PWM signals. From the previously determined potential measurement windows, some measurement windows are determined as valid measurement windows depending on the thus specified or permitted duty cycles (TW1, TW2, TW3). This then also determines the corresponding measurement times (SP1, SP2) depending on the specified or permitted duty cycles (TW1, TW2, TW3).For the purposes of this disclosure, a determination of associated measurement times (SP1, SP2) depending on the specified or permitted duty cycles (TW1, TW2, TW3) automatically also always represents the determination of the corresponding measurement windows as valid measurement windows depending on the thus specified or permitted duty cycles (TW1, TW2, TW3). The measurement of n-1 motor phase currents of the n motor phase currents (I. U , I V , I W ) by measuring the shunt current (I S ) by a common shunt resistor (R SS ) or by measuring a quantity derived from it (V Shunt ) in these previously determined valid measurement windows at the typically determined measurement times (SP1, SP2).

[0075] A further refinement of the method comprises the modulation of the phase voltages of the motor phases (U, V, W) in such a way that between successive falling edges of the PWM signals of the phase voltages of the n motor phases (U, V, W) of at least two different motor phases of the n motor phases (U, V, W) within one PWM period (T P ) a temporally constant time offset at least in the temporal size of a minimum measurement window size (ΔT min ) or, in the case of the other-sided PWM alignment, that between successive rising edges of the PWM signals of the phase voltages of the n motor phases (U, V, W) of at least two different motor phases of the n motor phases (U, V, W) within one PWM period (T P ) a temporally constant time offset at least in the temporal size of a minimum measurement window size (ΔT min ) consists.

[0076] In a modified variant of the method, n+1 potential measurement windows (1, 2, 3, 4) are determined depending on the specified or permitted duty cycles.

[0077] In a further modified method variant, n-1 measuring windows of the n+1 potential measuring windows are determined as valid measuring windows depending on the specified or permitted duty cycles.

[0078] Preferably, the number of valid measurement windows in a PWM period (T P ) at least n-1.

[0079] In a procedure based on this for a motor with n=3 motor phases, the two valid measurement windows are selected for a first measurement (measurement 1) and a second measurement (measurement 2) of the motor phase currents (I U , I V , I W) from the four possible measurement windows (1, 2, 3, 4) according to the following table depending on the duty cycles (TW1, TW2, TW3) of the PWM signals of the motor phases (U, V, W) and the threshold values ​​(SW uU , SW oU , SW uV , SW oV , SW uW , SW oW ): TV1 TV2 TV3 Messung1 Messung2 SW uU <TV1<SW oU SW uV <TV2<SW oV SW uW <TV3<SW oW 2 3 SW uU >TV1 SW uV <TV2<SW oV SW uW <TV3<SW oW 3 4 SW uU >TV1 TV2>SW oV SW uW <TV3<SW oW 3 4 SW uU >TV1 SW uV <TV2<SW oV TV3>SW oW 2 3 TV1>SW oU SW uV <TV2<SW oV SW uW <TV3<SW oW 2 3 TV1>SW oU SW uV >TV2 SW uW <TV3<SW oW 3 4 TV1>SW oU SW uV <TV2<SW oV SW uW >TV3 2 3 SW uU <TV1<SW oU SW uV >TV2 SW uW <TV3<SW oW 3 4 SW uU <TV1<SW oU SW uV >TV2 TV3>SW oW 1 2 SW uU <TV1<SW oU TV2>SW oV SW uW <TV3<SW oW 1 2 SW uU <TV1<SW oU TV2>SW oV TV3>SW oW 1 2

[0080] In a method based on this for a motor with n=3 motor phases, an offset measurement window is selected from the four potential measurement windows (1, 2, 3, 4) to measure an offset voltage of the measuring device for measuring the shunt current (I S ) by a common shunt resistor (R SS ) according to the following table: TV1 TV2 TV3 OffsetMessung SW uU <TV1<SW oU SW uV <TV2<SW oV SW uW <TV3<SW oW 1 oder 4 SW uU >TV1 SW uV <TV2<SW oV SW uW <TV3<SW oW 1 oder 2 SW uU >TV1 TV2>SW oV SW uW <TV3<SW oW SW uU >TV1 SW uV <TV2<SW oV TV3>SW oW TV1>SW oU SW uV <TV2<SW oV SW uW <TV3<SW oW 4 TV1>SW oU SW uV >TV2 SW uW <TV3<SW oW TV1>SW oU SW uV <TV2<SW oV SW uW >TV3 SW uU <TV1<SW oU SW uV >TV2 SW uW <TV3<SW oW 1 SW uU <TV1<SW oU SW uV >TV2 TV3>SW oW SW uU <TV1<SW oU TV2>SW oV SW uW <TV3<SW oW 4 SW uU <TV1<SW oU TV2>SW oV TV3>SW oW

[0081] In a further variant of the process, the shunt current (I S ) by the common shunt resistance (R SS ) or a quantity derived from it (V Shunt) in a time measurement window within the PWM period (T P ), in which the ideal shunt current (I S ) should be 0A, as a measurement error and the use of the detected measurement error for correction or as another control parameter to correct the detected errors.

[0082] According to the invention, it is proposed to use an engine control system which is provided and designed to carry out a method as described above.

[0083] The method described above and the motor control mentioned above are particularly suitable for use in noise reduction when controlling a motor. Advantage of the invention

[0084] The described methods enable the measurement of the motor phase currents (I U , I V , I W ) with only one common shunt resistor (R SS) between the reference potential (GND) and the common base point (FP) of the half-bridges. This refined method also has the advantage of generating virtually no noise. List of reference symbols ADC Analog-to-digital converter. The analog-to-digital converter is the common amplifier (V S ) preferably downstream. It preferably converts the output signal, the analog measured value signal (MW) of the common amplifier (V S ) into a measured value for the shunt current (I S ) around; BLDC brushless DC motor; ΔT min fixed time interval of the falling edge of the half-bridge, ie the rising edges of the control signals of the low-side switches within one PWM period (T P ). Preferably, this fixed time interval corresponds to the minimum measurement window size; ΔT SDToriginal time measurement window according to the state of the art (in the example of Fig. 5a for measuring the first motor phase current (I U ) without shifting the temporal PWM block of the exemplary second motor phase (V)); ΔT SDTimp extended time measurement window according to the state of the art (in the example of Fig. 5b for measuring the first motor phase current (I U ) after shifting the temporal PWM block of the exemplary second motor phase (V)); ΔT Umin Minimum switch-on time interval of the first motor phase (U) (see Fig. 14); FOC Field-Oriented Control (FOC) is a method for commutation of permanent magnet synchronous machines (PMSMs); FP common base point of the first low-side switch (LS U ) and the second low-side switch (LS V ) and the third low-side switch (LS W ); GND reference potential (ground); HS U first high-side switch for the first half-bridge to supply the first motor phase (U) with positive supply voltage (V BAT ). Preferably, the switch is a transistor; HS V second high-side switch for the second half-bridge to supply the second motor phase (V) with positive supply voltage (V BAT ). Preferably, the switch is a transistor; HS W third high-side switch for the third half-bridge to supply the third motor phase (W) with positive supply voltage (V BAT ). Preferably, the switch is a transistor; I S Shunt current through the common resistance (R SS ); I U Motor phase current of the first motor phase (U) into the first half bridge (HS U , LS U ) into it; IV Motor phase current of the second motor phase (V) into the second half bridge (HS V , LS V ) into it; I W Motor phase current of the third motor phase (W) into the third half bridge (HS W , LS W ) into it; LS U First low-side switch for the first half-bridge to supply the first motor phase (U) with a negative supply voltage from the reference potential (GND). The switch is preferably a transistor; LS V Second low-side switch for the second half-bridge to supply the second motor phase (V) with a negative supply voltage from the reference potential (GND). The switch is preferably a transistor; LS WThird low-side switch for the third half-bridge to supply the third motor phase (W) with negative supply voltage from the reference potential (GND). The switch is preferably a transistor; MW analog measured value of the shunt voltage (V Shunt ), which are connected to the common shunt resistor (R SS ) due to the shunt current (I S ) drops. The analog measured value of the shunt voltage (V Shunt ) is the output signal of the common amplifier (V S ); PMSM permanent magnet synchronous machine; PWM pulse width modulation; R SU first shunt resistor preferably between first low-side switch (LS U ) and reference potential (GND); R SV second shunt resistor preferably between second low-side switch (LS V ) and reference potential (GND); R SWthird shunt resistor preferably between third low-side switch (LS W ) and reference potential (GND); R SS common shunt resistor preferably between the common base point of the first low-side switch (LS U ) and the second low-side switch (LS V ) and the third low-side switch (LS W ) on the one hand and the reference potential (GND) on the other; SP1 first time measurement point. (In the example of Fig. 12 for measuring the third motor phase current (I W )); SP2 second time measurement point. (In the example of Fig. 12 for measuring the first motor phase current (I U )); SW uU first lower threshold value for the first duty cycle (TV1) of the PWM signal of the first motor phase (U). SW uV second lower threshold for the second duty cycle (TV2) of the PWM signal of the second motor phase (V). SWuW third lower threshold for the third duty cycle (TV3) of the PWM signal of the third motor phase (W). SW oU first upper threshold value for the first duty cycle (TV1) of the PWM signal of the first motor phase (U). SW oV second upper threshold value for the second duty cycle (TV2) of the PWM signal of the second motor phase (V). SW oW third upper threshold value for the third duty cycle (TV3) of the PWM signal of the third motor phase (W). T P temporal period, PWM period, of the PWM excitation of the motor phases (U, V, W); TV1 first duty cycle of the PWM signal of the first motor phase (U); TV2 second duty cycle of the PWM signal of the second motor phase (V); TV3 third duty cycle of the PWM signal of the third motor phase (W); U first motor phase; V second motor phase; V BAT positive supply voltage; V U first amplifier for detecting the first motor phase current (I U ) of the first motor phase (U) by means of an associated first shunt resistor (R SU ) between the first low-side switch (LS U ) and reference potential (GND); V V second amplifier for detecting the second motor phase current (I V ) of the second motor phase (V) by means of an associated second shunt resistor (R SV ) between the second low-side switch (LS V ) and reference potential (GND); V W third amplifier for detecting the third motor phase current (I W ) of the third motor phase (W) by means of an associated third shunt resistor (R SW ) between the third low-side switch (LS W ) and reference potential (GND); V S common amplifier for detecting the shunt current (I S ) by the common shunt resistance (R SS) between the common base point of the first low-side switch (LS U ) and the second low-side switch (LS V ) and the third low-side switch (LS W ) on the one hand and the reference potential (GND) on the other; V Shunt Shunt voltage V Shunt at the shunt resistance (R SS ) W third motor phase;

Claims

[1] Method for measuring the phase currents of a motor with n motor phases (U, V, W), where n is a positive integer greater than or equal to 3, comprising the steps - Generating a PWM signal for each of the n motor phases (U, V, W), • where the PWM signals have a common PWM period (T P ) and • where each of the PWM signals is generated at a duty cycle different from 0% and 100% within a common PWM period (T P ) has a rising edge and a falling edge and • where the phase voltages for the corresponding n motor phases (U, V, W) are generated using the PWM signals and • where each of these PWM signals has a duty cycle and • where each of these PWM signals is a left-aligned PWM signal with a left temporal alignment side or a right-aligned PWM signal with a right temporal alignment side and • where all PWM signals are aligned in the same way and • where in the case of left-aligned PWM signals, the PWM signals have a minimum time offset (ΔT min ) of the rising edge signals within the PWM period (T P ) to each other and in the case of right-aligned PWM signals, the PWM signals have a minimum temporal offset (ΔT min ) of the falling edge signals within the PWM period (T P ) to each other; - specifying the measuring windows (1, 2, 3, 4) and specifying the associated measuring times (SP1, SP2) depending on the specified or permitted duty cycles; - Specifying or allowing a duty cycle for each motor phase of the n motor phases (U, V, W) by specifying or allowing a duty cycle for the respective associated PWM signal of this respective motor phase, • whereby the duty cycle is limited so that the measurement is possible at the specified measuring times; - Modulation of the PWM signals with the respective specified or permitted duty cycle of the respective PWM signals; - Measurement of n-1 motor phase currents of the n motor phase currents (I U , I V , I W ) by measuring the shunt current (I S ) by a common shunt resistor (R SS ) or a quantity derived from it (V Shunt ) in these valid measurement windows. [2] Method according to claim 1 - Modulation of the phase voltages of the motor phases (U, V, W) in such a way that between successive falling edges of the PWM signals of the phase voltages of the n motor phases (U, V, W) of at least two different motor phases of the n motor phases (U, V, W) within one PWM period (T P ) a temporally constant time offset at least in the temporal size of a minimum measurement window size (ΔT min ) consists. [3] Method according to claim 1 - Modulation of the phase voltages of the motor phases (U, V, W) in such a way that between successive rising edges of the PWM signals of the phase voltages of the n motor phases (U, V, W) of at least two different motor phases of the n motor phases (U, V, W) within one PWM period (T P ) a temporally constant time offset at least in the temporal size of a minimum measurement window size (ΔT min ) consists. [4] Method according to one of the preceding claims, characterized in that - that the number of measurement windows in one PWM period (T P ) is at least n-1. [5] Engine control system which is intended and designed to carry out a method according to one or more of the preceding claims. [6] Use of a method according to one or more of the preceding method claims and / or an engine control according to the preceding claim for noise reduction when controlling an engine.

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