ISOLATED POWER CONVERTER

Abstract

The disclosed concepts relate to an isolated power converter circuit. The circuit includes a transformer comprising a primary winding and a secondary winding. An intermediate tap of the primary winding is coupled to a source. The circuit further includes a first power amplifier coupled to a first end of the primary winding, a second power amplifier coupled to a second end of the primary winding, a first rectifier coupled to a first end of the secondary winding, and a second rectifier coupled to a second end of the secondary winding. A capacitance provided at the first and second power amplifiers, and an inductance provided by the transformer, form a primary resonant circuit.A capacitance provided at the first rectifier and the second rectifier, and an inductance provided by the transformer, form a secondary resonant circuit.

Description

TECHNICAL AREA The present disclosure relates to isolated power converter circuits, systems incorporating the isolated power converter circuits, and methods for transmitting power via an isolation transformer. BACKGROUND Current integrated solutions for isolated gate drivers face the challenge that a significant portion of the space is occupied by an isolation transformer, which is necessary to provide an isolated power supply for the secondary side's power domain. To reduce both the space and cost required for this component, high-frequency switching topologies that utilize reduced inductance, such as air-core magnet structures, are frequently employed in isolated power conversion. However, the use of transformers with an air core (or cores with similar magnetic permeability) results in a reduced coupling factor, which typically decreases the efficiency of power transfer from the primary to the secondary side in commonly used power conversion topologies. Furthermore, operation at frequencies in the range of several tens of MHz or higher increases driver losses. Furthermore, the pursuit of space savings, minimal material usage, and reduced costs means that isolated converters, which can be integrated into standardized IC (integrated circuit) packages, are more attractive. Commercial solutions typically implement step-down or 1:1 conversion, where the input voltage is higher than or equal to the output voltage. However, there may be situations where step-up conversion is desired (where the output voltage is higher than the input voltage). In this case, the limited available space, especially within an IC package, may not permit power conversion with voltage boost solely through the transformer winding turns ratio, as increasing the number of windings on the secondary side can require more area than can be provided. Accordingly, there is a need for an improved isolated power converter. SUMMARY According to one aspect, an isolated power converter circuit is provided. The circuit comprises a transformer containing a primary winding and a secondary winding. An intermediate tap of the primary winding is coupled to a power source. The circuit further includes a first power amplifier coupled to the first end of the primary winding, a second power amplifier coupled to the second end of the primary winding, a first rectifier coupled to the first end of the secondary winding, and a second rectifier coupled to the second end of the secondary winding. A capacitance provided at the first and second power amplifiers, and an inductance provided by the transformer, form a primary resonant circuit.A capacitance provided at the first rectifier and the second rectifier, and an inductance provided by the transformer, form a secondary resonant circuit. According to another aspect of the present disclosure, a system is provided comprising a first load terminal for connection to a power source, a second load terminal connected to a load, and an isolated power converter circuit as described above. The first load terminal is coupled to the source of the isolated power converter circuit, and the second load terminal is coupled to the first rectifier and the second rectifier of the isolated power converter circuit. According to yet another aspect of the present disclosure, a method for transferring power from a first side of an isolation transformer to a second side of the isolation transformer is provided. The method comprises providing power to an intermediate tap of a primary winding on the first side of the isolation transformer, generating a first resonance in the primary winding using a first power amplifier coupled to a first end of the primary winding and a second power amplifier coupled to a second end of the primary winding, rectifying a current caused in a secondary winding on the second side of the isolation transformer, which is inductively coupled to the primary winding, using a first rectifier coupled to a first end of the secondary winding and a second rectifier coupled to a second end of the secondary winding, and providing the rectified current for an output. The expert will recognize additional features and advantages upon reading the following detailed description and upon examining the accompanying drawings. BRIEF DESCRIPTION OF THE DRAWINGS The present disclosure is illustrated by way of example and without limitation in the figures of the accompanying drawings, in which the same reference numerals refer to similar or identical elements. The elements of the drawings are not necessarily to scale relative to one another. The features of the various illustrated examples may be combined, provided they are not mutually exclusive. Fig. 1 represents a generalized circuit diagram of an isolated power transformer according to a prior art embodiment; Fig. 2 represents an implementation of the circuit diagram of Fig. 1 according to a prior art embodiment; Fig. 3 represents a generalized circuit diagram of an isolated power transformer according to one embodiment; Fig. 4 represents an implementation of the circuit diagram of Fig. 3 according to another embodiment; Fig. 5 represents an implementation of the circuit diagram of Fig. 6.3 including a gate driver circuit according to another embodiment; Fig. 6 shows an implementation of the circuit diagram of Fig. 5 including a specific implementation of the gate driver circuit according to a further embodiment; Fig. 7 shows various waveforms while the circuit shown in Fig. 6 is operated; Fig. 8 shows a circuit diagram of an isolated power converter including a specific implementation of the power amplifiers according to a further embodiment; Fig. 9 shows a timing diagram with respect to the circuit of Fig. 8; Fig. 10 shows a system including isolated power converter circuits according to one embodiment; and Fig. 11 shows a flowchart of a method for transferring power from a first side of an isolation transformer to a second side of the isolation transformer according to one embodiment. DETAILED DESCRIPTIONThe disclosed concepts relate to an isolated power converter circuit. The circuit includes a transformer comprising a primary winding and a secondary winding. An intermediate tap of the primary winding is coupled to a source. The circuit further includes a first power amplifier coupled to a first end of the primary winding, a second power amplifier coupled to a second end of the primary winding, a first rectifier coupled to a first end of the secondary winding, and a second rectifier coupled to a second end of the secondary winding. A capacitance provided at the first and second power amplifiers, and an inductance provided by the transformer, form a primary resonant circuit.A capacitance provided at the first rectifier and the second rectifier, and an inductance provided by the transformer, form a secondary resonant circuit. Since the first power amplifier, the second power amplifier, and the transformer (which provides primary leakage inductance and magnetizing inductance) form a primary resonant circuit, there is potential to excite a resonant voltage on the primary side of the transformer, resulting in an initial voltage spike. That is, if the first and second power amplifiers are controlled according to a nested operating mode with a switching frequency close to the resonant frequency of the primary resonant circuit, a voltage spike can be generated by the residual energy stored in the primary side of the resonant circuit when each power amplifier is switched off. Additionally, since the first rectifier, the second rectifier, and the transformer (which provides a secondary leakage inductance) form a secondary resonant circuit, there is the potential to excite a resonant voltage on the secondary side of the transformer, resulting in a second voltage spike. In other words, if the first power amplifier and the second power amplifier are controlled according to a nested operating mode with a switching frequency close to the resonant frequency of the secondary resonant circuit, another voltage spike can be generated on the secondary side. Naturally, if the resonant frequencies of the primary and secondary resonant circuits are similar, a voltage boost can be generated simultaneously on both sides. These voltage boosts can increase the voltage provided at the output without requiring an increase in the transformer's turns ratio (and the resulting increase in the space required for such turns). Accordingly, a power converter circuit is provided that can facilitate a voltage boost while requiring only a relatively small footprint. Furthermore, due to the interleaved operation of the first and second power amplifiers, electromagnetic emissions (EM) at a distance from the device (e.g., a few centimeters from the transformer) can be reduced because the two magnetic fluxes imparted to the transformer windings are canceled out, provided that a planar transformer with counter-wound windings is used. Additionally, placing the power amplifiers and rectifiers at opposite ends of the primary and secondary windings, respectively, makes voltage amplification on the primary and secondary sides more effective. Finally, the voltage load on the rectifier devices of the secondary side is halved compared to using a single full-bridge rectifier. It should be noted that the proposed power converter circuit operates efficiently even if the coupling of the transformer windings is not particularly strong. In fact, the topology can utilize the leakage inductance of the primary and secondary windings to form the primary and secondary resonant circuits, which in turn can provide a voltage boost to overcome the efficiency loss due to poor magnetic coupling. Of course, poor magnetic coupling is not critical for the power converter circuit. However, if manufacturing constraints do not permit good magnetic coupling (e.g., due to space limitations, inefficient dimensioning, or material costs), the proposed power converter aims to mitigate the efficiency loss. In specific embodiments, the first power amplifier may include a first switch connected between the first end of the primary winding and a first reference potential. The second power amplifier may include a second switch connected between the second end of the primary winding and a second reference potential. Then, either a primary capacitor may be provided between the first and second ends of the primary winding, or a first capacitor may be provided between the first end of the primary winding and the first reference potential, and a second capacitor may be provided between the second end of the primary winding and the second reference potential. To be clear, the first reference potential and the second reference potential can be the same potential (e.g., both can be mass, or both can be a positive or negative reference potential). Furthermore, the first rectifier can comprise a first rectifier device connected between the first end of the secondary winding and a third reference potential, and a second rectifier device connected between the first end of the secondary winding and a load. The second rectifier can comprise a third rectifier device connected between the second end of the secondary winding and a fourth reference potential, and a fourth rectifier device connected between the second end of the secondary winding and the load. The third reference potential and the second reference potential can be the same potential (e.g., both can be ground). The rectifier devices can be passive devices, such as diodes, or active devices, such as switches. In either case, the rectifier devices work together to rectify the current on the secondary side, so that a direct current can be provided at the output load. In some embodiments, either a secondary capacitor can be provided between the first end of the secondary winding and the second end of the secondary winding, or a third capacitor can be provided between the first end of the secondary winding and the third reference potential, and a fourth capacitor can be provided between the second end of the secondary winding and the fourth reference potential. The above specific implementation of the circuit provides a power converter that can be small in size and relatively simple in implementation, while offering several advantages over state-of-the-art power converters. In fact, increased efficiency, increased voltage output, small size, low cost, and reduced voltage stress can all be achieved with this embodiment. In some embodiments, the circuit may further include a gate driver circuit configured to control the first power amplifier and the second power amplifier in a nested operating mode and synchronously with a switching frequency. In particular, the switching frequency can be based on a resonant frequency of the primary resonant circuit. Additionally or alternatively, the switching frequency can be based on a resonant frequency of the secondary resonant circuit. Thus, the switching frequency can be selected such that at least one of the resonant circuits is excited by the switching, and therefore provides a voltage increase. The switching frequency may not be exactly equal to one of the resonant frequencies. In fact, the switching frequency may be less than or equal to the resonant frequency of the primary resonant circuit. This prevents the accidental excitation of higher-frequency resonances of the primary resonant circuit. In other words, a small time delay can be imposed on the self-resonant nature of the system to prevent the higher-frequency circuit from being excited, ultimately improving circuit efficiency. In some cases, the resonant frequency of the primary and / or secondary resonant circuits can be configured based on the switching frequency. That is, the capacitances provided at or through the power amplifiers and rectifiers, and / or the inductances provided by the transformer, can be selected to provide a resonant frequency at or near a desired switching frequency. Naturally, the interrelationship between the switching frequency, the resonant frequencies of the circuit, and their effect on circuit operation is fully understood, and the resulting circuit and its operation can be adjusted based on the intended application. In further embodiments, the gate driver circuit can be configured to detect a voltage spike at the first end of the primary winding and to switch on the second power amplifier in response to the detection of the voltage spike. Likewise, the gate driver can be configured to detect a voltage spike at the second end of the primary winding and to switch on the first power amplifier in response to the detection of the voltage spike. To be clear, the voltage spike occurs due to the resonant response of the primary resonant circuit on the first or second side of the primary winding when either the first or the second power amplifier is switched off. In other words, the voltage spike is a temporary increase in the magnitude of the voltage on the first or second side of the primary winding. Capturing the peak voltage can be achieved using standard logic. Additionally, the gate driver circuit can be configured to switch off the second power amplifier once a predetermined time has elapsed since the second power amplifier was switched on. Likewise, the gate driver can be configured to switch off the first power amplifier once a predetermined time has elapsed since the first power amplifier was switched on. The predetermined time can be based on the switching frequency (e.g., be inversely related to it). In this interleaved operation, whenever one power switch is switched off, the opposite one is switched on. In some cases, a large amount of the energy required for switching on can be provided by the resonant voltage spike determined at the high-voltage terminal of the first switch. Essentially, this provides an operating mode that synchronizes the switching of the power amplifiers. When a voltage spike occurs at the respective end of the primary winding, it indicates that the corresponding power amplifier has been switched off (since this results in a voltage spike at that end due to the resonance effect of the primary resonant circuit). This information can then be used to synchronize the switching on of the other power amplifier. Furthermore, the gate driver circuit can be configured to switch on the second power amplifier using the voltage at the first end of the primary winding. Likewise, the gate driver circuit can be configured to switch on the first power amplifier using the voltage at the second end of the primary winding. As a result, the peak voltage present at one end of the primary winding can be recovered and used to switch on the power amplifier connected to the opposite end of the primary winding. As a result, driver losses can be reduced. That is, driver losses can be reduced by using self-oscillating drivers, which can also recover power to supply the switch-on power for the power amplifiers driving the primary winding. In some embodiments, the first power amplifier may comprise a first switch connected between the first end of the primary winding and a first reference potential, and a first auxiliary switch connected between the first end of the primary winding and the first reference potential. Likewise, the second power amplifier may comprise a second switch connected between the second end of the primary winding and a second reference potential, and a second auxiliary switch connected between the second end of the primary winding and the second reference potential. In other words, the power amplifiers can include more than one switch. The first / second switch and the first / second auxiliary switch can be controlled to be turned on at different times. In particular, the gate driver circuit can be configured to detect one end of a voltage spike at the second end of the primary winding, to turn on the second auxiliary switch in response to the detection of the end of the voltage spike at the second end of the primary winding, and to turn on the second switch in response to the detection of the voltage spike at the first end of the primary winding. Similarly, the gate driver circuit can be configured to detect one end of a voltage spike at the first end of the primary winding, to turn on the first auxiliary switch in response to the detection of the end of the voltage spike at the first end of the primary winding, and to turn on the first switch in response to the detection of the voltage spike at the second end of the primary winding. Accordingly, the first auxiliary switch can be turned on / enabled in response to the detection of the end of the voltage spike (i.e., the end of the resonant voltage rise) at the first end of the primary winding. The second auxiliary switch can be turned on in response to the detection of the end of the voltage spike at the second end of the primary winding. For example, when the voltage spike at one end of the primary winding drops back to 0 V, the respective auxiliary switch is turned on. As soon as a voltage spike is then detected at the second end of the primary winding, the first switch can be turned on. Likewise, as soon as a voltage spike is detected at the first end of the primary winding, the second switch can be turned on. Furthermore, the gate driver circuit can be configured to turn off the second switch and the second auxiliary switch once a predetermined time has elapsed since the second switch was turned on. Similarly, the gate driver circuit can also be configured to turn off the first switch and the first auxiliary switch once a predetermined time has elapsed since the first switch was turned on. As above, the predetermined time is based on the switching frequency. Therefore, the first auxiliary switch turns on before the first switch, but they turn off simultaneously, and the second auxiliary switch turns on before the second switch, but they turn off simultaneously.The simultaneous switching off of a switch and its auxiliary switch, both connected to the respective ends of the same transformer winding, determines the start of the voltage spike at the end of that transformer winding, thereby setting the switching frequency. As a result, the transferred power can be increased if the auxiliary switch is activated for a larger part of the switching cycle. This can lead to improved conversion efficiency. In yet another embodiment, the first power amplifier can also include a first boost switch connected between the first end of the primary winding and the first reference potential, and the second power amplifier can also include a second boost switch connected between the second end of the primary winding and the second reference potential. In this case, the gate driver circuit can further be configured to turn on the first boost switch in response to the detection of the voltage spike at the second end of the primary winding, to detect one end of the voltage spike at the second end of the primary winding, and to turn off the first boost switch in response to the detection of the end of the voltage spike at the second end of the primary winding. Similarly, the gate driver circuit can also be configured to turn on the second boost switch in response to the detection of the voltage spike at the first end of the primary winding, to detect one end of the voltage spike at the first end of the primary winding, and to turn off the second boost switch in response to the detection of the end of the voltage spike at the first end of the primary winding. In other words, both the first and second power amplifiers can include a third switch. The boost switch can be synchronized to turn on when the voltage spike occurs at the opposite end of the primary winding. Specifically, the first boost switch can turn on when the voltage spike is detected on the second side of the primary winding (i.e., when the first switch turns on) and can turn off when the voltage spike on the second side of the primary winding ends. The second boost switch can turn on when the voltage spike on the first side of the primary winding is detected (i.e., when the second switch turns on) and can turn off when the voltage spike on the first side of the primary winding ends. As a result, it can be ensured that the power converter is not synchronized with disruptive higher frequencies, while at the same time an increase in power transmission and efficiency is achieved. In some embodiments, the first auxiliary switch can be much smaller than the first switch and the first boost switch. That is, the first auxiliary switch can have a much lower current capacity and / or a much lower on-resistance than the first switch and the first boost switch. Likewise, the second auxiliary switch can be much smaller than the second switch and the second boost switch. That is, the second auxiliary switch can have a much lower current capacity and / or a much lower on-resistance than the second switch and the second boost switch. This can ensure that higher resonant frequencies of the first resonant circuit are not excited by the first auxiliary switch and / or the second auxiliary switch being turned on for a larger portion of the switching cycle. In some embodiments, the transformer includes a core containing a material designed to provide galvanic isolation between the primary and secondary windings. In some cases, the material has a relative magnetic permeability of less than 10. For example, the transformer may have an air core or a core with a magnetic permeability close to that of air. Such a core may provide poor magnetic coupling between the primary and secondary windings. However, as described above, the proposed power converter can utilize this poor magnetic coupling to provide inductance for the resonant circuit on both the primary and secondary sides. Alternatively, the transformer can include a core containing a magnetic material. For example, the transformer can include one or more cores made of materials that provide a magnetic permeability much greater than that of air. This would improve the magnetic coupling between the primary and secondary windings. In some embodiments, the circuit may further include a tuning inductor provided in series with the secondary winding. In this case, the tuning inductor may form part of the secondary resonant circuit. This means the tuning inductor can provide a certain amount of inductance as part of the secondary resonant circuit. This can be desirable if the leakage inductance on the secondary side is particularly low (e.g., due to good magnetic coupling between the windings). Alternatively or additionally, it can be used to tune the resonant frequency of the secondary resonant circuit to a desired value. In some embodiments, the number of turns in the primary winding is the same as the number of turns in the secondary winding. That is, the turns ratio of the transformer can be 1:1. Accordingly, no voltage increase is provided due to the turns ratio, but can be provided by resonances as described above. Of course, if desired, the number of turns in the primary winding can be greater or less than the number of turns in the secondary winding. According to other aspects of one embodiment, a system is provided comprising a first load terminal for connection to a power source, a second load terminal for connection to a load, and an isolated power converter circuit according to one of the disclosed embodiments. In this case, the first load terminal is connected to the source of the isolated power converter circuit, and the second load terminal is connected to the first and second rectifiers of the isolated power converter circuit. For example, the second load terminal can be connected to the gate of a circuit breaker that requires galvanic isolation from the power source (e.g., for safety). To briefly explain, current isolated power converters require large isolation transformers to provide an isolated supply for the secondary-side power domain. To reduce both the space and cost required for this component, high-frequency switching topologies utilizing low-inductance air-core magnet structures are frequently employed in isolated power conversion. These can be commonly used for gate driver supplies. However, the use of air-core transformers results in a reduced coupling factor, which typically makes it more difficult to efficiently transfer power from the primary to the secondary side. Furthermore, operating the switches at high frequencies increases driver losses. Therefore, it was recognized that there is a need for an isolated power converter that has minimal power losses (to achieve high conversion efficiency), provides voltage boosting (if required), and operates via a transformer-based galvanic isolation barrier, incorporating a small transformer with a near 1:1 turns ratio and potentially poor magnetic coupling between the primary and secondary windings. Therefore, disclosed embodiments provide a power conversion topology that can utilize non-ideal coupling effects of the isolation transformer to create a resonant converter. This can help reduce power losses while requiring little space and not complex manufacturing processes. To be clear, the dimensions of the isolation transformer may have a minimum required size depending on the power conversion needs, since the magnetic flux and the energy that can be coupled from the primary to the secondary winding of the transformer also scale with the winding area. Therefore, it may be the case that the transformer needs to be too large (to provide the required power conversion) to be easily and cost-effectively integrated onto a silicon chip. Furthermore, silicon circuit integration processes typically impose limits on the maximum available thickness and width of the metal layers on the silicon chip (which are fabricated as part of the chip die). This, in turn, limits the resistivity of the integrated transformer metal windings, directly impacting the power conversion efficiency. For these reasons, it is generally more convenient for the isolation transformer to deliver relatively high power for a given integrated circuit (IC) size not to be implemented through silicon chip integration, but rather by other means, such as a printed circuit board (PCB) laminate. This PCB laminate is then embedded, in some way, in a standardized IC package. Conductive metallization wiring, insulating layers, and interlayer interconnect options are selected on the PCB laminate based on trade-offs regarding geometric dimensions, resistivity, power conversion efficiency, and cost. Nevertheless, even with trade-offs in power conversion efficiency, delivered power, and voltage gain, such PCB laminate solutions are much cheaper than solutions fully integrated on a silicon chip.The disclosed isolated power converter can be particularly well implemented using such PCB laminate solutions. As discussed, implementing the converter using PCB laminate solutions can provide further advantages, especially if the isolation transformer is coreless or includes a core without a significant impact on the absolute magnetic permeability (e.g., an air core). However, it should still be noted that the proposed isolated power converter can also be implemented by other means. Embodiments of the proposed power converter include at least some of the following aspects: (i) An isolation transformer with a primary winding having a center tap connected to an input / load. For example, the input may be connected to a center-tapped input winding; (ii) Two power amplifiers, each connected to opposite ends of the isolation transformer's primary winding and time-interlocked on the primary side, are used for efficiency-targeted resonant power driving of the primary winding and possible voltage boosting.This can be a dual-operated / differential / time-nested Class E power amplifier; (iii) Two rectifiers on the secondary side, each connected to opposite ends of the secondary winding of the isolation transformer, are used for efficiency-targeted voltage resonant rectification of the secondary winding and possible voltage boosting. These can be dual-operated / time-nested Class E rectifiers; (iv) A driver circuit designed to facilitate gate charge recovery for switching the power amplifiers driving the input windings. This also allows for power conversion matching in terms of deliverable output power, output voltage, and efficiency; (v) Power amplifiers that are segmented so that multiple switches with slightly different switching regimes are used.This can enable further optimization of the power conversion and efficiency of the power amplifiers. A specific aspect of the power converter is the resonant circuits implemented on the primary and secondary sides. Specifically, a resonant circuit on the primary side is formed by capacitances provided by (or across) the two power amplifiers and an inductance provided by the transformer (e.g., the magnetizing inductance and / or leakage inductance of the primary winding). Another resonant circuit is formed on the secondary side by capacitances provided by (or across) the two rectifiers and an inductance provided by the transformer (e.g., the magnetizing inductance and / or leakage inductance of the secondary winding). These resonant circuits can be used to increase the input output voltage and improve conversion efficiency. On the primary side, one embodiment implements a quasi-self-oscillating topology using two class E power amplifiers controlled according to a nested operating mode. Switch-gate charge recovery can be applied in the power amplifiers, thus minimizing driver losses. Additionally, resonance is induced between the magnetizing inductance of the transformer and the parasitic capacitance of the switch(es) of each power amplifier. This can provide an initial voltage boost on the primary side. On the secondary side, in one embodiment using two Class E rectifiers, the resonance between the leakage inductance on the secondary side (e.g., due to imperfect coupling between the primary and secondary transformer windings) and the capacitance provided at or by the rectifying components can be exploited. This can lead to an additional voltage boost, effectively achieving step-up conversion when required. It should be noted that this mechanism can exploit what is normally considered a weakness of transformers that provide low coupling between input and output windings. Air-core transformers, or transformers that do not rely on insulating materials that provide a significant increase in magnetic permeability compared to air, are typically used to reduce both the space and cost required for this component, which can operate at high frequencies. However, the use of such transformers results in a reduced coupling factor, which typically decreases the efficiency of power transfer from the primary to the secondary side in commonly used power conversion topologies.This means that air-coupled transformers can exhibit a relatively large uncoupled or lost magnetic flux between transformer windings. This magnetic flux loss can be modeled by an additional inductance (i.e., a leakage inductance) proportional to the magnitude of the lost magnetic field, placed in series with the transformer's secondary windings. This inductance (in addition to the capacitance provided on the secondary side by or at the rectifiers) can be used to create a further resonance. Thus, a second voltage rise can be obtained via a mechanism normally considered a disadvantage (i.e., via the lost magnetic flux). However, it is understood that a core providing low coupling is not a necessary part of a power converter. A power converter incorporating a transformer with good magnetic coupling (e.g., featuring magnetic cores, etc.) can be advantageous depending on the circuit parameter selection and product requirements. The inductance on the secondary side can be increased, if necessary, by using an additional inductor connected in series with the secondary winding. It should also be noted that poor magnetic coupling can arise from other transformer parameters, such as its shape, size, and the relative positioning of the primary and secondary windings. Furthermore, the use of two rectifiers allows the components of each individual rectifier (e.g., silicon or Schottky diodes) to withstand only half the voltage load that a single rectifier would have, since the voltage load can be shared and carried jointly by the rectifiers connected to the first and second ends of the secondary winding. While this results in an additional voltage drop across the rectifiers, it is generally negligible compared to the rectified output voltage. It is worth reiterating that a voltage increase can be achieved if required while maintaining a 1:1 turns ratio between the primary and secondary windings. In fact, depending on the parameters governing the primary-side resonant conversion and the secondary-side resonant rectification, conversion factors equal to or higher / lower than 1:1 can be obtained without requiring any variation in the turns ratio. To understand how the invention works, it is best to describe the operation of a prior art power converter 1, as shown in Fig. 1. The power converter 1 comprises a transformer 10, a power amplifier 20, and a rectifier 30. The transformer 10 forms a galvanic isolation barrier between the primary and secondary sides, and thus between the connected power source and load. The transformer 10 is shown according to its equivalent circuit. That is, the primary winding 12 is connected in parallel with a magnetizing inductance, Lm, and in series with a primary-side leakage inductance, Lk,pr. The secondary winding 14 is connected in series with a secondary-side leakage inductance, Lk,sec. One end of the primary winding 12 is connected to the power source, and the other end of the primary winding 12 is connected to the power amplifier 20. One end of the secondary winding 14 is connected to the load, and the other end of the secondary winding 14 is connected to the rectifier 30. The transformer 10 reflects the voltage across the primary winding 12 to the secondary winding 14.The voltage across the secondary winding 14 can depend, among other factors, on the turns ratio between the primary winding 12 and the secondary winding 14. For example, this ratio can be 1:1. The power amplifier 20 is designed to increase the voltage on the primary side of the transformer 10. For this purpose, the power amplifier 20 may include one or more switching elements. For example, the power amplifier 20 may be a Class E power amplifier to provide highly efficient RF power amplification by optimizing switching transitions (ZVS) and reducing power losses. A Class E power amplifier includes switches that operate at radio frequencies and under zero-current or zero-voltage switching schemes. Any known power amplifier can be used in the circuit if it provides a capacitor, or a capacitor is provided alongside the power amplifier 20. Rectifier 30 is designed to convert the alternating current at the secondary winding 14 into direct current. Rectifier 30 can be implemented with passive elements, active elements, or a combination thereof. Rectifier 30 can be a Class E rectifier to ensure efficient AC-DC conversion with minimal switching noise and reduced peak currents, resulting in a corresponding reduction in RMS power loss. Any known rectifier can be used in the circuit if it provides capacitance, or a capacitor can be placed alongside rectifier 30. An implementation of the circuit of Fig. 1 is shown in Fig. 2. In particular, a specific power amplifier 20 and rectifier 30 are provided. The power amplifier comprises a switch 22 and a capacitor 24 connected in parallel between the second end of the primary winding 12 and a reference potential (e.g., ground). The rectifier comprises a rectifier device 32 (e.g., a diode) and a capacitor 34 connected between the second end of the secondary winding 14 and a reference potential (e.g., ground), the rectifier device 32 blocking current flowing from the second end of the secondary winding 14 to the reference potential. Combining these power converter and rectifier topologies results in a voltage boost architecture that maximizes efficiency, reduces power losses, and minimizes the number of turns required on the secondary winding 14. The circuit of Fig. 2 operates as follows: (i) The gate of switch 22 turns on, and the inductors, Lk, pr and Lm, begin to charge; (ii) The gate of switch 22 turns off. Current flows into capacitor 24 when the switch is released, and resonance begins between the inductors Lk, pr and Lm and capacitor 24, during which the inductors, Lk, pr and Lm, discharge into capacitor 24; (iii) The voltage is reflected to the secondary winding 14, and the alternating voltage causes resonance between inductor, Lk, sec and capacitor 34; and (iv) The rectifier device 32 equalizes the resulting current. Accordingly, instead of a simple voltage source at the rectifier input, a resonant system is established through resonance between the total inductance present at the secondary winding 14 (thus utilizing the leakage inductance of the transformer 10 as a result of the uncoupled magnetic flux) and a capacitance provided at or by the rectifier. In other words, a resonant circuit is formed on the primary side by the leakage inductance and the capacitor 24. This resonant operation on the secondary side results in a further voltage increase, which is added to that obtained by the resonant mode on the primary side. To be clear, the further voltage increase across the isolation transformer 10 does not require an increase in the turns ratio of the transformer 10 winding, but instead utilizes the resonance generated on the secondary side. Therefore, the system operates in a multiple resonance mode, characterized by two different resonant circuits on the primary and secondary sides, which are crucial for achieving the voltage increase. In fact, the primary side can naturally operate in a quasi-self-resonant mode, allowing zero-voltage switching on the primary side due to the resonance generated by the inductance and capacitance on the primary side. The quasi-self-resonance mode functions as follows. By adjusting the on-time of switch 22 and extending this on-time beyond half the period of the natural oscillation of the primary-side resonant circuit (formed by the magnetizing inductance of transformer 10, the leakage inductance of the primary winding 12, and capacitor 22), it is possible to increase the voltage peak across the resonant capacitor when the switch is turned off, thus increasing the amplitude of the voltage harmonic component transmitted to the secondary side. In this case, the switching period of the primary side depends on the on-time, which becomes longer than that produced by the natural oscillation period of the primary resonant circuit. If this modified switching frequency is set to coincide with the resonant frequency of the secondary resonant circuit (formed by the leakage inductance on secondary winding 14 and the capacitor), a further increase in efficiency can be obtained. However, this increase in voltage does not necessarily lead to an increase in efficiency. Depending on the objectives, different trade-offs may be found between delivered output voltage, converted output power, or power conversion efficiency, which may also depend on the time delay added to the natural resonant period. Adding a time delay to the self-resonant switching period to determine an extended turn-on time for the primary winding 12 circuit breaker 22 introduces a further advantage. In fact, the resonant frequency of a system using a transformer as part of the resonant circuit cannot be uniquely determined by the reactive components connected only to the primary winding 12. A second, higher-frequency resonant peak may occur (or a peak split may occur), also depending on the equivalent impedance apparently connected to the secondary winding 14. This secondary impedance can also vary depending on the operating conditions on the secondary side. This higher-frequency resonant operation would be detrimental to conversion efficiency and power output and should be avoided.Without countermeasures, and depending on the operating conditions, the converter can begin to randomly jump between the two split resonant frequencies, introducing unpredictable converter behavior. Introducing a suitable time delay, added to the higher (lower frequency) resonant period, prevents the higher-frequency resonance from being excited, as the system then necessarily always operates at a frequency lower than the lower-split resonance. In this sense, this control scheme is quasi-resonant, in that the long-period, lower-frequency natural resonance is not perfectly achieved, but a slightly lower switching frequency is enforced by the added time delay without noticeable power or efficiency losses. In fact, depending on the adjustment of the added time delay, efficiency can even be improved. Fig. 3 shows a generalized circuit diagram of an isolated power converter 100 according to one embodiment. The circuit of Fig. 3 is similar to the circuit of Fig. 1. However, the circuit further includes a second power amplifier 20b and a second rectifier 30b, and the input is connected to an intermediate tap of the primary winding 12 of the transformer 10. In particular, the circuit comprises a first power amplifier 20a connected to a first end of the primary winding 12 of the transformer 10, and a second power amplifier 20b connected to a second end of the primary winding 12 of the transformer 10. The first end of the second power amplifier is opposite the second end of the primary winding 12. The circuit also comprises a first rectifier 30a connected to a first end of the secondary winding 14 of the transformer 10, and a second rectifier 30b connected to a second end of the secondary winding 14 of the transformer 10, each rectifier being connected to an output node. The input node is connected to an intermediate tap of the primary winding 12. For example, the input node could be connected to a center tap of the primary winding 12. A capacitance provided at or through the first power amplifier 20a, a capacitance provided at or through the second power amplifier 20b, and an inductance provided by the primary winding 12 of the transformer 10 form a first resonant circuit. That is, a first resonant circuit is provided on the primary side of the circuit by the transformer 10 and the power amplifiers. Similarly, a capacitance provided at or through the first rectifier 30a, a capacitance provided at or through the second rectifier 30b, and an inductance provided by the secondary winding 14 of the transformer 10 form a second resonant circuit. In other words, a second resonant circuit is provided on the secondary side of the circuit by the transformer 10 and the power rectifiers. For explanation, the operating principle is the same as described with reference to Figures 1 and 2. However, as shown, the architecture is duplicated. The power amplifiers can therefore be driven with a 180° phase shift. That is, the first and second power amplifiers 20a and 20b can be controlled according to a nested operating mode, whereby when one power amplifier is switched on, the other is switched off. This can be done at a predetermined switching frequency. In some embodiments, this predetermined switching frequency can be slightly lower than the resonant frequency of the first resonant circuit. As described above, this will result in a voltage increase on the primary side and will also avoid exciting the potential second (higher) resonant frequency of the resonant circuit. Furthermore, the predetermined switching frequency can additionally or alternatively be substantially equal to the resonant frequency of the second resonant circuit. As described above, this can result in a voltage increase on the secondary side. Of course, the first and second resonant circuits can be tuned to have resonant frequencies based on a desired switching frequency (e.g.,by changing the capacitance provided by the power amplifiers and rectifiers, and / or by changing the inductance provided by the transformer 10). This means that the sum of the voltages across the secondary winding 14 excites the second resonant circuit, consisting of the leakage inductance of the transformer 10 and the capacitances provided at or by the rectifiers (e.g., parasitic capacitances in parallel with the diodes of a rectifier). Thanks to the self-resonance of the secondary resonant circuit, an additional voltage boost can be obtained without relying on a turns ratio of the transformer 10, which would require more surface area / volume due to the necessary spacing between conductor tracks for PCB fabrication. It should also be noted that the interleaved operation on the primary side allows for a further improvement in the achievable conversion efficiency. As stated above, the second power amplifier 20b is switched off while the first power amplifier 20a is switched on. The voltage at the second end of the primary winding 12 can therefore reach a peak when this switching occurs, due to the quasi-resonant operation described above. This provides the possibility of using this strong voltage spike to synchronize the switch-on of the first power amplifier 20a with the switch-off of the second power amplifier 20b and to recover some of the voltage spike energy to drive the switch-on of the first power amplifier 20a. This solution would thus increase the architectural efficiency. When the second power amplifier 20b is switched off, a voltage spike begins at the second end of the primary winding 12 due to the interrupted inductive current. The start of the spike can be detected and used to synchronously determine the interleaved switch-on of the first power amplifier 20a. As the voltage spike at the second end of the primary winding 12 decreases, the first power amplifier 20a cannot, of course, lose its power supply (which would lead to an undesired switch-off).For this reason, a diode (or another type of controlled switch) is used to prevent this. While the voltage at the second end of the primary winding 12 is low, a main power supply can ensure that the first power amplifier 20a remains switched on until a programmable switch-off command is received. Of course, additionally or alternatively, a similar control of the second power amplifier 20b can be achieved using a voltage at the first end of the primary winding. This circuit offers several advantages over the state-of-the-art converter. First, driver losses can be reduced by using self-oscillating drivers, which can also recover power to supply the inrush power for the power amplifiers driving the primary winding 12. Second, due to the interleaved operation, "distant" electromagnetic emissions are reduced because of the cancellation of the two magnetic fluxes, assuming that the transformer windings 10 are wound in opposite directions. "Far away" means that the EM emissions are measured at a distance much greater than the winding dimensions of transformer 10 (e.g., a few centimeters away from transformer 10). Additionally, the dual design of the power amplifiers and rectifiers achieves the voltage increase on both the primary and secondary sides more effectively. Finally, the voltage load on the rectifier components of the rectifiers on the secondary side is halved and thus reduced compared to the prior art. Fig. 4 shows an implementation of the circuit from Fig. 3. As shown, the first power amplifier 20a includes a first switch 22a connected between the first end of the primary winding 12 and a first reference potential. The second power amplifier 20b includes a second switch 22b connected between the second end of the primary winding 12 and a second reference potential. A first capacitor 24a is provided between the first end of the primary winding 12 and the first reference potential, and a second capacitor 24b is provided between the second end of the primary winding 12 and the second reference potential. However, it is equally possible that the first 24a and the second capacitor 24b are replaced by a single primary capacitor provided between the first and second ends of the primary winding 12. Additionally, the first and second reference potentials can be the same or different. Furthermore, as shown, the first rectifier 30a comprises a first rectifier device 32a connected between the first end of the secondary winding 14 and a third reference potential, and a second rectifier device 36a connected between the first end of the secondary winding 14 and a load. The second rectifier 30b comprises a third rectifier device 32b connected between the second end of the secondary winding 14 and a fourth reference potential, and a fourth rectifier device 36b connected between the second end of the secondary winding 14 and the load. The rectifier devices shown are diodes, but each can be replaced by a switch or another rectifier device. A third capacitor 34a is provided between the first end of the secondary winding 14 and the third reference potential, and a fourth capacitor 34b is provided between the second end of the secondary winding 14 and the fourth reference potential. However, it is equally possible that the third capacitor 34a and the fourth capacitor 34b are replaced by a single secondary capacitor provided between the first and second ends of the secondary winding 14. Additionally, the third and fourth reference potentials can be the same or different. In some embodiments, the first through fourth reference potentials can all be ground potentials, and in particular, they can all be the same ground potential. Additionally, the circuit can optionally include a tuning inductor 40, which is provided in series with the secondary winding 14. Although it is shown connected to the first side of the secondary winding 14, the tuning inductor 40 can equally be provided connected to the second side of the secondary winding 14. The tuning inductor 40 forms part of the secondary resonant circuit. The tuning inductor 40 can be used when it is desired to change the resonant frequency of the secondary resonant circuit. This can be advantageous, for example, when the leakage inductance of the secondary winding 14 is very low. In this case, switching the first power amplifier 20a on and off includes switching the first switch 22a on and off, and switching the second power amplifier 20b on and off includes switching the second switch 22b on and off. As shown in Fig. 5, this control of the switches can be carried out by a gate driver circuit 50 which is configured to drive the switches of the power amplifiers in a nested operating mode. The gate driver circuit 50 is configured to control the first power amplifier 20a and the second power amplifier 20b in a nested operating mode and synchronously with a switching frequency. That is, the gate driver circuit 50 switches the first switch 22a on when the second switch 22b is off, and switches the second switch 22b on when the first switch 22a is off. This is done based on a switching frequency. The switching frequency, at which the interleaved operating mode is synchronized, can be based on a resonant frequency of the primary resonant circuit. This allows a resonance to be excited on the primary side of the circuit, resulting in a voltage increase. In particular, the switching frequency can be (only slightly) lower than the resonant frequency of the primary resonant circuit. This can prevent the excitation of other, higher-frequency resonant peaks of the primary resonant circuit. Furthermore, in this case, the resonant frequency of the secondary resonant circuit can be set based on the switching frequency. That is, the capacitance provided by the rectifiers and / or the inductance provided by the transformer 10 (and the inductance provided by the optional tuning inductor) can be selected such that the resonant frequency of the secondary resonant circuit is similar to the switching frequency. This can induce a further voltage increase on the secondary side by exciting the secondary resonant circuit. Alternatively, in some cases, the switching frequency to which the interleaved operating mode is synchronized can be based on the resonant frequency of the secondary resonant circuit. This allows a resonance to be excited on the secondary side of the circuit, resulting in a voltage increase—but it prevents a resonance from being excited on the primary side if the resonant frequency of the primary resonant circuit differs from the switching frequency. It is understood that there is an interdependence between the resonant frequencies of the resonant circuits and the switching frequency—each of which can be tailored and set based on the specific application of the circuit. In some embodiments, the resonant peak voltages present at the first end of the primary winding 12 and the second end of the primary winding 12 (due to resonance when the first switch 22a and the second switch 22b respectively turn off) can be used to synchronize the turning on of the opposite switch and possibly to help control the turning on. For this purpose, the gate driver circuit 50 can be configured to detect a voltage spike at the first end of the primary winding 12. Then, when the voltage spike is detected, the gate driver circuit 50 can switch on the second power amplifier 20b (e.g., switch on the second switch 22b). Additionally or alternatively, the gate driver circuit 50 can be configured to detect a voltage spike at the second end of the primary winding 12. Then, when the voltage spike is detected, the gate driver circuit 50 can switch on the first power amplifier 20a (e.g., switch on the first switch 22a). To be clear, the voltage spike can be detected by the gate driver circuit 50 using known logic. Of course, adding any logic components to the signal path introduces a delay, and therefore, in practice, the detection of the voltage spike may not be instantaneous. Nevertheless, it can be ensured that the delay introduced by such detection is consistent with the system operating frequency. That is, the delay introduced by the detection logic can be negligible relative to the switching frequency of the power amplifiers 20a, 20b. Furthermore, in some cases a small delay between the detection of the voltage spike and the switching on of the first / second power amplifier 20a, 20b may be desirable to avoid exciting further resonant frequencies of the resonant circuit. This delay can therefore be implemented by a programmable element. The delay introduced by the detection logic may be negligible compared to the delay introduced by the programmable element, or it may be compensated for by the programmable element. The gate driver circuit 50 can switch off the second power amplifier 20b as soon as a predetermined time has elapsed since the second power amplifier 20b was switched on, and / or can switch off the first power amplifier 20a as soon as a predetermined time has elapsed since the first power amplifier 20a was switched on. This predetermined time can be based on the switching frequency. That is, the predetermined time can be set to ensure that switching occurs at the switching frequency. Accordingly, this predetermined time can be pre-programmed based on a desired switching frequency. The gate driver circuit 50 can also be configured to switch on the second power amplifier 20b using the voltage at the first end of the primary winding 12. That is, the voltage at the drain of the first switch 22a can be used to drive (at least initially) the gate of the second switch 22b to switch on the second switch 22b. Likewise, the gate driver circuit 50 can also be configured to switch on the first power amplifier 20a using the voltage at the second end of the primary winding 12. That is, the voltage at the drain of the second switch 22b can be used to drive (at least initially) the gate of the first switch 22a to switch on the first switch 22a. This therefore enables self-synchronous operation of the power amplifiers. This self-oscillation can be achieved using the circuit shown in Fig. 6. This shows one implementation of the gate driver circuit integrated with the power converter. Of course, other implementations are possible, and any modifications are readily apparent to a person skilled in the art. As shown, the SR latch defines the on state of the gate driver switches based on voltages at the first and second ends of the primary winding 12. When the voltage on the first side of the primary winding 12 reaches a peak but then approaches 0 V, a pulse is generated to trigger the SR latch. The off command is fed back to the driver after a digitally or analogously programmable delay. The driver turns off the first switch in a dissipative manner and uses the connection to the voltage at the first end of the primary winding 12 via the diode to charge the gate of the second switch with energy recovery. The operation of the second switch is performed similarly. Fig. 7 shows various waveforms while the circuit shown in Fig. 6 is operating. The waveforms are provided under the assumption that the switching frequency of the first and second power amplifiers 20a, 20b coincides with the resonant frequency of the secondary resonant circuit. Diagram 210 shows the voltage, VD1, at the first end of primary winding 12 and the voltage, VD2, at the second end of primary winding 12, superimposed with the gate driver voltage, VG1, of the first switch 22a and the gate driver voltage, VD2, of the second switch 22b. Diagram 220 shows the pulses, S and R, provided to the SR latch as a result of the detected voltage spikes at the first and second ends of primary winding 12. Diagram 230 shows the resulting output, Q, of the SR latch. Diagram 240 shows the voltage, Vs, at secondary winding 14. Diagram 250 shows the current, ILr, through the tuning inductor 40, which is nearly sinusoidal. Diagram 260 shows the voltage, VB1, seen at the second rectifier device, and the voltage, VB2, seen at the fourth rectifier device.That is, VB1 is the voltage at the first end of the secondary winding 14, and VB2 is the voltage at the second end of the secondary winding 14. As shown, the current flowing through the tuning inductor is nearly sinusoidal, and the voltage swing across the third and fourth capacitors causes the two rectifier devices to conduct. Under these resonant conditions, the output voltage Vout can exceed the peak-to-peak amplitude of the voltage reflected from the secondary winding 14, thus introducing a further voltage boost if required. Fig. 8 shows another implementation of the circuit in which both power amplifiers include three switches. In contrast to circuits 101-103 of Fig. 4-6, the first switch 22a and the second switch 22b are each segmented into three different switches. That is, the first power amplifier comprises the first switch 22a, the first auxiliary switch 26a, and the first boost switch 28a. The second power amplifier comprises the second switch 22b, the second auxiliary switch 26b, and the second boost switch 28a. As shown, the first switch 22a, the first auxiliary switch 26a, and the first boost switch 28a can each be connected between the first side of the primary winding 12 and a reference potential. Likewise, the second switch 22b, the second auxiliary switch 26b, and the second boost switch 28b can each be connected between the second side of the primary winding 12 and the reference potential. In some cases, the switches may have different current ratings. For example, the first / second switch 22a / 22b may have a first current rating, the first / second auxiliary switch 26a / 26b may have a second current rating, and the first / second boost switch 28a / 28b may have a third current rating. The first and second auxiliary switches 26a, 26b may have the lowest current ratings. This means that the first and second auxiliary switches 26a, 26b may have the smallest active area. The first switch 22a, the second switch 22b, the first boost switch 28a, and the second boost switch 28b may each have the same current rating.In one example, the first auxiliary switch 26a can provide 10% of the total current handling capacity of the first power amplifier, the first switch 22a can provide 45% of the total current handling capacity of the first power amplifier, and the first boost switch 28a can provide 45% of the total handling capacity of the first power amplifier. Similarly, the second auxiliary switch 26b can provide 10% of the total current handling capacity of the second power amplifier, the second switch 22b can provide 45% of the total current handling capacity of the second power amplifier, and the second boost switch 28b can provide 45% of the total current handling capacity of the second power amplifier. However, embodiments are not limited to these, and the switches can have different current-relative current handling capacities compared to this example (e.g., they can all have the same current handling capacity). Each of the switches is controlled by a gate driver circuit 50. Each of the switches can be shifted with respect to its turn-on time to further increase the power output and efficiency of the converter. That is, unlike the previously described operation in which one power amplifier is turned on when the other is turned off, the switch, auxiliary switch, and boost switch of each power amplifier can be turned on and / or off at different times. For example, the switches of the first power amplifier can be controlled as follows: (i) The first auxiliary switch 26a can be turned on in response to the detection of the end of the resonant voltage spike at the first end of the primary winding 12. That is, the first auxiliary switch 26a can be turned on when the resonant voltage spike (which is detected at the same end of the primary winding 12 as the first power amplifier 20a) falls back to near 0 V. The turn-on energy for the first auxiliary switch 26a can be taken from an internal power supply; (ii) The first auxiliary switch 26a can be turned off together with the first switch 22a (i.e., after a predetermined time interval from the time the first switch 22a is turned on); (iii) The first switch 22a is operated as described above.This means that the first switch 22a is turned on when a voltage spike is detected at the second end of the primary winding 12 and is turned off after a delay; (iv) The first boost switch 28a can be operated with synchronized turning on of the first switch 22a. This means that the first boost switch 28a can also turn on when a voltage spike is detected at the second end of the primary winding 12. (v) The first boost switch 28a can be turned off in response to the detection of the end of the resonant voltage spike at the second end of the primary winding 12. This means that the first boost switch 28a can be turned off when the resonant voltage spike (which is detected at the opposite end of the primary winding 12 as the first power amplifier 20a) falls back to near 0 V.Thus, the operation of the boost switch 28a can be completely determined by the rise and fall of the voltage spike on the second side of the primary winding 12. Of course, the second power amplifier 20b, which includes the second switch 22b, the second auxiliary switch 26b and the boost switch 28b, can be controlled in a similar way. This operation can ensure that the circuit does not lock onto higher resonant frequencies, as previously discussed, while increasing the input-to-output transfer power and conversion efficiency. It should also be noted that in some embodiments the boost switches can be omitted. That is, the first power amplifier may include only the first switch and the first auxiliary switch, and / or the second power amplifier may include only the second switch and the second auxiliary switch. This can still provide an improvement in the power delivered and the conversion efficiency compared to power amplifiers that include only the first / second switch. The operation of the power amplifiers of Fig. 8 is demonstrated by the timing diagrams of Fig. 9. In particular, diagram 310 shows the voltage at the first end of the primary winding (i.e., the drain of each of the first switch, the first auxiliary switch, and the first boost switch). Diagram 320 shows the voltage at the second end of the primary winding (i.e., the drain of each of the second switch, the second auxiliary switch, and the second boost switch). As can be seen, the first and second ends experience nested voltage spikes due to the resonant response of the primary resonant circuit. Diagram 330 shows the voltage applied to the gate of the first switch, and diagram 340 shows the voltage applied to the gate of the second switch. The first switch turns on when the start of the voltage spike at the second end of the primary winding is detected, and the second switch turns on when the start of the voltage spike at the first end of the primary winding is detected. The first switch turns off after a predetermined time delay, and the second switch turns off after a predetermined time delay. This turn-off induces a voltage spike at the opposite end of the primary winding from the switch by exciting the primary resonant circuit. Diagram 350 shows the voltage applied to the gate of the first auxiliary switch, and diagram 360 shows the voltage applied to the gate of the second auxiliary switch. The first auxiliary switch turns on when a voltage spike ends at the first end of the primary winding, and the second auxiliary switch turns on when a voltage spike ends at the second end of the primary winding. The first auxiliary switch turns off at the same time as the first switch, and the second auxiliary switch turns off at the same time as the second switch (i.e., it has a synchronized turn-off with the first / second switch). Finally, diagram 370 shows the voltage applied to the gate of the first boost switch, and diagram 380 shows the voltage applied to the gate of the second boost switch. As can be seen, the first boost switch turns on when it detects the beginning of the voltage spike at the second end of the primary winding and turns off when it detects the end of the voltage spike at the second end of the primary winding (i.e., it only turns on if there is a voltage spike at the second end of the primary winding). The second boost switch turns on when it detects the beginning of the voltage spike at the first end of the primary winding and turns off when it detects the end of the voltage spike at the first end of the primary winding (i.e., it only turns on if there is a voltage spike at the first end of the primary winding). Fig. 10 shows a system 300 including one of the isolated power converter circuits 100, 101, 102, 103, 104 described above. System 400 comprises a first load terminal 410 configured for connection to a power source 412. The power source 412 may be provided or be a separate component from System 400. For example, the power source 412 may be any constant voltage power source. The first load terminal 410 is coupled to the source of the isolated power transformer circuit. That is, the first load terminal 410 is electrically coupled to an intermediate tap of the primary winding 12 of the transformer 10. Furthermore, the system includes a second load terminal 420, which is connected to a load 422. The load 422 can be, for example, a gate driver for a power switch, an integrated logic circuit (IC), a sensor device, or a communication device. The second load terminal 420 is coupled to the first and second rectifiers of the isolated power converter circuit. That is, the second load terminal 420 is electrically coupled to an output of both the first and second rectifiers. For example, the load can be a gate driver, and the gate driver and the isolated power converter circuit 100-104 can be arranged in a common package. The common package can provide external connections (e.g., connection 410 to receive power, and other connections to receive and / or provide control signals, such as a driver output or a control signal input, or the like). In another example, the system 400 can implement an integrated power module that integrates, in a common package, the isolated power converter circuit 100-104, at least one isolated gate driver powered by the isolated power converter 100-104, and at least one power transistor switched between states by the isolated gate driver. Fig. 11 shows a flow diagram of a method 500 for transferring power from a first side of an isolation transformer to a second side of the isolation transformer according to one embodiment. In step 510, power is provided for an intermediate tap of a primary winding on the first side of the isolation transformer. This power can be supplied by a single source. The intermediate tap can be a center tap of the primary winding. In step 520, a first resonance is generated in the primary winding using a first power amplifier coupled to a first end of the primary winding and a second power amplifier coupled to a second end of the primary winding. This means that the first and second power amplifiers can be controlled according to a nested operating mode (where one amplifier is switched on, the other is switched off, and vice versa, at a switching frequency). If this switching frequency is close to the resonant frequency of a primary resonant circuit formed by the first power amplifier, the second power amplifier, and the primary winding (e.g., the magnetizing inductance of the transformer and the leakage inductance of the primary winding), then resonance can be generated. In step 530, a current induced in a secondary winding on the second side of the isolation transformer, which is inductively coupled to the primary winding, is rectified using a first rectifier coupled to the first end of the secondary winding and a second rectifier coupled to the second end of the secondary winding. The current is induced due to the oscillating voltage across the primary winding, resulting in a current through the secondary winding. This current is then rectified using two rectifiers provided on opposite sides of the secondary winding. In some cases, the current oscillation can be close to the resonant frequency of a secondary resonant circuit on the secondary side, formed by the first rectifier, the second rectifier, and the leakage inductance of the secondary winding. This can lead to a further increase in the resonant voltage on the secondary side. Finally, in step 540, the rectified current is provided at an output. The output can be connected to a load to be powered. Although specific examples have been illustrated and described herein, the person skilled in the art will recognize that a multitude of alternative and / or equivalent implementations can replace the specific examples shown and described without deviating from the scope of this disclosure. This application is intended to cover any adaptations or variations of the specific examples discussed herein. Therefore, it is intended that this disclosure is limited only by the claims and their equivalents. It should be noted that the methods and devices, including their preferred embodiments, as set forth in this document, can be used alone or in combination with the other methods and devices disclosed herein. Furthermore, the features set forth in connection with a device are also applicable to a corresponding method and vice versa. Moreover, all aspects of the methods and devices set forth in this document can be combined as desired. In particular, the features of the claims can be combined with one another in any way desired. It should be noted that the description and drawings merely illustrate the principles of the proposed methods and systems. A person skilled in the art will be able to implement various arrangements which, although not expressly described or shown herein, embody the principles of the invention and are contained within its spirit and scope. Furthermore, all examples and embodiments set forth in this document are expressly intended primarily for illustrative purposes only, to assist the reader in understanding the principles of the proposed methods and systems. Moreover, all statements herein that provide principles, aspects, and embodiments of the invention, as well as specific examples thereof, are intended to include equivalents thereof. The following embodiments are disclosed: 1. Isolated power converter circuit comprising: a transformer comprising a primary winding and a secondary winding, wherein an intermediate tap of the primary winding is coupled to a source; a first power amplifier coupled to a first end of the primary winding; a second power amplifier coupled to a second end of the primary winding; a first rectifier coupled to a first end of the secondary winding; and a second rectifier coupled to a second end of the secondary winding, wherein a capacitance provided at the first power amplifier and the second power amplifier, and an inductance provided by the transformer, form a primary resonant circuit, and wherein a capacitance provided at the first rectifier and the second rectifier, and an inductance,which is provided by the transformer, form a secondary resonant circuit. 2. Circuit according to embodiment 1, wherein the first power amplifier comprises a first switch connected between the first end of the primary winding and a first reference potential, the second power amplifier comprises a second switch connected between the second end of the primary winding and a second reference potential, and a primary capacitor is provided between the first end of the primary winding and the second end of the primary winding, or a first capacitor is provided between the first end of the primary winding and the first reference potential, and a second capacitor is provided between the second end of the primary winding and the second reference potential. 3. Circuit according to embodiment 1 or 2, wherein the first rectifier comprises a first rectifier device,which is connected between the first end of the secondary winding and a third reference potential, and a second rectifier device connected between the first end of the secondary winding and a load, the second rectifier comprising a third rectifier device connected between the second end of the secondary winding and a fourth reference potential, and a fourth rectifier device connected between the second end of the secondary winding and the load, and a secondary capacitor is provided between the first end of the secondary winding and the second end of the secondary winding, or a third capacitor is provided between the first end of the secondary winding and the third reference potential, and a fourth capacitor is provided between the second end of the secondary winding and the fourth reference potential. 4. Circuit according to one of embodiments 1-3,further comprising a gate driver circuit configured to control the first power amplifier and the second power amplifier in a nested operating mode and synchronously with a switching frequency. 5. Circuit according to embodiment 4, wherein the switching frequency is based on a resonant frequency of the primary resonant circuit. 6. Circuit according to embodiment 5, wherein the switching frequency is less than or equal to the resonant frequency of the primary resonant circuit. 7. Circuit according to embodiment 5 or 6, wherein a resonant frequency of the secondary resonant circuit is configured based on the switching frequency. 8. Circuit according to any one of embodiments 4-7, wherein the switching frequency is based on a resonant frequency of the secondary resonant circuit. 9. Circuit according to any one of embodiments 4-8,wherein the gate driver circuit is configured to: detect a voltage spike at the first end of the primary winding; and switch on the second power amplifier in response to the detection of the voltage spike. 10. Circuit according to embodiment 9, wherein the gate driver circuit is configured to: switch on the second power amplifier using the voltage at the first end of the primary winding. 11. Circuit according to embodiment 9 or 10, wherein the gate driver circuit is configured to: switch off the second power amplifier once a predetermined time period has elapsed since the second power amplifier was switched on, the predetermined time period being based on the switching frequency. 12. Circuit according to embodiment 9, wherein the second power amplifier comprises: a second switch connected between the second end of the primary winding and a second reference potential; a second auxiliary switch,which is connected between the second end of the primary winding and the second reference potential; wherein the gate driver circuit is configured to: detect one end of a voltage spike at the second end of the primary winding; turn on the second auxiliary switch in response to the detection of the end of the voltage spike at the second end of the primary winding; and turn on the second switch in response to the detection of the voltage spike at the first end of the primary winding. 13. Circuit according to embodiment 12, wherein the gate driver circuit is further configured to turn off the second switch and the second auxiliary switch once a predetermined time interval has elapsed since the second switch was turned on, the predetermined time interval being based on the switching frequency. 14. Circuit according to embodiment 12 or 13, wherein the second power amplifier further comprises: a second boost switch,which is connected between the second end of the primary winding and the second reference potential, and wherein the gate driver circuit is further configured to: turn on the second boost switch in response to the detection of the voltage spike at the first end of the primary winding; detect one end of the voltage spike at the first end of the primary winding; and turn off the second boost switch in response to the detection of the end of the voltage spike at the first end of the primary winding. 15. Circuit according to one of embodiments 4-14, wherein the gate driver circuit is configured to: detect a voltage spike at the second end of the primary winding; and turn on the first power amplifier in response to the detection of the voltage spike. 16. Circuit according to embodiment 15,wherein the gate driver circuit is configured to: switch on the first power amplifier using the voltage at the second end of the primary winding. 17. Circuit according to embodiment 15 or 16, wherein the gate driver circuit is configured to: switch off the first power amplifier once a predetermined time period has elapsed since the first power amplifier was switched on, the predetermined time period being based on the switching frequency. 18. Circuit according to embodiment 15, wherein the first power amplifier comprises: a first switch connected between the first end of the primary winding and a first reference potential; a first auxiliary switch,which is connected between the first end of the primary winding and the first reference potential; wherein the gate driver circuit is configured to: detect one end of a voltage spike at the first end of the primary winding; turn on the first auxiliary switch in response to the detection of the end of the voltage spike at the first end of the primary winding; and turn on the first switch in response to the detection of the voltage spike at the second end of the primary winding. 19. Circuit according to embodiment 18, wherein the gate driver circuit is further configured to turn off the first switch and the first auxiliary switch once a predetermined time interval has elapsed since the first switch was turned on, the predetermined time interval being based on the switching frequency. 20. Circuit according to embodiment 18 or 19, wherein the first power amplifier further comprises: a first boost switch,which is connected between the first end of the primary winding and the first reference potential, and wherein the gate driver circuit is further configured to: turn on the first boost switch in response to the detection of the voltage spike at the second end of the primary winding; detect one end of the voltage spike at the second end of the primary winding; and turn off the first boost switch in response to the detection of the end of the voltage spike at the second end of the primary winding. 21. Circuit according to any one of embodiments 1-20, wherein the transformer comprises a core including a material configured to provide galvanic isolation between the primary winding and the secondary winding. 22. Circuit according to embodiment 20, wherein the material has a relative magnetic permeability of less than 10. 23. Circuit according to any one of embodiments 1-22, wherein the transformer comprises a core,which includes a magnetic material. 24. Circuit according to one of embodiments 1-23, further comprising a tuning inductor provided in series with the secondary winding, the tuning inductor forming part of the secondary resonant circuit. 25. Circuit according to one of embodiments 1-24, wherein the number of turns of the primary winding is the same as the number of turns of the secondary winding. 26. System comprising: a first load terminal for connection to a power source; a second load terminal connected to a load; and an isolated power converter circuit according to one of embodiments 1-25.wherein the first load terminal is coupled to the source of the isolated power converter circuit and the second load terminal is coupled to the first rectifier and the second rectifier of the isolated power converter circuit. 27. Method for transferring power from a first side of an isolation transformer to a second side of the isolation transformer, the method comprising: providing power to an intermediate tap of a primary winding on the first side of the isolation transformer; generating a first resonance in the primary winding using a first power amplifier coupled to a first end of the primary winding and a second power amplifier coupled to a second end of the primary winding; rectifying a current induced in a secondary winding on the second side of the isolation transformer, which is inductively coupled to the primary winding.using a first rectifier coupled to a first end of the secondary winding and a second rectifier coupled to a second end of the secondary winding; and providing the rectified current at an output.

Claims

Isolated power converter circuit (100) comprising: a transformer (10) comprising a primary winding (12) and a secondary winding (14), wherein an intermediate tap of the primary winding is coupled to a source; a first power amplifier (20a) coupled to a first end of the primary winding; a second power amplifier (20b) coupled to a second end of the primary winding; a first rectifier (30a) coupled to a first end of the secondary winding;and a second rectifier (30b) coupled to a second end of the secondary winding, wherein a capacitance provided at the first power amplifier and the second power amplifier and an inductance provided by the transformer form a primary resonant circuit, and wherein a capacitance provided at the first rectifier and the second rectifier and an inductance provided by the transformer form a secondary resonant circuit.; Circuit according to claim 1, wherein the first power amplifier (20a) comprises a first switch (22a) connected between the first end of the primary winding (12) and a first reference potential, the second power amplifier (20b) comprises a second switch (22b) connected between the second end of the primary winding and a second reference potential, and a primary capacitor (24a, 24b) is provided between the first end of the primary winding and the second end of the primary winding, or a first capacitor (24a) is provided between the first end of the primary winding and the first reference potential, and a second capacitor (24b) is provided between the second end of the primary winding and the second reference potential. Circuit according to claim 1 or 2, wherein the first rectifier (30a) comprises a first rectifier device (32a) connected between the first end of the secondary winding (14) and a third reference potential, and a second rectifier device (36a) connected between the first end of the secondary winding and a load, the second rectifier (30b) comprises a third rectifier device (32b) connected between the second end of the secondary winding and a fourth reference potential, and a fourth rectifier device (36b) connected between the second end of the secondary winding and the load, and a secondary capacitor (34a, 34b) is provided between the first end of the secondary winding and the second end of the secondary winding, or a third capacitor (34a) is provided between the first end of the secondary winding and the third reference potential.and a fourth capacitor (34b) is provided between the second end of the secondary winding and the fourth reference potential. Circuit according to one of claims 1-3, further comprising a gate driver circuit (50) configured to control the first power amplifier (20a) and the second power amplifier (20b) in a nested operating mode and synchronously with a switching frequency. Circuit according to claim 4, wherein the switching frequency is based on a resonant frequency of the primary resonant circuit and / or the switching frequency is based on a resonant frequency of the secondary resonant circuit. Circuit according to claim 5, wherein a resonant frequency of the secondary resonant circuit is set up based on the switching frequency. Circuit according to one of claims 4-6, wherein the gate driver circuit (50) is configured to: detect a voltage spike at the first end of the primary winding (12); and switch on the second power amplifier (20b) in response to the detection of the voltage spike. Circuit according to claim 7, wherein the gate driver circuit (50) is configured to: switch on the second power amplifier (20b) using the voltage at the first end of the primary winding (12). Circuit according to claim 7 or 8, wherein the gate driver circuit (50) is configured to: switch off the second power amplifier (20b) as soon as a predetermined time period has elapsed since the switch-on of the second power amplifier, wherein the predetermined time period is based on the switching frequency. Circuit according to claim 7, wherein the second power amplifier (20b) comprises: a second switch (22b) connected between the second end of the primary winding (12) and a second reference potential; a second auxiliary switch (26b) connected between the second end of the primary winding and the second reference potential; wherein the gate driver circuit (50) is configured to: detect one end of a voltage spike at the second end of the primary winding; turn on the second auxiliary switch in response to the detection of the end of the voltage spike at the second end of the primary winding; and turn on the second switch in response to the detection of the voltage spike at the first end of the primary winding. Circuit according to claim 10, wherein the gate driver circuit (50) is further configured to switch off the second switch (22b) and the second auxiliary switch (26b) as soon as a predetermined time period has elapsed since the second switch was switched on, wherein the predetermined time period is based on the switching frequency. Circuit according to claim 10 or 11, wherein the second power amplifier (20b) further comprises: a second boost switch (28b) connected between the second end of the primary winding (12) and the second reference potential, and wherein the gate driver circuit (50) is further configured to: turn on the second boost switch in response to the detection of the voltage spike at the first end of the primary winding; detect one end of the voltage spike at the first end of the primary winding; and turn off the second boost switch in response to the detection of the end of the voltage spike at the first end of the primary winding. Circuit according to one of claims 4-12, wherein the gate driver circuit (50) is configured to: detect a voltage spike at the second end of the primary winding (12); and switch on the first power amplifier (20a) in response to the detection of the voltage spike. Circuit according to claim 13, wherein the gate driver circuit (50) is configured to: switch on the first power amplifier (20a) using the voltage at the second end of the primary winding (12). Circuit according to claim 13 or 14, wherein the gate driver circuit (50) is configured to: switch off the first power amplifier (20a) as soon as a predetermined time period has elapsed since the first power amplifier was switched on, wherein the predetermined time period is based on the switching frequency. Circuit according to claim 13, wherein the first power amplifier (20a) comprises: a first switch (22a) connected between the first end of the primary winding (12) and a first reference potential; a first auxiliary switch (26a) connected between the first end of the primary winding and the first reference potential; wherein the gate driver circuit (50) is configured to: detect one end of a voltage spike at the first end of the primary winding; turn on the first auxiliary switch in response to the detection of the end of the voltage spike at the first end of the primary winding; and turn on the first switch in response to the detection of the voltage spike at the second end of the primary winding. Circuit according to claim 16, wherein the gate driver circuit (50) is further configured to switch off the first switch (22a) and the first auxiliary switch (26a) as soon as a predetermined time period has elapsed since the first switch was switched on, wherein the predetermined time period is based on the switching frequency. Circuit according to claim 16 or 17, wherein the first power amplifier (20a) further comprises: a first boost switch (28a) connected between the first end of the primary winding (12) and the first reference potential, and wherein the gate driver circuit (50) is further configured to: turn on the first boost switch in response to the detection of the voltage spike at the second end of the primary winding; detect one end of the voltage spike at the second end of the primary winding; and turn off the first boost switch in response to the detection of the end of the voltage spike at the second end of the primary winding. System (400) comprising: a first load terminal (410) for connection to a power source (412); a second load terminal (420) connected to a load (422); and an isolated power converter circuit (100, 101, 102, 103, 104) according to any one of claims 1-18, wherein the first load terminal is coupled to the source of the isolated power converter circuit and the second load terminal is coupled to the first rectifier and the second rectifier of the isolated power converter circuit. Method (500) for transferring power from a first side of an isolation transformer to a second side of the isolation transformer, the method comprising: providing (510) power to an intermediate tap of a primary winding on the first side of the isolation transformer; generating (520) a first resonance in the primary winding using a first power amplifier coupled to a first end of the primary winding and a second power amplifier coupled to a second end of the primary winding; rectifying (530) a current caused in a secondary winding on the second side of the isolation transformer, which is inductively coupled to the primary winding, using a first rectifier coupled to a first end of the secondary winding and a second rectifier coupled to a second end of the secondary winding;and providing (540) the rectified current at an output.;

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