Engine control device and power conversion system

Abstract

A motor control device is provided capable of reducing loss by controlling a current phase angle based on a change characteristic of inductance. A motor control device is connected to a power converter that performs power conversion from DC power to three-phase AC power and drives a motor with the three-phase AC power to control the power conversion of the power converter. The motor control device includes: an inductance change detection unit configured to calculate an inductance change characteristic indicative of a change in the inductance of the motor in a first phase range including a phase where an absolute value of one of the three-phase AC currents is a maximum; and a current phase angle control unit configured to control a current phase angle based on the inductance change characteristic.

Description

Technical area

[0001] The present invention relates to an engine control device and a power conversion system. State of the art

[0002] Motor control devices determine three-phase AC voltage commands that satisfy torque commands or the like and reduce losses (copper losses and iron losses) in three-phase synchronous motors, and control inverters based on the three-phase AC voltage commands.

[0003] PTL 1 discloses a technique for suppressing such loss. As disclosed in PTL 1, "the carrier wave frequency adjustment unit adjusts a phase difference between the voltage command and the carrier wave to reduce an eddy current loss occurring in a magnet of a rotor of the AC motor according to a d-axis current applied to the AC motor and a rotation speed of the AC motor." In claim 1, "an eddy current loss We is indicated by a proportional relationship expressed by the following formula (11)." In paragraph 0061, "formula (11) can be expressed by replacing it with the proportional relationship expressed in formula (12) below." In paragraph 0063, "the fixed triangular wave phase determination unit 1633 determines a value of the carrier wave phase difference Δθcarr based on the d-axis current sum calculated by the d-axis current sum calculation unit 1632."Here, the value of the carrier wave phase difference Δθcarr is determined so that the value of the d-axis current sum is a minimum.", in paragraph 0072.

[0004] As disclosed in PTL 1, in paragraph 0076, "the voltage phase error calculation unit 163 determines the carrier wave phase difference Δθcarr and calculates the voltage phase error Δθv as described above. Accordingly, the voltage phase error Δθv can be determined so that the d-axis current sum is a minimum according to the d-axis current Id and the motor rotation speed ωr. As a result, the carrier wave frequency fc can be adjusted by changing the phase difference between the voltage command for the inverter 3 and the carrier wave used for pulse width modulation, so as to reduce the eddy current loss occurring in the magnet of the rotor of the motor 2. As a result, an increase in the magnet temperature Tmag can be suppressed, and the occurrence of irreversible demagnetization can be prevented." Citation listPatent literature

[0005] PTL 1: JP 2022-18168 A Summary of the inventionTechnical problem

[0006] However, iron loss, such as eddy current loss, depends on the magnetic flux generated in a coil, and this magnetic flux can be reduced by increasing the d-axis current. Therefore, iron loss can be suppressed if the d-axis current is not minimal. Accordingly, one problem is that the d-axis current must be selected appropriately to minimize the loss.

[0007] In addition, since the iron loss varies depending on the inductance, a problem is that it is necessary to detect a change in inductance and control the current according to the change in inductance to suppress the loss.

[0008] Therefore, it is an object of the present invention to provide a motor control apparatus and a power conversion system capable of reducing a loss by controlling a current phase angle based on a change characteristic of inductance. Solution to the problem

[0009] To solve the above problem, a motor control device according to the present invention is a motor control device connected to a power converter that performs power conversion from DC power to three-phase AC power and drives a motor with the three-phase AC power to control the power conversion of the power converter. The motor control device includes: an inductance change detection unit configured to calculate an inductance change characteristic indicative of a change in the inductance of the motor in a first phase range including a phase where an absolute value of one of the three-phase AC currents is a maximum; and a current phase angle control unit configured to control a current phase angle based on the inductance change characteristic.

[0010] A power conversion system according to the present invention includes, for example, the motor control device and the power converter. Advantageous effects of the invention

[0011] According to the present invention, it is possible to provide a motor control apparatus and a power conversion system capable of reducing a loss by controlling a current phase angle based on a change characteristic of inductance. Short description of the drawings [ Fig. 1] Fig. 1 is an overall configuration diagram of a motor drive system including a motor control device. [ Fig. 2] Fig. 2 is a block diagram illustrating a functional configuration of a motor control device according to an embodiment. [ Fig. 3] Fig. 3 is a diagram illustrating a sampling period of an average value of a current. [ Fig. 4] Fig. Figure 4 is a diagram illustrating one sampling period of the average value of the current. [ Fig. 5] Fig. 5 is a diagram illustrating an example of a relationship between a current I flowing in a coil and a magnetic flux Φ generated in the coil. [ Fig. 6] Fig. 6 is a diagram illustrating a sampling period of a current used to calculate an inductance change characteristic. [ Fig. 7] Fig. 7 is a diagram illustrating a sampling period of a current used to calculate an inductance change characteristic. [ Fig. 8] Fig. Figure 8 is a diagram illustrating a relationship between a carrier wave frequency and a current ripple. [ Fig. 9] Fig. Figure 9 is a diagram illustrating a relationship between a three-phase alternating current and a carrier wave frequency. [ Fig. 10] Fig. Figure 10 is a diagram illustrating a relationship between a loss and a current phase angle. Description of embodiments

[0012] The present invention will be described in detail below with reference to the drawings. The present invention is not limited to the embodiments described below. The embodiments are merely examples, and the present invention can be implemented in various modified and improved forms based on the skill of the art. In the drawings used in the following description, common devices and operations are denoted by the same reference numerals, and the description of the devices, devices, and operations described above may be omitted.

[0013] Fig. 1 is an overall configuration diagram of a motor drive system including a motor control device. Fig. 1, a motor drive system 100 according to the embodiment includes a motor control device 1, a motor 2, an inverter (power converter) 3, a rotational position detector 4, a high-voltage battery 5, a current detection unit 7, and a rotational position sensor 8.

[0014] A rotational position θr of the motor 2 is input from the rotational position detector 4 to the motor control device 1. Iu, Iv, and Iw, which denote three-phase alternating currents flowing in the motor 2, are input from the current detection unit 7, and a torque command T* is input from a host control device (not shown). The motor control device 1 generates a gate signal for controlling the drive of the motor 2 based on input information and outputs the gate signal to the inverter 3. Accordingly, an operation of the inverter 3 is controlled, and the drive of the motor 2 is controlled. Details of the motor control device 1 will be described below.

[0015] The inverter 3 includes an inverter circuit 31, a PWM signal drive circuit 32, and a smoothing capacitor 33. The PWM signal drive circuit 32 generates a PWM signal for controlling each switching element included in the inverter circuit 31 based on the gate signal input from the motor control device 1, and outputs the generated PWM signal to the inverter circuit 31. The inverter circuit 31 includes switching elements corresponding to upper and lower arms of the U-phase, the V-phase, and the W-phase, respectively. By controlling each of these switching elements according to the PWM signal input from the PWM signal drive circuit 32, the DC power supplied from the high-voltage battery 5 is converted into AC power to be output to the motor 2.The smoothing capacitor 33 smoothes the DC power supplied from the high-voltage battery 5 to the inverter circuit 31.

[0016] The high-voltage battery 5 is a DC power source of the motor drive system 100 and outputs a power supply voltage Hvdc to the inverter 3. The power supply voltage Hvdc of the high-voltage battery 5 is converted by the inverter circuit 31 and the PWM signal drive circuit 32 of the inverter 3 into a pulsed three-phase AC voltage with a variable voltage and a variable frequency, which is to be applied to the motor 2 as the grid voltage. Accordingly, AC power is supplied from the inverter 3 to the motor 2 based on the DC power of the high-voltage battery 5. The power supply voltage Hvdc of the high-voltage battery 5 varies according to a state of charge of the high-voltage battery 5.

[0017] The motor 2 is a three-phase motor rotationally driven by AC power supplied from the inverter 3, and includes a stator and a rotor. In the embodiment, an example in which a permanent magnet synchronous motor is used as the motor 2 is described, but the present invention is not limited thereto. When the AC power input from the inverter 3 is applied to the three-phase coils Lu, Lv, and Lw provided in the stator, three-phase alternating currents Iu, Iv, and Iw are electrified in the motor 2, and thus a magnetic flux is generated in each coil. When an attractive force and a repulsive force are generated between the magnetic flux of each coil and the magnetic flux of the permanent magnet disposed in the rotor, a torque is generated in the rotor, and the motor 2 is rotationally driven.

[0018] The rotational position sensor 8, which detects a rotational position θr of the rotor, is fitted into the motor 2. The rotational position detector 4 calculates a rotational position θr from an input signal of the rotational position sensor 8. A calculation result of the rotational position θr by the rotational position detector 4 is input to the motor control device 1 and is used for phase control of the AC power performed by the motor control device 1, which generates a pulsed gate signal according to a phase of an induced voltage of the motor 2.

[0019] Here, a resolver including an iron core and a winding is more suitable than the rotational position sensor 8, but there is no problem even if a magnetoresistive element such as a giant magnetoresistance (GMR) sensor or a sensor using a Hall element is applied. Any sensor can be used as the rotational position sensor 8 as long as a magnetic pole position of the rotor can be measured. The rotational position detector 4 can estimate the rotational position θr using the three-phase alternating currents Iu, Iv, and Iw flowing in the motor 2 and the three-phase alternating voltages Vu, Vv, and Vw applied to the motor 2 from the inverter 3, without using the input signal from the rotational position sensor 8.

[0020] The current detection unit 7 is arranged in a current path between the inverter 3 and the motor 2. The current detection unit 7 detects three-phase alternating currents Iu, Iv, and Iw (the U-phase alternating current Iu, the V-phase alternating current Iv, and the W-phase alternating current Iw) that electrify the motor 2. For the current detection unit 7, for example, a Hall current sensor or the like can be used. Detection results of the three-phase alternating currents Iu, Iv, and Iw by the current detection unit 7 are input to the motor control device 1 and are used to generate a gate signal, which is performed by the motor control device 1. Although Fig. 1 illustrates an example in which the current detection unit 7 includes three current detectors. Two current detectors may be provided, and an alternating current of the remaining one phase can be calculated from the fact that the sum of the three-phase alternating currents Iu, Iv, and Iw is zero. The pulsed direct current flowing from the high-voltage battery 5 into the inverter 3 can be detected by a shunt resistor or the like inserted between the smoothing capacitor 33 and the inverter 3, and the three-phase alternating currents Iu, Iv, and Iw can be obtained based on the direct current and the three-phase alternating voltages Vu, Vv, and Vw applied from the inverter 3 to the motor 2.

[0021] Next, details of the engine control device 1 will be described. Fig. 2 is a block diagram illustrating a functional configuration of the engine control device 1 according to the embodiment.

[0022] The motor control device 1 includes functional blocks of a current command generation unit 10, a rotational speed calculation unit 11, a current conversion unit 12, a current control unit 13, a carrier wave frequency adjustment unit 14, a carrier wave generation unit 15, a phase calculation unit 16, a three-phase voltage conversion unit 18, a gate signal generation unit 19, an inductance change detection unit 21, a current phase angle control unit 22, and a carrier wave frequency switching unit 23. Here, the current command generation unit 10, the current conversion unit 12, the current control unit 13, and the three-phase voltage conversion unit 18 function as a three-phase AC voltage command generation unit. A configuration of the three-phase AC voltage command generation unit is not limited to this and may include other functions.The motor control device 1 includes, for example, a microcomputer and can implement these functional blocks by executing a predetermined program in the microcomputer. Alternatively, some or all of the functional blocks can be implemented using a hardware circuit such as a logic IC or a field-programmable gate array (FPGA).

[0023] The rotation speed calculation unit 11 calculates a motor rotation speed ωr, which indicates a rotation speed (rotational speed) of the motor 2, from a temporal change in the rotational position θr of the motor 2. The motor rotation speed ωr can be a value expressed either by an angular velocity (rad / s) or a rotational speed (rpm). The values ​​can be mutually converted and used.

[0024] The carrier wave frequency setting unit 14 determines a carrier wave frequency fc, which indicates a frequency of the carrier wave used to generate a gate signal, based on the rotational speed ωr obtained by the rotational speed calculation unit 11. For example, the carrier wave frequency fc is determined so that the number Nc of synchronous PWM carrier waves, which indicates the number of carrier waves for one cycle of a voltage waveform in synchronous PWM control, is a predetermined integer. Here, the number Nc of synchronous PWM carrier waves can be set, for example, as a number that satisfies a conditional formula of Nc = 3 × (2 × n-1) among multiples of 3. In this conditional formula, n indicates any natural number. For example, n = 1 (Nc = 3), n = 2 (Nc = 9), n = 3 (Nc = 15), or the like can be selected. The number Nc of synchronous PWM carrier waves can also be changed according to the rotation speed ωr.Then, the carrier wave frequency setting unit 14 determines fc based on the rotation speed ωr and the number Nc of synchronous PWM carrier waves.

[0025] The carrier wave frequency setting unit 14 sets the carrier wave frequency fc according to a switching flag FlgFrqSw from the carrier wave frequency switching unit 23, which will be described below. For example, an example will be described in which the carrier wave frequency setting unit 14 sets the carrier wave frequency fc = K*wr*Nc / 2n using a coefficient K and changes K according to a value of the switching flag FlgFrqSw, but the present invention is not limited to this. If the coefficient K is set to be larger than the coefficient K during the switching flag FlgFrqSw = 1, the carrier wave frequency can be switched so that a carrier wave is a high-frequency carrier wave during the switching flag FlgFrqSw = 1 and a low-frequency carrier wave during the switching flag FlgFrqSw = 0. A method for switching the carrier wave frequency fc is not limited to this.The carrier wave frequency setting unit 14 can use the rotational position θr of the motor 2 as an input, so that the carrier wave frequency can be switched only when the rotational position θr of the motor 2 is within a specific range. Accordingly, the carrier wave frequency switching unit 23 performs a calculation only when the rotational position θr of the motor 2 is within a specific range, and a processing load of the motor control device 1 can be reduced.

[0026] The carrier wave generation unit 15 generates a carrier wave signal Sc for each of the three-phase AC voltage commands Vu*, Vv* and Vw* based on the carrier wave frequency fc determined by the carrier wave frequency setting unit 14.

[0027] The phase calculation unit 16 calculates an estimated rotational position θe of the motor 2 by the following formulas (1) to (3) based on the rotational position θr, the rotational speed ωr and the carrier wave frequency fc. θe=θr+φv φv=ωr⋅1.5Tc Tc=1 / fc

[0028] Here, φv indicates a calculation delay compensation value of a voltage phase, and Tc indicates a carrier wave period. The calculation delay compensation value φv is a value that compensates for the occurrence of a calculation delay corresponding to 1.5 control cycles from the time when the rotational position detector 4 detects the rotational position θr to the time when the motor control device 1 outputs the gate signal to the inverter 3.

[0029] Fig. 3 and Fig. 4 are diagrams illustrating sampling periods of average current values. In Fig. 3 and Fig. In Figure 4, the vertical axis represents current, the horizontal axis represents time, and examples of a current waveform of one phase in the three-phase alternating current are illustrated. A current generates a current ripple and oscillates at a frequency close to the carrier wave frequency fc. To facilitate the description, a frequency of the current ripple is described below assuming that the frequency is equal to the carrier wave frequency fc, but the present invention is not limited to this. As shown in Fig. As illustrated in Figure 3, the current conversion unit 12 acquires an average current value for each sampling period Ts, which is determined based on the carrier wave frequency fc, for example, Ts = 1 / fc. When the carrier wave frequency fc is high, the average current value can be acquired every integer multiple of 1 / fc, for example, every Ts = 2 / fc, as shown in Fig. 4. As the sampling period becomes longer, the frequency with which the average value is acquired becomes smaller. Therefore, a processing load of the engine control device 1 can be reduced.

[0030] Fig. 5 is a diagram illustrating an example of a relationship between a current I flowing in a coil and a magnetic flux Φ generated in the coil. The horizontal axis represents the current I, and the vertical axis represents the magnetic flux Φ. When the current I is small, the current I and the magnetic flux Φ have a linear relationship. When the current I is large, the current I and the magnetic flux Φ have a nonlinear relationship. Hereinafter, in a graph representing the relationship between the current I and the magnetic flux Φ, a portion where the current I and the magnetic flux Φ have a linear relationship is called a linear portion, and a portion where the current I and the magnetic flux Φ have a nonlinear relationship is called a nonlinear portion.

[0031] The inductance L of the coil is defined by Φ = LI. Therefore, if the inductance L is calculated at each of a point P1 on the linear section and a point P2 on the nonlinear section, the inductance L at point P1 is an inclination L0 of a straight line connecting the origin and P1, and the inductance L at point P2 is an inclination L2 of a straight line connecting the origin and P2. As described above, the inductance L takes a constant value L0 at each point on the linear section, but the inductance L takes a value other than L0 at each point on the nonlinear section, as shown in the equation shown in Fig. 5, and varies at different points. That is, when the current I and the magnetic flux Φ have a linear relationship, the inductance L assumes a constant value. When the current increases and the current I and the magnetic flux Φ have a nonlinear relationship, the inductance L changes. This is because magnetic saturation or the like occurs as the current increases. A difference between the inductance L at a point on the nonlinear portion and the inductance L0 at the linear portion increases as the current increases. In order to detect such a change in inductance from the inductance L0, it is desirable to calculate the inductance L directly from the magnetic flux Φ and the current I. However, there is a problem that it is difficult to detect the magnetic flux Φ. Accordingly, in the embodiment, a tangent of the Fig. 5, that is, dΦ / dI, is defined as an inductance change characteristic ΔL (= dΦ / dI), and a change in inductance from the inductance L0 is detected using the inductance change characteristic ΔL.

[0032] The inductance change characteristic ΔL at point P1 on the linear section matches the inductance L0 in the linear section. Similarly, the inductance change characteristic ΔL is also L0 at any point on the linear section. Accordingly, the inductance change characteristic ΔL at any point on the linear section is constant at ΔL = L0. On the other hand, the inductance change characteristic ΔL at any point on the nonlinear section, such as point P2, takes a value different from the inductance change characteristic ΔL = L0 in the linear section and approaches 0 as the current increases.Such a change characteristic of the inductance change characteristic ΔL, that is, a change characteristic in which the inductance change characteristic ΔL is constant at L0 in the linear section and deviates from the inductance change characteristic L0 in the linear section as the current in the nonlinear section increases, is similar to the change characteristic of the inductance L. Therefore, by monitoring the inductance change characteristic ΔL, it is possible to detect a change in inductance. As described below, the inductance change characteristic ΔL can be easily calculated without using the magnetic flux Φ, unlike the inductance. Accordingly, in the embodiment, the inductance change characteristic ΔL is calculated by the inductance change detection unit 21 to detect a change in inductance.

[0033] As in Fig. As shown in Figure 2, the three-phase alternating currents Iu, Iv, and Iw and previous values ​​Vu*_z, Vv*_z, and Vw*_z of the three-phase alternating voltage command are input to the inductance change detection unit 21. The inductance change detection unit 21 calculates the inductance change characteristic ΔL using the formula V = ΔL × (dI / dt) for a phase with a largest absolute value of current among the three-phase alternating currents Iu, Iv, and Iw. V = ΔL × (dI / dt) can be derived using the above-described dΦ / dI = ΔL with respect to a formula V = dΦ / dt established between the voltage V applied to the coil and the magnetic flux Φ generated in the coil.

[0034] Fig. 6 and Fig. 7 are diagrams illustrating a sampling period of a current used to calculate an inductance change characteristic. In Fig. 6 and Fig. In Figure 7, the vertical axis represents current, the horizontal axis represents time, and an example of a current waveform of one phase of three-phase alternating current is illustrated. As shown in Fig. As illustrated in Figure 6, the inductance change detection unit 21 obtains dI / dt using a minimum value I_min and a maximum value I_max of the current, which are indicated by a square diagram in a current current. Therefore, the inductance change detection unit 21 detects the current value at a time point when a current ripple becomes the minimum value and the maximum value. For example, as described above, when detecting the average value (pie chart) of the current for each sampling period Ts, the inductance change detection unit 21 detects a minimum value of the current ripple at a time point of (4N + 1) × Ts / 4 (where N is an integer) and detects a maximum value of the current ripple at a time point of (4N + 3) × Ts / 4. In this case, a time interval between the minimum value I_min and the maximum value I_max of the current ripple detected within the sampling period Ts of the average value of the current is approximately Ts / 2.Accordingly, the inductance change detection unit 21 calculates dI / dt as (I_max - I_min) / (Ts / 2). The inductance change detection unit 21 substitutes the calculated dI / dt and a previous value of the AC voltage command of the phase for which dI / dt was calculated into V = ΔL × (dI / dt) to calculate the inductance change characteristic ΔL, and outputs the inductance change characteristic ΔL to the current phase angle control unit 22.

[0035] In the above description, the inductance change characteristic ΔL is calculated for a phase having the largest absolute current value among the currents Iu, Iv, and Iw of each phase, but the present invention is not limited to this. For example, the inductance change characteristic ΔL may be calculated for each phase regardless of the magnitude of the absolute values ​​of the currents Iu, Iv, and Iw of each phase, and the inductance change characteristic ΔL that deviates most from L0 may be output to the current phase angle control unit 22.

[0036] When the carrier wave frequency fc is high, the inductance change detection unit 21 can alternately detect the minimum value I_min of the current ripple and the maximum value I_max of the current ripple for every 1 / fc, as shown in Fig. 7. In this case, the inductance change detection unit 21 calculates dI / dt by considering the detected minimum value I_min (solid square plot) of the current ripple as the minimum value (dotted square plot) of the current ripple of a next cycle.

[0037] As in Fig. As illustrated in Figure 2, the current phase angle control unit 22 determines a current phase angle θcrr according to the inductance change characteristic ΔL calculated by the inductance change detection unit 21. Here, the current phase angle θcrr is a leading phase angle from the q-axis of a current vector determined by the three-phase alternating currents Iu, Iv, and Iw. The current phase angle control unit 22 preferably detects and stores in advance the current phase angle θcrr at which a loss is minimal for each inductance change characteristic ΔL. For example, the current phase angle θcrr at which the loss is minimal is detected in advance for each inductance change characteristic ΔL through experiment, analysis, or the like. When characteristics or the like of the motor are clear, the current phase angle θcrr at which the loss is minimal can be calculated for each inductance change characteristic ΔL based on the mathematical formula and acquired in advance.The current phase angle control unit 22 outputs the current phase angle θcrr corresponding to the input inductance change characteristic ΔL with reference to the current phase angle θcrr for each inductance change characteristic ΔL detected in advance.

[0038] The carrier wave frequency switching unit 23 determines the switching of the carrier wave frequency based on the current phase angle θcrr and the estimated rotational position θe of the motor 2. Therefore, a direction of a current vector is obtained from the estimated rotational position θe and the current phase angle θcrr of the motor 2, and a phase angle of the U-phase current, a phase angle of the V-phase current, and a phase angle of the W-phase current are obtained. The carrier wave frequency switching unit 23 outputs a switching flag FlgFrqSw based on the phase angle of the U-phase current, the phase angle of the V-phase current, and the phase angle of the W-phase current, as described below. For example, two types of frequencies, a high frequency and a low frequency, are prepared as the carrier wave frequency. The switching flag FlgFrqSw is set to 1 when the carrier wave frequency is set to the high frequency.The switching flag FlgFrqSw is set to 0 when the carrier wave frequency is set to the low frequency. Details of an operation of the carrier wave frequency switching unit 23 will be described below.

[0039] The current command generation unit 10 calculates a d-axis current command Id* and a q-axis current command Iq* based on the input torque command T*, the power supply voltage Hvdc, and the current phase angle θcrr. Here, for example, the d-axis current command Id* and the q-axis current command Iq* are obtained according to the torque command T*, the power supply voltage Hvdc, and the current phase angle θcrr using a preset current command map, a mathematical formula representing a relationship between the d-axis current Id and the q-axis current Iq, and the motor torque, or the like.After the d-axis current command Id* and the q-axis current command Iq* that satisfy a requirement for the torque command T* and the power supply voltage Hvdc are obtained, a final d-axis current command Id* and q-axis current command Iq* can be obtained by correcting the d-axis current command Id* and the q-axis current command Iq* using the current phase angle θcrr.

[0040] The current conversion unit 12 performs dq conversion based on the rotational position θr obtained by the rotational position detector 4 on the three-phase alternating currents Iu, Iv and Iw detected by the current detection unit 7, and calculates a d-axis current value Id and a q-axis current value Iq.

[0041] Based on the deviations between the d-axis current command Id* and the q-axis current command Iq* output by the current command generation unit 10, and the d-axis current value Id and the q-axis current value Iq output by the power conversion unit 12, the current control unit 13 calculates a d-axis voltage command Vd* and a q-axis voltage command Vq* in response to the torque command T* so that these values ​​coincide with each other. Here, for example, according to a control method such as PI control, the d-axis voltage command Vd* is obtained according to the deviation between the d-axis current command Id* and the d-axis current value Id, and the q-axis voltage command Vq* is obtained according to the deviation between the q-axis current command Iq* and the q-axis current value Iq.

[0042] The three-phase voltage conversion unit 18 calculates and outputs the three-phase AC voltage commands Vu*, Vv*, and Vw* (a U-phase voltage command Vu*, a V-phase voltage command Vv*, and a W-phase voltage command Vw*) based on the d-axis current command Id* and the q-axis current command Iq* output from the current command generation unit 10 and the estimated rotation position θe calculated by the phase calculation unit 16.

[0043] Using the carrier wave signal Sc output from the carrier wave generation unit 15, the gate signal generation unit 19 performs pulse width modulation on each of the three-phase AC voltage commands Vu*, Vv*, and Vw* output from the three-phase voltage conversion unit 18, and generates a gate signal for controlling the operation of the inverter 3. Specifically, a pulsed voltage is generated for each of the U-phase, the V-phase, and the W-phase based on a comparison result between the three-phase AC voltage commands Vu*, Vv*, and Vw* output from the three-phase voltage conversion unit 18 and the carrier wave signal Sc output from the carrier wave generation unit 15. Then, a pulsed gate signal is generated for the switching element of each phase of the inverter 3 based on the generated pulsed voltage.At this time, gate signals Gup, Gvp, and Gwp of the upper arms of the phases are logically inverted to generate gate signals Gun, Gvn, and Gwn of the lower arms. The gate signal generated by the gate signal generation unit 19 is output from the motor control device 1 to the PWM signal drive circuit 32 of the inverter 3 and is converted into a PWM signal by the PWM signal drive circuit 32. Accordingly, each switching element of the inverter circuit 31 is controlled between ON and OFF, and the output voltage of the inverter 3 is adjusted.

[0044] Fig. Figure 8 is a diagram illustrating a relationship between a carrier wave frequency and a current ripple. The vertical axis represents current, the horizontal axis represents time, and a portion of a current waveform corresponding to a phase of the three-phase alternating current within a time width T1 is extracted. It is assumed that a carrier wave frequency of Fig. 8b higher than a carrier wave frequency of Fig. 8a is. In Fig. 8a, a current ripple corresponding to two cycles is contained in the time width T1. On the other hand, in Fig. 8b contains a current ripple corresponding to four cycles in the same time width T1. The current ripple contained in the same time width T1 is larger than that in Fig. 8b, in which a carrier wave frequency is higher. Therefore, a current difference ΔI and a time width ΔT used to calculate dI / dt are smaller than those in Fig. 8b, in which a carrier wave frequency is higher. As ΔI and ΔT decrease, the accuracy of calculating dI / dt also improves. Therefore, dI / dt can be calculated with a higher accuracy than that in Fig. 8b, in which a carrier wave frequency is higher. Accordingly, the accuracy of the inductance change characteristic ΔL calculated using dI / dt is also improved. Thus, the motor control device according to the embodiment increases the carrier wave frequency in a phase range where the inductance changes, that is, a phase range including a phase where an absolute value of one of the three-phase alternating currents is a maximum. Accordingly, it is possible to accurately detect a change in the inductance.

[0045] Fig. Figure 9 is a diagram illustrating a relationship between a three-phase alternating current and a carrier wave frequency. Fig. Figure 9 is a diagram in which a gate signal is superimposed on a current waveform of a three-phase alternating current, and the horizontal axis represents a phase. Fig. 9, PR1 is a first phase range and PR2 is a second phase range. A linewidth of the gate signal indicates a level of the carrier wave frequency. The lower the carrier wave frequency, the thicker the linewidth of the gate signal. As shown in Fig. As illustrated in Figure 9, the carrier wave frequency adjustment unit 14 increases the frequency of a carrier wave in the first phase range PR1, which includes a phase where the absolute value of one of the three-phase alternating currents is a maximum. Within a range of one period (2π) of the phase of the three-phase alternating current, there are three phases where one of Iu, Iv, and Iw is a maximum, and there are three phases where one of Iu, Iv, and Iw is a minimum. Therefore, the first phase range PR1 is set in advance to include six phases where one of Iu, Iv, and Iw is a maximum or minimum. Fig. 9 illustrates an example in which regions near six phases where one of Iu, Iv, and Iw is a maximum or minimum, that is, six regions where the line width of the gate signal is thin, are set as the first phase regions PR1, but the present invention is not limited to this. When the phase of the three-phase alternating current calculated based on the current phase angle θcrr and the estimated rotational position θe of the motor 2 is within the first phase region PR1, the carrier wave frequency switching unit 23 sets the switching flag FlgFrqSw to 1. The carrier wave frequency setting unit 14 sets the frequency of the carrier wave to a high frequency when the switching flag FlgFrqSw is 1. In this way, the change in inductance can be accurately detected by increasing the carrier wave frequency in the phase region where the change in inductance occurs.

[0046] As in Fig. 9, the carrier wave frequency adjusting unit 14 lowers the frequency of the carrier wave in the second phase range PR2 including the phase where the absolute value of one of the three-phase alternating currents becomes 0. There are six phases where one of Iu, Iv, and Iw becomes 0 within a range of one period (2π) of the phase of the three-phase alternating current. Therefore, the second phase range PR2 is set in advance to include six phases where one of Iu, Iv, and Iw becomes 0. Fig. 9 illustrates an example in which regions near six phases where one of Iu, Iv, and Iw is 0, that is, six regions where the line width of the gate signal is thick, are set as the second phase regions PR2, but the present invention is not limited to this. When the phase of the three-phase alternating current calculated based on the current phase angle θcrr and the estimated rotational position θe of the motor 2 is within the second phase region PR2, the carrier wave frequency switching unit 23 sets the switching flag FlgFrqSw to 0. Then, the carrier wave frequency setting unit 14 sets the frequency of the carrier wave to a low frequency when the switching flag FlgFrqSw is 0. As described above, the switching loss can be suppressed by lowering the carrier wave frequency in the phase region where the inductance is unlikely to change.

[0047] Fig. Figure 10 is a diagram illustrating a relationship between a loss and a current phase angle. As shown in Fig. As illustrated in Figure 10, a current phase angle at which loss is minimal differs from a current phase angle at which current is minimal. In the present invention, loss can be minimized by calculating the inductance change characteristic from the measured current value and controlling the current phase angle based on the calculated inductance change characteristic, rather than controlling the current phase angle to minimize current. List of reference symbols 1 engine control device 2 engines 3 inverters 4 rotation position detector 5 high-voltage battery 7 Current measuring unit 8 Rotation position sensor 10 Current command generation unit 11 Speed ​​calculation unit 12 Power conversion unit 13 Power control unit 14 Carrier wave frequency adjustment unit 15 Carrier wave generation unit 16 Phase calculation unit 18 Three-phase voltage conversion unit 19 Gate signal generation unit 21 Inductance change detection unit 22 Current phase angle control unit 23 Carrier wave frequency switching unit 31 Inverter circuit 32 PWM signal control circuit 33 smoothing capacitor 34 Voltage detection unit 100 Motor drive system QUOTES CONTAINED IN THE DESCRIPTION

[0000] This list of documents submitted by the applicant was generated automatically and is included solely for the convenience of the reader. This list is not part of the German patent or utility model application. The DPMA assumes no liability for any errors or omissions. Cited patent literature

[0000] JP 2022-18168 A

[0005]

Claims

[1] A motor control device connected to a power converter that performs power conversion from DC power to three-phase AC power and drives a motor with the three-phase AC power to control the power conversion of the power converter, the motor control device comprising: an inductance change detection unit configured to calculate an inductance change characteristic indicative of a change in the inductance of the motor in a first phase range including a phase at which an absolute value of one of the three-phase alternating currents is a maximum; and a current phase angle control unit configured to control a current phase angle based on the inductance change characteristic. [2] The engine control device according to claim 1, further comprising: a three-phase AC voltage command generation unit configured to generate a three-phase AC voltage command, wherein the inductance change detection unit calculates the inductance change characteristic based on the three-phase AC voltage command and a three-phase AC current of the motor. [3] The motor control device according to claim 2, wherein the inductance change detecting unit calculates the inductance change characteristic from the time differentiation of the three-phase AC current and the three-phase AC voltage command. [4] The engine control device according to claim 2, further comprising: a gate signal generating unit configured to perform pulse width modulation on the three-phase AC voltage command and generate a gate signal for controlling the power conversion of the power converter; and a carrier wave frequency setting unit configured to set a frequency of a carrier wave used for the pulse width modulation based on the current phase angle and a rotational position of the motor, wherein the carrier wave frequency setting unit sets the frequency of the carrier wave to be higher than the frequency of the carrier wave in a second phase range including a phase at which an absolute value of one of the three-phase alternating currents in the first phase range becomes zero. [5] Power conversion system comprising: the engine control device according to one of claims 1 to 4; and the power converter.

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