RADAR SENSOR WITH SYNCHRONIZED HIGH-FREQUENCY COMPONENTS

Description

[0001] The invention relates to a radar sensor with at least two synchronously operating high-frequency components, each having at least one signal path in which the phase of the transmitted high-frequency signal is changed by a temperature-dependent phase difference. State of the art

[0002] In the pursuit of fully autonomous driving, radar sensors for motor vehicles are becoming increasingly complex, particularly with an increase in the number of transmit and receive channels, for example, to implement MIMO (multiple input multiple output) or digital beamforming concepts. From a reliability and power dissipation perspective, it is desirable to limit the size of the high-frequency components (MMICs; Monolithic Microwave Integrated Circuits) used in radar sensors and instead employ a large number of MMICs, preferably of the same design, that are synchronized with each other so that the phase relationships between the signals transmitted in all transmit channels are known and can be appropriately taken into account when evaluating the received signals.

[0003] Due to the space requirements of the individual MMICs, the distances between them become so large as the number of components increases that the propagation times of the signals used for synchronization cannot be neglected. A particular difficulty arises from the fact that with larger spatial distances between the individual MMICs, it can no longer be assumed that all MMICs have the same temperature, so propagation time and phase differences of unknown magnitude can occur due to the temperature response of the electronic components. Disclosure of the invention

[0004] The object of the invention is to create a possibility to achieve precise synchronization of the high-frequency components even in the case of temperature-dependent phase differences without impairing the function of the radar sensor.

[0005] This object is achieved according to the invention in that in each high-frequency component a phase detector is connected in parallel to the signal path, which phase detector supplies a signal which assumes an extremum at a certain known phase difference, independent of temperature, and in that a phase shifter is arranged in the signal path, with which the phase difference can be adjusted such that the signal of the phase detector assumes the extremum, and in that a control device is designed to adjust the phase differences in the signal paths of the at least two high-frequency components such that the signal of the phase detector assumes the extremum, and to synchronize the high-frequency components with one another on the basis of the phase differences which can be determined under these conditions.

[0006] The invention offers the advantage that the inherently unknown temperature-dependent phase differences can be adjusted to known values ​​using the phase shifter and the phase detector, so that they can be taken into account when synchronizing the high-frequency components. Although the phase detectors used according to the invention do not allow for a quantitative measurement of the phase difference, they do have the advantage that the point at which their signal reaches its extreme is independent of temperature, so that no disruptive temperature influences can occur when adjusting the phase difference. The components to be added to the individual high-frequency components, i.e. the phase detectors and phase shifters, can be easily integrated into the components without significantly impairing the power dissipation or the measurement accuracy of the components.

[0007] Advantageous embodiments and further developments of the invention are specified in the subclaims.

[0008] The phase detectors can, for example, be rectifier diodes which, when supplied with signals from opposite ends of the signal path, produce a DC voltage as an output signal which is proportional to the superimposed amplitude of the signals and thus varies depending on the phase difference between a maximum (in the case of constructive superposition) and a minimum (zero in the case of complete cancellation of the signals).

[0009] An IQ modulator, for example, can be used as a phase shifter in the signal path.

[0010] The synchronization of the various high-frequency components can be achieved via special synchronization signals exchanged between the components. In another embodiment, the transmission signal of one high-frequency component or a transmission channel of this component (the master) is simultaneously used as a synchronization signal for another component (the slave). Each individual component can have multiple signal paths in which potentially temperature-dependent phase differences can occur. In this case, each of these signal paths is assigned a phase detector and a phase shifter.

[0011] Each high-frequency component can have its own voltage-controlled local oscillator for generating the high-frequency signal. A common reference signal, which is fed to all components, can then be used to synchronize the oscillators in the various components. Since the transmission path for this reference signal is not subject to temperature variation, the phase differences resulting from the different propagation times of the reference signal to the individual oscillators are known or can be adjusted to known values ​​by selecting the appropriate line lengths.

[0012] In an FMCW (Frequency Modulated Continuous Wave) radar, each radio-frequency component has a mixer for each receiving channel. This mixer mixes the received signal with a portion of the signal transmitted at the same time, generating an intermediate-frequency signal whose spectrum provides information about the distances and relative speeds of the detected objects during the radar sensor's measurement mode. In one or more calibration modes, these mixers can be used to measure the phase differences in various closed signal path chains. Since at least some of these chains also contain the path of the synchronization signal from one radio-frequency component to another, once the phase differences for the remaining signal paths have been calibrated using phase detectors and phase shifters, all relevant phase differences can be determined, allowing the radio-frequency components to be correctly synchronized with each other.

[0013] In one embodiment, each radio frequency component may include an additional mixer that can monitor the complex amplitude (magnitude and phase) of the transmit signal.

[0014] In the following, an embodiment example is explained in more detail using the drawing.

[0015] They show: Fig. 1 is a simplified circuit diagram of a radar sensor according to the invention with two synchronized high-frequency components; and Fig. 2 is a signal characteristic of a phase detector as used in the high-frequency components according to Fig. 1 is used.

[0016] The Fig. 1 The radar sensor shown has two radio-frequency components 10, 12, which are designed, for example, as MMICs and can be arranged on a common circuit board. Each radio-frequency component has several transmit channels and several receive channels. However, for the sake of simplicity, only a single transmit channel (TX) and a single receive channel (RX) are shown for each radio-frequency component.

[0017] To synchronize the high-frequency components 10, 12, a signal path L is provided that couples the outputs of the transmission channels TX of the two high-frequency components. For example, the radar sensor can be operated with the high-frequency component 10 acting as the master and the high-frequency component 12 acting as the slave, which uses the signal transmitted on the signal path L as the synchronization signal. Since the two high-frequency components 10, 12 must necessarily be arranged at a certain distance from each other on the board, the signal transmitted on the signal path L has a certain signal propagation time, which leads to an a priori unknown phase difference L, which must be compensated for correct synchronization of the components.

[0018] Each of the two high-frequency components 10, 12 contains a local oscillator 16 with a downstream amplifier 18. The output signal of amplifier 18 is fed into the transmit channel TX via a coupler 20 and transmitted as a transmit signal, and is also fed to a first input of a mixer 22. The signal received in the receive channel RX is fed to a second input of mixer 22, so that mixer 22 delivers the mixed product of the signals present at both inputs as an output signal.

[0019] In the high-frequency component 10, a signal path EXT1 leads from the output of the transmit channel TX back to the coupler 20. This makes it possible to use a signal arriving via the signal path L as the external transmit signal instead of the signal from this component's own local oscillator 16. A signal path from the coupler 20 to the output of the transmit channel TX is designated TX1. A signal path from the coupler 20 to the mixer 22 is designated LO1. These two signal paths each contain a phase shifter 24, which is formed, for example, by an IQ modulator.

[0020] If no signal is received in the receive channel RX, the output signal of amplifier 18 can be coupled directly to the corresponding input of mixer 22 via a signal path TST1. Signal path TST1 also contains a phase shifter 24. The output of this phase shifter is connected to an input of another mixer 26. The other input of this mixer can receive the transmit signal, which is tapped at the output of the transmit channel TX.

[0021] A phase detector 28, formed, for example, by a rectifier diode, is connected between this output of the transmit channel TX and the output of amplifier 18. Another phase detector 28 is connected between the output of amplifier 18 and the receive input of mixer 22.

[0022] The inputs of the local oscillators 16 of the two high-frequency components 10, 12 are connected to a common reference signal source 30, which serves to synchronize the oscillators with each other. The connections between the two oscillators 16 and the reference signal source 30 can be considered as a further signal path DPH, on which a phase difference DPH which depends on the respective line lengths. For example, the arrangement can be chosen so that the lines from the reference signal source 30 to each of the two oscillators 16 have the same length. In this case, the phase relationship DPH = 0.

[0023] The high-frequency component 12 has the same structure as the high-frequency component 10. The various signal paths are designated with the same letter code as in the high-frequency component 10, but with the index "2" instead of "1".

[0024] In the signal path TX1 a phase difference occurs TX1 which may have a component dependent on the local temperature. The same applies to the signal paths LO1, EXT1, and TST1. The corresponding phase differences are designated here with the same reference symbols, but with an underscore. The same applies to the signal paths in the high-frequency module 12. The phase difference L in the signal path L is generally temperature dependent.

[0025] To synchronize the two high-frequency components 10 and 12, the temperature-dependent phase differences must be calibrated. For this purpose, the radar sensor can be operated in four different calibration modes.

[0026] In calibration mode 1, the high-frequency component 10 provides its transmission signal (output of the transmission channel TX) to the high-frequency component 12 as a synchronization and transmission signal. The signal then travels via the signal paths TX1, L, EXT2, and LO2 to the mixer 22. There, the measurement signal is formed by the mixing product with the signal supplied to the mixer 22 in the high-frequency component 12 via the signal path TST2. The output signal of the mixer then represents the following phase difference: D 1 = DPH ¯ + TX 1 ¯ + L _ + EXT 2 ¯ + LO 2 ¯ − TST 2 ¯

[0027] In calibration mode 2, the high-frequency module 12 provides its transmission signal to the mixer of the high-frequency module 10. The signal runs via the signal paths TX2, L, EXT1 and LO1 to the mixer 22 of the high-frequency module 10. There, the mixing product is formed with the signal supplied via the signal path TST1. The output signal of the mixer 22 in this case is D 2 = − DPH ¯ + TX 2 ¯ + L _ + EXT 1 ¯ + LO 1 ¯ − TST 1 ¯

[0028] In calibration mode 3, the high-frequency component 10 provides the transmission signal, which is transmitted via signal path LO1, to its own mixer. There, the mixing product is formed with the signal transmitted as a reference signal via signal path TST1. The output signal of the mixer then corresponds to the phase difference. D 3 = LO 1 ¯ − TST 1 ¯

[0029] In calibration mode 4, the high-frequency component 12 provides the transmission signal to its own mixer 22, which is fed to the mixer via signal path LO2. There, the mixing product is formed with the reference signal, which is fed via signal path TST2. The output signal of the mixer then represents the phase difference. D 4 = LO 2 ¯ − TST 2 ¯

[0030] The measurements in the four calibration modes result in a system of equations with the ten variables: DPH , TX1 , L , EXT2 , LO2 , TST2 , TX2 , EXT1 , LO1 and TST1 .

[0031] This includes DPH can be determined based on the cable lengths and can therefore be assumed to be known. For the calibration of the two high-frequency components, the phase difference L not directly relevant, but only the sum of the phase differences L + EXT2 or the sum L + EXT1 , so that L as an independent variable can be eliminated and instead only the independent variables ( L + EXT2 ) and L + EXT1 ) need to be considered. This leaves eight unknowns, but the system of equations is still underdetermined.

[0032] To solve the system of equations, four of the eight unknowns must therefore be determined in another way. However, instead of measuring these unknowns directly (which would again require active components with their own unknown temperature response), according to the invention, the unknown phase differences are adjusted to known values ​​with the aid of the phase detectors 28 and the phase shifters 24. For this purpose, phase detectors 28 are used which, although they do not allow an absolute measurement of the phase difference, due to their specific characteristic curve, allow a temperature-independent determination of a phase difference at which the characteristic curve assumes a minimum (or maximum). An example of such a characteristic curve is shown in Fig. 2 shown. Here, a DC voltage U, which drops across the rectifier diode 28 serving as phase detector, is plotted as a function of the phase difference ϕ of the signals that are superimposed in the rectifier diode. With a sinusoidal course of the superimposed signals, the characteristic curve has maxima at the phase differences of 0 and 360°, and at the phase difference of 180° there is a pronounced minimum that can be sharply localized due to the characteristic curve (which cannot be differentiated at this point). With the help of the associated phase shifter 24, the phase difference can then be adjusted so that the output signal U supplied by the phase detector assumes this minimum. It is then known that the phase difference between the compared signals is 180°.

[0033] In Fig. 1 In this way, with the help of the phase detector 28, which is connected in the high-frequency module 10 between the amplifier 18 and the output of the transmission channel TX, and with the help of the phase shifter 24 in the signal path TX1, the phase difference TX1 to 180°. With the help of the phase detector 28, which is connected between the amplifier 18 and the mixer 22, and with the help of the phase shifter 24 in the signal path TST1, the phase difference TST1 to 180°. The same applies to the phase differences TX2 and TST2 in the high-frequency module 12. This then allows equations (3) and (4) to LO1 and LO2 so that the remaining system of equations for the unknowns ( L + EXT2 ) and ( L + EXT1) can be resolved. Since all relevant phase shifts can be determined independently of temperature, it is possible to precisely synchronize the high-frequency components 10 and 12.

[0034] In the example shown, the mixer 26 can also be used to directly compare the signal transmitted in the respective transmission channel TX with the signal that is fed to the mixer 22 in calibration mode 1 or 2.

[0035] Any deviations can then be compensated using the phase shifter 24 in the signal path LO1 or LO2.

[0036] The functions of the above-described components of the high-frequency modules 10, 12 in the measuring mode and in the various calibration modes are controlled by an electronic control device (not shown).

Claims

1. Radar sensor with at least two synchronously operating radio-frequency modules (10, 12), each having at least one signal path (TX1, TST1; TX2, TST2), in which the phase of the transmitted radio-frequency signal is changed by a temperature-dependent phase difference, characterized in that a phase detector (28) is connected in parallel with the signal path (TX1, TST1; TX2, TST2) in each radio-frequency module (10, 12) and provides a signal (U) which assumes an extreme value regardless of the temperature for a certain phase difference, in that a phase shifter (24) is arranged in the signal path (TX1, TST1; TX2, TST2) and can be used to adjust the phase difference such that the signal (U) from the phase detector assumes the extreme value, and in that an electronic control device is designed to adjust the phase differences in the signal paths of the at least two radio-frequency modules such that the signal (U) from the phase detector assumes the extreme value, and to synchronize the radio-frequency modules with each other on the basis of the phase differences that can be determined under these conditions.

2. Radar sensor according to Claim 1, in which the phase detector (28) is formed by a rectifier diode, at which the signals to be compared are superimposed on each other.

3. Radar sensor according to Claim 1 or 2, in which the phase shifter (24) is formed by an IQ modulator.

4. Radar sensor according to one of the preceding claims, in which each radio-frequency module (10, 12) has a local oscillator (16) and the local oscillators (16) of the various radio-frequency modules (10, 12) are connected to a common reference signal source (30) via lines of a known length.

5. Radar sensor according to one of the preceding claims, in which each radio-frequency module (10, 12) has a mixer (22) which is designed to mix a signal received in a receiving channel (RX) with a signal transmitted in a transmitting channel (TX).

6. Radar sensor according to Claim 5, in which a synchronization signal output of each radio-frequency module (10, 12) can be coupled to a synchronization signal input of each other radio-frequency module (12, 10) via a signal path (L) and the signal received via the synchronization signal path (L) in each radio-frequency module can be supplied as a transmission signal to the mixer (22) of this radio-frequency module via a further signal path (EXT1, EXT2).

7. Radar sensor according to Claim 6, in which, in each of the radio-frequency modules (10, 12), an output of at least one transmitting channel (TX) forms the synchronization input and the synchronization output of this radio-frequency module.

8. Radar sensor according to Claim 6 or 7, in which the mixer (22) of each radio-frequency module (10, 12) can be supplied with the output signal from its own local oscillator (16) as a test signal via a signal path (TST1).

9. Radar sensor according to one of Claims 5 to 8, in which each radio-frequency module (10, 12) has a further mixer (26) which is designed to monitor the complex amplitude of the signal transmitted in the transmitting channel (TX) of this radio-frequency module.