A motor control circuit

Abstract

A motor control circuit for a permanent magnet motor comprises a motor demand signal generator that outputs an intermediate motor current demand signal IDQ* dependent on requested current demand and e

Description

This invention relates to motor control circuit for Pulse Width modulated drive of permanent magnet motors. It is well known in the art of motor control circuits for permanent magnet motors that at a certain velocity and operating point (current demand) a certain torque ripple is produced in a motor due to cogging, current measurement noise, inverter bridge nonlinearities and the like. This is especially significant at low motor speeds in automotive power steering systems because the ripple will be felt by the driver at the handwheel. To ameliorate this it is known to include within a motor control circuit a torque ripple cancellation stage, sometimes referred to as a low speed compensation stage (LSCS) because may be active only at low speeds in some applications. This stage generates an appropriate torque ripple cancellation waveform. Based on current demand correct coefficients are selected, which serve as sine and cosine magnitude for the waveform. The waveform frequency will depend on the motor velocity as it is known that the ripple frequency is a harmonic of the motor velocity. The resulting sine and cosine waveforms are summed together, which produces the torque ripple cancellation waveform and this in then added to (or subtracted from) the motor control waveforms to produce an overall motor demand signal. The effect is to reduce ripple for a specific order or multiple order of harmonic for which it was tuned. Each order requires a separate component in the compensation waveform. Tuning of the current demand ripple compensation can be performed by taking three measurements at a certain magnitude (used for scaling) for each operating point (current demand). Those measurements are used to calculate an optimal cancellation magnitude and phase of the torque ripple cancellation waveform. The tune is typically produced by a software tool during the design of the motor control circuit, which does the calculation. The coefficients are then saved in a memory of the motor control circuit for use during operation of the motor control circuit. The tune works most effectively at the demanded velocity and bandwidth it was tuned at. Various demands are measured, but typically only 1 speed, e.g. 25 rpm, and 1 bandwidth, e.g. 100 Hz, partly, due to a large number of measurements necessary. As such the tuning is only partly effective when a different bandwidth is used for the motor control. Consider an example where an LSCS operates between 0 - 220 rpm, which means there is a certain range of frequency of the LSCS compensation signal present in the system, depending on the order compensated. For example, for 3rd electrical order in a motor with 3 pole-pairs at 100 rpm: f = 100 / 60*3*3 = 15 Hz LSCS signal frequency. This frequency will vary with speed. Depending what bandwidth is selected for the motor current control circuit the presence of the LSCS compensation signal can have a significant effect on phase delay in the system. Phase delay means that the cancellation waveform is not completely in anti-phase to the harmonic, which needs to be cancelled. This can be seen in Figure 1 and Figure 2 of the accompanying drawings in which the bottom plot represents phase delay vs frequency. The higher the frequency of the signal, higher the delay. Additionally, the effect is stronger when bandwidth of the controller is lower, most typically because the operation of the motor current controller is slower or due to delays introduced by the current measurement circuit). During the tuning process, any phase delay due to the bandwidth is automatically taken into the account by the tune. However, if the motor was tuned at 100Hz, 25 rpm and operates at 220 rpm, the difference in signal phase between 25 rpm and 220 rpm for 6th electrical ripple order, for example, is ~60 degrees. The higher the order, the bigger the difference. Therefore, the compensation signal might not be the most effective at that speed. In this particular case shown in Figure 2 the compensation completely loses its effectiveness at 60 degrees delay. For even higher delays now the ripple will be reinforced even further until 180 degrees. The applicant has appreciated that it would be desirable to further reduce the presence of torque ripple harmonics beyond that achievable using a prior art LSCS as described above in particular at low speeds. The invention aims to ameliorate the effect of the cancellation signal on the ripple felt at a handwheel of a vehicle. According to a first aspect the invention provides a motor control circuit for a permanent magnet motor, the control circuit comprising: a motor demand signal generator that outputs an intermediate motor current demand signal IDQ* that is dependent on the requested current demand and the electrical position of the motor; a torque ripple compensation stage which outputs a torque ripple compensation waveform that contains a frequency having a magnitude that matches at least harmonic frequency of a torque ripple that would be generated by the motor if the motor was driven in response to the intermediate signal, the torque ripple compensation waveform being in anti-phase with the torque ripple at a set frequency and control circuit bandwidth at which the torque ripple compensation stage is tuned; and a phase error compensator stage that receives as an input the torque ripple compensation waveform and outputs a modified torque ripple compensation waveform in which any unwanted phase delay or phase advance of the torque ripple compensation signal has been substantially removed when the motor control bandwidth and frequency differ from that used in the tuning of the torque ripple compensation stage such that the modified waveform is in anti-phase to the torque ripple, and in which the motor control circuit combines the intermediate current demand signal IDQ* with the modified torque ripple compensation waveform to produce a compensated current demand signal, and a PI controller which receives an input signal that is dependent on value of the compensated current demand signal. By implementing the present invention a set of motor control circuits can be tuned at a standard motor current control circuit bandwidth and speed - for example 500Hz and 25 rpm or even lower velocity with the lead compensator circuit ensuring that any phase error in the torque ripple compensation signal is removed when the circuit is in use at a bandwidth that differs from the tuned bandwidth. Furthermore, the user can switch to any bandwidth and operate within the full speed range of the torque ripple compensator stage without worrying of degraded current demand signal performance as this will automatically be corrected by the motor control circuit. The phase error compensator stage may apply a phase compensation to the torque ripple compensation signal based on a transfer function having at least one scaling coefficient, the value of the or each constant being stored in an electronic memory of the control circuit and in which the appropriate scaling coefficient or co-efficients are selected as a function of the motor velocity, the bandwidth and the frequency. By bandwidth here we mean the bandwidth of the motor current closed loop circuit that include the motor and the PI controller. The stored information may comprise a multi-dimensional map of one or more scaling coefficients that are used by the transfer function. The motor control circuit may be configured such that the output of the torque ripple compensation stage is input to the transfer function and the output of the transfer function is the combined with the intermediate torque demand signal. The transfer function may be of the following form: L (s; = —— = — 1------I =----- a \ 1 -¼ Ts J s -r Where: X is the input to the function; Y is the output of the function C(s) is the output of the transfer function that defines the modified torque ripple compensation waveform; s is the complex laplace transform variable coo- is the filter zero crossover frequency, Ts is a time constant, and alpha is a scaling coefficient that is stored in the memory. In this transfer function 6¾ moves with operating point, so $ — & This gain may be set by appropriate selection of the value of the coefficient “alpha” so as to compensate for any gain loss due to the application of the transfer function to the torque ripple compensation waveform. The transfer function also applies the following phase shift to the input signal: Phase: =                 | — tan-1              = tan-2 |   1 — tan-1 (—J = = -tan-1 (a) —tan-Hl) From gain and phase equations the required value of "a" can be determined depending on the phase shift needed. The skilled person will understand that the phase error compensator stage may use a different transform which may have two or more scaling co-efficient. The system may include a memory in which the transfer function is stored and in which the or each scaling coefficient are stored, optionally as map as suggested above. This may be the same memory in which the parameters used by the torque ripple compensation stage that are set during tuning are stored. The phase error compensator stage may comprise two functional blocks. A first block determines which value or values to use for the scaling coefficient or coefficients as a function of the bandwidth and motor rotor velocity. As such the function of this block may be considered to be a prediction of the phase error. A second block receives at an input the torque ripple compensation waveform and modifies this using the transfer function to remove the predicted phase error. In an alternative, the torque ripple compensation stage and the phase error compensator stage may be combined such that a modified torque ripple compensation waveform is derived directly as a function of the requested current demand, the motor velocity, the bandwidth and the frequency without first generating an unmodified torque demand signal. The motor control circuit may form a part of a motor circuit according to a second aspect of the invention that comprises a permanent magnet motor, a motor drive circuit that applies voltage waveforms to each phase of the motor, and the motor control circuit as defined by the first aspect of the invention. There will now be described by way of example only one embodiment of the present invention with reference to and as illustrated in the accompanying drawings of which: Figure 1 is plot of a single torque ripple harmonic and a torque compensation signal waveform that is in perfect anti-phase together with the resulting torque generated by the motor that has the ripple removed; Figure 2 is a plot of the same waveforms as Figure 1 but where the torque ripple compensation waveform is 60 degrees out of phase with the ideal anti-phase waveform of Figure 1 showing how the ripple is not removed from the motor torque in this situation; Figure 3 is a schematic representation of an electric steering system that includes an embodiment of a motor control circuit in accordance with a first aspect of the invention; Figure 4 is detailed schematic of the motor circuit part of the electric steering system showing various signals present within the circuit; Figure 5 is bode plot the for the output of the phase error compensation stage with the transfer function set to apply phase shifts from -40 to +40 degrees as a frequency of 80Hz for different values of “a” in the transfer function. Figure 6 is a bode plot corresponding to Figure 5 showing the magnitude of the output of the phase error compensation stage when the value of “a” is selected so that the frequency is varied whilst maintaining a constant 45 degree phase shift. Figure 7 is a bode plot corresponding to Figure 5 showing the magnitude of the output of the phase error compensation stage when the scaling factor “a” is chosen so that the frequency is fixed but the phase shift is varied; Figure 8 is schematic showing in detail the functional blocks within the motor control circuit that generate a modified torque ripple compensation waveform which is in perfect anti-phase with the torque ripple; Figure 9 is a circuit diagram of a suitable inverter bridge and the motor phase windings that may be used in the electric steering system of Figure 3; and Figure 10 shows in more detail the blocks that make up the motor control circuit of the electric steering system of Figure 3. Figure 3 is a schematic representation of the key parts of an electric steering system for a vehicle which incorporates an embodiment of a motor control circuit in accordance with the present invention. The electric steering system includes a three phase brushless motor 1 that is connected through a gearbox to a steering wheel. In use the motor 1 applies a Motor Torque that either assists the driver in turning the wheel or resists the driver turning the wheel. The former mode is typically used where there is a direct mechanical connection from the steering wheel to the road wheels of the vehicle and makes the steering lighter. The later mode is typically employed where the steering wheel has no mechanical connection to the wheels, an arrangement commonly referred to as a steer by wire system. The electric steering system includes a torque sensor 26 that measures the torque in a steering column, and feeds this to a torque controller 28. The torque controller calculates a requested torque demand signal T*, shown in Figure 4, that is indicative of the Motor Torque that is to be produced by the motor. This is fed into a current controller 21, which converts the torque demand into a requested current demand according to known characteristics of the motor 1 and feeds this into a PI controller. The control circuit 21 causes currents to flow in the motor 1 to make the motor operate to provide the demanded torque. The system in the example of Figure 3 therefore has two closed loops- the torque controller loop and the current control circuit loop. As shown in Figure 4 the motor control circuit 21 takes the demanded torque T* and a measure of the motor rotor position w and generates a requested current demand IDQ* signal in the fixed d-q frame. This signal is then fed into a subtractor block of the motor current circuit 21 along with a measure of the actual current iDQ flowing in the motor, expressed in the d-q frame and derived from the output of the current sensor 24. The output of the subtractor is an error signal eDQ indicative of the difference between the requested current demand IDQ* and the actual measured d-q current iDQ. The error signal eDQ is fed to the input of a proportional-integral (PI) controller stage that converts the current error signal eDQ which is in the d-q frame into a voltage signal in the d-q frame, and this is then converted into three phase voltages UVW by a dq-UVW converter. This is then converted into the required PWM voltage signals for each phase using the chosen SVM modulation technique by a PWM converter. The motor control circuit constantly attempts to drive the value of the error signal eDQ to zero, which ensures that the current in the motor phases matches the demanded current. The signals take time to travel round the closed loop formed by the motor control circuit 21, the motor 1 and the current measurement path and this determines the bandwidth, referred to in this document as the motor current control circuit bandwidth. Note that there will be a second closed loop in most implementations at the torque demand level which will have its own independent bandwidth. The motor 1 in the example of Figures 3 and 4 comprises three motor windings generally designated as phases A, B and C. This can be seen in Figure 9. The phases may be connected in a star network or could be connected in another arrangement, such as a delta topology. One end of each coil is connected to a respective terminal. The other ends of the coils are connected together to form the star centre 7. The free ends are connected to a motor drive circuit that includes a set of high power switches arranged as an inverter bridge 8. An example inverter bridge 8 for use in the electric steering system of Figures 3 and 4 is shown in Figure 9. The inverter bridge comprises a three arm bridge 8, one arm for each phase of the motor. Each arm 10, 12, 14 of the bridge comprises a pair of switches in the form of a top transistor 16 and a bottom transistor 18 connected in series between a supply rail 20 and ground line 22. The motor windings 2, 4, 6 are each tapped off from between a respective complementary pair of transistors 16, 18. The transistors 16, 18 are turned on and off in a controlled manner in response to the signal output from the motor control circuit 21 of Figure 3 of the drawings, to provide pulse width modulation of the potential applied to each of the phase windings, thereby to control the potential difference applied across each of the windings 2, 4, 6 and hence also the current flowing through the windings. This in turn controls the strength and orientation of the magnetic field produced by the windings, which sets the torque produced by the motor. The current sensor 24 is provided in the ground line 22 between the motor 1 and ground so that the controller 21 can measure the total current flowing though all of the windings 2, 4, 6. In order to measure the current in each of the windings the total current has to be sampled at precise instances within the PWM period where the voltage applied to each terminal of the winding (and hence the conduction state of a particular phase) is known. If preferred a separate current sensor could be provided for each phase. The controller in this example uses a Space Vector Modulation (SVM) algorithm although any modulation technique can equally be used within the scope of the present invention and this should not be construed as limiting. Each winding 2, 4, 6 in a three phase system can only be connected to either the supply rail 20 or the ground line 22 and there are therefore eight possible states of the switches of the control circuit. Using 1 to represent one of the phases being at positive voltage and 0 to represent a phase connected to ground, state 1 can be represented as

[100] indicating phase A at 1, phase B at 0 and phase C at 0, State 2 is represented as

[110] , state 3 as

[010] , state 4 as [Oil], state 5 as

[001] , state 6 as

[101] , state 0 as

[000] and state 7 as

[111] . Each of states 1 to 6 is a conducting state in which current flows through all of the windings 2, 4, 6, flowing in one direction through one of them and in the other direction through the other two. State 0 is a zero volt state in which all of the windings are connected to ground and state 7 is a zero volt state in which all the windings are connected to the supply rail. During normal operation when the inverter bridge switching circuit is being controlled by the motor control circuit 21 to produce pulse width modulation, each of the phases 2,4,6 will normally be turned on and off once in each PWM period. The relative lengths of time that are taken up in each state will determine the magnitude and direction of the magnetic field produced in each winding, and hence the magnitude and direction of the total torque applied to the rotor. In a real world application, the motor 1 when rotating will generate a torque that varies periodically from the requested torque demand due to a phenomena known as cogging torque. The variation manifests as a harmonic component overlaid onto the desired motor torque and this will vary with the motor speed. This phenomena is well known and is caused by interaction between the permanent magnets of the rotor and the teeth of the stator. To reduce the effect of cogging torque it is known to add to the motor current demand signal IDQ* that is generated by the the motor control circuit 21 an extra torque ripple compensation waveform that has the same frequency as a torque ripple harmonic but is in anti-phase to the torque ripple. The motor control circuit generates a current demand signal in the usual way but this is combined with the torque ripple compensation waveform before being passed to the motor drive stage as a modified IDQ* signal. As shown in Figure 10 the motor control circuit includes a torque ripple compensation stage 30. In this example the stage is active only at low motor speeds and is often referred to as a low speed compensation stage (LSCS) for that reason. Based on the requested motor current demand IDQ* the torque ripple compensation stage selects two coefficients which serve as sine and cosine magnitude of a compensation waveform. The waveform frequency will depend on the motor velocity. The resulting sine and cosine waveforms are summed together which produces a torque ripple cancellation waveform in the DQ domain and reduces ripple for specific orders it was tuned. Each order requires separate coefficients map. The torque ripple cancellation stage may be tuned to remove a single harmonic or several different harmonics. This signal is the output of the LSCS. Tuning of the torque ripple compensation stage 30 is performed prior to use of the motor control circuit with a motor, typically during the design and development phase of a project. For each operating point (requested current demand) three measurements are taken at a certain magnitude (used for scaling). Those measurements are used to calculate the required cancellation magnitude and phase of the waveform and to set the values stored as the map. The tune is produced by internal tool, which does the calculation. The applicant has appreciated that the tuning only provides effective compensation when the motor circuit is operated at the tuned bandwidth. Outside of this the cancellation in ineffective due to phase offsets that mean the compensation signal will not be in perfect anti-phase with the torque ripple. To ameliorate this effect the motor control circuit also includes a phase error compensation stage 32 which applies a transfer function to the output of the LCSC to produce a modified torque ripple compensation stage. Figure 8 shows schematically how this stage and the preceding torque ripple compensation stage may be implement within the motor current control circuit 21. In figure 8 the first block shows the LSCS stage 30 receiving the demand current IDQ* and a measure of the motor electrical position. From this a transfer function is applied according to the tuning of the LSCS. This is then fed to an input of the phase error compensation 32 stage that applies a further transfer function 36 that alters the phase and the magnitude of the signal. The parameters of this transfer function are received from a phase lag prediction circuit 34 whose role is to feed to the compensation stage the parameters of the transfer function. These may be selected from a map or look up table stored in memory 38. The selection is made as a function of the actual control circuit bandwidth and the frequency. The phase modified signal output from the phase error compensation stage is then combined using an addition of subtraction block with the current demand signal IDQ* generated by the motor control stage and a measurement of the actual current IDQ to provide a compensated current demand signal. This is then combined with the measured current signal IDQ to form the error signal eDQ that is fed to the PI controller. As shown in Figures 8 and 10 this means that the compensation is applied to IDQ* before subtracting the measured IDQ signal to form the error signal eDQ, but it could also be applied to the result of this subtraction. Note that this subtraction is not shown in Figure 4 of the drawings. The transfer function used in the phase error compensation stage of this example is in the form of a lead lag compensator that has the following form; , ton ) = = - ----- =----X(s) a V 1 4- Ts / s 4 - mprti         a — scaling coe / / tei«wt Gain and phase cont. time In this implementation to0 moves with operating point, so to 5 — to (this will be necessary to compensate LSCS signal gain loss due to the current controller) Phase: <C( / to) =                 | — tan-1              = tan-2 |   1 — tan-1 (—) = /             x              kJ.? >             * — /            x&v = tan"2(a) -tan"2(l) from this ”a” can be determined depending on the phase shift needed. Figure 5 shows the lead compensator bode plot with phase shifts from -40 to +40 degrees as a Wgwre i. Compensator plot tot?; pfewe at S3 to              Vporonjeter. frequency of 80Hz for different values of "a" in the transfer function. <C( / to) = tan' 2 (a) — 45’ => tan (tan X^)) = tan (<£( / &>) + 45’) = > a = tan         + 45 ) (In Matlab ®convert to radians) ,           .                sitotoCtoj S+45'S Due to implementation use a = —;—.... .. . cos':         *+5 ) The behaviour of the transfer function may also be considered in the discrete time domain. Discretizing the system using "Tustin" method gives: 1 + z = es'5 = e'l-”1- = cost to 72 J 4- j smtto72.,5 1 - 2 2 - 1 72z + .1 Algorithm Implementation C( ~ j = —=-----— => x » Zsffc +11 - v S’ = m * Oscecs +15 - n »=> " '      ms — s     ...... x ■>      411 — v * la(fc) 4 »i * 0«tffe5 + 1) = ---------------:---------------------tn Gain and phase disc, time A<3 — y- Gain: = VFIS- — ?S JI )4 / SSS51^-¾. Phase: To avoid complicated mathematical transformations phase is kept the same as continuous time: ¢1 = tan (<C( / &s) + 45^) As co moves with the operating frequency of the LSCS harmonic order that is being compensated, then the system is always in the same operating point of the bode plot, i.e., same gain and phase as shown in Figure 6. However, if we choose different phase shift, “a” parameter will change and gain plot will change as well as shown in Figure 7. Therefore, new gain compensation needs to be calculated to keep output signals as close as possible to the original magnitude. Overall, here it is demonstrated that modifying “a” it is possible to shift phase for different frequencies. However, it is also observed that shifting frequency for the same “a”, keeps phase shift the same. It means that if there is input of sinusoids with different frequencies (i.e. a torque ripple compensation signal at various motor velocities), a phase error compensation stage that employs this transfer function can independently control the phase shift applied to the input of the transfer function and keep the same magnitude of the sinusoid. The present invention overcomes shortcomings in the prior art design of motor control circuit which uses Low Speed Current Shaping (LSCS) that is tuned at single bandwidth and speed. The calculation algorithm for LSCS implicitly accounts for any system lag present as it optimises for the minimum torque ripple. However, at different speeds system lag changes even within the same bandwidth of the current controller. Therefore, LSCS does not operate at its most optimal point outside of the speed at which it was tuned. Furthermore, if application requires different bandwidth from what original LSCS was tuned at, it will affect the whole “slope” of how lag changes with the speed. Therefore, LSCS would need to be re-tuned for various operating points and it is a lot of work. The arrangement of the present invention compensates for this phase shift allowing a simple design of tuned motor control circuit to be used in a range of different applications through selection of the appropriate mapped coefficient values for the phase error compensation stage. This removes the need to retune the motor control circuit for each application as the impact of varying bandwidth and frequency is automatically removed by the phase error compensation stage.

Claims

1. A motor control circuit for a permanent magnet motor, the control circuit comprising:a motor demand signal generator that outputs an intermediate motor current demand signal IDQ* that is dependent on the requested current demand and the electrical position of the motor;a torque ripple compensation stage which outputs a torque ripple compensation waveform that contains a frequency having a magnitude that matches at least harmonic frequency of a torque ripple that would be generated by the motor if the motor was driven in response to the intermediate signal, the torque ripple compensation waveform being in anti-phase with the torque ripple at a set frequency and control circuit bandwidth at which the torque ripple compensation stage is tuned; anda phase error compensator stage that receives as an input the torque ripple compensation waveform and outputs a modified torque ripple compensation waveform in which any unwanted phase delay or phase advance of the torque ripple compensation signal has been substantially removed when the motor control bandwidth and frequency differ from that used in the tuning of the torque ripple compensation stage such that the modified waveform is in anti-phase to the torque ripple,and in which the motor control circuit combines the intermediate current demand signal IDQ* with the modified torque ripple compensation waveform to produce a compensated current demand signal, anda PI controller which receives an input signal that is dependent on value of the compensated current demand signal.

2. A motor control circuit according to claim 1 in which the phase error compensator stage applies a phase compensation to the torque ripple compensation signal based on a transfer function having at least one scaling coefficient, the value of the or each constant being stored in an electronic memory of the control circuit and in which the appropriate scaling coefficient or co-efficients are selected as a function of the motor velocity, the bandwidth and the frequency.

3. A motor control circuit according to claim 2 in which the stored information comprises a multi-dimensional map of one or more scaling coefficients that are used by the transfer function.

4. A motor control circuit according to claim 2 or claim 3 configured such that the output of the phase compensator stage function is added to the intermediate motor current demand signal to produce the overall current demand signal.

5. A motor control circuit according to any one of claim 2 to 4 in which the transfer function of the phase error compensator stage has the form:• -.. 'F(s) 1 / 14-aT.sX s +Cfs) = —= - -----J =---—X(x)   a x 1 4- Ts > s -rWhere:Cfs) is the output of the transfer function that defines the modified torque ripple compensation waveform;co_0- is the input frequency, anda is a scaling coefficient that is stored in the memory.

6. A motor control circuit according to claim 5 in which the gain of the transfer function is set by appropriate selection of the value of the coefficient “a” so as to compensate for any gain loss due to the application of the transfer function to the torque ripple compensation waveform.

7. A motor control circuit according to any preceding claim in which the phase error compensator stage comprises:a first functional block that determines which value or values to use for the scaling coefficient or coefficients as a function of the bandwidth and motor rotor velocity; anda second functional block that receives at an input the torque ripple compensation waveform and modifies this using the transfer function to remove the predicted phaseerror.

8. A motor circuit comprising a permanent magnet motor and a motor control circuit according to any one of claims 1 to 7.

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