Current control module

Abstract

A current control module is for controlling an in-rush current resulting from the initiation of an electrical connection of a battery to an electrical load, wherein a metal-oxide-semiconductor field-e

Description

Field of the Invention The present invention relates to a current control module for controlling an in-rush current resulting from the initiation of an electrical connection of a battery to an electrical load; a battery management system including the control module; a method of controlling the in-rush current; and a computer-readable medium for executing the method. Background Typically, when a battery is initially connected to a load (or source), such as a device to be powered by the battery or a charger for charging the battery, an input capacitance of the load / battery is initially discharged. This causes an initial surge of high current, known as an inrush current, to be generated between the battery and the load / source. This can cause damage to the battery and / or load. Existing electronic devices incorporate methods and circuits to reduce the potentially damaging effects of the in-rush current and existing battery systems may incorporate circuitry to protect against short-circuits. Recent innovations in battery technology have resulted in batteries which allow for faster charging and discharging than traditional batteries. These fast-(dis)charging batteries typically have a lower internal impedance than traditional batteries which can result in higher peak inrush currents to be generated following connection to a load or source. In some circumstances, damage to the battery or connected load may occur owing to the much higher in-rush current. Moreover, the present inventors have observed that conventional short-circuit protection in batteries may be too slow to react or shut down a connection between the battery and the load / source entirely in the presence of these high in-rush currents. US6067239 proposes a smoothing circuit that can operate at high temperatures despite the fact that a capacitor’s internal resistance varies with temperature. It incorporates a MOSFET connected in series with the capacitor and fed by a voltage that is divided by a resistor and a thermistor. The thermistor reduces the voltage input to the MOSFET such that the MOSFET’s Rds increases over the course of the connection. However, thermistors typically have a slow reaction time to changing temperature. Therefore, controlling the MOSFET based on the temperature in this way can result in a lag in controlling the in-rush current which could result in damaging current spikes, particularly when used with low impedance batteries. EP3080540 proposes a circuit breaker with a switch that opens automatically in response to fault events. The switch incorporates a FET connected in series with a bi-metallic strip. The bimetallic strip deforms by different amounts at different temperatures and therefore if the circuit begins to overheat it will physically disconnect the FET. The reaction time of the bimetallic strip to changing temperatures can be slow and can cause the connection between battery and load to be inconveniently broken in response to an in-rush current. The present invention has been devised in light of the above considerations. Summary of the Invention In a general sense, the present invention provides a device for controlling in-rush current between a battery and a load (or a source) by gradually adjusting a resistance to the in-rush current in a feedback loop until the current reaches a steady state. This resistance is applied by adjusting an effective drain-source resistance (Rds) of a transistor, such as a MOSFET (Metal Oxide Semi-conductor Field Effect Transistor), between the battery and the load. The use of a MOSFET and feedback loop to control in-rush current results in quicker and more adjustable control of the in-rush current. By employing a MOSFET in this way, existing circuitry provided in a battery management system (BMS) may be adapted to regulate the current, thereby reducing the circuit complexity and cost compared to more widely used methods for protecting against high in-rush currents. Accordingly, the present invention provides a control module for controlling a MOSFET in response to a current signal, to control the (Rds) of the MOSFET such that the current sense signal is kept below an acceptable level. The MOSFET is controlled to have a high Drain Source ON Resistance (Rds) when the battery is connected to a load, to prevent a high in-rush current, and to have a progressively reduced Rds over time as the connection persists and the in-rush current subsides. In general, the controller is configured to run a feedback loop wherein a current sense signal is compared to a threshold level and the control signal (which controls the Rds of the MOSFET is updated based on the comparison. The present invention also provides a battery management system (BMS) connected to a battery and incorporating a MOSFET which is controlled to have a high Drain Source ON Resistance (Rds) when the battery is connected to a load and to have a progressively reduced Rds over time as the connection persists. The load could be a device to be powered by the battery, a device to charge the battery or some test equipment, for example. A method of controlling an in-rush current and a control algorithm carried out by the BMS to control the Rds of the MOSFET over the course of a connection between the battery and the load (or source) is also provided. Control of the Rds is affected by adjusting a gate voltage (Vgs) of the MOSFET. The control algorithm may advantageously be carried out on a discharge MOSFET already present in the battery system to reduce additional cost and complexity while still providing the benefits of the invention. This invention is beneficial because it provides protection from short-circuits while still permitting rapid charging and discharging of the battery. Upon first connection, the increased Rds of the MOSFET reduces the initial discharge of the battery, allowing for initial discharge to be relatively rapid while avoiding triggering short-circuit protection. No amendment to short-circuit thresholds or delay times are required, so protection against a short across the battery output terminals is still available. The reduction in the MOSFET Rds then allows full advantage to be taken of fast charging and -discharging battery technology once the in-rush current is reduced to an acceptable level. Accordingly, the proposed system may be quicker to react and more adjustable than existing solutions which rely on temperature sensing, or which may disconnect the battery and load completely in the present of high currents. The invention is set out in the appended set of claims. In a first aspect, there is provided a current control module for controlling an in-rush current resulting from the initiation of an electrical connection of a battery to an electrical load, wherein a metal-oxide-semiconductor field-effect transistor (MOSFET) is electrically connectable between the battery and the load, and an effective drain-source resistance (Rds) of the MOSFET regulates the current between the battery and the electrical load; and the control module is configured to: receive a current sense signal indicative of a current between the battery and the electrical load, compare the current sense signal to a threshold level, and adjust a control signal for the MOSFET in response to the comparison, to control the Rds of the MOSFET such that the current sense signal moves towards or below the threshold level or is maintained below the threshold level. Advantageously, by adjusting the control of the MOSFET in response to sensing the current between the battery and load / source, a more effective limiting of in-rush current can be achieved which is adaptable and quicker to adjust to the changing current compared to conventional current limiters which may comprise fixed resistances or NTC (negative temperature coefficient) devices such as thermistors. Consequently, the control module of the first aspect is particularly suited for use with low-impedance batteries. Accordingly, by implementing more effective in-rush current protection, the likelihood of damage to the battery is reduced and the lifetime of the battery can be extended. The MOSFET may be a high-side MOSFET positioned between a high-side of the load / source and the battery. A high-side MOSFET may be more preferably owing to the permanent ground connection maintained during operation. Moreover, a high-side BMS drive is more common in the market facilitating the modification of existing BMS circuits to include the control module of the present invention. However, the techniques discussed herein may also be applied using a low-side MOSFET. The electrical load may be an electrical load or source which is configured to draw current from the battery or provide current to the battery (i.e., for charging). Accordingly, when the electrical load is a current source it may be considered as a “negative electrical load”. For example, the load may be a device which is powered by the battery or a second battery (such as an EV battery) for charging. Therefore, in this example, the current may flow from the battery to the load. Alternatively, when the electrical load is a source (i.e., a negative load) it may be a device which is configured to charge the battery. Therefore, in this example, the current may flow from the source to the battery, which, in this case, may be considered as an electrical load. The load (or source) may have an input capacitance which causes an initial surge of high-in-rush current to be drawn from the battery until the input capacitance reaches a charged state. The techniques and systems for limiting in-rush current discussed herein may be applied to both examples, except where one of these cases is clearly excluded. Accordingly, the electrical source and / or load may be simply referred to as a “load” herein, wherein the load may be interpreted as a load for drawing current or as a “negative load” for providing current. Generally, the present disclosure discuses three embodiments of control module, and variants upon those embodiments for limiting an in-rush current according to the present invention. In particular, the present disclosure provides: a first embodiment wherein the control module is implemented using a microcontroller to adjust the control signal; a second embodiment wherein the control module is implemented using analogue circuitry to adjust the control signal; and a third embodiment wherein the control signal is adjusted using pulse width modulation (PWM). However, in each of these three embodiments, a feedback loop is provided wherein the current sense signal is compared to a threshold level and the control signal is updated based on the comparison. The current sense signal may be a sense voltage indicative of the current flowing between the battery and the load. In other examples, the current sense signal may be the current itself, or a digital representation of the current received by e.g., a microcontroller (MCU). The threshold level may be a voltage level for comparing to the current sense signal, for example, using analogue circuitry as discussed in more detail below. In other examples, the threshold level may be a threshold current for limiting the in-rush current to. In further examples, the threshold level may be a digital value stored in memory of an MCU. By continuously comparing the current signal to the threshold level, the control of the MOSFET may be dynamically adjusted in a feedback loop so that the in-rush current is adaptively regulated thereby providing smoother current regulation than existing solutions. The metal-oxide-semiconductor field-effect transistor (MOSFET) may be a three-terminal transistor having a gate (G), drain (D) and source (S) terminals. The MOSFET may have a drain-source resistance (Rds) which is controllable. The Rds is typically controlled using a control signal applied to the gate of the MOSFET. Any suitable MOSFET may be used. For example, the MOSFET may be an N-channel FET which is more common in high power BMS applications. However, in other examples the MOSFET may be a P-channel FET wherein pull-down resistors may be used to drive the gate of the p-channel MOSFET. However, the techniques and control circuitry discussed herein for controlling the MOSFET may be the same in each case. An electrical connection may refer to a connection permitting the flow of electrical energy between two terminals or nodes. The electrically connectability of two components refers to where two components may be electrically connected (e.g., by a conductor such as a wire, cable, or trace). The components may be releasably electrically connected. The current between the battery and the load maybe considered as a current flow through the electrical connection via the MOSFET. The current may be indicated by the current sense signal. The current sense signal may be compared to a threshold level. The comparison may be a direct comparison or an indirect comparison, where the current sense signal and / or threshold level is converted to a comparable unit. The threshold value may be selected depending on the battery and load / source. For example, the threshold value may be or correspond to a current value, such as 300A, such as 250A, 200A, 150A, 100A, or 50A. The controller may be configured to adjust a control signal for the MOSFET in response to the comparison, to control the RDs of the MOSFET such that the current sense signal moves towards or below the threshold level or is maintained below the threshold level. The control module may be configured to set the Rds of the MOSFET to an initial effective resistance when electrical connection of the battery to the electrical load is initiated. The initial effective resistance may refer to a pre-determined resistance. The initial effective resistance may be a resistance which is configured to prevent the in-rush current from exceeding a threshold value. For example, a MOSFET used in the simulation examples discussed herein (IRF2805S) has a Rds fully ON resistance of 3.6mQ and a Rds fully OFF resistance of several MQ. The initial effective resistance may therefore be a Rds anywhere within this range which is adjustable according to the current sense signal so that the current sense signal is maintained below the threshold level. In other examples, discussed herein, the initial effective resistance may correspond to an average resistance between an “on” resistance of the MOSFET and an “off” resistance of the MOSFET when the MOSFET is being switched by a PWM signal having a predetermined frequency and duty cycle. The electrical connection of the battery to the load may be initiated when current starts to flow between the battery and load / source. This may be when or in response to an electrical being formed between the battery and load / source. The control module may be configured to: set the Rds of the MOSFET to a first effective resistance when the current sense signal is above the threshold level, and decrease the Rds of the MOSFET to a second effective resistance, which is lower than the first effective resistance, when the current sense signal has reduced to lower than the threshold level. ln this context, the current sense signal being “above” the threshold level may be taken to mean that a voltage or value of the current sense signal is higher than a value of the threshold level. The first effective resistance may refer to a predetermined resistance configured to limit the inrush current. The predetermined resistance may be configured to regulate the in-rush current to below the threshold value. For example, the first effective resistance may be the initial effective resistance discussed above. The second effective resistance may refer to a predetermined resistance. The second effective resistance is less than the first effective resistance. The second effective resistance may be a resistance which is configured to regulate the current between the battery and the load / source to a current which is below the threshold value. In some examples, the second effective resistance may be configured to be a minimum resistance of RDS to thereby maximise the current flow between the load / source and the battery during normal operation (i.e., after the inrush current has settled to a steady-state). The second effective resistance may be an “on” resistance of the MOSFET. In other examples, the second effective resistance may be an intermediate resistance which is lower than the first effective resistance. Accordingly, the effective Rds may be decreased (gradually or in steps) as the in-rush current decreases. The effective RDs of the MOSFET may be controllable by controlling a gate-source voltage for driving a MOSFET. The gate-source voltage of the MOSFET may be dependent on the control signal. Accordingly, the skilled person would understand that the first effective resistance may correspond to a first voltage of the control signal and the second effective resistance may correspond to a second voltage of the control signal. In some examples, the MOSFET may be adjusted from the first Rds to the second Rds in a stepchange, upon determining that the current sense signal has reduced to lower than the threshold level. In other examples, the effective Rds may be gradually reduced as the in-rush current reduces towards a steady state. The second resistance may be an “on” resistance of the MOSFET (i.e., wherein the MOSFET is operated in its saturation region). The control signal may be configured to operate the MOSFET in its linear region when the current sense signal is higher than the threshold level. The linear region may refer to an operation region of the gate source voltage of the MOSFET wherein the MOSFET is between a fully ON state and a fully OFF state. Preferably, the linear region refers to where the MOSFET is ON and the effective Rds is substantially linearly dependent on the gate-source voltage. When the MOSFET is operated in its linear region the effective Rds of the high-side MOSFET may be (substantially linearly) dependent on the gate-source voltage thereby facilitating fine control of the current regulation which can be adjusted in response to the changing in-rush current. In other examples (discussed below) the MOSFET may be switched between an on-state and an off-state using a PWM signal to adjust the effective Rds. The control signal may be an analogue voltage which is variable between a minimum and a maximum voltage, wherein the minimum voltage corresponds to a maximum Rds of the MOSFET, and the maximum voltage corresponds to minimum a Rds of the MOSFET. Accordingly, on this example the control signal may be adjustable between a minimum and a maximum analogue voltage. Therefore, the minimum voltage may correspond to a maximum Rds of the MOSFET, and the maximum voltage may correspond to minimum a Rds of the MOSFET. When the MOSFET has the maximum Rds the MOSFET may be in an “off” state. When the MOSFET has the minimum Rds the MOSFET may be in an “on” state and operating in its saturation region. The control signal may be dependent on (and, optionally, proportional to) a difference between the current sense signal and the threshold level. For example, the control signal may be generated by amplifying a difference between the current sense signal and the threshold level (e.g., using analogue circuitry) or by subtracting the threshold level from the current sense signal (e.g., in software). In an embodiment, the current control module may comprise a microcontroller and a digital-to-analogue converter, DAC. The microcontroller may be configured to receive the current sense signal, compare the current sense signal to a threshold level, adjust a digital control signal in response to the comparison, and provide the digital control signal to the DAC. The DAC may be configured to generate the control signal based on the digital signal. Accordingly, the digital control signal may be adjusted so as to move the current sense signal towards or below the threshold level or maintain the current sense signal below the threshold level. In another embodiment, which includes analogue circuitry for controlling the in-rush current, the current sense signal may be a voltage input signal which is indicative of the current between the battery and load, and the threshold level may be a reference voltage. The control module may, comprise an error amplifier configured to receive the current sense signal and the reference voltage as inputs, and output the control signal. Accordingly, the control signal is this example may be dependent on the difference between the voltage input signal and the reference voltage. An inverting input of the error amplifier may be connected to the reference voltage and a noninverting input of the error amplifier may be connected to the current sense signal. In this context connected to may be considered to mean that an electrical connection between the respective terminal of the error amplifier may be formed between a rail or conductor carrying the reference voltage or the current sense signal. The resulting control signal, which is output by the error amplifier, may be connected to a gate-drive circuit for controlling the MOSFET. For example, when the MOSFET is a high-side MOSFET a gate drive circuit may be provided to increase the voltage of the control signal to a gate voltage which is higher than the battery voltage so that the gate-source voltage of the MOSFET is sufficiently high enough to switch the MOSFET. ln another embodiment, the control signal may be a PWM signal, and the control module may be configured to adjust a duty cycle of the PWM signal based on the comparison between the current sense signal and the threshold level. Accordingly, the MOSFET may be configured to be switched between an “on” state and “off” state at a rate determined by the duty cycle of the PWM signal such that the effective Rds of the MOSFET, and hence the output current of the battery, is dependent on the duty cycle of the PWM signal. In this example, the effective Rds of the MOSFET may be defined as an average Rds of the MOSFET over time as determined by the “off” and “on” times of the MOSFET. The control module may comprise a signal generator for generating the PWM signal. Alternatively, the control module may comprise a microcontroller for generating the PWM signal. The PWM signal may be based on the comparison of the current sense signal to the threshold value. Accordingly, the PWM signal may be based on the current sense signal. In some examples, if the current sense signal is above the threshold level, the control module may be configured to generate the PWM signal at a first duty cycle. Accordingly, when the PWM signal is operated at a first duty cycle, the MOSFET may have a corresponding first effective Rds. The duty cycle of the PWM signal may be fixed for all values of the current sense signal above the threshold level so that a fixed amount of current regulation is applied to the in-rush current until it decreases to a steady-state. Alternatively, the control module may be configured to drive the PWM signal at a variable duty cycle which is dependent on a difference between the current sense signal and the threshold level. The control module may, therefore, be configured to gradually adjust the duty cycle of the PWM signal based on the current sense signal. When the current sense signal is above the threshold level, a switching frequency of the PWM signal (e.g., at the first duty cycle) may be configured to be higher than a rate parameter of the MOSFET, such that the MOSFET is operated in its linear region. Accordingly, when there is a high current detected between the battery and the load, the MOSFET is operated in its linear region thereby presenting a constant, effective Rds in series with the battery and thereby limiting the current. The rate parameter may be a slew rate of the MOSFET defined by a transition time of the MOSFET between its on-state and off-state. If the current sense signal is below the threshold level (or decreases to below the threshold level), the control module may be configured to adjust the duty cycle of the PWM signal to 100% such that the MOSFET is in an “on” state. Accordingly, when the in-rush current reduces to a steady-state, the PWM control of the MOSFET, and hence control of the in-rush current, may be ceased. When the MOSFET is in the “on” state it may be operating in its saturation region and therefore have a lower “on” Rds. The switch from the first duty cycle to the 100% duty cycle may be performed as a step-change to the PWM control signal. Accordingly, the MOSFET in this example, the MOSFET may have a fixed effective Rds which is dependent on the first duty cycle for limiting the in-rush current until the current decreases sufficiently and the MOSFET is turned “on” long-term. In other examples discussed herein, the control module may be configured to gradually increase the duty cycle of the PWM control signal as the in-rush current decreases, such that the MOSFET is operated in the “on-state” for increasingly longer, thereby gradually decreasing the effective Rds of the MOSFET until the current decreases below a threshold and the duty cycle of the PWM control signal reaches 100%. In further examples, if the current sense signal is higher than the threshold level and lower than a second threshold level, the control module may be configured to vary the duty cycle of the PWM signal dependent on a difference between the current sense signal and the threshold level. Accordingly, as the in-rush current decreases to below the second threshold, but is still higher than the threshold level, the duty cycle of the PWM signal may begin to be gradually increased, thereby gradually decreasing the effective Rds of the MOSFET until the current has reached a steady state (wherein the current sense signal is lower than the threshold level). In a further aspect there is provided a battery management system for controlling an in-rush current resulting from the initiation of an electrical connection of a battery to an electrical load, wherein the battery management system comprises: a MOSFET, wherein the MOSFET is electrically connectable between the battery and the load, and an effective drain-source resistance (Rds) of the MOSFET regulates the current between the battery and the load, a gatedrive circuit for controlling the Rds of the MOSFET in response to a control signal, and a current control module according to the first aspect for providing the control signal. The current control module of the first aspect is configured to provide the control signal for controlling a drainsource resistance, Rds, of the MOSFET. The gate-drive circuit may comprise a gate-drive transistor connected between a gate of the MOSFET and electrical ground. The control signal may be provided to a gate of the gate-drive transistor such that a gate-source voltage, and thereby Rds of the gate-drive transistor is adjustable by the control signal such that an intermediate control voltage provided to the gate of the MOSFET is dependent on the control signal. The intermediate control voltage may be a gate-source volage of the MOSFET. In other examples, the intermediate control voltage may be stepped-up by a gate driver to a gate-source voltage for controlling the MOSFET. As mentioned above, the MOSFET may be a low-side MOSFET or a high-side MOSFET. When the MOSFET is a high-side MOSFET, the gate-drive circuit may comprise a high-side driver for stepping up the intermediate control voltage to a gate-source voltage for driving the high-side MOSFET. The gate-source voltage may be higher than a supply voltage of the battery when the MOSFET is in an “on” state. Accordingly, when the source voltage is near to the battery voltage, owing to the MOSFET being located on a high-side of the load (or battery), the gate-source voltage may be stepped up to exceed a switching threshold of the MOSFET. For example, the high-side driver may comprise a charge-pump for stepping up the intermediate control voltage to the gate-source voltage. The battery management system may further comprise a current sense circuit for detecting the current between the battery and load, generating the current sense signal, and providing the current sense signal to the current control module. The current sense circuit may be electrically connectable between a low-side of the load (or source) and a negative terminal of the battery. For example, the current sense circuit may comprise a sense resistor connectable in series with the load and a current sense amplifier configured to amplify a voltage drop across the sense resistor. The battery (for connecting to the load / source via the BMS) may have an internal resistance of 10mQ or less, 5mD or less, or 1mQ or less. This low internal resistance can result in particularly high in-rush currents being drawn when the battery is connected to a load. This can cause conventional protection circuits to break the connection between battery and load entirely or cause damage to the battery and / or load because conventional protections circuits may be too slow to react to the high current. Accordingly, the feedback loop provided in the control module of the present invention enables the provision of more adaptable and quicker current protection which can increase or decrease the amount of protection provided accordingly. The threshold level may depend on the battery, or the load (or source). For example, the threshold level may depend on the battery size and / or the battery internal resistance. The threshold level, for comparing the current sense signal to, may correspond to a current from 50A to 300A, more preferably from 100A to 250A, yet more preferably from 150A to 200A. The threshold level may correspond to a current of 50A or more, preferably 100A or more, yet more preferably 150A or more, even more preferably 200A or more. The threshold level may correspond to a current of about 50A, preferably about 100A, yet more preferably about 150A, even more preferably about 200A. The threshold level may correspond to an in-rush current between the battery and the load, which is from 100A to 300A higher than an expected steady-state current between the battery and the load, more preferably 250A higher that the expected steady-state. Therefore, the BMS of the present invention is particularly suited to high current applications such as EV (electric vehicle) charging. The expected steady-state current between the battery and the load is typically the current between the battery and the load when the MOSFET is ON and the current is constant. Steadystate current may refer to a period after the in-rush current has subsided, and may be taken as the average current flow for 5 second or more after connection of the battery and load is initiated, preferably 10 second or more, more preferably 30 second or more. The expected current flow may be predicted based off previous connections, may be provided by a look-up table or pre-set by a user. As mentioned above, the load may be a current source for providing power to the battery, i.e., to charge the battery. In this example, the MOSFET may be a charge MOSFET which is part of the source, wherein, in this example, the source may be a charging device. Alternatively, the load may be a device for receiving power from the battery thereby discharging the battery In a second aspect there is provided a method of controlling an in-rush current resulting from the initiation of an electrical connection of a battery to an electrical load, wherein a MOSFET is electrically connectable between the battery and the load, and an effective drain-source resistance (Rds) of the MOSFET regulates the current between the battery and the load; the method comprising the steps of: receiving a current sense signal indicative of a current between the battery and the load, comparing the current sense signal to a threshold level, and adjusting a control signal for the MOSFET in response to the comparison, to control the Rds of the MOSFET such that the current sense signal moves towards or below the threshold level or is maintained below the threshold level. For example, the method of the present aspect may be performed by a microcontroller provided in a BMS to regulate the current. The microcontroller may be an existing microcontroller provided in a battery pack which is adapted to perform the method of the present aspect to regulate. As discussed above for the first aspect, the electrical load may refer to a load or source wherein the load may be a positive or negative load and therefore may source or draw current to / from the battery. The method may further comprise a step of: connecting the battery to the load / source via the MOSFET. The method may further comprise a step of repeating the receiving, comparing, and adjusting steps until the current sense signal is below the threshold level (and the output current has reached a steady state). Accordingly, the method may be repeated as part of a continuous feedback loop for regulating the in-rush current. The adjusting step may include adjusting the control signal to set the effective Rds of the MOSFET to an initial resistance when the electrical connection of the battery and the electrical load is initiated. The adjusting step may include, adjusting the control signal to set the Rds of the MOSFET to a first effective resistance when the current sense signal is above the threshold level, and adjusting the control signal to set the RDS of the MOSFET to a second resistance, which is lower than the first resistance, when the current sense signal has reduced to lower than the threshold level. In the adjusting step, the control signal may be adjusted to: set the effective Rds of the MOSFET to a first resistance when the battery is first connected to the load, and decrease the effective Rds of the MOSFET to a second resistance, which is lower than the first resistance, when the current sense signal has reduced to lower than the threshold level. Accordingly, in this example, the effective resistance provided by the MOSFET may be decreased in a step change after the in-rush current decreases below a threshold. For example, as described above in relation to driving the MOSFET with a PWM control signal, a duty cycle of the PWM signal may be adjusted between a predetermined duty cycle which is between 0% and 100% for regulating the current between the battery and the load and a duty cycle of 100% wherein the MOSFET is in an “on-state” and is not being used to regulate the current. In other examples, the control signal may be adjusted to gradually adjust the effective Rds of the MOSFET in response to the current sense signal. In a further aspect, there is provided a computer-readable medium comprising instructions that, when executed by a processor, cause the processor to carry out the methods described herein. In other words, the methods disclosed herein may be implemented by a computer or a processor to control the in-rush current. As such, the present invention encompasses computer-readable media, and computer program products that comprise logic and / or instructions that, when executed by the processor of a computer, cause said computer to implement the methods disclosed herein. In a further aspect of the present invention there is provided a battery comprising a battery management system according to the second aspect. The battery may comprise a plurality of electrochemical cells. The battery management system may be in communication with the electrochemical cells. The battery (e.g., electrochemical cells) may have an internal resistance of 1mQ or less. The electrochemical cells may each have an internal resistance of 1mQ or less. In a further aspect of the present invention there is provided a use of the current control module of the first aspect or the battery management system of the second aspect, for controlling an inrush current resulting from the initiation of an electrical connection of a battery to a load (or source). The invention includes the combination of the aspects and preferred features described except where such a combination is clearly impermissible or expressly avoided. Summary of the Figures Embodiments and experiments illustrating the principles of the invention will now be discussed with reference to the accompanying figures in which: Figure 1 shows a block diagram of a battery-powered system according to the current art; Figure 2 shows a block diagram of a battery-powered system according to aspects of the present invention; Figure 3 shows a graph of MOSFET drain-source resistance (ds) against MOSFET gate voltage (Vgs); Figure 4 shows a simplified process for controlling in-rush current according to aspects of the present invention; Figure 5 shows a graph of in in-rush current draw from the battery over time alongside Rds and Vgs of the MOSFET. Figures 6-10 show example battery management systems for controlling in-rush current according to aspects of the present invention; Figure 11 shows another example battery management system for controlling in-rush current according to aspects of the present invention; Figure 12 shows a process for controlling in-rush current using PWM; Figure 13 shows example simulation results for an in-rush current which is not limited using the system of the present invention; Figure 14 shows a simulation circuit for controlling in-rush current according to aspects of the present invention; Figures 15 shows simulation results for the simulation circuit of Figure 14; Figure 16 shows more simulation results for an in-rush current which is not limited using the system of the present invention; Figure 17 shows another simulation circuit for controlling in-rush current according to aspects of the present invention; and Figures 18 shows simulation results for the simulation circuit of Figure 17. Detailed Description of the Invention Aspects and embodiments of the present invention will now be discussed with reference to the accompanying figures. Further aspects and embodiments will be apparent to those skilled in the art. All documents mentioned in this text are incorporated herein by reference. In this description, the term “battery” may refer to any number of battery cells arranged in one or more packs or modules and a battery may be made up of one or more sub-batteries connected to provide power as appropriate. The battery may further be a modular system such that subbatteries may be removed, replaced, and added as required to maintain and increase the functionality of the battery. The cells comprised in the battery preferably include a negative active material comprising a metal oxide. The metal oxide comprising the negative active material is preferably a niobium-based material, such as niobium oxide, or a niobium metal oxide such as niobium nickel oxide, niobium tungsten oxide, niobium titanium oxide, niobium molybdenum oxide, niobium aluminium oxide, niobium gallium oxide, niobium germanium oxide, niobium copper oxide, or niobium zinc oxide for example, as described in WO 2019 / 234248, the content of which is incorporated herein by reference in its entirety. Accordingly, the negative active material may comprise Nb2O5, Nb2NiO6, Nb12WO33, Nb26W4O77, Nb14W3O44, Nb16W5O55, Nb18W8O69, Nb2WO8, Nb18W16O93, Nb22W20O115, Nb8W9O47, Nb54W82O381, Nb20W31O143, Nb4W7O31, Nb2W15O50, Nb2WO8, Nb2TiO7, Nb10Ti2O29, Nb24TiO62, Nb2Mo3O14, Nb14Mo3O44, Nb12MoO44, Nb11AIO29, Nb11GaO29 Nb49GaO124, Nb18GeO47, Nb34Cu2O87, or Nb34Zn2O8. This material has favourable lithium ion diffusion properties and thus exhibits superior performance even where micron-sized particles of the niobium-based material are used. Accordingly, a negative electrode comprising a niobium-based material, such as niobium oxide or a niobium metal oxide, exhibits extremely high volumetric energy density and high capacity at high rates of charging and discharging. Alternatively, the metal oxide comprising the negative active material could comprise another metal oxide that displays similar electrochemical properties such as lithium titanium oxide, titanium dioxide, silicon oxide, or vanadium oxide. The metal oxide could be combined with other suitable active materials such as carbon, graphite, and other metal oxides. Further alternatively, the cells comprised in the battery may include other negative active materials which do not incorporate metal oxides, such as carbon and graphite. In addition to the negative electrode, an electrochemical cell comprises a positive electrode, an electrolyte, and a separator, such as a microporous polyethylene film, between the negative electrode and positive electrode. Suitable materials for the positive electrode include lithium-containing or lithium-intercalated material, such as a lithium metal oxide, wherein the metal may be a transition metal such as Co, Fe, Ni, V, or Mn, or combination thereof. Some examples of positive electrode materials include lithium cobalt oxide (LiCoO2) lithium nickel manganese cobalt oxide (NMC, LiNiMnCoO2, e.g., LiNiO.6CoO.2MnO.202), lithium vanadium fluorophosphate (LiVPO4F), lithium nickel cobalt aluminium oxide (NCA, LiNiCoAI2), lithium iron phosphate (LFP, LiFePO4) and manganese-based spinels (e.g. LiMn2O4). In one embodiment, the positive electrode is substantially free of binders. In an alternative embodiment, the positive electrode is admixed with a binder or adhesive. Some examples of binders or adhesives include PVDF, PTFE, CMC, PAA, PMMA, PEO, SBR and co-polymers thereof. The positive electrode may be fixed to a current collecting substrate, such as an aluminium plate. The electrolyte comprises lithium salts, such as lithium (bis(trifluoromethane)sulfonimide (LiTFSI), LiPF6, LiBF4, LiCIO4, lithium triflate (LiTF), or lithium bis(oxalate)borate (LiBOB). The electrolyte may be a liquid electrolyte, such as a liquid at ambient temperature, for example at 25°C. The electrolyte may be a non-aqueous electrolyte. The electrolyte may comprise a polar aprotic solvent, such a cyclic or linear carbonate, such as ethylene carbonate, dimethyl carbonate, or ethyl methyl carbonate. Figure 1 shows a simplified block diagram of a battery-powered system according to the current art comprising a battery pack 11 connected to a battery-powered device 12. The battery pack 11 and the battery-powered device 12 may be co-located in a single casing so they appear to be a single device or may be separate devices connected via an appropriate power transfer mechanism and protocol. Accordingly, the battery-powered device 12 may be an industrial device or tool such as an industrial robot, drill, etc.; a household device such as a vacuum cleaner, food processor, etc.; or a personal device such as a mobile telephone, toothbrush, etc.; or a larger device such as a battery powered vehicle or other mobile device. The battery pack 11 comprises a battery 13 connected to a battery management system (BMS) 14 which carries out balancing of the cells that make up the battery 13 based on detected voltage, current, and temperature data and may further include a protection circuit containing fuses, which acts as a buffer to protect the battery 13 from possible spikes during rapid charging and discharging. The battery-powered device 12 comprises the load 16, the nature of which depends on the particular embodiment. For example, it may be a motor to drive wheels for self-propulsion or it may be computer processors in a device such as a personal computer. The load 16 is connected to a capacitor 15. Such capacitors are commonly used in electronic circuits to minimise voltage ripple caused by switching in the power supply connection and, accordingly, the capacitor 15 is connected to the battery 13 and the BMS 14 via such a power supply connection between the battery pack 11 and the battery-powered device 12. When the battery pack 11 is first connected to the battery-powered device 12, the capacitor 15 is usually fully discharged. Upon connection, an initial high current known as “in-rush current” flows from the battery 13 into the capacitor 15. The magnitude and duration of the in-rush current depends on the input capacitance and output impedance of the input source. Increased in-rush current levels and duration may have adverse impact on the battery safety as well as the cycle life. Moreover, it may cause thermal or overcurrent stress on the input stage components of the device 12 if they are designed for a lower in-rush current. A battery 13 with fast-charging and discharging capability has low internal resistance resulting in high in-rush current for a longer duration. As a result, it may trigger the short-circuit and overcurrent protection mechanisms in the BMS 14, causing the BMS 14 to stop current flow and therefore stop the battery 13 from providing power to the load 16. There are several conventional methods to limit in-rush current, but they are not suitable for high-power and high-current circuits involving low-impedance battery technology. As described above, some conventional systems use negative temperature coefficient (NTC) devices such as thermistors, which change their internal resistance with ambient temperature such that as temperature rises the resistance increases. This is effective because when a high current flows through the NTC its temperature is likely to increase. However, this solution is unsuitable for high-power and high-current circuits due to significant power loss and potential overheating of the NTC during operation. Other conventional systems use a separate pre-charging circuit or soft-start mechanism to limit the in-rush current. However, these mechanisms require new control schemes, power resistors and additional circuits in the recipient battery-powered device 12. Finally, a BMS 14 can be modified to increase appropriate thresholds to accommodate increased in-rush current levels. However, this reduces the effectiveness of the short-circuit protection, leading to increased transient thermal stress on the battery, reduced cycle life, and potential damage to the battery, possibly including catastrophic damage such as fire or explosion. Figure 2 shows a simplified block diagram of a system according to the invention. Similar to the system described in Figure 1, a battery pack 21 is connected to a battery-powered device 22, the battery-powered device 22 comprising a load 27 and a capacitor 26 which acts as a filter on the incoming power supply. The battery pack 21 comprises a battery 23 connected to a BMS 24, which is in turn connected to an output MOSFET 25. The MOSFET 25 is here shown as a separate device for clarity but may be incorporated into the BMS 24. It may also be one of the MOSFETs currently present in many BMSs. The BMS 24 includes a controller 28, which receives information on the battery 23 such as its temperature, state of charge, and the output current. It is connected to a gate drive 29 within the BMS 24, which is further connected to the MOSFET 25 to control its Rds. This control is carried out by the gate drive 29 applying a variable gate voltage (Vgs) to the MOSFET 25 such that when the Vgs applied increases the RDs decreases and vice versa. The controller is described in more detail below in reference to Figure 6. Figure 3 shows the relationship between Rds (on the Y axis) and Vgs (on the X axis) during discharge at two different temperatures. The light-coloured curve 31 shows the change in Rds at different Vgs levels at 125°C and the dark-coloured curve 32 shows the change in Rds at different Vgs levels at 25°C. In both cases, when the Vgs rises above 4V the Rds falls significantly, which allows for a higher current to pass through the MOSFET. Accordingly, during the initial in-rush current period the Vgs could be kept low and the battery-powered device 22 will be protected from potential damage from high in-rush current regardless of temperature, which is an improvement over conventional NTC-based systems as described above. Figure 4 shows an example process followed by the controller 28. At Step S41, the battery pack 21 is connected to the battery-powered device 22 to supply power to the load 27. At Step S42, two things occur almost simultaneously. At Step S42A, power is transferred from the battery pack 21 to the battery-powered device 22 and the capacitor 26 begins to charge. At Step S42B, the controller 28 in the BMS 24 detects the level of in-rush current and signals the gate drive 29, which lowers the Vgs applied to the MOSFET 25. This results in an increase in the Rds of the MOSFET 25, as shown in Figure 3. At Step S43, the controller 28 determines whether the in-rush current is higher than the desired threshold. If so, the process follows the branch to the right, beginning at “Yes”, and returns to Step S42B where the gate drive 29 maintains the low Vgs and accordingly the high Rds of the MOSFET 25. If the in-rush current is lower than the desired threshold than the process follows the branch to the left, beginning at “No”, to Step S45. This corresponds to the second timepoint 53 in Figure 5 as described below. At Step S45, the controller 28 signals the gate drive 29, which raises the Vgs applied to the MOSFET 25, as shown at point 55 of Figure 5. As a result, at Step S46, the MOSFET’s 25 Rds decreases. The process ends here, but it may continue to be applied at any point during the connection between the battery and the load to regulate current, for example during power surges. Since at this point the initial in-rush current has passed, the Vgs and, accordingly, the Rds, can remain steady and power supply continues as appropriate with minimal power dissipation, possibly at a higher rate than is common in conventional systems depending on the nature of the battery 23. Unlike conventional methods of solving the problems of in-rush current, there is no need to reduce short-circuit protection or necessarily add additional components, especially resistors that result in excessive heating and power dissipation in normal use. This process is further illustrated in Figure 5, which shows the correspondence of RDs, the inrush current, and Vgs. Rds is indicated by the curve with a dashed line 56, Vgs is indicated by the curve with a dotted line 58 and the in-rush current is indicated by the line with alternating dots and dashes 57. Although the three curves use different Y-axis units, they are shown on the same graph to clarify their behaviour at different timepoints. The corresponding timepoints and time periods are indicated with arrows and brackets. For example, the first arrow 51 indicates the point at which the connection between the battery and the load device is formed, corresponding to Step S41 of Figure 4. The first bracket 52 indicates the time during which the peak in-rush current is charging the capacitor 26. During this time, as previously described with reference to Step 42 of Figure 4, Vgs is held low resulting in a high Rds of the MOSFET. At time point 53, the in-rush current decreases to the desired threshold. The second bracket 54 indicates a transition region during which the peak in-rush current has reduced to a steady-state. During this time, as previously described with reference to Step 45 of Figure 4, Vgs is increased resulting in a decreased Rds of the MOSFET. The third bracket 55 indicates a steady-state region wherein Vgs and, accordingly, Rds is held in a steady “on” state for transferring power between the battery and load device during typical operation. Figure 6 shows an example battery management system (BMS) 100 for connecting a battery 120 to a load 122 for regulating in-rush current between the battery 120 and the load 122. The load 122 has an associated input capacitance C1 and the BMS includes a controller 128 (which may also be referred to as a control module), a gate driver 112, a high-side MOSFET 110 and current sense circuitry 130, 132, 136. The battery 120 is electrically connected to one side of the MOSFET 110 (i.e., the drain), and the load 122 is electrically connected to the other side of the MOSFET 110 (i.e., the source). The gate driver 112 is configured to operate the gate of the MOSFET 110 according to a control signal provided by the controller 128. The controller 128 is configured to receive a current sense signal from the current sense circuitry 130, 132, 136 and adjust the control signal to the gate driver 122 based on a comparison between the current sense signal and a threshold level. The gate driver 112 is configured to apply a variable gate voltage (Vgs) to the MOSFET 110, based on the control signal from the controller 128. For example, the gate driver 112 may comprise a charge-pump for driving the high-side MOSFET 110 by stepping up the control signal to a gate voltage (Vgs) which is above the battery voltage. The control signal is configured to operate the MOSFET 110 in its linear region so that when the Vgs is increased, the Rds of the MOSFET 110 decreases and vice-versa. In Figure 6, the current sense circuitry 130, 132, 136 comprises a sense resistor 136 connected in series with the load 122 and battery 120, a current sense amplifier 130 and an ADC 132. The current between the load 122 and the battery 120 therefore flows through the sense resistor 136 (which has a low value resistance, e.g., 1mO). The resulting potential difference across the sense resistor 136 is amplified by the sense amplifier 130 to provide a current sense signal, which in this case is a voltage input signal to the ADC 132. The ADC 132 converts the input voltage to a digital signal for processing by the controller 128 to generate the MOSFET control signal. In some examples, the ADC 132 may be included in the controller 128 (e.g., in a microcontroller package) or may not be included at all (for example when the voltage input is compared to the threshold level using analogue components as discussed below in reference to Figures 8 and 9). In this way, the controller 128 can sample the current sense signal and adjust the MOSFET control signal to regulate the in-rush current in a continuous feedback loop. The controller 128 may be configured to generate and adjust the MOSFET control signal in a variety of ways including using a processor, such as a microcontroller (e.g., as discussed below in relation to Figures 7-8), or using analogue circuitry (e.g., as discussed below in relation to Figures 9-10), or by adjusting a duty cycle of a PWM control signal (e.g., as discussed below in relation to Figures 11-13). Figures 7 and 8 show embodiments of a system 200 for connecting a battery 220 to a capacitive load 222 wherein the controller comprises a microcontroller (MCU) 228. In these examples, the MCU 228 is configured to provide a digital control signal to a DAC 225 which generates an analogue control signal for controlling the MOSFET 210 via a gate driver. In Figures 7 and 8, the gate driver comprises a drive transistor 234 connected between the gate of the high-side MOSFET 210 and electrical ground. The gate of the drive transistor 234 is under the control of the analogue control signal. Accordingly, as the control signal adjusts a gate voltage of the drive transistor 234, to adjust a resistance of the drive transistor 234 and therefore modifying Vgs of the high-side MOSFET 210. In Figure 7 the gate driver comprises a power supply 211 for stepping up Vgs to a voltage which is higher than the battery voltage in order to operate the high-side MOSFET 210. Additionally, the current sense circuitry of Fig 7 comprises a sense resistor 236 and sense amplifier 230 as described above. The resulting sense voltage is provided to the MCU 228 for adjusting the control signal in a continuous feedback loop. In Figure 8 the gate driver comprises a charge-pump for stepping up the Vgs of the MOSFET 210. In this example, the charge-pump is part of an existing BMS 212 which is already provided with the battery 220 and is adapted to regulate the current using the techniques discussed herein. Additionally, in this example, current sense circuitry for measuring a voltage drop over the sense resistor 236 is also part of the existing BMS 212. Figure 9 shows a battery management system 300 wherein the control signal for controlling the MOSFET 310 is generated using analogue components. The BMS 300 is configured to connect a battery pack 320 (via terminals B+ and B-) to a load 322 (via terminals P+ and P-). As described above for Figure 6, current sense circuitry is provided comprising a sense resistor 323, and a sense amplifier 330 for generating a current sense signal which is indicative of a current flowing between the battery 320 and the load 322. The current sense signal is provided to the non-inverting input of an error amplifier 332 (i.e., an operational amplifier (op-amp). A voltage reference signal (Vref) is provided to the inverting input of the error amplifier 332 as a threshold level for comparing to the current sense signal. For example, Vref may be generated using any suitable method for providing a voltage reference such as a voltage reference IC, a zener diode, a resistor divider, a voltage regulator, etc. The output of the error amplifier 332 is provided as a control signal to drive the gate of the high-side MOSFET 310. As described above in relation to Figures 7 and 8, the control signal is used to operate a drive transistor 334 which, in-turn, is configured to control the gate of the high-side MOSFET 310. Similarly, a charge-pump 312 is provided to step up the Vgs of the high-side MOSFET 310. As a general rule, an op-amp in a closed loop such as the error amplifier 332 of Figure 9, will adjust its output to reduce a voltage difference between its inputs to zero. Accordingly, the error amplifier 332 of Figure 9 will act to equalize the current sense signal (i.e., Vsense) to the threshold level (i.e., Vref) by adjusting its output (the control signal) and therefore Rds of the high-side MOSFET 310 in order to regulate the in-rush current until the current sense signal is equal to the threshold level. Accordingly, as the in-rush current decreases over time to a steadystate, the error amplifier 332 will adjust the control signal accordingly to decrease Rds of the high-side MOSFET 310 until the MOSFET 310 is operated in a fully “on” state. Figure 10 shows another embodiment of a battery management system 300 comprising analogue components for controlling a high-side MOSFET 310. This embodiment is very similar to the embodiment of Figure 9 wherein like components have like reference numbers. However, in this embodiment, the battery management system 300 is for connecting the battery 320 to a battery charger 360. Accordingly, the battery charger 360 is a current source and the in-rush current surge, observed when the charger 360 is first connected to the battery 320, flows in the opposite direction to the system in Figure 9. In this example, the source of the MOSFET 310 is connectable to the battery 320, which is the low-side of the MOSFET 310 in this example, and the drain of the MOSFET 310 is connectable to the battery charger 360. Accordingly, in this example, the MOSFET 310 may be viewed as a charge MOSFET 310 (rather than a discharge MOSFET as in Fig. 9). Figure 11 shows another battery management system 400 wherein the in-rush current is mitigated using a method called pulse charging, which involves the use of a Pulse Width Modulation (PWM) gate driver 432. Like components to the embodiments of Figures 9 and 10 have like reference numerals. Broadly, the PWM control signal can be used to reduce in-rush current by increasing the voltage rise time on the load capacitance and slowing down the rate at which the load capacitance charges. This can be achieved by controlling the duty cycle and / or frequency of the PWM signal, effectively controlling the average power delivered to the load capacitors. In Figure 11 the BMS 500 comprises a current sense amplifier 430, a sense resistor 436, a gate drive transistor 434, a charge-pump 412, and a high-side MOSFET 410, each of which operate as described above for the embodiments of Figures 6 to 10. However, in this embodiment, the BMS 400 comprises a PWM driver 432 for providing a PWM control signal to the gate drive transistor 534. The PWM driver 432 is configured to generate the PWM control signal for a period of time upon detecting an in-rush current, until the load capacitance is charged, and the in-rush current has decreased to a steady-state. The current sense circuitry detects the level of in-rush current as voltage drop on the low value sense resistor 436 and the current sensing amplifier 430 amplifies the voltage across the sense resistor 436 to be detected by the PWM driver 432. The PWM driver 432 may be a Microcontroller (MCU) or a controllable PWM signal generator. As soon as an in-rush current is detected, the PWM driver 432 is configured to generate a high frequency PWM control signal with fixed switching frequency (and a fixed or variable duty cycle) until the in-rush current drops down to a certain point (i.e., when the current sense signal has decreased to below a threshold level) and the load capacitance is charged to a safe voltage level. For example, the switching frequency of the PWM signal may be between 100kHz and 500kHz. In the simulation examples discussed herein in relation to Figs. 17 and 18, the switching frequency of the PWM control signal was set to 250kHz. However, the particular switching frequency may be varied with the particular model of MOSFET and the inrush current level. In some examples, a fixed duty cycle may be used for the PWM control signal. However, in other examples, a varying duty cycle may be used. For example, the duty cycle of the PWM control signal may be increased over time so that the MOSFET is held “ON” for longer as the inrush current reduces, thereby reducing the effective (average) Rds of the MOSFET. After the in-rush current has reduced to below the threshold level (which may, for example, be a predetermined value stored on the PWM driver 432) the control signal provided to the gate drive transistor 534 is switched from a PWM control signal to a fully active mode signal (i.e., a control signal having a 100% duty cycle which is not pulse width modulated) such that the high-side MOSFET 410 is held in an “on-state” and the load 422 is powered continuously. The switching frequency and duty cycle of the PWM control signal can be configured based on parameters of the high-side MOSFET 410 (e.g., an on-off transition time or slew rate) so that the high-side MOSFET 410 is operated in its lineal region at a relatively high Rds during the current surges. In particular, the switching frequency of the PWM control signal may be configured to be faster than a slew rate of the MOSFET 410. Figure 12 shows a process for controlling the in-rush current using the system of Figure 11. For example, the process may be followed by a control module comprising the PWM driver 532. First, at step S501, the battery pack 420 is connected to the load 422 via the BMS 400. At Step S402, two things occur almost simultaneously. Firstly, at Step S402A, power is transferred from the battery pack 420 to the load 422 and begins to charge the load capacitance. Secondly, at Step S502B, the PWM driver 432 detects a high surge of current, i.e., the in-rush current, and provides a PWM control signal with a fixed frequency and duty cycle to the gate drive transistor 434. This causes the high-side MOSFET 410 to switch on and off at a rate determined by the PWM, which in-turn causes the load capacitance to charge more slowly (i.e., pulse charging) and results in an effective increase in the Rds of the high-side MOSFET 410. At Step S503, the voltage across the load capacitance begins to increase as the capacitor charges, therefore causing the in-rush current to decrease. At Step S504, the PWM driver 432 determines whether the in-rush current is higher than a desired threshold. If so, the process follows the branch to the right, beginning at “Yes”, and returns to Step S502B where the gate driver 432 maintains the pulse width modulation of the control signal and the effective high Rds of the MOSFET. However, if the current is lower than the desired threshold in Step S504, then the process follows the branch to the left, beginning at “No”, to Steps S505 and S506 where the PWM driver ceases the pulse width modulation of the control signal. At Step S506, the control signal is configured to operate the high-side MOSFET 410 in an “on-state”. The process ends here, but it may continue to be applied at any point during the connection between the battery 420 and the load 422 to regulate current, for example during power surges. Accordingly, in Figure 12 the PWM driver 428 is configured to switch the control signal between a fixed PWM control signal and a constant, on-state control signal depending on if the in-rush current has reduced below a threshold level. Therefore, in this example, a fixed level of regulation is applied to the in-rush current. However, in other examples, the PWM driver 428 may be configured to adjust the duty cycle of the PWM control signal depending on a difference between the current sense signal and the threshold level. For example, as the in-rush current decreases, the duty cycle of the PWM control signal may be increased so that the MOSFET 410 is held on for longer periods of time until the current has reached a steady-state. Simulations Figure 13 shows example simulation results for an in-rush current resulting from the initiation of an electrical connection of a battery to a load. In this simulation, the in-rush current is not limited using the system of the present invention, (a) shows the in-rush current flowing through a sense resistor (i.e., R2 in Figure 13) against time, (b) shows Vds (Drain-Source voltage) of the MOSFET, (c) shows the output voltage (i.e., which is charging the capacitive load), and (d) is the power draw from the battery (i.e., P = l*V). In this simulation, the capacitive load is connected to the battery at 10ms. Accordingly, a high current surge (0.6KA) and a high power surge (1.5KW) are observed upon initialisation of the connection between the battery and load at 10ms. This can be damaging to the load and / or battery. Figure 14 shows a simulation circuit for controlling in-rush current using analogue circuitry as described above in relation to Figure 9. Figure 15 shows simulation results for the simulation circuit of Figure 14 wherein the graphs (a) to (d) show the in-rush current, Vds, output voltage, and power draw from the battery as discussed above for Figure 13. As shown in Figure 15, the current and power draw from the battery is now much lower and smoother than in Figure 13. For example, in Figure 15 the current (see (a)) peaks at just 250A, and the power (see graph (d)) peaks at just 4KW. Figure 16 shows more simulation results for an in-rush current (generated using the simulation circuit of Figure 17) wherein the current is not limited using the system of the present invention (i.e., when the MOSFET M1 is held “on”). The graphs (a) to (d) show the in-rush current, Vds, output voltage, and power draw from the battery as discussed above for Figure 13 wherein the capacitive load and the battery are connected 10ms into the simulation. As shown in (c), the output voltage of the battery pack is 20V when the battery is connected to the load. As shown in (a), the current between the battery and the load surges, in this example, to ~1000A upon connecting the battery to the load, and as shown in (d) the power draw from the battery peaks at ~2kW. Figure 17 shows a simulation circuit for controlling in-rush current using a PWM control signal as described above in reference to Figures 11 and 12. In this simulation, the switching frequency of the PWM control signal was 250kHz. The duty cycle of the PWM signal initialised at 1% and was increased over time to 100%, wherein the PWM signal did not switch at all. Figures 18 shows simulation results from the simulation circuit of Figure 17. As shown in Figure 18, the current (a), drain-source voltage (b), output voltage (c) and the power (d) drawn from the battery each pulse in time with the pulse width modulated signal. Accordingly, an average of the current and power drawn from the battery is much reduced. After ~430ps the current has reduced to below a threshold level and the MOSFET is no longer switched using PWM. Other Preferences Each and every compatible combination of the embodiments described above is explicitly disclosed herein, as if each and every combination was individually and explicitly recited. Various further aspects and embodiments of the present invention will be apparent to those skilled in the art in view of the present disclosure. “and / or” where used herein is to be taken as specific disclosure of each of the two specified features or components with or without the other. For example “A and / or B” is to be taken as specific disclosure of each of (i) A, (ii) B and (iii) A and B, just as if each is set out individually herein. 5 Unless context dictates otherwise, the descriptions and definitions of the features set out above are not limited to any particular aspect or embodiment of the invention and apply equally to all aspects and embodiments which are described. Certain aspects and embodiments of the invention will now be illustrated by way of example and with reference to the figures described above. 10 References All documents mentioned in this specification are incorporated herein by reference in their entirety. US6067239 EP3080540 15

Claims

1. A current control module for controlling an in-rush current resulting from the initiation of an electrical connection of a battery to an electrical load,wherein a metal-oxide-semiconductor field-effect transistor (MOSFET) is electrically connectable between the battery and the electrical load, and an effective drain-source resistance (Rds) of the MOSFET regulates the current between the battery and the electrical load; andthe control module is configured to:receive a current sense signal indicative of a current between the battery and the electrical load,compare the current sense signal to a threshold level, andadjust a control signal for the MOSFET in response to the comparison, to control the Rds of the MOSFET such that the current sense signal moves towards or below the threshold level or is maintained below the threshold level.

2. The current control module of claim 1 wherein the control module is configured to set the Rds of the MOSFET to an initial effective resistance when electrical connection of the battery to the electrical load is initiated.

3. The current control module of claim 1 or 2 wherein the control module is configured to: set the Rds of the MOSFET to a first effective resistance when the current sense signal is above the threshold level, anddecrease the Rds of the MOSFET to a second effective resistance, which is lower than the first effective resistance, when the current sense signal has reduced to lower than the threshold level.

4. The current control module of any preceding claim wherein the control signal is configured to operate the MOSFET in its linear region, when the current sense signal is above the threshold level.

5. The current control module of claim 4 wherein the control signal is an analogue signal adjustable between a minimum and a maximum voltage,wherein the minimum voltage corresponds to a maximum Rds of the MOSFET, and the maximum voltage corresponds to minimum a Rds of the MOSFET.

6. The current control module of claim 5 wherein the control signal is dependent on a difference between the current sense signal and the threshold level.

7. The current control module of any preceding claim wherein the current control module comprises a microcontroller and a digital-to-analogue converter, DAC,wherein the microcontroller is configured to receive the current sense signal, compare the current sense signal to a threshold level, adjust a digital control signal in response tothe comparison, and provide the digital control signal to the DAC; andthe DAC is configured to generate the control signal based on the digital signal.

8. The current control module of any preceding claim wherein the current sense signal is a voltage input signal which is indicative of the current between the battery and the electrical load,the threshold level is a reference voltage, andthe control module comprises an error amplifier configured to receive the current sense signal and the reference voltage as inputs, and output the control signal.

9. The current control module of claim 8 wherein an inverting input of the error amplifier receives the reference voltage, and a non-inverting input of the error amplifier receives the current sense signal,and the control signal output by the error amplifier is connected to a gate-drive circuit for controlling the MOSFET.

10. The current control module of any one of claims 1 to 4 wherein the control signal is a PWM signal, and the control module is configured to adjust a duty cycle of the PWM signal based on the comparison between the current sense signal and the threshold level.

11. The current control module of claim 10 wherein the control module comprises a signal generator for generating the PWM signal.

12. The current control module of claim 10 wherein the control module comprises a microcontroller for generating the PWM signal.

13. The current control module of any one of claims 10 to 12 wherein, if the current sense signal is above the threshold level, the control module is configured to generate the PWM signal at a first duty cycle.

14. The current control module of claim 13 wherein a switching frequency of the PWM signal at the first duty cycle is higher than a rate parameter of the MOSFET, such that the MOSFET is operated in its linear region.

15. The current control module of any one of claims 10 to 14 wherein, if the current sense signal is below the threshold level, the control module is configured to adjust the duty cycle of the PWM signal to 100% such that the MOSFET is in an “on” state.

16. A battery management system for controlling an in-rush current resulting from the initiation of an electrical connection of a battery to a electrical load, wherein the battery management system comprises:a MOSFET, wherein the MOSFET is electrically connectable between the battery and the load, and an effective drain-source resistance (Rds) of the MOSFET is configured toregulate the current between the battery and the load ,a gate-drive circuit for controlling the Rds of the MOSFET in response to a control signal, anda current control module according to any one of claims 1 to 15 for providing the control signal.

17. The battery management system of claim 16, wherein the gate-drive circuit comprises a gate-drive transistor connected between a gate of the MOSFET and electrical ground,wherein the control signal is provided to a gate of the gate-drive transistor such that a gate-source voltage, and thereby Rds of the gate-drive transistor, is adjustable by the control signal,such that: an intermediate control voltage provided to the gate of the MOSFET is dependent on the control signal.

18. The battery management system of claim 17 wherein the MOSFET is a high-side MOSFET electrically connected to a high-side of the load and the gate-drive circuit comprises a high-side driver for stepping up the intermediate control voltage to a gatesource voltage for driving the high-side MOSFET, andoptionally, wherein the high-side driver comprises a charge-pump for stepping up the intermediate control voltage.

19. The battery management system of any one of claims 16 to 19 further comprising a current sense circuit for detecting the current between the battery and the load, generating the current sense signal, and providing the current sense signal to the current control module, andoptionally, wherein the current sense circuit comprises a sense resistor connectable in series with the load, and a current sense amplifier configured to amplify a voltage drop across the sense resistor20. The battery management system of any of claims 16 to 19 wherein the battery management system is for a battery having an internal resistance of 1mQ or less.

21. The battery management system of any of claims 16 to 20 wherein the threshold level corresponds to an in-rush current between the battery and the load of 250A or more.

22. A method of controlling an in-rush current resulting from the initiation of an electrical connection of a battery to an electrical load,wherein a MOSFET is electrically connectable between the battery and the electrical load, and an effective drain-source resistance (Rds) of the MOSFET regulates the current between the battery and the electrical load;the method comprising the steps of:receiving a current sense signal indicative of a current between the battery and theelectrical load,comparing the current sense signal to a threshold level, andadjusting a control signal for the MOSFET in response to the comparison, to control the Rds of the MOSFET such that the current sense signal moves towards or below the threshold level or is maintained below the threshold level.

23. The method of claim 22, further comprising a step of:connecting the battery to the electrical load via the MOSFET, andadjusting the control signal to set the effective RDs of the MOSFET to an initial resistance when the electrical connection of the battery and the electrical load is initiated.

24. The method of either claim 22 or claim 23, further comprising a step of:adjusting the control signal to set the Rds of the MOSFET to a first effective resistance when the current sense signal is above the threshold level, andadjusting the control signal to set the Rds of the MOSFET to a second resistance, which is lower than the first resistance, when the current sense signal has reduced to lower than the threshold level.

25. A computer-readable medium comprising code that when executed by a processor causes the processor to perform the method of any of claims 22 to 24.

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