Control device, control method, and control system

Abstract

To improve the stability of control system of a motor.SOLUTION: A control device is a device that sets a δ axis current command to be zero when the direction of an output voltage vector of an inverter for energizing an AC current of a permanent magnet synchronous motor is the γ axis direction, and the axis perpendicular to the γ axis is the δ axis, controlling the inverter using a δ axis voltage command and a γ axis voltage command, which are set on the basis of a rotational speed command. The control device also includes: setting the δ axis voltage command on the basis of the deviation between the δ axis current command and a δ axis current; performing a prescribed phase compensation to a γ axis current when setting the γ axis voltage command on the basis of the rotational speed command, a back electromotive force constant of the permanent magnet synchronous motor, an inverter frequency command, and an offset voltage based on the integral value of the δ axis currents; and calculating the inverter frequency command by further subtracting the γ axis current value multiplied by a prescribed frequency adjustment gain from the rotational speed command.SELECTED DRAWING: Figure 2

Description

[Technical field]

[0001] The present disclosure relates to a control device, a control method, and a control system. [Background technology]

[0002] In the V / f control of a brushless DC (direct current) motor described in Patent Document 1 and Patent Document 2, the rotation speed control of the motor is stabilized against load fluctuations by subtracting a value obtained by multiplying the γ-axis current, which changes according to the output torque, by a frequency adjustment gain from a rotation speed command. In this case, the attenuation rate of the rotation speed control increases as the inertia (moment of inertia) of the motor and the frequency adjustment gain increase, and decreases as the inductance of the motor (armature winding) increases. [Prior art documents] [Patent documents]

[0003] [Patent Document 1] Patent No. 4764124 [Patent Document 2] JP 2009-124801 A Summary of the Invention [Problem to be solved by the invention]

[0004] Incidentally, the resonance frequency caused by the V / f control of the motor is proportional to the product of the number of pole pairs and the back electromotive force constant, and inversely proportional to the square root of the product of the inertia and inductance. For this reason, for example, in a motor that is integrated with a compressor, as the compressor driven by the motor becomes smaller and lighter, the inertia of the motor decreases and the back electromotive force constant increases, which increases the resonance frequency of the motor and raises the risk of the resonance frequency falling within the normal operating range. To address this issue, the damping rate can be increased by increasing the frequency adjustment gain, thereby stabilizing the control system. However, if the frequency adjustment gain is made too large, the feedback amount of the speed system increases, and there is a risk that the actual rotation speed of the compressor will fluctuate significantly.

[0005] The present disclosure has been made in consideration of the above circumstances, and has an object to provide a control device, a control method, and a control system that can improve the stability of a motor control system. [Means for solving the problem]

[0006] In order to achieve the above object, a control device according to the present disclosure is a control device which sets a δ-axis current command to zero, with the direction of an output voltage vector of an inverter which supplies AC current to a permanent magnet synchronous motor being the γ-axis direction and the axis perpendicular to the γ-axis being the δ-axis, and controls the inverter based on a δ-axis voltage command and a γ-axis voltage command which are set based on a rotation speed command, and which sets the δ-axis voltage command based on a deviation between the δ-axis current command and the δ-axis current, and which sets the γ-axis voltage command based on the rotation speed command, a back electromotive force constant of the permanent magnet synchronous motor, an inverter frequency command, and an offset voltage based on an integral value of the δ-axis current, and which performs a predetermined phase compensation on the γ-axis current and calculates the inverter frequency command by subtracting a value multiplied by a predetermined frequency adjustment gain from the rotation speed command.

[0007] A control method according to the present disclosure is a control device which sets a δ-axis current command to zero, defines a direction of an output voltage vector of an inverter which supplies an AC current to a permanent magnet synchronous motor as a γ-axis direction, and defines an axis perpendicular to the γ-axis as a δ-axis direction, and controls the inverter based on a δ-axis voltage command and a γ-axis voltage command which are set based on a rotation speed command, and in this case, sets the δ-axis voltage command based on a deviation between the δ-axis current command and the δ-axis current, and sets the γ-axis voltage command based on the rotation speed command, a back electromotive force constant of the permanent magnet synchronous motor, an inverter frequency command, and an offset voltage based on an integral value of the δ-axis current, performs a predetermined phase compensation on the γ-axis current, and calculates the inverter frequency command by subtracting a value multiplied by a predetermined frequency adjustment gain from the rotation speed command.

[0008] A control system according to the present disclosure includes: a permanent magnet synchronous motor; a control device which sets a δ-axis current command to zero, with a γ-axis direction being the direction of an output voltage vector of an inverter which supplies an AC current to the permanent magnet synchronous motor, and an axis perpendicular to the γ-axis being the δ-axis, and controls the inverter based on a δ-axis voltage command and a γ-axis voltage command which are set based on a rotation speed command, and which sets the δ-axis voltage command based on a deviation between the δ-axis current command and the δ-axis current, and which performs a predetermined phase compensation on the γ-axis current and calculates the inverter frequency command by subtracting a value multiplied by a predetermined frequency adjustment gain from the rotation speed command when setting the γ-axis voltage command based on the rotation speed command, a back electromotive force constant of the permanent magnet synchronous motor, an inverter frequency command, and an offset voltage based on an integral value of the δ-axis current; and the inverter. Effect of the Invention

[0009] According to the control device, control method, and control system disclosed herein, it is possible to improve the stability of the motor control system. [Brief description of the drawings]

[0010] [Figure 1] FIG. 1 is a block diagram showing a configuration example of a control system according to an embodiment of the present disclosure. [Diagram 2] 1 is an example of a vector diagram of V / f control according to an embodiment of the present disclosure. [Diagram 3] FIG. 2 is a block diagram showing an example configuration of a frequency command / position estimated value calculation unit according to an embodiment of the present disclosure. [Figure 4] FIG. 4 is a Bode plot illustrating frequency characteristics of a phase compensator according to an embodiment of the present disclosure. [Diagram 5] 11 is a schematic diagram showing an example of the operation of a Kω switching unit according to the present disclosure. FIG. [Figure 6] FIG. 2 is another block diagram illustrating a configuration example of a control system according to an embodiment of the present disclosure. [Figure 7] FIG. 4 is a Bode plot illustrating an example of frequency characteristics of a control system according to an embodiment of the present disclosure. [Figure 8] FIG. 1 is a schematic block diagram illustrating a configuration of a computer according to an embodiment of the present disclosure. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0011] Hereinafter, a control device, a control method, and a control system according to an embodiment of the present disclosure will be described with reference to FIGS. 1 to 8. FIG. 1 is a block diagram showing a configuration example of a control system according to an embodiment of the present disclosure. FIG. 2 is an example of a vector diagram of V / f control according to an embodiment of the present disclosure. FIG. 3 is a block diagram showing a configuration example of a frequency command / position estimated value calculation unit according to an embodiment of the present disclosure. FIG. 4 is a Bode plot showing the frequency characteristic of a phase compensator according to an embodiment of the present disclosure. FIG. 5 is a schematic diagram showing an operation example of a Kω switching unit according to an embodiment of the present disclosure. FIG. 6 is another block diagram showing a configuration example of a control system according to an embodiment of the present disclosure. FIG. 7 is a Bode plot showing an example of the frequency characteristic of a control system according to an embodiment of the present disclosure. FIG. 8 is a schematic block diagram showing a configuration of a computer according to an embodiment of the present disclosure. Note that the same or corresponding configurations in each drawing are designated by the same reference numerals and the description will be omitted as appropriate.

[0012] A control system 10 according to the present disclosure shown in FIG. 1 is a control system for a motor 5, which is a permanent magnet synchronous motor configured coaxially with a compressor, which is a load. The control system 10 includes an AC (alternating current)-DC converter 2, an inverter 3, a motor 5, current detectors 6u, 6v, and 6w, a control device 8, and a voltage detection circuit 9. The AC-DC converter 2 converts an AC voltage VAC supplied from an AC power source 1 into a DC voltage VDC and supplies the DC voltage to a voltage-type inverter 3. The inverter 3 generates a three-phase drive voltage from the DC voltage VDC and supplies the three-phase drive voltage to a u-phase armature winding, a v-phase armature winding, and a w-phase armature winding of the motor 5 via a u-phase power line 4u, a v-phase power line 4v, and a w-phase power line 4w. The motor 5, which is supplied with the respective voltages, generates a rotating magnetic field in the armature winding from the supplied three-phase drive voltage.

[0013] In order to realize sensorless control of the motor 5, the u-phase power line 4u, the v-phase power line 4v, and the w-phase power line 4w are provided with current detectors 6u, 6v, and 6w, respectively, which measure the u-phase current iu flowing through the u-phase armature coil, the v-phase current iv flowing through the v-phase armature coil, and the w-phase current iw flowing through the w-phase armature coil. The measured u-phase current iu, v-phase current iv, and w-phase current iw are used for PWM (pulse width modulation) control (i.e., V / f control) that outputs a pseudo sine wave. However, in another configuration, only two of the u-phase current iu, v-phase current iv, and w-phase current iw may be measured, and the other current may be calculated from the two measured currents. In addition, these u-phase current iu, v-phase current iv, and w-phase current iw can also be measured by a current detection circuit (a so-called three-shunt current detection circuit) arranged in the lower arm portion of inverter 3, or a current detection circuit (a so-called one-shunt current detection circuit) arranged between inverter 3 and AC-DC converter 2.

[0014] The voltage detection circuit 9 is connected to a power supply line that supplies a DC voltage VDC from the AC-DC converter 2 to the inverter 3, and measures the DC voltage VDC.

[0015] The current detectors 6u, 6v, and 6w and the voltage detection circuit 9 are connected to the control device 8, which generates a PWM signal SPWM for controlling the inverter 3 based on the δ-axis current iδ and the γ-axis current iγ, which are three-phase / two-phase converted from the armature currents iu, iv, and iw measured by the current detectors 6u, 6v, and 6w, and the δ-axis voltage command vδ* and the γ-axis voltage command vγ*. At this time, the control device 8 sets the δ-axis current command iδ* to zero, with the direction of the output voltage vector V of the inverter 3, which supplies AC current to the motor 5 (permanent magnet synchronous motor) as the γ-axis direction, and the axis perpendicular to the γ-axis as the δ-axis, and controls the inverter 3 based on the δ-axis voltage command δv* and the γ-axis voltage command γv*, which are set based on the rotation speed command ωm*. FIG. 2 shows an example of the relationship between the d-axis and the q-axis and the δ-axis and the γ-axis. The direction of the d-axis is the direction of the magnetic flux generated by the magnet. The direction of the q-axis is the direction in which the back electromotive force is generated. The direction of the γ-axis is the direction of the output voltage vector V of the inverter 3. The direction of the δ-axis is perpendicular to the γ-axis. In FIG. 2, φv is the phase difference between the d-axis and the γ-axis. φ is the phase difference between the output voltage vector V and the output current vector I of the inverter 3. Λd is the back electromotive force constant (also called the induced voltage constant (coefficient)) of the motor 5. n is the number of pole pairs of the motor 5.

[0016] The PWM signal SPWM is generated so that the rotor speed is controlled based on a rotation speed command ω* given to the control device 8. Switching transistors (not shown) that constitute the inverter 3 are turned on and off in response to the PWM signal SPWM, and a three-phase drive voltage is supplied to the armature windings of the motor 5.

[0017] The control device 8 can be configured using a computer such as a microcontroller. The control device 8 has the following units as a functional configuration constituted by a combination of the computer constituting the control device 8, its peripheral circuits and peripheral devices, and software such as a program executed by the computer. That is, the control device 8 has, as its functional configuration, an A / D (analog / digital) conversion unit 31, a three-phase / two-phase conversion unit 32, a frequency command / position estimated value calculation unit 33, a voltage command calculation unit 34, a two-phase / three-phase conversion unit 35, a PWM calculation unit 36, and an A / D conversion unit 37.

[0018] The A / D converter 31 acquires and A / D converts the output signals from the current detectors 6u, 6v, and 6w. The three-phase / two-phase converter 32 performs three-phase / two-phase conversion on the u-phase current iu, v-phase current iv, and w-phase current iw acquired by the A / D converter 31, using the rotor position estimate θ* calculated in the previous clock cycle, and calculates and outputs the δ-axis current iδ and the γ-axis current iγ.

[0019] The frequency command / position estimate value calculation unit 33 receives, for example, a rotation speed command ωm* (or a rotation speed command nωm* multiplied by the number of pole pairs n) of the motor 5 received from a higher-level control device (not shown) and the γ-axis current iγ output from the three-phase / two-phase conversion unit 32, generates an inverter frequency command (angular frequency command) ωf*, which is the frequency of a three-phase AC voltage supplied to the motor 5, and outputs the generated command to a voltage command calculation unit 34. The frequency command / position estimate value calculation unit 33 also generates a rotor position estimate θ* and outputs the rotor position estimate θ* to the three-phase / two-phase conversion unit 32. The frequency command / position estimate value calculation unit 33 also sets a power factor control angle φ(i) using the value of the γ-axis current iγ, and outputs a value (=θ*+φ(i)) obtained by adding the power factor control angle φ(i) to the position estimate θ* to the two-phase / three-phase conversion unit 35. The control device 8 of this embodiment performs power factor angle control, which is a control for adjusting the power factor by providing a phase difference φ between the coordinates used by the three-phase / two-phase conversion unit 32 for current detection and the coordinates used by the two-phase / three-phase conversion unit 35. In this embodiment, the power factor angle control of the motor is performed by adjusting the power factor control angle φ(i). How the inverter frequency command ωf*, the position estimate θ*, and the power factor control angle φ(i) are generated will be described later with reference to FIG. 3.

[0020] A voltage command calculation unit 34 receives the inverter frequency command ωf* output by the frequency command / position estimate value calculation unit 33 and the δ-axis current iδ output by the 3-phase / 2-phase conversion unit 32, calculates a δ-axis voltage command vδ* and a γ-axis voltage command vγ* in accordance with the following equations (1) and (2), and outputs the calculated values ​​to a 2-phase / 3-phase conversion unit 35.

[0021]

number

[0022]

number

[0023] In equation (1), vδ* is the δ-axis voltage command, Kpd is the δ-axis current control gain, iδ* is the δ-axis current command, and iδ is the δ-axis current. Also, (iδ*-iδ) represents the deviation between the δ-axis current command iδ* and the δ-axis current iδ. Also, in equation (2), vγ* is the γ-axis voltage command, Λd is the induced voltage coefficient of the motor 5, ωf* is the inverter frequency command, and Vγofs is the γ-axis adjustment voltage offset (hereinafter referred to as the offset voltage).

[0024] In this embodiment, the voltage command calculation unit 34 adjusts the γ-axis voltage command vγ* using the offset voltage Vγofs calculated by the following equation (3) according to equation (2), thereby setting the δ-axis current command iδ* to zero as shown in the following equation (4), and controls the δ-axis current iδ to zero.

[0025]

number

[0026]

number

[0027] Here, Kvq is a voltage control gain. The offset voltage Vγofs is a value based on the integral value of the δ-axis current iδ. In this embodiment, the V / f control is performed based on equations (1) and (2), so that the motor terminal voltages (Vδ, Vγ) are controlled to be approximately proportional to the rotation speed ωm.

[0028] The two-phase / three-phase conversion unit 35 performs two-phase to three-phase conversion on the γ-axis voltage command vγ* and the δ-axis voltage command vδ*, ​​calculates a u-phase voltage vu, a v-phase voltage vv, and a w-phase voltage vw of the three-phase drive voltages to be supplied to the motor 5, and outputs them to the PWM calculation unit 35. In this two-phase to three-phase conversion, a value obtained by adding a power factor control angle φ(i) derived from the value of the γ-axis current iγ to a position estimate value θ* for inverter drive is used for power factor angle control of the motor.

[0029] The PWM calculation unit 36 ​​generates a PWM signal SPWM for controlling the inverter 3 so that a three-phase drive voltage having a u-phase voltage vu, a v-phase voltage vv, and a w-phase voltage vw calculated by the two-phase / three-phase conversion unit 35 is supplied to the motor 5, and outputs the PWM signal SPWM to the inverter 3. The PWM signal SPWM is generated with reference to the magnitude of the DC voltage VDC supplied from the AC-DC converter 2 to the inverter 3, and the A / D conversion unit 37 performs A / D conversion on the voltage notification signal output by the voltage detection circuit 9, and the PWM signal SPWM is generated with reference to the acquired DC voltage VDC. A switching transistor (not shown) of the inverter 3 is turned on or off in response to the generated PWM signal SPWM.

[0030] Next, with reference to Fig. 3, a method for generating the inverter frequency command ωf*, the position estimate θ*, and the power factor control angle φ(i) by the frequency command / position estimate calculation unit 33 shown in Fig. 1 will be described. The frequency command / position estimate calculation unit 33 shown in Fig. 3 includes a phase compensator 331, an amplifier (or a multiplier) 332, a Kω switching unit 333, a subtractor 334, an integrator 335, a power factor angle calculation unit 336, and an adder 337.

[0031] The phase compensator 331 receives the γ-axis current iγ, performs a predetermined phase compensation, and then outputs the phase-compensated γ-axis current iγ to the amplifier 332. The phase compensation for the γ-axis current iγ can be, for example, phase lead compensation using a transfer function Fc(s) shown in equation (5).

[0032]

number

[0033] Here, α is a constant greater than 1, s is a complex variable, and T is one of the parameters of phase compensation and the time constant in the primary system element. Figure 4 shows the Bode plot of the transfer function Fc(s). The horizontal axis is the angular frequency [rad / s] (logarithmic scale), and the vertical axis is the gain [dB] and phase [deg]. The gain characteristic is 20 dB / dec with two corner frequencies 1 / (αT) and 1 / T. The phase characteristic has an upwardly convex, symmetrical characteristic with the maximum phase lead Ψmax at the frequency ωmax, which is the midpoint between the two corner frequencies 1 / (αT) and 1 / T. The frequency ωmax can be calculated using equation (6), and the maximum phase lead Ψmax can be calculated using equation (7).

[0034]

number

[0035]

number

[0036] It should be noted that the frequency ωmax at which the maximum phase lead Ψmax occurs can be positioned, for example, in the resonance frequency band of the motor 5. Equation (8) is a formula for calculating the resonance frequency ωn [rad / s].

[0037]

number

[0038] Here, J is the inertia [kg m^2], L is the inductance [mH], and Λd is the induced voltage coefficient [V / (rad / s)]. Since J, L, and Λd vary to some extent depending on the individual motor, and L changes with temperature, the resonant frequency of the motor 5 has a certain width (called the resonant frequency band in this embodiment). In this embodiment, in order to stabilize the control system, phase compensation is performed on the γ-axis current iγ to change the phase in the resonant frequency band.

[0039] Moreover, the amplifier 332 calculates Kω·Fc(iγ) by multiplying the phase-compensated γ-axis current iγ(Fc(iγ)) output by the phase compensator 331 by the frequency adjustment gain Kω, and outputs the result to the subtractor 334. Note that the frequency adjustment gain Kω is assumed to be positive.

[0040] A subtractor 334 outputs, as the inverter frequency command ωf*, a value obtained by subtracting Kω·Fc(iγ) output by the amplifier 332 from a rotation speed command nωm* obtained by multiplying the rotation speed command ωm* by the number of pole pairs n. In this case, the inverter frequency command ωf* is calculated by the following equation (9).

[0041]

number

[0042] According to equation (9), when the γ-axis current iγ increases, i.e., when the output torque increases, the frequency command ωf* decreases, and when the γ-axis current iγ decreases, i.e., when the output torque decreases, the frequency command ωf* increases. By using equation (9), stall is prevented by decreasing the frequency command ωf* when the output torque increases, and when the output torque decreases, the frequency command ωf* is increased to control the rotor so that it does not accelerate.

[0043] Moreover, the integrator 335 integrates the inverter frequency command ωf* according to the following equation (10), and calculates and outputs the position estimate value θ*.

[0044]

number

[0045] Further, the power factor angle calculation unit 336 calculates the power factor control angle φ(i) by, for example, the following equation (11), and outputs it to the adder 337.

[0046]

number

[0047] Here, Ld is the d-axis inductance, and I is the output current vector of the inverter 3, which has the γ-axis current iγ and the δ-axis current iδ as components. In this embodiment, since id*=0, I=iγ.

[0048] In addition, the adder 337 adds the power factor control angle φ(i) output by the power factor angle calculation unit 336 to the position estimate value θ* output by the integrator 335, and outputs the sum (=θ*+φ(i)).

[0049] Further, the Kω switching unit 333 generates a control signal for switching the value of the frequency adjustment gain Kω based on the rotation speed command ωm* (or the rotation speed command nωm*), and outputs it to the amplifier 332. The amplifier 332 changes the value of the frequency adjustment gain Kω based on the control signal output by the Kω switching unit 333. FIG. 5 shows an example of the change in the value of the frequency adjustment gain Kω under the control of the Kω switching unit 333. In FIG. 5, the horizontal axis represents the compressor rotation speed (motor rotation speed) [rps], and the vertical axis represents the value of the frequency adjustment gain Kω. For example, if the band where resonance occurs (resonance frequency band) is equal to or lower than the rotation speed N1, the frequency adjustment gain Kω is set to a relatively large constant value at or below the rotation speed N2 (>N1). Furthermore, the value of the frequency adjustment gain Kω is gradually decreased from the rotation speed N2 to the rotation speed N3 (>N2), and is set to a relatively small constant value at or above the rotation speed N3.

[0050] The damping rate ζ of the entire system in the control system 10 can be calculated by the following formula (12). In this case, the damping rate ζ can be increased by increasing the frequency adjustment gain Kω. However, when the frequency adjustment gain Kω is increased, there are disadvantages as described above. In this embodiment, the disadvantages are minimized by switching the frequency adjustment gain Kω according to the rotation speed, and the control system is stabilized.

[0051]

number

[0052] Next, with reference to Figs. 6 and 7, the improvement of the stability of the control system according to this embodiment will be described. Fig. 6 is a block diagram in which the control system 10 shown in Fig. 1 is equivalently converted so that the input is the compressor rotation speed [rps] and the output is the motor current [A]. In Fig. 6, the rotation speed to voltage conversion unit 100 is newly defined, and among the elements shown in Fig. 1, the elements other than the amplifier 332, the phase compensator 331, and the motor 5 are moved into the rotation speed to voltage conversion unit 100. Fig. 7 is a Bode diagram of the open loop transfer function of the control system 10 shown in Fig. 6. In Fig. 7, the frequency characteristic of the control system 10 of this embodiment shown in Fig. 6 is shown by a solid line, and the frequency characteristic in the case where the phase compensator 331 is omitted from the configuration shown in Fig. 6 is shown by a dashed line. The phase margin with the phase compensator is larger than the phase margin without the phase compensator, and it was confirmed that the stability of the control system is improved by performing phase compensation by the phase compensator 331.

[0053] (Action and effect) As described above, the control device 8 of this embodiment sets the direction of the output voltage vector V of the inverter 3 that supplies AC current to the motor 5 (permanent magnet synchronous motor) as the γ-axis direction and the axis perpendicular to the γ-axis as the δ-axis, sets the δ-axis current command iδ* to zero (Equation (4)), and controls the inverter 3 based on the δ-axis voltage command vδ* and the γ-axis voltage command vγ* that are set based on the rotation speed command nωm*. The control device 8 sets the δ-axis voltage command vδ* based on the deviation between the δ-axis current command iδ* and the δ-axis current iδ (Equation (1)), and sets the γ-axis voltage command vγ* based on the rotation speed command ωm*, the back electromotive force constant Λd of the motor 5, the inverter frequency command ω*f, and the offset voltage Vγofs based on the integral value of the δ-axis current iδ (Equation (2)). In this case, a predetermined phase compensation Fc(s) is performed on the γ-axis current iγ, and the inverter frequency command ω*f is multiplied by a predetermined frequency adjustment gain Kω and the inverter frequency command ω*f is subtracted from the rotation speed command nωm* to calculate the inverter frequency command ω*f (Equation (9)). According to this configuration, the influence of the resonance frequency of the motor can be easily suppressed by phase compensation, so that the stability of the control system of the motor can be improved.

[0054] (Other embodiments) Although the embodiment of the present disclosure has been described above in detail with reference to the drawings, the specific configuration is not limited to this embodiment, and design changes within the scope of the present disclosure are also included. For example, the phase compensation by the phase compensator 331 is not limited to phase lead compensation, and may be phase lead / lag compensation, etc. Also, the power factor angle control that controls the power factor by providing a phase difference φ between the coordinates used by the three-phase / two-phase conversion unit 32 for current detection and the coordinates used by the two-phase / three-phase conversion unit 35 may be omitted. In this case, the control method is one in which the power factor control angle φ(i)=0.

[0055] Computer Configuration FIG. 8 is a schematic block diagram showing the configuration of a computer according to this embodiment. The computer 90 comprises a processor 91, a main memory 92, a storage 93, and an interface 94. The above-mentioned control device 8 is implemented in a computer 90. The operations of the above-mentioned processing units are stored in the form of a program in a storage 93. The processor 91 reads the program from the storage 93, loads it in a main memory 92, and executes the above-mentioned processing in accordance with the program. The processor 91 also secures storage areas in the main memory 92 corresponding to the above-mentioned storage units in accordance with the program.

[0056] The program may be for realizing a part of the functions to be performed by the computer 90. For example, the program may be for realizing the functions by combining with other programs already stored in the storage or by combining with other programs implemented in other devices. In another embodiment, the computer may include a custom LSI (Large Scale Integrated Circuit) such as a PLD (Programmable Logic Device) in addition to or instead of the above configuration. Examples of PLDs include PAL (Programmable Array Logic), GAL (Generic Array Logic), CPLD (Complex Programmable Logic Device), and FPGA (Field Programmable Gate Array). In this case, some or all of the functions to be realized by the processor may be realized by the integrated circuit.

[0057] Examples of the storage 93 include a hard disk drive (HDD), a solid state drive (SSD), a magnetic disk, a magneto-optical disk, a compact disc read only memory (CD-ROM), a digital versatile disc read only memory (DVD-ROM), and a semiconductor memory. The storage 93 may be an internal medium directly connected to the bus of the computer 90, or an external medium connected to the computer 90 via an interface 94 or a communication line. In addition, when the program is distributed to the computer 90 via a communication line, the computer 90 that receives the program may load the program into the main memory 92 and execute the above-mentioned process. In at least one embodiment, the storage 93 is a non-transitory tangible storage medium.

[0058] <Additional Notes> The control device 8 described in each embodiment can be understood, for example, as follows.

[0059] (1) A control device 8 according to a first aspect is a control device 8 that sets a δ-axis current command iδ* to zero with a direction of an output voltage vector V of an inverter 3 that supplies an AC current to a permanent magnet synchronous motor (motor 5) as a γ-axis direction and an axis perpendicular to the γ-axis as a δ-axis direction, and controls the inverter 3 based on a δ-axis voltage command vδ* and a γ-axis voltage command vγ* that are set based on a rotation speed command ωm*(nωm*), and that sets the δ-axis voltage command vδ* based on a deviation between the δ-axis current command iδ* and the δ-axis current iδ. and when setting the γ-axis voltage command vγ* based on the rotation speed command ωm*(nωm*), the back electromotive force constant Λd of the permanent magnet synchronous motor, the inverter frequency command ωf*, and an offset voltage Vγofs based on the integral value of the δ-axis current iδ, a predetermined phase compensation Fc(s) is performed on the γ-axis current iγ, and the inverter frequency command ωf* is calculated by subtracting a value (Kω·Fc(iγ)) obtained by multiplying the rotation speed command (nωm*) by a predetermined frequency adjustment gain Kω. According to this aspect and each of the following aspects, it is possible to improve the stability of the control system of the motor.

[0060] (2) The control device 8 of a second aspect is the control device 8 of (1), in which the γ-axis voltage command is set by multiplying the inverter frequency command by the back electromotive force constant and adding the offset voltage to the result.

[0061] (3) A control device 8 of a third aspect is the control device 8 of (1) or (2), wherein the phase compensation is phase lead compensation in which the frequency producing the maximum phase lead is located in the resonant frequency band of the permanent magnet synchronous motor.

[0062] (4) A fourth aspect of the control device 8 is the control device 8 of (1) to (3), in which the frequency adjustment gain is set to a first value (a relatively large value) when the rotational speed of the permanent magnet synchronous motor is within a first range (a range equal to or less than rotational speed N1) corresponding to the resonant frequency band, and is set to a second value (a relatively small value) smaller than the first value when the rotational speed is within a second range (a range equal to or greater than rotational speed N2) larger than the first range.

[0063] (5) The control device 8 of a fifth aspect is the control device 8 of (1) to (4), and controls the phase difference between the output current vector and the output voltage vector of the inverter in response to a γ-axis current. [Explanation of symbols]

[0064] 10. Control system 3…Inverter 5. Motor 8...Control device 331...Phase compensator 332…Amplifier 333...Kω switching section 336...Power factor angle calculation section

Claims

1. A control device for controlling a permanent magnet synchronous motor, the control device defining a γ-axis direction of an output voltage vector of the inverter and a δ-axis direction orthogonal to the γ-axis direction, the control device setting a δ-axis current command to zero, and controlling the inverter based on a δ-axis voltage command and a γ-axis voltage command set based on a rotation speed command, the control device comprising: when setting the δ-axis voltage command based on a deviation between the δ-axis current command and the δ-axis current, and setting the γ-axis voltage command based on the rotation speed command, the back electromotive force constant of the permanent magnet synchronous motor, an inverter frequency command, and an offset voltage based on an integral value of the δ-axis current, A predetermined phase compensation is performed on the γ-axis current, and the inverter frequency command is calculated by multiplying the γ-axis current by a predetermined frequency adjustment gain and subtracting the resultant value from the rotation speed command. Control device.

2. The γ-axis voltage command is set by multiplying the inverter frequency command by the back electromotive force constant and adding the offset voltage to the product. The control device according to claim 1 .

3. The phase compensation is a phase lead compensation in which a frequency causing a maximum phase lead is located in a resonant frequency band of the permanent magnet synchronous motor. The control device according to claim 2.

4. The frequency adjustment gain is set to a first value when the rotation speed of the permanent magnet synchronous motor is within a first range corresponding to the resonant frequency band, and is set to a second value smaller than the first value when the rotation speed is within a second range larger than the first range. The control device according to claim 3.

5. A phase difference between an output current vector of the inverter and an output voltage vector of the inverter is controlled in accordance with a γ-axis current. The control device according to any one of claims 1 to 3.

6. A control device for controlling a permanent magnet synchronous motor, the control device defining a γ-axis direction of an output voltage vector of the inverter and a δ-axis direction orthogonal to the γ-axis direction, the control device setting a δ-axis current command to zero, and controlling the inverter based on a δ-axis voltage command and a γ-axis voltage command set based on a rotation speed command, the control device comprising: when setting the δ-axis voltage command based on a deviation between the δ-axis current command and the δ-axis current, and setting the γ-axis voltage command based on the rotation speed command, the back electromotive force constant of the permanent magnet synchronous motor, an inverter frequency command, and an offset voltage based on an integral value of the δ-axis current, A predetermined phase compensation is performed on the γ-axis current, and the inverter frequency command is calculated by multiplying the γ-axis current by a predetermined frequency adjustment gain and subtracting the resultant value from the rotation speed command. Control methods.

7. A permanent magnet synchronous motor; a control device which sets a γ-axis direction of an output voltage vector of an inverter which supplies an AC current to the permanent magnet synchronous motor and an axis perpendicular to the γ-axis as a δ-axis, sets a δ-axis current command to zero, and controls the inverter based on a δ-axis voltage command and a γ-axis voltage command which are set based on a rotation speed command, when setting the δ-axis voltage command based on a deviation between the δ-axis current command and the δ-axis current, and setting the γ-axis voltage command based on the rotation speed command, the back electromotive force constant of the permanent magnet synchronous motor, an inverter frequency command, and an offset voltage based on an integral value of the δ-axis current, A predetermined phase compensation is performed on the γ-axis current, and the inverter frequency command is calculated by multiplying the γ-axis current by a predetermined frequency adjustment gain and subtracting the resultant value from the rotation speed command. A control device; The inverter; A control system comprising:

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