Current sensor and measuring device

Abstract

To suppress detection errors caused by a coil in a high-frequency band.SOLUTION: A current sensor 1 includes: a magnetic core 11 surrounding a measurement target in which a current flows; a detection coil 12 wound around the magnetic core 11; a feedback coil 13 wound around the magnetic core 11, the feedback coil receiving a current output from an integration circuit which integrates voltages between both ends of the detection coil 12; and a capacitive element 14. The current sensor 1 adds a current output from the detection coil 12 through the capacitive element 14 and a current output from the feedback coil 13, and outputs a converted voltage from an operation circuit 22 which converts the added current into a voltage.SELECTED DRAWING: Figure 2

Description

[Technical field]

[0001] The present invention relates to a current sensor and a measurement device. [Background technology]

[0002] Patent Document 1 discloses a circuit having an AC pickup winding capable of detecting AC currents with frequencies exceeding 1 Hz, and an auxiliary winding that generates an auxiliary current that attempts to counteract the electromotive force induced by the main current flowing in a conductor. [Prior art documents] [Patent documents]

[0003] [Patent Document 1] Special Publication No. 2008-519269 Summary of the Invention [Problem to be solved by the invention]

[0004] In a current sensor having the circuit described above, in the low frequency band of approximately 1 kHz or less, zero flux operation causes the magnetic flux generated in the magnetic core wound around both the AC detection winding corresponding to the AC pickup winding and the feedback winding corresponding to the auxiliary winding to converge to approximately zero.

[0005] However, as the frequency of the measurement current flowing through the measurement target becomes higher than the low frequency band, the magnetic permeability of the magnetic core decreases and the effects of the parasitic capacitance of each winding and the line capacitance between the windings increase, which results in a weakening of the magnetic flux cancellation action by the feedback winding, causing magnetic flux to remain in the magnetic core and an induced current to flow through the AC detection winding.

[0006] Furthermore, in the high-frequency band of approximately 10 kHz or more, an induced current flows in the AC detection winding, causing the AC detection winding and feedback winding wound around the magnetic core to function as a current transformer, resulting in a large detection error in the measured current.

[0007] The present invention has been made in view of the above-mentioned problems, and has an object to suppress detection errors caused by the windings in the high frequency band. [Means for solving the problem]

[0008] According to one aspect of the present invention, the current sensor includes a magnetic core surrounding a measurement object through which a current flows, an AC detection winding wound around the magnetic core, a feedback winding wound around the magnetic core and receiving a current output from an integrating circuit that integrates the voltage between both ends of the AC detection winding, and a capacitive element, and adds the current output from the AC detection winding via the capacitive element and the current output from the feedback winding, and outputs the converted voltage from an arithmetic circuit that converts the added current into a voltage (converts it into a voltage proportional to the magnitude of the added current). Effect of the Invention

[0009] According to this aspect, in the high frequency band, the impedance of the capacitance element becomes low, and the current output from the AC detection winding through the capacitance element and the current output from the feedback winding are added together, so that the AC detection winding operates as a second feedback winding. Therefore, the induced current of the AC detection winding becomes the current of the feedback winding and reinforces the current of the feedback winding, so that detection errors caused by the windings in the high frequency band can be suppressed. [Brief description of the drawings]

[0010] [Figure 1] FIG. 1 is a diagram showing the configuration of a measurement device including a current sensor in the first embodiment. [Diagram 2] FIG. 2 is a schematic diagram showing a configuration example of a current sensor. [Diagram 3] FIG. 3 is a circuit diagram showing a detailed configuration example of the current sensor. [Figure 4] FIG. 4 is a diagram for explaining the operation of the current sensor. [Diagram 5]FIG. 5 is a diagram showing a simulation result of the gain frequency characteristic of the current sensor. [Figure 6] FIG. 6 is a circuit diagram showing a detailed configuration example of the current sensor in the second embodiment. [Figure 7] FIG. 7 is a schematic diagram showing a configuration example of a current sensor according to the third embodiment. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

[0011] Hereinafter, embodiments of the present invention will be described with reference to the accompanying drawings. In this specification, the same reference numerals are used throughout to refer to the same or equivalent elements.

[0012] First embodiment FIG. 1 is a diagram showing the configuration of a measurement device in the first embodiment.

[0013] The measuring device 100 is a device for measuring a physical quantity of a measuring object Lm. The measuring object Lm is, for example, an electric circuit through which a measuring current Im of DC or AC flows, and an example of this electric circuit is a single-phase or three-phase power cable. Moreover, an example of the physical quantity of the measuring object Lm is a current value, a magnetic flux value, a power value, or the like.

[0014] The measurement device 100 in the first embodiment measures the magnitude of a measurement current Im flowing through a measurement object Lm. The measurement device 100 includes a current sensor 1 and a measurement unit 2.

[0015] The current sensor 1 is a sensor that detects magnetic flux generated by an AC measurement current Im. The current sensor 1 detects the magnetic flux generated by the measurement current Im and outputs a voltage as a detection amount. The current sensor 1 is composed of a magnetic core with multiple windings.

[0016] For example, the current sensor 1 is realized by a current sensor that employs a winding detection type AC zero flux method (magnetic balance method), or a current sensor that includes a magnetic detection element such as a fluxgate or a Hall element in addition to this method. The current sensor 1 may have a through-type structure that surrounds the measurement object Lm, or may have an openable and closable clamp-type structure.

[0017] In the first embodiment, the current sensor 1 detects the detection amount of the measurement current Im by the AC zero flux method when the frequency of the measurement current Im is within a first frequency band equal to or lower than the cutoff frequency of the feedback winding 13 (see FIG. 2). On the other hand, when the measurement current Im is within a second frequency band higher than the first frequency band, the current sensor 1 detects the detection amount of the measurement current Im by a CT (current transformer) operation of the measurement object Lm wound around the magnetic core 11 and the feedback winding 13.

[0018] Hereinafter, the first frequency band is referred to as a “negative feedback operating region,” and the second frequency band is referred to as a “CT operating region.” The current sensor 1 outputs to the measurement unit 2 a detection amount related to the detected measurement current Im.

[0019] The measurement unit 2 calculates the current value of the measurement current Im for the measurement object Lm based on a detection amount indicating the magnitude of the voltage obtained by the current sensor 1. The measurement unit 2 is realized by, for example, a power analyzer, a current meter, and a magnetic field meter.

[0020] Next, the circuit configuration of the current sensor 1 will be described with reference to FIG.

[0021] FIG. 2 is a circuit diagram illustrating a schematic circuit configuration of the current sensor 1 according to the first embodiment.

[0022] The current sensor 1 comprises a sensor unit 10 for detecting a measurement magnetic flux generated by a measurement current Im, and a circuit unit 20 for converting an output signal of the sensor unit 10 into a voltage having a magnitude proportional to the magnitude of the measurement magnetic flux.

[0023] The sensor unit 10 includes a magnetic core 11, a detection winding 12, a feedback winding 13, and a capacitance element 14. The circuit unit 20 includes an integral amplifier circuit 21 and an arithmetic circuit 22.

[0024] The magnetic core 11 is formed to surround the measurement object Lm. The overall shape of the magnetic core 11 is formed to be annular so that the measurement object Lm can be inserted therethrough. The annular shape here includes a circular shape, an elliptical shape, a rectangular shape, a polygonal shape, and the like.

[0025] The magnetic core 11 in the first embodiment is made of an iron material such as permalloy. The magnetic core 11 may be realized by a hollow structure instead of an iron material.

[0026] The detection winding 12 is an AC detection winding wound around the magnetic core 11. The detection winding 12 functions as a magnetic flux detection winding in the negative feedback operating region. For example, the detection winding 12 converts a differential magnetic flux (φm-φr) between a measurement magnetic flux φm based on a measurement current Im in the magnetic core 11 and an anti-magnetic flux φr generated by the feedback winding 13 into a voltage V12 whose magnitude is proportional to the time derivative of the differential magnetic flux. Hereinafter, the differential magnetic flux is referred to as a residual magnetic flux.

[0027] One end of the detection winding 12, which is an output end, is connected to an input end of the integral amplifier circuit 21, and the other end of the detection winding 12 is connected to a reference potential G. The detection winding 12 is wound around the entirety or a part of the magnetic core 11.

[0028] The feedback winding 13 is wound around the magnetic core 11 and is supplied with current by the integral amplifier circuit 21. A current I2 flows through the feedback winding 13 so as to cancel the measurement magnetic flux φm generated in the magnetic core 11 by the measurement current Im due to the electromagnetic induction phenomenon. At this time, an anti-magnetic flux φr having a direction opposite to that of the measurement magnetic flux φm is generated in the magnetic core 11.

[0029] The feedback winding 13 functions as a current transformer in the CT operation region. At this time, the feedback winding 13 alone converts the measurement magnetic flux φm generated in the magnetic core 11 into a current I2 whose magnitude is proportional to the magnitude of the magnetic flux φm.

[0030] One end of the feedback winding 13 is connected to an output terminal of the integral amplifier circuit 21, and the other end, which is an output terminal, of the feedback winding 13 is connected to a first input terminal of the arithmetic circuit 22. The feedback winding 13, like the detection winding 12, is wound around all or a part of the magnetic core 11.

[0031] The capacitive element 14 is a capacitive element that bypasses the current flowing through the detection winding 12 to the arithmetic circuit 22 by AC coupling without passing through the integral amplifier circuit 21. As the capacitive element 14, a capacitor is used, for example.

[0032] One end of the capacitance element 14 is connected between one end of the detection winding 12 and an input end of the integral amplifier circuit 21, and the other end of the capacitance element 14 is connected to a second input end of the arithmetic circuit 22. The capacitance value of the capacitance element 14 is determined so that the current I1 flowing through the detection winding 12 passes through the capacitance element 14 in the CT operating region.

[0033] When the frequency of the measurement current Im is within the negative feedback operating region, the capacitive element 14 prevents the current I1 flowing through the detection winding 12 from passing directly to the arithmetic circuit 22 via the capacitive element 14 itself.

[0034] On the other hand, when the frequency of the measurement current Im is within the CT operation region, the capacitance element 14 directly outputs the current I1 flowing through the detection winding 12 to the arithmetic circuit 22. At this time, the higher the frequency of the measurement current Im, the greater the proportion of the current I1 flowing through the detection winding 12 that passes through the capacitance element 14.

[0035] The circuit section 20 is, for example, a circuit board on which an integral amplifier circuit 21 and an arithmetic circuit 22 are mounted.

[0036] The integrating amplifier circuit 21 functions as an integrating circuit that integrates the voltage V12 across the detection winding 12. In the first embodiment, the integrating amplifier circuit 21 amplifies and integrates the voltage V12 across the detection winding 12. The integrating amplifier circuit 21 supplies the amplified and integrated current to the feedback winding 13.

[0037] The arithmetic circuit 22 functions as an adder circuit that adds up the current I1 flowing from the detection winding 12 via the capacitive element 14 and the current I2 flowing through the feedback winding 13. The arithmetic circuit 22 also functions as an IV conversion circuit that converts the current obtained by the addition into a voltage whose magnitude is proportional to the magnitude of the current.

[0038] The arithmetic circuit 22 outputs to the measurement unit 2 a detection amount whose amplitude changes in proportion to the current value of the measurement current Im based on the current I2 flowing through the feedback winding 13 and the current I1 from the detection winding 12 passing through the capacitance element .

[0039] For example, when the winding directions of the detection winding 12 and the feedback winding 13 are the same, the arithmetic circuit 22 adds the current I2 flowing through the feedback winding 13 to the current I1 flowing through the detection winding 12. On the other hand, when the winding directions of the detection winding 12 and the feedback winding 13 are opposite to each other, the arithmetic circuit 22 subtracts the current I2 flowing through the feedback winding 13 from the current I1 flowing through the detection winding 12.

[0040] Next, a detailed configuration of the current sensor 1 will be described with reference to FIG.

[0041] 3 is a circuit diagram showing a detailed configuration example of the current sensor 1 in the first embodiment. In the first embodiment, both the detection winding 12 and the feedback winding 13 are wound in the same direction around a circular magnetic core 11, and the detection winding 12 and the feedback winding 13 have the same number of turns.

[0042] In this example, for convenience, the detection winding 12 and the feedback winding 13 are wound around different parts of the magnetic core 11, but the detection winding 12 and the feedback winding 13 may be wound around the same part of the magnetic core 11.

[0043] In the feedback winding 13, due to the electromagnetic induction phenomenon, an anti-magnetic flux φr is generated which cancels the measurement magnetic flux φm generated in the magnetic core 11 by the measurement current Im, and an induced current flows.

[0044] The detection winding 12 converts the residual magnetic flux (φm-φr), which is the difference between the measurement magnetic flux φm and the antimagnetic flux φr generated in the magnetic core 11, into a current I1 whose magnitude is proportional to its magnitude so that the residual magnetic flux (φm-φr) converges to zero through zero flux operation.

[0045] In the first embodiment, one end of the capacitive element 14 is connected to one end 12a of the detection winding 12 which is connected to the input terminal of the integral amplifier circuit 21, and the other end of the capacitive element 14 is connected to the other end 13b of the feedback winding 13 which is connected to the input terminal of the arithmetic circuit 22.

[0046] The integral amplifier circuit 21 includes a resistive element 211 , a capacitive element 212 , a resistive element 213 , and an operational amplifier 214 .

[0047] In the integral amplifier circuit 21, one end 12a of the detection winding 12 is connected to one end of a resistive element 211, and the other end of the resistive element 211 is connected to an inverting input terminal (-) of an operational amplifier 214. A capacitive element 212 is connected between the output terminal and the inverting input terminal (-) of the operational amplifier 214, and a resistive element 213 is connected in parallel to the capacitive element 212.

[0048] The other end 12b of the detection winding 12 is connected to the non-inverting input terminal (+) of the operational amplifier 214 and the reference potential G.

[0049] The arithmetic circuit 22 includes a resistive element 221 for converting the flowing current into a voltage proportional to the magnitude of the current, and outputs to the measurement unit 2 the voltage generated across the resistive element 211 by the current flowing into it.

[0050] In the arithmetic circuit 22, the other end 13b of the feedback winding 13 is connected to one end of a resistive element 221, and the other end of the resistive element 221 is connected to a reference potential G. One end of the resistive element 221 is connected to an input end of the measuring unit 2 as an output end that outputs a voltage proportional to an inflow current.

[0051] In the first embodiment, the detection winding 12 and the feedback winding 13 have the same number of turns, so that the detection sensitivity of both the detection winding 12 and the feedback winding 13 to changes in the magnetic flux generated in the magnetic core 11 is equal.

[0052] Therefore, in the CT operation region of the current sensor 1, it is possible to directly add the current I1 flowing through the detection winding 12 to the current I2 flowing through the feedback winding 13 without adjusting the output levels of the detection winding 12 and the feedback winding 13. Therefore, the arithmetic circuit 22 can be configured with one resistance element 221.

[0053] Next, the operation of the current sensor 1 will be described with reference to FIGS.

[0054] 4 is a diagram showing the gain frequency characteristics of the current sensor 1 in an ideal state, where the vertical axis represents the gain of the current sensor 1, and the horizontal axis represents the frequency f of the measurement current Im.

[0055] In FIG. 4, the gain frequency characteristic of the detection winding 12 is indicated by a solid line, the gain frequency characteristic of the feedback winding 13 is indicated by a dashed line, and the gain frequency characteristic of the entire current sensor 1 combining these is indicated by a dashed line.

[0056] The maximum gain Gmax shown on the vertical axis is the maximum gain, and means the maximum conversion efficiency of converting the magnetic flux interlinked with the windings into a current. Ideally, the maximum gain Gmax is a value obtained by dividing the resistance value R of the load resistor by the number of turns N of the detection winding 12 and the feedback winding 13.

[0057] Furthermore, the negative feedback operating region defined by the horizontal axis is the frequency band in which negative feedback operation is performed to converge the magnetic flux generated in the magnetic core 11 by the detection winding 12 to zero, and the CT operating region is the frequency band in which the feedback winding 13 operates as a current transformer.

[0058] In the first embodiment, the negative feedback operating region is a frequency band from the cutoff frequency fc of the current sensor 1 including the detection winding 12 and the feedback winding 13 to the cutoff frequency fc_ct of the feedback winding 13. The CT operating region is a frequency band higher than the cutoff frequency fc_ct of the feedback winding 13.

[0059] In the negative feedback operating region, the measurement magnetic flux φm, which is one of the magnetic fluxes produced by the measurement current Im and which interlinks with the feedback winding 13, generates an anti-magnetic flux φr in the magnetic core 11 in the opposite direction to the direction of the measurement magnetic flux φm. As a result, a current is generated in the feedback winding 13 so as to cancel out the measurement magnetic flux φm.

[0060] At this time, a differential magnetic flux between the measurement magnetic flux φm and the antimagnetic flux φr remains in the magnetic core 11, and this residual magnetic flux generates a voltage V12 between one end 12a and the other end 12b of the detection winding 12. At this time, the current passing through the capacitance element 14 is approximately zero, and the voltage V12 between one end 12a and the other end 12b of the detection winding 12 is input to the integral amplifier circuit 21.

[0061] A voltage V12 output between one end 12a and the other end 12b of the detection winding 12 is input to an integrating amplifier circuit 21, where the voltage V12 output from the detection winding 12 is integrated and amplified.

[0062] Then, the current output from the integral amplifier circuit 21 is input to one end 13a of the feedback winding 13, and the output current of the integral amplifier circuit 21 is added to the current generated in the feedback winding 13. As a result, a current I2 having a magnitude equal to the sum of the output current of the integral amplifier circuit 21 and the induced current of the feedback winding 13 is output from the other end 13b of the feedback winding 13 to the arithmetic circuit 22.

[0063] As shown in FIG. 4, as the frequency of the measurement current Im approaches the cutoff frequency fc of the current sensor 1 from a frequency lower than fc, the gain of the current sensor 1 increases monotonically, and the gain of the current sensor 1 reaches a maximum gain Gmax near the cutoff frequency fc.

[0064] In the negative feedback operating region, the gain of the current sensor 1 is maintained at the maximum gain Gmax, and as the frequency of the measurement current Im increases, the gain of the current transformer formed by the object to be measured Lm and the feedback winding 13 increases monotonically toward the maximum gain Gmax.

[0065] In the CT operating region, the measurement magnetic flux φm is detected only by the feedback winding 13. At this time, the feedback winding 13 converts the measurement magnetic flux φm into a current I2 whose magnitude is proportional to the magnitude of the measurement magnetic flux φm with higher accuracy than in the negative feedback operating region.

[0066] That is, a current flows through the feedback winding 13 so that no magnetic flux remains in the magnetic core 11, in other words so as to cancel all of the measurement magnetic flux φm. Then, only the induced current generated in the feedback winding 13 is output as a current I2 from the other end 13b, and the current I2 is input to one end of the resistor element 221 in the arithmetic circuit 22.

[0067] In the CT operating region, as the frequency of the measurement current Im increases, the permeability of the magnetic core 11 decreases and the influence of the parasitic capacitance of each of the detection winding 12 and the feedback winding 13 and the line capacitance generated between the detection winding 12 and the feedback winding 13 increases. As a result, the conversion efficiency of the feedback winding 13 decreases, and magnetic flux remains in the magnetic core 11. This causes an induced current to flow in the detection winding 12.

[0068] The current I2 flowing through the detection winding 12 causes the gain to drop significantly from the maximum gain Max near the upper limit of the CT operating region. At this time, the frequency of the measurement current Im is higher than the negative feedback operating region, so the integral amplifier circuit 21 shown in Fig. 3 functions as a low-pass filter, and therefore the integral amplifier circuit 21 does not operate, and the operational amplifier 214 does not integrate or amplify the current output from one end 12a of the detection winding 12. Therefore, the current flowing through the detection winding 12 is not superimposed on the feedback winding 13 through the integral amplifier circuit 21.

[0069] In the first embodiment, since the detection winding 12 and the feedback winding 13 have the same number of turns, the current I1 flowing through the detection winding 12 is directly added to the current I2 flowing through the feedback winding 13 via the capacitance element 14. This compensates for the decrease in the current I2 caused by the decrease in the conversion efficiency of the feedback winding 13.

[0070] Then, the current I1 flowing through the detection winding 12 is added to the current I2 flowing through the feedback winding 13 as an added current (I1+I2), and this current is input to one end of the resistance element 221 in the arithmetic circuit 22.

[0071] Here, the capacitance value of the capacitive element 14 will be briefly described.

[0072] It is desirable to set the capacitance value of the capacitance element 14 so that the current I1 flowing through the detection winding 12 passes through the capacitance element 14 in the CT operating region.

[0073] The capacitance value of the capacitive element 14 is determined based on the cutoff frequency of a high-pass filter configured by the series-connected capacitive element 14 and resistive element 221. The cutoff frequency fc_hpf of the high-pass filter is calculated using the capacitance value C of the capacitive element 14 and the resistance value R of the resistive element 221, as in the following equation (1).

[0074] fc_hpf = 1 / (2πCR) (1)

[0075] Specifically, the capacitance value C of the capacitive element 14 is determined so that the cutoff frequency fc_hpf of the high-pass filter is a predetermined frequency that is higher than the cutoff frequency fc_ct of the feedback winding 13 and lower than the upper limit of the CT operating region.

[0076] The cutoff frequency fc_ct of the feedback winding 13 is calculated by the following equation (2) using the inductance value L of the feedback winding 13 and the resistance value R of the resistive element 221 when the winding resistance value is not taken into consideration. fc_ct = R / (2πL) (2)

[0077] FIG. 5 is a diagram showing a simulation result of the gain frequency characteristic of the current sensor 1 in the first embodiment.

[0078] Here, the vertical axis indicates the gain of the current sensor 1 on a linear scale, and the horizontal axis indicates the frequency of the measurement current Im on a logarithmic scale. The gain frequency characteristic obtained by the simulation result of the current sensor 1 in the first embodiment is indicated by a solid line, and the gain frequency characteristic of a current sensor not including the capacitive element 14 as a comparative example is indicated by a dashed line.

[0079] As shown in FIG. 5, in the current sensor of the comparative example, as the frequency of the measurement current Im increases from around 1 MHz, the gain gradually decreases from the maximum gain Gmax of approximately -34 dB, and then decreases sharply from around 20 MHz.

[0080] In contrast, in the current sensor 1 of the first embodiment, even in the high frequency band where the gain of the current sensor of the comparative example is reduced, the current flowing through the detection winding 12 is supplied through the capacitive element 14 to the calculation circuit 22 which calculates the current flowing through the feedback winding 13.

[0081] As a result, the arithmetic circuit 22 outputs a voltage value corresponding to the magnitude of the current obtained by adding the current I1 flowing through the detection winding 12 to the current I2 flowing through the feedback winding 13. In other words, the detection amount of the arithmetic circuit 22 is corrected by the current I1 of the detection winding 12.

[0082] In this way, in the current sensor 1 of the first embodiment, the current I1 generated in the detection winding 12 in association with the decrease in the conversion efficiency of the feedback winding 13 is used to compensate for the decrease in the current I2 flowing through the feedback winding 13. Therefore, the decrease in the gain of the current sensor 1 in the high frequency band can be suppressed more effectively than in the current sensor of the comparative example.

[0083] Next, the effects of the current sensor 1 of the first embodiment will be described.

[0084] The current sensor 1 in the first embodiment includes a magnetic core 11 surrounding a measurement object Lm through which a measurement current Im flows, a detection winding 12 functioning as an AC detection winding wound around the magnetic core 11, and a feedback winding 13 functioning as a feedback winding wound around the magnetic core 11. A current is supplied to the feedback winding 13 by an integral amplifier circuit 21 functioning as an integrating circuit that integrates the current flowing through the detection winding 12.

[0085] Further, the current sensor 1 includes a capacitive element 14, and outputs a voltage output from an arithmetic circuit 22. The arithmetic circuit 22 adds the current output from the detection winding 12 via the capacitive element 14 and the current output from the feedback winding 13, and converts the sum into a voltage proportional to the magnitude of the current obtained by the addition.

[0086] According to this configuration, in the CT operation region where the feedback winding 13 operates as a current transformer, the higher the frequency of the measurement current Im, the greater the proportion of the current flowing through the detection winding 12 that is input to the arithmetic circuit 22 via the capacitance element 14.

[0087] At this time, as the frequency of the measurement current Im increases, the magnetic permeability of the magnetic core 11 decreases and the influence of the parasitic capacitance of each of the detection winding 12 and the feedback winding 13 and the line capacitance between the windings increases. As a result, the magnetic flux cancellation action by the feedback winding 13 weakens, magnetic flux remains in the magnetic core 11, and an induced current flows in the detection winding 12. Then, the induced current flows in the detection winding 12, and the detection winding 12 and the feedback winding 13 wound around the magnetic core 11 operate as a current transformer, resulting in a large detection error in the measurement current.

[0088] Therefore, in the CT operation region, the detection winding 12 operates as a second feedback winding by using the current passing through the capacitive element 14, so that it is possible to compensate for the decrease in the induced current in the feedback winding 13 caused by the decrease in the conversion efficiency of the feedback winding 13. Therefore, it is possible to suppress the detection error of the measurement current Im in the high frequency region.

[0089] In the first embodiment, the detection winding 12 and the feedback winding 13 have the same number of turns.

[0090] According to this configuration, by making the number of turns of the detection winding 12 and the number of turns of the feedback winding 13 equal, the degree of influence on the induced current due to the decrease in the conversion efficiency of the detection winding 12 and the feedback winding 13 becomes equal. Therefore, it is not necessary to adjust the output level of the current I2 flowing through the feedback winding 13 and the output level of the current I1 of the detection winding 12 passing through the capacitive element 14, and the current I2 and the current I1 can be directly added together.

[0091] Therefore, since it is possible to detect the added current in the arithmetic circuit 22 using one resistance element 221, the arithmetic circuit 22 can have a simple configuration.

[0092] In the first embodiment, the capacitive element 14 is connected between the output terminal of the detection winding 12 and the output terminal of the feedback winding 13 .

[0093] According to this configuration, since the capacitance element 14 is connected between the output terminal of the detection winding 12 and the output terminal of the feedback winding 13, it is possible to arrange the capacitance element 14 close to the detection winding 12 and the feedback winding 13. Therefore, the wiring distance from the output terminals of the detection winding 12 and the feedback winding 13 to the capacitance element 14 can be made shorter than in a configuration in which the wiring is made on the board on which the integral amplifier circuit 21 and the arithmetic circuit 22 are mounted. Therefore, noise mixed into the capacitance element 14 can be suppressed.

[0094] In the first embodiment, the arithmetic circuit 22 has a resistive element 221, and the series-connected capacitive element 14 and the resistive element 221 configure a filter. The capacitance value C of the capacitive element 14 is determined so that the cutoff frequency fc_hpf of the filter expressed by the formula (1) is a predetermined frequency higher than the cutoff frequency fc_ct of the feedback winding 13.

[0095] According to this configuration, the capacitive element 14 can supply the current I1 flowing through the detection winding 12 to the arithmetic circuit 22 in the high frequency band within the CT operating region.

[0096] Moreover, the current sensor 1 in the first embodiment includes an integral amplifier circuit 21 and an arithmetic circuit 22.

[0097] According to this configuration, the arithmetic circuit 22 can output a detection amount in which detection errors caused by a decrease in the conversion efficiency of the feedback winding 13 in the high frequency band are suppressed.

[0098] Moreover, the measurement device 100 in the first embodiment includes a current sensor 1 and a measurement unit 2 that calculates a measurement amount for the measurement object Lm based on a detection amount detected by the current sensor 1.

[0099] According to this configuration, it is possible to suppress detection errors caused by a decrease in the conversion efficiency of the feedback winding 13 in the high frequency band, and therefore it is possible to improve the accuracy of the measurement quantity in the high frequency band.

[0100] In the current sensor 1 of the first embodiment, the detection winding 12 and the feedback winding 13 have the same number of turns, but the numbers of turns may be different from each other. Therefore, a detailed configuration of the current sensor 1 in which the detection winding 12 and the feedback winding 13 have different numbers of turns from each other will be described below as a second embodiment.

[0101] Second Embodiment FIG. 6 is a circuit diagram showing a detailed configuration of a current sensor 1A according to the second embodiment.

[0102] In the second embodiment, the number of turns N1 of the detection winding 12 and the number of turns N2 of the feedback winding 13 are different from each other. Accordingly, the current sensor 1A differs from the first embodiment in that it includes an arithmetic circuit 22A instead of the arithmetic circuit 22 of the current sensor 1 shown in Fig. 3, and the connection location of the capacitive element 14 is changed. Therefore, only the points of configuration that are different from the first embodiment will be described, and the same reference numerals will be used to denote the other configurations, and description thereof will be omitted.

[0103] The arithmetic circuit 22A adjusts the output level of the current I2 flowing through the feedback winding 13 and the output level of the current I1 passing from the detection winding 12 to the capacitive element 14, and outputs a sum current value of the adjusted current I2 and current I1. The arithmetic circuit 22A includes a resistive element 221, a resistive element 222, and an adder 223.

[0104] The resistive element 221 is a second resistor that functions as a second detection element that detects the current I2 flowing through the feedback winding 13. The resistive element 221 outputs to the adder 223 a voltage whose magnitude is proportional to the magnitude of the current I2 flowing through the feedback winding 13.

[0105] The resistance value of the resistive element 221 is determined based on the ratio between the number of turns N1 of the detection winding 12 and the number of turns N2 of the feedback winding 13 so that the output levels of both the detection winding 12 and the feedback winding 13 are equal.

[0106] Specifically, the resistance value R2 of the resistive element 221 is determined using the number of turns N1 of the detection winding 12, the number of turns N2 of the feedback winding 13, and the resistance value R1 of the resistive element 222 so that the relationship of the following equation (3) is established.

[0107] R2 / R1 = N2 / N1 (3)

[0108] The resistive element 222 is a first resistor that functions as a first detection element for detecting a current passing through the capacitive element 14 out of the current I2 flowing through the detection winding 12. The resistive element 222 outputs a voltage having a magnitude proportional to the magnitude of the current flowing through the detection winding 12 to the adder 223. The resistance value R1 of the resistive element 222 is determined so that the relationship of the above formula (3) is established, similar to the resistance value R2 of the resistive element 221.

[0109] The adder 223 adds the voltage generated across the resistive element 221 and the voltage generated across the resistive element 222. The adder 223 outputs to the measurement unit 2 an added voltage of the voltage value obtained by the addition.

[0110] The voltage across resistor element 221 and the voltage across resistor element 222 indicate voltage values ​​in which the difference in output level between the detection winding 12 and the feedback winding 13 resulting from the difference between the number of turns N2 of the feedback winding 13 and the number of turns N1 of the detection winding 12 has been adjusted.

[0111] Therefore, even if the detection winding 12 and the feedback winding 13 have different numbers of turns, it is possible to compensate for the decrease in current I2 caused by a decrease in the conversion efficiency of the feedback winding 13 by using the current I2 of the detection winding 12 passing through the capacitance element 14.

[0112] In the arithmetic circuit 22A, the other end 13b of the feedback winding 13 is connected to one end of a resistive element 221, and the other end of the resistive element 221 is connected to a reference potential G. In addition, one end of the capacitive element 14 is connected to one end 12a of the detection winding 12, the other end of the capacitive element 14 is connected to one end of a resistive element 222, and the other end of the resistive element 222 is connected to the reference potential G.

[0113] One end of the resistive element 222 is connected to a first input terminal of the adder 223, and one end of the resistive element 221 is connected to a second input terminal of the adder 223. In addition, an input terminal of the measurement unit 2 is connected to an output terminal of the adder 223.

[0114] In this way, even if the detection winding 12 and the feedback winding 13 have different numbers of turns, by determining the resistance values ​​of the resistive elements 221 and 222 based on the ratio of the numbers of turns of each winding, the amount of decrease in current I2 can be appropriately compensated for using the current passing through the capacitive element 14.

[0115] The capacitance value of the capacitive element 14 is determined by using the above formula (1) in the same manner as in the first embodiment. Specifically, the capacitance value C of the capacitive element 14 is set so that the cutoff frequency fc_hpf of the high-pass filter having the capacitive element 14 and the resistive element 222 connected in series is higher than the cutoff frequency fc_ct of the feedback winding 13.

[0116] Next, the effects of the current sensor 1A in the second embodiment will be described.

[0117] In the second embodiment, the arithmetic circuit 22A includes a resistive element 222 that functions as a first resistor for detecting a current I1 flowing through the detection winding 12, and a resistive element 221 that functions as a second resistor for detecting a current I2 flowing through the feedback winding 13.

[0118] The arithmetic circuit 22A further includes an adder 223 that functions as an adder that adds up the detection amounts detected by the resistive element 222 and the resistive element 221. The capacitive element 14 is connected between the input terminal of the integral amplifier circuit 21 and one input terminal of the resistive element 222.

[0119] According to this configuration, by determining the resistance values ​​of the resistive elements 222 and 221 based on the ratio of the number of turns N1 and N2 of the detection winding 12 and the feedback winding 13, it is possible to adjust the output levels of the detection winding 12 and the feedback winding 13.

[0120] For example, one end of the resistive element 222 is connected to the output end of the detection winding 12, one end of the resistive element 221 is connected to the output end of the feedback winding 13, and the other end of the resistive element 222 and the other end of the resistive element 221 are each connected to a reference potential G. And, as in the above formula (3), the ratio of the number of turns N1 of the detection winding 12 to the resistance value R1 of the resistive element 222 is equal to the ratio of the number of turns N2 of the feedback winding 13 to the resistance value R2 of the resistive element 221.

[0121] According to this configuration, the output levels of the detection winding 12 and the feedback winding 13, which have different numbers of turns, can be adjusted so that the residual magnetic flux (φm-φr) converges to zero. Therefore, even if the detection winding 12 and the feedback winding 13 have different numbers of turns, the amount of decrease in the current I2 caused by the decrease in the conversion efficiency of the feedback winding 13 can be appropriately compensated for by using the current I2 passing from the detection winding 12 through the capacitive element 14 in the high frequency band.

[0122] Third embodiment FIG. 7 is a diagram illustrating a schematic configuration of a current sensor 1B according to the third embodiment.

[0123] The current sensor 1B includes a magnetic detection element in addition to the configuration of the sensor unit 10 shown in Fig. 2. The current sensor 1B in the third embodiment includes a sensor unit 10B and a circuit unit 20B.

[0124] The sensor unit 10B includes a fluxgate 110 as a magnetic detection element disposed in all or part of the annular magnetic core 11 in addition to the configuration of the sensor unit 10 shown in FIG.

[0125] The fluxgate 110 in the third embodiment includes annular magnetic cores 111 and 112 , an excitation winding 15 wound around the magnetic core 111 , and an excitation winding 16 wound around the magnetic core 112 .

[0126] The magnetic cores 111 and 112 are made of, for example, a permalloy material, and are arranged next to each other within the magnetic core 11 .

[0127] The excitation winding 15 and the excitation winding 16 are wound around the magnetic cores 111 and 112, respectively, so that the magnetic fluxes generated in the magnetic cores 111 and 112 cancel each other out.

[0128] The circuit section 20B includes an oscillator circuit 31, an excitation circuit 32, a detection circuit 33, an adder circuit 34, and a feedback circuit 35 in addition to the components of the circuit section 20 shown in FIG.

[0129] The oscillator circuit 31 generates an AC signal having a specific frequency f, and generates a synchronization signal having a double frequency 2f, which is twice the specific frequency. The oscillator circuit 31 then outputs the generated AC signal having the specific frequency f to the excitation circuit 32, and outputs the generated synchronization signal having the double frequency 2f to the detection circuit 33.

[0130] The excitation circuit 32 generates an AC excitation current of a frequency f based on the AC signal output from the oscillation circuit 31. The excitation circuit 32 outputs the generated AC excitation current to both the excitation windings 15 and 16.

[0131] As a result, a magnetic flux is generated in the magnetic core 111 which is the difference between a first magnetic flux generated by the excitation winding 15 and a second magnetic flux generated by the excitation winding 16 and having a direction opposite to that of the first magnetic flux.

[0132] The detection circuit 33 synchronously detects the difference signal between the voltage signal of the excitation winding 15 and the voltage signal of the excitation winding 16 based on the synchronization signal of the oscillation circuit 31. The detection circuit 33 outputs a signal proportional to the current to be measured generated in the magnetic cores 111 and 112.

[0133] The adder circuit 34 adds the output signal of the integral amplifier circuit 21 and the output signal of the detection circuit 33 together, and outputs the resulting signal to a feedback circuit 35 .

[0134] The feedback circuit 35 converts the sum signal obtained from the addition circuit 34 into a current and outputs it to the feedback winding 13 .

[0135] In this way, by including a flux gate 110 in addition to the configuration of current sensor 1 or current sensor 1A, current sensor 1B is able to obtain a detection quantity whose amplitude changes in proportion to the value of the measured current Im in a frequency band lower than the negative feedback operating region.

[0136] In the third embodiment, the fluxgate 110 has a configuration in which the two excitation windings 15 and 16 are wound around one magnetic core 111, but the present invention is not limited to this.

[0137] For example, the fluxgate 110 may have two annular magnetic cores, with the excitation winding 15 wound around one magnetic core and the excitation winding 15 wound around the other magnetic core. In this case, one magnetic core around which the excitation winding 15 is wound and the other magnetic core around which the excitation winding 16 is wound are arranged side by side within the magnetic core 11.

[0138] In addition, in the third embodiment, the two excitation windings 15 and 16 are wound around the magnetic core 111, but it may be configured so that only one of them is wound around the magnetic core 11.

[0139] In addition, although the flux gate 110 is used as the magnetic detection element in the third embodiment, a Hall element may be used instead. In this case, the Hall element is disposed in a part of the magnetic core 11, and an output signal of the Hall element is input to the adder circuit 34, where the output signal of the integral amplifier circuit 21 and the output signal of the Hall element are added together.

[0140] Next, the effects of the current sensor 1B in the third embodiment will be described.

[0141] A current sensor 1B in the third embodiment has the configuration of the first and second embodiments.

[0142] According to this configuration, similar to the effects of the first and second embodiments, in the CT operation region, it is possible to compensate for the decrease in induced current in the feedback winding 13 that accompanies a decrease in the conversion efficiency of the feedback winding 13 by using the current passing through the capacitive element 14. Therefore, it is possible to suppress detection errors for the measurement current Im in the high frequency region.

[0143] Moreover, the current sensor 1B in the third embodiment further includes a magnetic detection element that detects the magnetic flux generated in the magnetic core 11. This makes it possible to improve the detection accuracy in a frequency band lower than the CT operating region.

[0144] Although the embodiments of the present invention have been described above, the above-mentioned embodiments merely show some of the application examples of the present invention, and it is not intended that the technical scope of the present invention be limited to the specific configurations of the above-mentioned embodiments. [Explanation of symbols]

[0145] 1, 1A, 1B Current Sensor 2 Measuring part 11 Magnetic core 12 Detection winding (AC detection winding) 13 Feedback Winding 21 Integrating amplifier circuit (integrating circuit) 22 Arithmetic circuit 100 Measuring Equipment 110 Fluxgate (magnetic detection element) Lm Measurement target Im measurement current

Claims

1. a magnetic core surrounding a measurement object through which a current flows; an AC detection winding wound around the magnetic core; a feedback winding that is wound around the magnetic core and receives a current output from an integrating circuit that integrates a voltage between both ends of the AC detection winding; a capacitance element; The current output from the AC detection winding via the capacitance element and the current output from the feedback winding are added together, and the resulting sum is converted into a voltage by a calculation circuit, which outputs the converted voltage. Current sensor.

2. 2. The current sensor according to claim 1, The AC detection winding and the feedback winding have an equal number of turns. Current sensor.

3. 2. The current sensor according to claim 1, the capacitance element is connected between an output terminal of the AC detection winding and an output terminal of the feedback winding. Current sensor.

4. 2. The current sensor according to claim 1, the AC detection winding and the feedback winding have different numbers of turns; the arithmetic circuit includes a first resistor and a second resistor, One end of the first resistor is connected to the output end of the AC detection winding, one end of the second resistor is connected to the output end of the feedback winding; the other end of the first resistor and the other end of the second resistor are each connected to a reference potential; a ratio of the number of turns of the AC detection winding to the resistance value of the first resistor is equal to a ratio of the number of turns of the feedback winding to the resistance value of the second resistor; Current sensor.

5. 2. The current sensor according to claim 1, The magnetic core further includes a magnetic detection element for detecting a magnetic flux generated in the magnetic core. Current sensor.

6. 2. The current sensor according to claim 1, the arithmetic circuit has one or more resistance elements, a capacitance value of the capacitance element is determined so that a cutoff frequency of a filter formed of the capacitance element and at least one of the resistance elements connected in series becomes a predetermined frequency higher than a cutoff frequency of a current transformer formed of the feedback winding and the resistance element; Current sensor.

7. 2. The current sensor according to claim 1, The integration circuit and the arithmetic circuit Current sensor.

8. A current sensor according to any one of claims 1 to 7; a measurement unit that calculates a measurement amount for the measurement object based on the amount detected by the current sensor; A measuring device comprising:

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