Open winding motor drive device

Abstract

To provide an open winding motor drive device capable of generating a three-phase PWM signal by two inverters so as to surely detect a high frequency component of a three-phase current necessary for estimating a magnetic pole position within a carrier wave period.SOLUTION: An open winding motor drive device controls a primary-side inverter and a secondary-side inverter connected to both ends of an open winding structure motor by PWM. A PWM signal generation unit divides, into six sectors, a hexagonal spatial vector whose center is a point where two second switching patterns are located that generates no zero-phase voltage for sixty-four voltage vectors and generates no voltage acting between phases of the motor and whose vertex is a point where two first switching patterns are located that generates no zero-phase voltage and generates a voltage acting between phases of the motor. The PWM signal generation unit shifts phases by synchronizing on a primary side and a secondary side so that a current change amount detection unit provides a time for detecting a difference of a three-phase current value twice for each of the maximum phase, the middle phase, and the minimum phase that are obtained by classifying a duty ratio of three phases of the primary-side inverter and the secondary-side inverter of each sector by its magnitude.SELECTED DRAWING: Figure 1

Description

[Technical Field]

[0001] An embodiment of the present invention relates to an apparatus for estimating the magnetic pole position of a motor based on three-phase currents of a motor with an open winding structure driven by two inverters. [Background technology]

[0002] Conventionally, magnetic pole position has been detected to properly control DC brushless motors, which are a type of permanent magnet motor that uses a permanent magnet in the rotor. Detecting magnetic pole position means detecting the electrical angle phase, which is the position on the electrical angle coordinate system of the motor. Magnetic pole position can be detected using position sensors such as rotary encoders, resolvers, and Hall elements. However, due to cost and structural constraints, it is sometimes not possible to install a position sensor. For example, in the case of compressor motors for refrigeration cycles, the motor is built into the refrigerant-filled space inside the compressor, which is a sealed container, so a position sensor cannot be installed.

[0003] Therefore, there are methods for estimating the magnetic pole position from current and voltage information without using a position sensor. Such methods include, for example, an induced voltage type and an inductance type. The induced voltage type calculates an induced voltage proportional to the motor speed from the input voltage and current to the motor, and estimates the magnetic pole position based on this induced voltage. This method utilizes the fact that the induced voltage generated by the rotation of the motor changes depending on the electrical angle of the motor, which is the magnetic pole position.

[0004] The induced voltage type can achieve sufficient accuracy in the high-speed range where the motor rotation speed is high. However, in the low-speed range where the rotation speed is low, the amplitude of the induced voltage becomes small or does not occur at all, making it impossible to estimate the position accurately when the motor is stopped or moving at low speeds. On the other hand, the inductance-based estimation method calculates the motor inductance from current and voltage information and estimates the magnetic pole position. This method utilizes the fact that the motor inductance changes with a period that is twice as long as the motor's electrical angle. Several inductance-based estimation methods have been proposed, for example, which apply an AC voltage signal to the motor for sensing that is unrelated to the drive frequency, and estimate the magnetic pole position from the relationship between voltage and current.

[0005] The frequency of the AC voltage signal applied to find inductance using this method is about several hundred Hz to several kHz, which is below the carrier frequency. However, the motor current ripple frequency falls within the audible range for humans, which increases noise. To address this problem, Patent Document 1 proposes a method of generating high-frequency current amplitude and estimating the magnetic pole position while suppressing noise by increasing or decreasing the duty of one phase of a three-phase PWM signal in both directions, lagging or leading, based on an arbitrary phase of the carrier cycle; increasing or decreasing the duty of another phase in one direction, lagging or leading, based on an arbitrary phase of the carrier cycle; and increasing or decreasing the duty of the remaining phase in the opposite direction, based on an arbitrary phase of the carrier cycle.

[0006] This method detects each of the three phase currents twice at four fixed timing points within the carrier cycle, finds the difference between them, and calculates the estimated magnetic pole position of the motor based on the amount of current change.This method therefore has the advantage that it is possible to accurately calculate the estimated magnetic pole position using a position sensorless system even in extremely low speed regions, including zero speed, without relying on the motor's electrical characteristics, such as a small salient pole ratio or large inductance, and can be easily applied to a variety of motors. [Prior art documents] [Patent documents]

[0007] [Patent Document 1] Japanese Patent Application Laid-Open No. 2015-126565 Summary of the Invention [Problem to be solved by the invention]

[0008] The above technology is applicable when one motor is driven by one inverter, but it cannot be applied when two inverters are used to drive a motor with an open winding structure, where three independent phase windings are provided with six winding terminals. Therefore, we provide an open winding motor drive device that can generate three-phase PWM signals using two inverters so that the three-phase current change amounts required for magnetic pole position estimation can be reliably detected within a carrier wave period in a motor with an open winding structure. [Means for solving the problem]

[0009] The open winding motor drive device of the embodiment includes a primary side inverter connected to three output terminals of a motor having an open winding structure, in which three phase windings are independent from each other and the motor has six output terminals; a secondary-side inverter connected to the remaining three output terminals of the motor; a current detection unit that detects a three-phase current flowing through the motor; a current change amount detection unit that outputs the difference between the two detected three-phase current values ​​as a current change amount; a magnetic pole position estimating unit that estimates a magnetic pole position of the motor based on the amount of change in current; a control unit for controlling the current supplied to the motor and the rotation speed; a zero-phase current suppression unit that suppresses zero-phase currents flowing in the same direction through three-phase windings of the motor along paths between the primary and secondary inverters and the motor; a PWM signal generating unit that compares a three-phase duty ratio calculated based on a command voltage and a DC voltage obtained from the control unit and the zero-phase current suppressing unit with a carrier wave and generates three-phase PWM signals for each of the primary and secondary inverters; the PWM signal generating unit divides a hexagonal space vector into six sectors, with the 64 voltage vectors being combinations of on / off patterns of the primary and secondary side inverters, the hexagonal space vector having a center at a point where two second switching patterns that do not generate a zero-phase sequence voltage and do not generate a voltage acting between the phases of the motor are located and vertices at points where two first switching patterns that do not generate a zero-phase sequence voltage and generate a voltage acting between the phases of the motor are located, The phases of the primary and secondary inverters in each sector are shifted synchronously so that the current change amount detection unit detects the difference in the three-phase current values ​​twice for each of the maximum, medium, and minimum phases classified by their magnitude.

[0010] In addition, the open winding motor drive device of the embodiment similarly includes the above-mentioned configuration from the primary side inverter to the PWM signal generation unit, but excluding the zero-phase current suppression unit, the PWM signal generation unit increases or decreases the pulse width of a first phase of the three-phase PWM signal of the primary side inverter in both directions of the delay side and the advance side based on the phase of the center or the front and rear ends of a carrier cycle; The second phase increases or decreases the pulse width in one direction, either the leading or trailing side, based on the phase of the leading or trailing end of the carrier cycle. The third phase increases or decreases the pulse width in the opposite direction based on the phase opposite to the second phase, Regarding the three-phase PWM signal of the secondary side inverter, the first phase is based on a phase different from the first phase of the primary side inverter, and the pulse width is increased or decreased in both directions on the lagging side and the leading side; The second and third phases increase and decrease the pulse width in the opposite direction based on the same phases as the second and third phases of the primary side inverter, respectively; The current change amount detection unit detects the difference in current values ​​of three phases or two phases twice near the leading and trailing ends of a carrier cycle. [Brief explanation of the drawings]

[0011] [Figure 1]FIG. 1 is a functional block diagram showing a control device for an open-winding motor drive system according to a first embodiment. [Figure 2] Diagram showing an open winding motor drive system [Figure 3] Diagram showing the configuration of an air conditioner [Figure 4] Diagram showing the PWM signals of the primary and secondary inverters [Figure 5] A diagram showing three current change amounts including position information and the on / off patterns for detecting these current change amounts. [Figure 6] A graph showing the amount of current change obtained through simulation verification [Figure 7] Diagram showing the 64 voltage vectors that can be generated by an open-winding motor drive system [Figure 8] FIG. 2 is a functional block diagram showing a control device for an open-winding motor drive system according to a second embodiment. [Figure 9] Flowchart for calculating the phase shift amount of a PWM signal (part 1) [Figure 10] Flowchart for calculating the phase shift amount of a PWM signal (part 2) [Figure 11] Flowchart for calculating the phase shift amount of a PWM signal (part 3) [Figure 12] FIG. 10 is a diagram showing a PWM signal according to a second embodiment that can detect a current change amount while suppressing a zero-phase current of a carrier component. [Figure 13] Enlarged view of Figure 12 for sector 0 [Figure 14] Diagram showing the PWM signals of the primary and secondary inverters [Figure 15] FIG. 10 is a diagram showing the amount of current change obtained by simulating and verifying the second embodiment without operating the zero-phase current suppression unit. [Figure 16] FIG. 10 is a diagram showing the amount of current change obtained by simulating and verifying the second embodiment by operating the zero-phase current suppression unit. [Figure 17] FIG. 10 is a diagram showing a zero-phase voltage and a zero-phase current of a carrier frequency component due to a PWM signal of the first embodiment. [Figure 18]A diagram showing conventional PWM signal generation and space vector modulation to suppress zero-phase current of the carrier component. [Figure 19] FIG. 10 is a diagram showing a third embodiment of an open winding motor drive system. [Figure 20] Functional block diagram showing a control device for an open-winding motor drive system [Figure 21] A diagram showing three current change amounts including position information and the on / off patterns for detecting these current change amounts. [Figure 22] Diagram showing the PWM signals of the primary and secondary inverters [Figure 23] FIG. 10 is a diagram showing the amount of current change obtained by simulation verification of the third embodiment. [Figure 24] Diagram showing equations (1) and (2) [Figure 25] Diagram showing equations (3) to (5) DETAILED DESCRIPTION OF THE INVENTION

[0012] (First embodiment) A first embodiment will be described below with reference to Figs. 1 to 7. Fig. 2 is a diagram showing the configuration of a motor drive system according to this embodiment. Motor 1 is, for example, a three-phase permanent magnet synchronous motor, and the three phase windings are not connected to each other, with both terminals being open. In other words, motor 1 has six winding terminals.

[0013] The primary-side inverter 2 and the secondary-side inverter 3 are each configured by connecting switching elements, such as IGBTs U, V, W, and X, Y, and Z, in a three-phase bridge configuration, and are connected in parallel to a DC power supply 4. The DC power supply 4 may be one that converts AC power to DC. Each phase output terminal of the primary-side inverter 2 is connected to one winding terminal of the motor 1, and each phase output terminal of the secondary-side inverter 3 is connected to the other winding terminal. In this embodiment, the primary-side inverter 2 and the secondary-side inverter 3 share a DC link, so that a zero-phase current flows in the same direction through the three-phase windings of the motor 1. This zero-phase current is divided into a low-frequency current that flows at three times the fundamental frequency of the phase currents applied to the motor 1, and a current with a carrier frequency component that flows due to the switching of the primary-side and secondary-side inverters 2 and 3.

[0014] The current sensors 5U, 5V, and 5W are sensors that detect the phase currents Iu, Iv, and IW of the motor 1, and correspond to a current detection unit. The voltage sensor 6 detects the voltage Vdc of the DC power supply. The control device 10 shown in FIG. 1 generates switching signals to be applied to the gates of the IGBTs that make up the primary-side and secondary-side inverters 2 and 3, based on the phase currents Iu, Iv, and IW detected by the current sensors 5U, 5V, and 5W, and the DC voltage Vdc detected by the voltage sensor 6. The control device 10 corresponds to a control unit.

[0015] A speed control unit 11 constituting the control device 10 generates and outputs a q-axis current command Iqref, for example, by performing a PI (Proportional Integral) calculation on the difference between a speed command ωref input from a higher-level control device in the system that drives the motor 1 and the motor speed ω estimated by the magnetic pole position estimation unit 12 so that they coincide with each other. A flux-weakening control unit 13 similarly generates and outputs a d-axis current command Idref, for example, by performing a PI calculation on the difference between a DC voltage Vdc and a voltage amplitude Vdq of the dq axes.

[0016] The current control unit 14 generates and outputs d- and q-axis voltage commands Vd and Vq by, for example, performing PI calculation on the difference between the input q-axis current command Iqref and d-axis current command Idref and the currents Id and Iq detected by a coordinate conversion unit (uvw / dqz) 15. Based on the magnetic pole position θ estimated by the magnetic pole position estimation unit 12, the coordinate conversion unit (uvw / dqz) 15 converts the phase currents Iu_detect, Iv_detect, and IW_detect detected by a phase current and current change amount detection unit 16 via the current sensor 5 into currents Id and Iq on the d- and q-axis coordinates and a zero-phase current I0 used for vector control by using equation (1) shown in Fig. 24.

[0017] The zero-phase current suppression unit 17 generates and outputs a zero-phase voltage command V0ref, for example, by proportional resonance control, from the zero-phase current command I0ref, which is generally zero, the zero-phase current I0 input from the coordinate transformation unit (uvw / dqz) 15, and the motor speed ω input from the magnetic pole position estimation unit 12. The proportional resonance control suppresses the zero-phase current of the triple component by controlling so as to increase the responsiveness to the triple component of the fundamental frequency of the motor phase current.

[0018] A coordinate conversion unit (dqz / uvw) 18 converts the d- and q-axis voltage commands Vd, Vq and the zero-phase voltage command V0ref into three-phase voltage commands Vu1, Vv1, Vw1 and Vu2, Vv2, Vw2 for the primary and secondary inverters 2 and 3 using equation (2) shown in Fig. 24 based on the magnetic pole position θ. The three-phase voltage commands for the primary and secondary sides have a phase difference of 180 degrees from each other.

[0019] The magnetic pole position estimation unit 12 estimates the motor speed ω and the rotor magnetic pole position θ based on three current change amounts dIu_V52, dIv_V52, and dIw_V14, which have a phase difference of 120 degrees and are detected by the phase current and current change amount detection unit 16, based on a principle described below. Note that V52 and V14 are composite voltage vectors of the primary-side inverter 2 and the secondary-side inverter 3. FIG. 7 shows the 8 × 8 = 64 voltage vectors that can be output by two inverters in an open-winding motor drive system. For example, the combination in which the primary-side inverter 2 represents vector V1 and the secondary-side inverter 3 represents vector V4 is represented as vector V14. The magnetic pole position θ is input to a coordinate transformation unit (uvw / dqz) 15 and a coordinate transformation unit (dqz / uvw) 18.

[0020] The PWM signal generator 19 determines the duty ratio, which is the ratio of the pulse width of the PWM signal for each phase, based on the three-phase voltage commands Vu1, Vv1, Vw1, Vu2, Vv2, Vw2 and DC voltage Vdc for the primary and secondary inverters 2 and 3. Based on these duty ratios, the PWM signal generator 19 generates three-phase high-side and low-side PWM signals U1, V1, W1, X1, Y1, Z1 and U2, V2, W2, X2, Y2, Z2 for the primary and secondary inverters 2 and 3, and outputs gate drive signals to the primary and secondary inverters 2 and 3. The primary and secondary inverters 2 and 3 generate three-phase AC voltages for driving the motor 1 on their primary and secondary sides based on the supplied DC power source 4 and the gate drive signals provided, and apply these voltages to the winding terminals of the motor 1. The configuration of the PWM signal generator 19 will be described in detail later.

[0021] The current sensor 5 detects the three-phase currents flowing through the motor 1 and outputs a signal corresponding to the detection result; for example, a CT (Current Transistor) can be used. The phase current and current change amount detection unit 16 detects the three-phase current signals output by the current sensor 5 by A / D conversion at four fixed timing points at the leading and trailing ends of the PWM carrier cycle. The detection process for this phase current will be described in detail later.

[0022] FIG. 3 shows the configuration of an air conditioner, which is a heat pump device to which the motor-driven system of this embodiment can be applied. Heat pump devices to which the motor-driven system can be applied include, in addition to air conditioners, hot water generators serving as water heaters and hot and cold water generators such as chillers. The air conditioner 21 comprises refrigerant piping and signal communication lines connecting an indoor unit 24 and an outdoor unit 35. The indoor unit 24, which is installed indoors, houses an indoor heat exchanger 27 and an indoor fan 30 inside. Meanwhile, the outdoor unit 35 is located outdoors and houses devices such as the control unit 20, compressor 22, outdoor heat exchanger 29, four-way valve 26, pressure reducing device 28, outdoor fan 31, and outdoor fan motor 53.

[0023] Compressor 22 is configured by housing compression section 23 and motor 10 in the same iron hermetic container 25, and the rotor shaft of motor 10 is connected to compression section 23. Compressor 22, four-way valve 26, indoor heat exchanger 27, pressure reducing device 28, and outdoor heat exchanger 29 are connected to form a closed loop by pipes that serve as refrigerant passages. Note that compressor 22 is, for example, a single-cylinder rotary compressor, but is not limited to this and multi-cylinder rotary compressors, scroll compressors, and reciprocating compressors can also be used.

[0024] During heating, the four-way valve 26 is in the state shown by the solid line, and the high-temperature refrigerant compressed in the compression section 23 of the compressor 22 is supplied from the four-way valve 26 to the indoor heat exchanger 27, where it condenses and releases heat into the room, heating the room. It is then decompressed by the pressure reducing device 28, becomes colder, and flows to the outdoor heat exchanger 29, where it absorbs heat from the outside air, evaporates, and returns to the compressor 22.

[0025] On the other hand, during cooling, the four-way valve 26 is switched to the state shown by the dashed line. Therefore, the high-temperature refrigerant compressed in the compression section 23 of the compressor 22 is supplied from the four-way valve 26 to the outdoor heat exchanger 29, where it condenses by releasing heat to the outdoors. It is then decompressed by the pressure reducing device 28, becomes low temperature, and flows to the indoor heat exchanger 27, where the refrigerant evaporates by absorbing heat from the indoor air, cooling the room and returning to the compressor 22. Air is then blown to the indoor and outdoor heat exchangers 27, 29 by an indoor fan 30 and an outdoor fan 31, respectively, so that the air blown therethrough efficiently exchanges heat between the indoor air and the outdoor air in each of the heat exchangers 27, 29.

[0026] In this embodiment, the PWM signal generating unit 19 generates high-frequency current ripples that allow the magnetic pole position to be detected, as shown in Fig. 4. In the figure, the upper side shows the carrier waveforms and PWM signals for each phase of the primary-side inverter 2, and the lower side shows the carrier waveforms and PWM signals for each phase of the secondary-side inverter 3. Note that, hereinafter, the primary-side and secondary-side inverters 2 and 3 may be referred to as a dual inverter.

[0027] First, the on / off patterns of the dual inverter that enable detection of the magnetic pole position in an open-winding motor drive system will be explained using Fig. 5. The lower part of Fig. 5 shows the three-phase motor current in terms of the carrier cycle at a certain timing while the motor 1 is being driven by the PWM signal generator 19, around 50 ms as shown in Fig. 6. The upper part of the same figure shows the on / off patterns of the dual inverter corresponding to the motor current ripple when the composite voltage vector of the dual inverter is V52 or V14.

[0028] The open-winding motor 1 has three independent windings. By generating an on-off pattern for each phase winding, as shown in Figure 5, a high-frequency current ripple that allows magnetic pole position detection is obtained. Focusing on the U phase, for example, the high-side PWM signal of the primary inverter 2 is off and the high-side PWM signal of the secondary inverter 3 is on. This pattern results in a negative-sloping current gradient, assuming the current flow direction from the primary inverter 2 to the secondary inverter 3 is positive. Figure 4 shows a three-phase PWM signal pattern that can be generated for the V and W phases within the carrier cycle, with this negative-sloping current gradient. It is also possible to generate a three-phase pattern with a positive-sloping current gradient by turning on the high-side PWM signal of the primary inverter 2 and turning off the high-side PWM signal of the secondary inverter 3. Hereafter, we will explain the negative-sloping current gradient pattern.

[0029] As shown in Fig. 4, in order to generate the on / off patterns shown in Fig. 5 for three phases within a carrier cycle, the U-phase carrier of the primary-side inverter 2 is an inverted sawtooth wave, the V-phase carrier is a triangular wave, and the W-phase carrier is a sawtooth wave. Furthermore, the secondary-side inverter 3 uses carriers that are in opposite phase relationship to the carriers of each phase of the primary-side inverter 2. In this way, by changing the base point of the PWM signal for each phase and the direction of increase / decrease of the signal pulse according to the duty ratio, composite voltage vectors V52 and V14 are generated at the leading and trailing ends of the carrier cycle.

[0030] Focusing on the area enclosed by the dashed line, the U and V phases of vector V52 and the W phase of vector V14 have the on-off pattern shown in Figure 5. Therefore, for these vectors V52 and V14, A / D conversion triggers 0 to 3, which are the current detection timings shown in Figure 4, are set, and the three-phase current signals output by current sensor 5 are detected four times within the carrier period. This results in three current change amounts dIu_V52, dIv_V52, and dIW_V14, which allow the magnetic pole position to be detected. In addition, one of the four detected values ​​is used for the three-phase motor currents used for vector control. The magnetic pole position θ is estimated using equation (3) shown in Figure 25.

[0031] The magnetic pole position θ estimated by equation (3) is obtained by dividing 2θ, and in principle has an error of ±180 degrees. Therefore, in order to determine the above error when the motor is stopped before it is driven, an algorithm for identifying the initial position is required. This is determined using a method that uses magnetic saturation characteristics, which is a conventional well-known technique. Examples of this well-known technique include the following techniques: IEEJ Transactions on Industrial Applications, Vol. 125 (2005), No. 3, "Initial Rotational Position Estimation Method for Surface Permanent Magnet Synchronous Motors Using Pulse Voltage", Osamu Yamamoto, Takahiro Ara Then, the estimated speed ω is calculated by time-differentiating the estimated position with respect to the control period.

[0032] FIG. 6 shows the current change, motor current, and zero-phase current versus electrical angle obtained by simulation verification of this embodiment. It can be seen that the current change having a 2θ component is obtained versus the electrical angle, making it possible to estimate the magnetic pole position. This simulation verification targets the control device 10 shown in FIG. 1, but aims to confirm whether the current change can be obtained from the detected current signal; the speed and angle are detected by a position sensor. Therefore, feedback control based on the estimated value obtained by the magnetic pole position estimator 12 is not performed. Furthermore, the zero-phase current suppressor 17, which suppresses the zero-phase current of the 3f component shown in FIG. 4, is not operated, and the zero-phase current command V0ref is zero.

[0033] As described above, according to this embodiment, when the motor 1 with an open winding structure is driven by the primary side inverter 2 and the secondary side inverter 3, the control device 10 detects the phase currents Iu_detect, Iv_detect, and IW_detect flowing through the motor 1 using the current sensor 5 and the phase current and current change amount detector 16. Furthermore, the current change amount calculator 16 calculates the difference between the current values ​​of each phase detected twice and outputs the difference as current changes dIu_V52, dIv_V52, and dIW_V14. The magnetic pole position estimator 12 estimates the magnetic pole position θ of the motor 1 based on these current changes.

[0034] The speed control units 11 and 11 and the current control unit 14 control the current flowing through the motor 1 and the rotational speed, and the PWM signal generation unit 19 compares the three-phase duty ratios calculated based on the voltage commands Vu1, Vv1, Vw1 and Vu2, Vv2, Vw2 obtained from the current control unit 14 via the coordinate conversion unit 18 and the DC voltage Vdc with a carrier wave, and generates three-phase PWM signals for the primary and secondary side inverters 2 and 3, respectively.

[0035] The PWM signal generator 19 increases or decreases the pulse width of the V phase of the primary inverter 2 in both the leading and lagging directions relative to the center phase of the carrier cycle, the U phase in the lagging direction relative to the trailing end phase of the carrier cycle, and the W phase in the leading direction relative to the leading end phase of the carrier cycle. The secondary inverter 3 increases or decreases the pulse width of the V phase in both the leading and lagging directions relative to the leading end phase of the carrier cycle, the U phase in the leading direction relative to the trailing end phase of the carrier cycle, and the W phase in the lagging direction relative to the leading end phase of the carrier cycle. The current change detector 16 detects the differences in the current values ​​of the three phases twice, near the leading and trailing ends of the carrier cycle.

[0036] With this configuration, three-phase PWM signals for the two inverters 2 and 3 can be generated so that three-phase currents required for estimating the magnetic pole position θ can be reliably detected within the carrier period in PWM control.

[0037] (Second embodiment) In the following, the same parts as in the first embodiment are assigned the same reference numerals and their descriptions are omitted, and only the differences will be described. The second embodiment will be described with reference to Figures 8 to 18. In a motor drive system with a shared DC link and open winding structure, a PWM signal that generates the harmonic current amplitude required for magnetic pole position estimation generates a zero-phase voltage, which is the difference between the common-mode voltage of the primary and secondary inverters, that is, the average value of the three-phase output voltage. This zero-phase voltage causes a zero-phase current to flow in the same direction through the three-phase windings of the motor 1 in the path that returns current between the dual inverter and the motor 1, thereby reducing the accuracy of position estimation.

[0038] FIG. 17 shows the zero-phase-sequence voltage and zero-phase-sequence current of the carrier frequency component generated by the PWM signal of the first embodiment. The zero-phase-sequence voltage depends on the total number of on-states of the primary-side inverter and the total number of on-states of the secondary-side inverter, and occurs in positive and negative forms when the total number of on-states differs. Therefore, the zero-phase-sequence voltage waveform shown in FIG. 17 fluctuates in positive and negative forms depending on the switching state, and the zero-phase-sequence current increases when the zero-phase-sequence voltage is generated on the positive side and decreases when the zero-phase-sequence voltage is generated on the negative side. This zero-phase-sequence current of the carrier component disturbs the amount of change in the motor phase current, which provides position information, and therefore needs to be suppressed.

[0039] A method for suppressing the zero-phase current of the carrier component is disclosed in, for example, Japanese Patent Application Laid-Open No. 2022-173520, which is hereinafter referred to as the reference publication, in which only a triangular wave is used for the three-phase carrier to simultaneously suppress the zero-phase current of the carrier component and the 3f component.

[0040] Figure 18 shows conventional PWM signal generation to suppress the zero-phase current of the carrier component, and space vector modulation. The dq-axis voltage is converted to the αβ-axis voltage using equation (4) shown in Figure 25, and the generation time of the voltage vector for each sector is determined from the αβ-axis voltage, and a PWM signal is output.

[0041] Of the 64 voltage vector patterns shown in Figure 7, there are 12 voltage vectors that generate a motor voltage but do not generate a zero-phase sequence voltage. In other words, the total number of ON voltage vectors of the primary-side and secondary-side inverters 2 and 3 is equal. These are V15, V24, V26, V35, V31, V46, V42, V51, V64, and V13. These correspond to the first switching pattern. A regular hexagon is drawn with the vertices of the 12 voltage vectors arranged in pairs, centered on the voltage vectors V00 and V77 corresponding to the second switching pattern, and the hexagon is divided into six sectors. This is shown in Figure 18. Because the total number of ON voltage vectors of the primary-side inverter 2 and the secondary-side inverter 3 is the same, the zero-phase sequence voltage becomes zero, suppressing the ripple of the zero-phase sequence current of the carrier component.

[0042] However, although the technique of the reference publication can suppress the zero-phase current of the carrier component, it cannot obtain the on / off pattern that enables detection of the magnetic pole position shown in Fig. 5. Therefore, in the second embodiment, the on / off pattern shown in Fig. 5 required for magnetic pole position estimation is obtained while suppressing the zero-phase current of the carrier component within the carrier cycle in PWM control, and a PWM signal that enables detection of the amount of change in three-phase current is generated.

[0043] The control device 20 of the second embodiment shown in FIG. 8 differs from the control device 10 of the first embodiment in the following points. The coordinate transformation unit (dqz / uvw) 18 is changed to a coordinate transformation unit (dq / αβ) 21. The zero-phase voltage command V0ref, which suppresses the 3f component, which is three times the carrier frequency, is not input to the coordinate transformation unit (dq / αβ) 21.

[0044] The PWM signal generating unit 19 is changed to a PWM signal generating unit 22. In the PWM signal generating unit 22, the method shown in Fig. 12 is used instead of equation (1) shown in Fig. 4. Accordingly, the zero-phase sequence voltage command V0ref is input. The phase current and current change amount detection unit 16 is replaced with a phase current and current change amount detection unit 23. The phase current and current change amount detection unit 23 receives sector information and information on A / D conversion triggers 0 to 5 from the PWM signal generation unit 22. The method of reflecting the zero-axis voltage command V0ref in the PWM signal is described in the reference publication.

[0045] The magnetic pole position estimation unit 12 has been changed to a magnetic pole position estimation unit 24. In the phase current and current change amount detection unit 23, the relationship between the A / D conversion triggers 0 to 5 and the detected current phase changes depending on the sector. Therefore, the phase current input to the coordinate conversion unit (dq / αβ) 21 and the current change amount input to the magnetic pole position estimation unit 24 need to be distributed depending on the sector.

[0046] Figure 12 shows the three-phase PWM signals of the dual inverter according to the sector, with the left side showing the conventional method shown in Figure 18, and the right side showing the phase of the PWM signal shifted by the method described below based on the conventional method. Phase shift methods can be classified according to whether the sector is even or odd, and the specific processing method is shown in the flowcharts of Figures 9 to 11.

[0047] This flowchart begins after the PWM signal is calculated using space vector modulation, a conventional method that does not generate zero-phase sequence voltages. That is, the d-axis voltage Vd and q-axis voltage Vq are calculated using vector control, and the three-phase duty ratios of the primary and secondary inverters 2 and 3 are calculated using a conventional method that suppresses the zero-phase sequence current of the carrier component shown in Figure 17. This flowchart then shows the phase shift processing of the primary PWM signal, how to set A / D conversion triggers 0 to 5 according to the duty, and how to set the secondary PWM signal.

[0048] First, difference 1 between the middle and minimum phases of the PWM signal, and difference 2 between the middle and maximum phases are calculated (S1). Shift 1 of the middle phase of the even sector is calculated as the amount of phase shift to the right based on difference 1 and difference 2 and the setting value of the section for detecting the amount of current change (S2). Difference 2 is then updated by subtracting shift 1 from difference 2 (S3). Next, shift 2 of the maximum phase of the even sector (S4), shift 4 of the middle phase of the odd sector (S5), and shift 5 of the maximum phase are calculated (S6).

[0049] In the case of an even sector, such as sector 0 (Yes in S7), during the first half of the carrier period (Yes in S8), the maximum and middle phases are phase-shifted to the right based on shifts 2 and 1 to generate V13 and V24 (S9). Next, the difference 3 between the minimum phase and the maximum phase after the phase shift is calculated (S10), and shift 3 is calculated as the amount of phase shift to the left based on the setting value of the section for detecting the amount of current change (S11). Finally, based on shift 3, the minimum phase is phase-shifted to the left as necessary to generate V51 and V46 (S12). A / D conversion triggers 0 and 1 for the first half of the carrier are calculated based on the minimum phase after the phase shift (S13).

[0050] In the second half of the carrier cycle (No in S8), the maximum and middle phases are phase shifted to the right, and the minimum phase is phase shifted to the left as needed, based on shift 2, shift 1, and shift 3 calculated in the first half of the carrier (S15). Then, A / D conversion triggers 2 to 5 in the second half of the carrier are calculated based on the middle and maximum phases after the phase shift (S16).

[0051] In the case of an odd sector, for example sector 1 (No in S7), in the first half of the carrier period (Yes in S17), phase shift processing is performed to the right of the maximum phase and middle phase based on shifts 5 and 4 to generate V42 and V53 (S18). Next, calculations are performed on A / D conversion triggers 0 to 3 in the first half of the carrier period based on the minimum phase and the middle phase after the phase shift (S19). Similarly, in the second half of the carrier period (No in S17), phase shift processing is performed to the right of the maximum phase and middle phase based on shifts 5 and 4 to generate V26 and V15 (S20). Then, calculations are performed on A / D conversion triggers 4 and 5 in the second half of the carrier period based on the maximum phase after the phase shift (S21).

[0052] This completes the phase shift process for the PWM signal of the primary-side inverter 2 and the setting of A / D conversion triggers 0 to 5 according to the duty ratio. Finally, the PWM signal of the primary-side inverter for which phase shifting has been completed is allocated to the PWM signal of the secondary-side inverter according to the sector (S14). As a result, the phases of the three-phase PWM signals of the primary-side and secondary-side inverters 2 and 3 are shifted in synchronization with the maximum, middle, and minimum phases classified by magnitude, so as to provide time for detecting the amount of current change.

[0053] Although not shown in the flowchart, in odd-numbered sectors where the maximum and middle phases are phase-shifted to the right, the phase shift amount is larger than in even-numbered sectors. Therefore, if the PWM signal after phase shifting does not fit within the carrier period, all three phases can be finally phase-shifted to the left. This is shown as shift 6 in Figure 12. As a result, in an open-winding motor drive system, a PWM signal is generated that suppresses zero-sequence current due to zero-sequence voltage and provides a current change amount that allows the magnetic pole position to be detected.

[0054] Figures 13 and 14 are enlarged views of Figure 12 for sector 0. Focusing on the area enclosed by the dashed line, it can be seen that within the carrier cycle, the U phase follows the on-off pattern shown in Figure 2 during the first half of the carrier, and the V and W phases follow the on-off pattern shown in Figure 2 during the second half. By setting A / D conversion triggers 0 to 5 according to the duty ratio at these timings, the current change amounts dIu_VU, dIV_VV, and dIw_VW, which allow for the detection of the magnetic pole position, can be obtained. VU, VV, and VW are voltage vectors, where VU is V46, V31, V42, or V51; VV is V13, V62, V53, or V64; and VW is V24, V15, V26, or V35, which vary depending on the sector. The magnetic pole position θ is estimated using equation (5) shown in Figure 25.

[0055] FIG. 15 shows the current change, motor current, and zero-phase current versus electrical angle obtained by simulating the second embodiment. It can be seen that the current change, which has a 2θ component, is obtained versus the electrical angle, making it possible to estimate the magnetic pole position. However, when focusing on the PWM signals of the primary and secondary inverters for each phase, it can be seen that at the point where the sector switches, the phase relationship between the PWM signals changes significantly, causing current distortion and resulting in a section where the current change cannot be accurately detected. Therefore, for this section after the sector switch, it is sufficient to mask the position estimation function by interpolating based on the estimated speed and angle immediately before the sector switch.

[0056] In the above simulation, the zero-phase current suppression unit 17, which suppresses the zero-phase current of the 3f component, was not operated, as in Fig. 6 of the first embodiment. Fig. 16 shows the simulation results when the zero-phase current suppression unit 17 was operated. Compared with Fig. 15, it can be seen that the 3f component, which is three times the fundamental frequency of the motor current, was suppressed.

[0057] As described above, according to the second embodiment, the zero-phase current suppression unit 17 suppresses the zero-phase current flowing in the same direction through the three-phase windings of the motor 1, using the path between the primary-side and secondary-side inverters 2 and 3 and the motor 1. For the 64 voltage vectors that are combinations of the on / off patterns of the primary-side and secondary-side inverters 2 and 3, the PWM signal generation unit 22 divides a hexagonal space vector into six sectors, with the point where the second switching patterns V00 and V77 are located at the center and the points where two of the first switching patterns are located at each vertex.

[0058] The three-phase duty ratios of the primary and secondary inverters 2 and 3 of each sector are categorized by magnitude into maximum, medium, and minimum phases, and the phases are shifted synchronously on the primary and secondary sides so that the current change amount detection unit 23 has time to detect the difference in the three-phase current values ​​twice. With this configuration, the magnetic pole position θ can be estimated more accurately than in the first embodiment.

[0059] (Third embodiment) In the third embodiment shown in Fig. 19, the DC power supply 4 is a power supply dedicated to the primary-side inverter 2, and an independent DC power supply 7 is provided for the secondary-side inverter 3. In this configuration, no zero-phase current flows between the primary-side inverter 2 and the secondary-side inverter 3, so the zero-phase current suppression unit 17 is eliminated from the control device 30 shown in Fig. 20. Also, instead of the magnetic pole position estimator 12 and the phase current and current change amount detector 16, a magnetic pole position estimator 31 and a phase current and current change amount detector 32 are provided. The current change amount detector 32 calculates a current change amount dIv_V14 instead of the current change amount dIw_V14 and outputs it to the magnetic pole position estimator 31.

[0060] In the PWM signals of the primary and secondary inverters 2 and 3 shown in Figure 22, the on / off pattern for detecting the current change amount is V-phase pulses V1 and V2 instead of W-phase pulses W1 and W2, and the current change amount dIv_V14 is obtained by A / D conversion triggers 2 and 3.

[0061] The simulation results shown in FIG. 23 show that a current change amount having a 2θ component with respect to the electrical angle can be obtained from the three current change amounts dIu_V52, dIv_V52, and dIv_V14, and that the magnetic pole position can be estimated. Although several embodiments of the present invention have been described, these embodiments are presented as examples and are not intended to limit the scope of the invention. These novel embodiments can be embodied in various other forms, and various omissions, substitutions, and modifications can be made without departing from the spirit of the invention. These embodiments and their modifications are included within the scope and spirit of the invention, and are also included in the scope of the invention and its equivalents as defined in the claims. [Explanation of symbols]

[0062] In the drawing, 1 indicates a motor, 2 indicates a primary side inverter, 3 indicates a secondary side inverter, 4 indicates a DC power supply, 5 indicates a current sensor, 10 indicates a control device, 11 indicates a speed control unit, 12 indicates a magnetic pole position estimation unit, 14 indicates a current control unit, 16 indicates a phase current and current change amount detection unit, 17 indicates a zero-phase current suppression unit, and 19 indicates a PWM signal generation unit.

Claims

1. a primary-side inverter connected to three output terminals of a motor having an open winding structure, the three phase windings of which are independent from each other and have six output terminals; a secondary-side inverter connected to the remaining three output terminals of the motor; a current detection unit that detects a three-phase current flowing through the motor; a current change amount detection unit that outputs the difference between the two detected three-phase current values ​​as a current change amount; a magnetic pole position estimating unit that estimates a magnetic pole position of the motor based on the amount of change in current; a control unit for controlling the current supplied to the motor and the rotation speed; a zero-phase current suppression unit that suppresses zero-phase currents flowing in the same direction through three-phase windings of the motor along paths between the primary and secondary inverters and the motor; a PWM signal generating unit that compares a three-phase duty ratio calculated based on a command voltage and a DC voltage obtained from the control unit and the zero-phase current suppressing unit with a carrier wave, and generates three-phase PWM signals for each of the primary and secondary inverters, the PWM signal generating unit divides a hexagonal space vector into six sectors, with the 64 voltage vectors being combinations of on / off patterns of the primary and secondary inverters, the hexagonal space vector having a center at a point where two second switching patterns that do not generate a zero-phase sequence voltage and do not generate a voltage acting between the phases of the motor are located and vertices at points where two first switching patterns that do not generate a zero-phase sequence voltage and generate a voltage acting between the phases of the motor are located; An open winding motor drive device that shifts the phases synchronously on the primary and secondary sides so that the current change amount detection unit detects the difference in the three-phase current values ​​twice for each of the maximum, intermediate, and minimum phases classified by the duty ratio of the three phases of the primary and secondary inverters in each sector.

2. a primary-side inverter connected to three output terminals of a motor having an open winding structure, the three phase windings of which are independent from each other and have six output terminals; a secondary-side inverter connected to the remaining three output terminals of the motor; a current detection unit that detects a three-phase current flowing through the motor; a current change amount detection unit that outputs the difference between the two detected three-phase current values ​​as a current change amount; a magnetic pole position estimating unit that estimates a magnetic pole position of the motor based on the amount of change in current; a control unit for controlling the current supplied to the motor and the rotation speed; a PWM signal generating unit that compares a three-phase duty ratio calculated based on the command voltage and the DC voltage obtained from the control unit with a carrier wave and generates three-phase PWM signals for each of the primary and secondary inverters, the PWM signal generation unit increases or decreases a pulse width of a first phase of the three-phase PWM signal of the primary side inverter in both directions of a delay side and a lead side based on a phase of the center or front and rear ends of a carrier cycle; The second phase is a phase in which the pulse width is increased or decreased in one direction, either the delayed side or the advanced side, based on the phase of the leading end or the trailing end of the carrier cycle; The third phase increases or decreases the pulse width in one direction opposite to the second phase, based on a phase opposite to the second phase. Regarding the three-phase PWM signal of the secondary side inverter, the first phase has a pulse width increased or decreased in both directions of the delay side and the advance side based on a phase different from the first phase of the primary side inverter; The second and third phases increase and decrease the pulse width in the opposite direction with respect to the same phases as the second and third phases of the primary side inverter, respectively; The current change amount detection unit detects the difference in current values ​​of three or two phases twice near the leading and trailing ends of a carrier cycle.

3. a first power source that supplies driving power to the primary side inverter; a second power supply that supplies driving power to the secondary-side inverter, 3. The open winding motor drive device according to claim 2, wherein the current change amount detector detects the difference between the two phase current values ​​twice, near the leading and trailing ends of the carrier cycle.

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